This is a semi modular chassis which is mounted on the left side of the cabinet.
The power supply unit is mounted on the bottom floor of the cabinet.
All functions of the receiver are summed up in the left chassis the GRUNDIG CUC70KT which is the successor of the CUC720 seen HERE
A very compact CHASSIS THE GRUNDIG CUC70KT for that era and was used even for models of 26 Inches (66Cm) sets with different wooden cabinet colors types.
Highly and friendly serviceability we can see here, and it's not highly complex.
This is a semi modular chassis which is mounted on the left side of the cabinet.
The power supply unit is mounted on the bottom floor of the cabinet.
All functions of the receiver are summed up in the left chassis the GRUNDIG CUC70KT which is the successor of the CUC720 seen HERE
A very compact CHASSIS THE GRUNDIG CUC70KT for that era and was used even for models of 26 Inches (66Cm)and smaller sets and portable types with different wooden cabinet colors types.
Highly and friendly serviceability we can see here, and it's not highly complex.
GRUNDIG SUPER COLOR C8443 SERIE F3026 " Berlino 26 " CHASSIS CUC70KT PCB (29701-029.01) (EINBAU CHASSIS) VIEW
It's a TDA4600 (SIEMENS) BASED SMPS Circuit which delivers +151V and St-by supply.
Power supply Description based on TDA4601d (SIEMENS)
TDA4601 Operation. * The TDA4601 device is a single in line, 9 pin chip. Its predecessor was the TDA4600 device, the TDA4601 however has improved switching, better protection and cooler running. The (SIEMENS) TDA4601 power supply is a fairly standard parallel chopper switch mode type, which operates on the same basic principle as a line output stage. It is turned on and off by a square wave drive pulse, when switched on energy is stored in the chopper transformer primary winding in the form of a magnetic flux; when the chopper is turned off the magnetic flux collapses, causing a large back emf to be produced. At the secondary side of the chopper transformer this is rectified and smoothed for H.T. supply purposes. The advantage of this type of supply is that the high chopping frequency (20 to 70 KHz according to load) allows the use of relatively small H.T. smoothing capacitors making smoothing easier. Also should the chopper device go short circuit there is no H.T. output. In order to start up the TDA4601 I.C. an initial supply of 9v is required at pin 9, this voltage is sourced via R818 and D805 from the AC side of the bridge rectifier D801, also pin 5 requires a +Ve bias for the internal logic block. (On some sets pin 5 is used for standby switching). Once the power supply is up and running, the voltage on pin 9 is increased to 16v and maintained at this level by D807 and C820 acting as a half wave rectifier and smoothing circuit. PIN DESCRIPTIONS Pin 1 This is a 4v reference produced within the I.C. Pin 2 This pin detects the exact point at which energy stored in the chopper transformer collapses to zero via R824 and R825, and allows Q1 to deliver drive volts to the chopper transistor. It also opens the switch at pin 4 allowing the external capacitor C813 to charge from its external feed resistor R810. Pin 3 H.T. control/feedback via photo coupler D830. The voltage at this pin controls the on time of the chopper transistor and hence the output voltage. Normally it runs at Approximately 2v and regulates H.T. by sensing a proportion of the +4v reference at pin 1, offset by conduction of the photo coupler D830 which acts like a variable resistor. An increase in the conduction of transistor D830 and therefor a reduction of its resistance will cause a corresponding reduction of the positive voltage at Pin 3. A decrease in this voltage will result in a shorter on time for the chopper transistor and therefor a lowering of the output voltage and vice versa, oscillation frequency also varies according to load, the higher the load the lower the frequency etc. should the voltage at pin 3 exceed 2.3v an internal flip flop is triggered causing the chopper drive mark space ratio to extend to 244 (off time) to 1 (on time), the chip is now in over volts trip condition. Pin 4 At this pin a sawtooth waveform is generated which simulates chopper current, it is produced by a time constant network R810 and C813. C813 charges when the chopper is on and is discharged when the chopper is off, by an internal switch strapping pin 4 to the internal +2v reference, see Fig 2. The amplitude of the ramp is proportional to chopper drive. In an overload condition it reaches 4v amplitude at which point chopper drive is reduced to a mark-space ratio of 13 to 1, the chip is then in over current trip. The I.C. can easily withstand a short circuit on the H.T. rail and in such a case the power supply simply squegs quietly. Pin 4 is protected by internal protection components which limit the maximum voltage at this pin to 6.5v. Should a fault occur in either of the time constant components, then the cho
pper transistor will probably be destroyed. Pin 5 This pin can be used for remote control on/off switching of the power supply, it is normally held at about +7v and will cause the chip to enter standby mode if it falls below 2v. Pin 6 Ground. Pin 7 Chopper switch off pin. This pin clamps the chopper drive voltage to 1.6v in order to switch off the chopper. Pin 8 Chopper base current output drive pin. Pin 9 L.T. pin, approximately 9v under start-up conditions and 16v during normal running, Current consumption of the I.C. is typically 135mA. The voltage at this pin must reach 6.7v in order for the chip to start-up.
Semiconductor circuit for supplying power to electrical equipment, comprising a transformer having a primary winding connected, via a parallel connection of a collector-emitter path of a transistor with a first capacitor, to both outputs of a rectifier circuit supplied, in turn, by a line a-c voltage; said transistor having a base controlled via a second capacitor by an output of a control circuit acted upon, in turn by the rectified a-c line voltage as actual value and by a reference voltage; said transformer having a first secondary winding to which the electrical equipment to be supplied is connected; said transformer having a second secondary winding with one terminal thereof connected to the emitter of said transistor and the other terminal thereof connected to an anode of a first diode leading to said control circuit; said transformer having a third secondary winding with one terminal thereof connected, on the one hand, via a series connection of a third capacitor with a first resistance, to the other terminal of said third secondary winding and connected, on the other hand, to the emitter of said transistor, the collector of which is connected to said primary winding; a point between said third capacitor and said first resistance being connected to the cathode of a second diode; said control circuit having nine terminals including a first terminal delivering a reference voltage and connected, via a voltage divider formed of a third and fourth series-connected resistances, to the anode of said second diode; a second terminal of said control circuit serving for zero-crossing identification being connected via a fifth resistance to said cathode of said second diode; a third terminal of said control-circuit serving as actual value input being directly connected to a divider point of said voltage divider forming said connection of said first terminal of said control circuit to said anode of said second diode; a fourth terminal of said control circuit delivering a sawtooth voltage being connected via a sixth resistance to a terminal of said primary winding of said transformer facing away from said transistor; a fifth terminal of said control circuit serving as a protective input being connected, via a seventh resistance to the cathode of said first diode and, through the intermediary of said seventh resistance and an eighth resistance, to the cathode of a third diode having an anode connected to an input of said rectifier circuit; a sixth terminal of said control circuit carrying said reference potential and being connected via a fourth capacitor to said fourth terminal of said control circuit and via a fifth capacitor to the anode of said second diode; a seventh terminal of said control circuit establishing a potential for pulses controlling said transistor being connected directly and an eighth terminal of said control circuit effecting pulse control of the base of said transistor being connected through the intermediary of a ninth resistance to said first capacitor leading to the base of said transistor; and a ninth terminal of said control circuit serving as a power supply input of said control circuit being connected both to the cathode of said first diode as well as via the intermediary of a sixth capacitor to a terminal of said second secondary winding as well as to a terminal of said third secondary winding.
Description:
The
invention relates to a blocking oscillator type switching power supply
for supplying power to electrical equipment, wherein the primary winding
of a transformer, in series with the emitter-collector path of a first
bipolar transistor, is connected to a d-c voltage obtained by
rectification of a line a-c voltage fed-in via two external supply
terminals, and a secondary winding of the transformer is provided for
supplying power to the electrical equipment, wherein, furthermore, the
first bipolar transistor has a base controlled by the output of a
control circuit which is acted upon in turn by the rectified a-c line
voltage as actual value and by a set-point transmitter, and wherein a
starting circuit for further control of the base of the first bipolar
transistor is provided. Such a blocking oscillator switching
power supply is described in the German periodical, "Funkschau" (1975)
No. 5, pages 40 to 44. It is well known that the purpose of such a
circuit is to supply electronic equipment, for example, a television
set, with stabilized and controlled supply voltages. Essential for such
switching power supply is a power switching transistor i.e. a bipolar
transistor with high switching speed and high reverse voltage. This
transistor therefore constitutes an important component of the control
element of the control circuit. Furthermore, a high operating frequency
and a transformer intended for a high operating frequency are provided,
because generally, a thorough separation of the equipment to be supplied
from the supply naturally is desired. Such switching power supplies may
be constructed either for synchronized or externally controlled
operation or for non-synchronized or free-running operation. A blocking
converter is understood to be a switching power supply in which power is
delivered to the equipment to be supplied only if the switching
transistor establishing the connection between the primary coil of the
transformer and the rectified a-c voltage is cut off. The power
delivered by the line rectifier to the primary coil of the transformer
while the switching transistor is open, is interim-stored in the
transformer and then delivered to the consumer on the secondary side of
the transformer with the switching transistor cut off. In the blocking converter described in the aforementioned reference in the literature, "Funkschau" (1975), No. 5, Pages 40 to 44, the power switching transistor is connected in the manner defined in the introduction to this applic
ation. In addition, a so-called starting
circuit is provided. Because several diodes are generally provided in
the overall circuit of a blocking oscillator according to the definition
provided in the introduction hereto, it is necessary, in order not to
damage these diodes, that due to the collector peak current in the case
of a short circuit, no excessive stress of these diodes and possibly
existing further sensitive circuit parts can occur. Considering the operation of a blocking oscillator, this means that, in the event of a short circuit, the number of collector current pulses per unit time must be reduced. For this purpose, a control and regulating circuit is provided. Simultaneously, a starting circuit must bring the blocking converter back to normal operation when the equipment is switched on, and after disturbances, for example, in the event of a short circuit. The starting circuit shown in the literature reference "Funkschau" on Page 42 thereof, differs to some extent already from the conventional d-c starting circuits. It is commonly known for all heretofore known blocking oscillator circuits, however, that a thyristor or an equivalent circuit replacing the thyristor is essential for the operation of the control circuit.
It is accordingly an object of the invention to provide another starting circuit. It is a further object of the invention to provide a possible circuit for the control circuit which is particularly well suited for this purpose. It is yet another object of the invention to provide such a power supply which is assured of operation over the entire range of line voltages from 90 to 270 V a-c, while the secondary voltages and secondary load variations between no-load and short circuit are largely constant.
With
the foregoing and other objects in view, there is provided, in
accordance with the invention, a blocking oscillator-type switching
power supply for supplying power to electrical equipment wherein a
primary winding of a transformer, in series with an emitter-collector
path of a first bipolar transistor, is connected to a d-c voltage
obtained by rectification of a line a-c voltage fed-in via two external
supply terminals, a secondary winding of the transformer being
connectible to the electrical equipment for supplying power thereto, the
first bipolar transistor having a base controlled by the output of a
control circuit acted upon, in turn, by the rectified a-c line voltage
as actual value and by a set-point transmitter, and including a starting
circuit for further control of the base of the first bipolar
transistor, including a first diode in the starting circuit having an
anode directly connected to one of the supply terminals supplied by the
a-c line voltage and a cathode connected via a resistor to an input
serving to supply power to the control circuit, the input being directly
connected to a cathode of a second diode, the second diode having an
anode connected to one terminal of another secondary winding of the
transformer, the other secondary winding having another terminal
connected to the emitter of the first bipolar transmitter. In accordance with another feature of the invention, there is provided a second bipolar transistor having the same conduction type as that of the first bipolar transistor and connected in the starting circuit with the base thereof connected to a cathode of a semiconductor diode, the semiconductor diode having an anode connected to the emitter of the first bipolar transistor, the second bipolar transistor having a collector connected via a resistor to a cathode of the first diode in the starting circuit, and having an emitter connected to the input serving to supply power to the control circuit and also connected to the cathode of the second diode which is connected to the other secondary winding of the transformer.
In accordance with a further feature of the invention, the base of the second bipolar transistor is connected to a resistor and via the latter to one pole of a first capacitor, the anode of the first diode being connected to the other pole of the first capacitor.
In accordance with an added feature of the invention, the input serving to supply power to the control circuit is connected via a second capacitor to an output of a line rectifier, the output of the line rectifier being directly connected to the emitter of the first bipolar transistor.
In accordance with yet another feature of the invention, the control circuit has a control output connected via a fifth capacitor to the base of the first bipolar transistor for conducting to the latter control pulses generated in the control circuit.
In accordance with a concomitant feature of the invention, there is provided a sixth capacitor shunting the emitter-collector path of the first transistor.
Other features which are considered as characteristic for the invention are set forth in the appended claim.
Although the invention is illustrated and described herein as embodied in a blocking oscillator type switching power supply, it is nevertheless not intended to be limited to the details shown, since various modifications and structural changes may be made therein without departing from the spirit of the invention and within the scope and range of equivalents of the claims. The construction and method of operation of the invention, however, together with additional objects and advantages thereof will be best understood from the following description of specific embodiments when read in connection with the accompanying drawings, in which:
FIGS. 1 and 2 are circuit diagrams of the blocking oscillator type switching power supply according to the invention; and
FIG. 3 is a circuit diagram of the control unit RS of FIGS. 1 and 2.
Referring now to the drawing and, first, particularly to FIG. 1 thereof, there is shown a rectifier circuit G in the form of a bridge current, which is acted upon by a line input represented by two supply terminals 1' and 2'. Rectifier outputs 3' and 4' are shunted by an emitter-collector path of an NPN power transistor T1 i.e. t
he
series connection of the so-called first bipolar transistor referred to
hereinbefore with a primary winding I of a transformer Tr. Together
with the inductance of the transformer Tr, the capacitance C1 determines
the frequency and limits the opening voltages of the switch embodied by
the first transistor T1. A capacitance C2, provided between the base of
the first transistor T1 and the control output 7,8 of a control circuit
RS, separates the d-c potentials of the control or regulating circuit
RS and the switching transistor T1 and serves for addressing this
switching transistor T1 with pulses. A resistor R1 provided at the
control output 7,8 of the control circuit RS is the negative-feedback
resistor of both output stages of the control circuit RS. It determines
the maximally possible output pulse current of the control circuit RS. A
secondary winding II of the transformer Tr takes over the power supply
of the control circuit, in steady state operation, via the diode D1. To
this end, the cathode of this diode D1 is directly connected to a power
supply input 9 of the control circuit RS, while the anode thereof is
connected to one terminal of the secondary winding II. The other
terminal of the secondary winding II is connected to the emitter of the
power switching transistor T1.
The nine terminals of the control circuit RS have the following purposes or functions:
Terminal 1 supplies the internally generated reference voltage to ground i.e. the nominal or reference value required for the control or regulating process;
Terminal 2 serves as input for the oscillations provided by the secondary winding III, at the zero point of which, the pulse start of the driving pulse takes place;
Terminal 3 is the control input, at which the existing actual value is communicated to the control circuit RS, that actual value being generated by the rectified oscillations at the secondary winding III;
Terminal 4 is responsive to the occurrence of a maximum excursion i.e. when the largest current flows through the first transistor T1 ;
Terminal 5 is a protective input which responds if the rectified line voltage drops too sharply; Terminal 6 serves for the power supply of the control process and, indeed, as ground terminal;
Terminal 7 supplies the d-c component required for charging the coupling capacitor C2 leading to the base of the first transistor T1 ;
Terminal 8 supplies the control pulse required for the base of the first transistor T1 ; and
Terminal 9 serves as the first terminal of the power supply of the control circuit RS.
Further details of the control circuit RS are described hereinbelow.
The capacity C3 smoothes the positive half-wave pulses which are provided by the secondary winding II, and simultaneously serves as an energy storage device during the starting time. The secondary winding III generates the control voltage and is simultaneously used
as
feedback. The time delay stage R2 /C4 keeps harmonics and fast
interference spikes away from the control circuit RS. The resistor R3 is
provided as a voltage divider for the second terminal of the control
circuit RS. The diode D2 rectifies the control pulses delivered by the
secondary winding III. The capacity C5 smoothes the control voltage. A
reference voltage Uref, which is referred to ground i.e. the potential
of terminal 6 is present at the terminal 1 of the control circuit RS.
The resistors R4 and R5 form a voltage divider of the input-difference
control amplifier at the terminal 3. The desired secondary voltage can
be set manually via the variable resistor R4. A time-delay stage R6 /C6
forms a sawtooth rise which corresponds to the collector current rise of
the first bipolar transistor T1 via the primary winding I of the
transformer Tr. The sawtooth present at the terminal 4 of the control
circuit RS is limited there between the reference voltage 2 V and 4 V.
The voltage divider R7 /R8 (FIG. 2), brings to the terminal 5 of the
control circuit RS the enabling voltage for the drive pulse at the
output 8 of the control circuit RS.
The diode D4, together with the resistor R9 in cooperation with the diode D1 and the secondary winding II, forms the starting circuit provided, in accordance with the invention. The operation thereof is as follows:
After the switching power supply is switched on, d-c voltages build up at the collector of the switching transistor T1 and at the input 4 of the control circuit RS, as a function in time of the predetermined time constants. The positive sinusoidal half-waves charge the capacitor C3 via the starting diode D4 and the starting resistor R9 in dependence upon the time constant R9.C3. Via the protective input terminal 5 and the resisto
r
R11 not previously mentioned and forming the connection between the
resistor R9 and the diode D1, on the one hand, and the terminal 5 of the
control circuit RS, on the other hand, the control circuit RS is biased
ready for switching-on, and the capacitor C2 is charged via the output
7. When a predetermined voltage value at the capacitor C3 or the power
supply input 9 of the control circuit RS, respectively, is reached, the
reference voltage i.e. the nominal value for the operation of the
control voltage RS, is abruptly formed, which supplies all stages of the
control circuit and appears at the output 1 thereof. Simultaneously,
the switching transistor T1 is switched into conduction via the output
8. The switching of the transistor T1 at the primary winding T of the
transformer Tr is transformed to the second secondary winding II, the
capacity C3 being thereby charged up again via the diode D1. If
sufficient energy is stored in the capacitor C3 and if the re-charge via
the diode D1 is sufficient so that the voltage at a supply input 9 does
not fall below the given minimum operating voltage, the switching power
supply then remains connected, so that the starting process is
completed. Otherwise, the starting process described is repeated several
times.In FIG. 2, there is shown a further embodiment of the circuit for a blocking oscillator type switching power supply, according to the invention, as shown in FIG. 1. Essential for this circuit of FIG. 2 is the presence of a second bipolar transistor T2 of the type of the first bipolar transistor T1 (i.e. in the embodiments of the invention, an npn-transistor), which forms a further component of the starting circuit and is connected with the collector-emitter path thereof between the resistor R9 of the starting circuit and the current supply input 9 of the control circuit RS. The base of this second transistor T2 is connected to a node which leads, on the one hand, via a resistor R10 to one electrode of a capacitor C7, the other electrode of which is connected to the anode of the diode D4 of the starting circuit and, accordingl
y,
to the terminal 1' of the supply input of the switching power supply G.
On the other hand, the last-mentioned node and, therefore, the base of
the second transistor T2 are connected to the cathode of a Zener diode
D3, the anode of which is connected to the output 3' of the rectifier G
and, whereby, to one pole of the capacitor C3, the second pole of which
is connected to the power supply input 9 of the control circuit RS as
well as to the cathode of the diode D1 and to the emitter of the second
transistor T2. In other respects, the circuit according to FIG. 2
corresponds to the circuit according to FIG. 1 except for the resistor
R11 which is not necessary in the embodiment of FIG. 2, and the missing
connection between the resistor R9 and the cathode of the diode D1,
respectively, and the protective input 5 of the control circuit RS.
Regarding the operation of the starting circuit according to FIG. 2,
it can be stated that the positive sinusoidal half-wave of the line
voltage, delayed by the time delay stage C7, R10 drives the base of the
transistor T2 in the starting circuit. The amplitude is limited by the
diode D3 which is provided for overvoltage protection of the control
circuit RS and which is preferably incorporated as a Zener diode. The
second transistor T2 is switched into conduction. The capacity C3 is
charged, via the serially connected diode D4 and the resistor R9 and the
collector-emitter path of the transistor T2, as soon as the voltage
between the terminal 9 and the terminal 6 of the control circuit RS i.e.
the voltage U9, meets the condition U9 <[UDs -UBE (T2)].Because of the time constant R9.C3, several positive half-waves are necessary in order to increase the voltage U9 at the supply terminal 9 of the control circuit RS to such an extent that the control circuit RS is energized. During the negative sine half-wave, a partial energy chargeback takes place from the capacitor C3 via the emitter-base path of the transistor T2 of the starting circuit and via the resistor R10 and the capacitor C7, respectively, into the supply network. At approximately 2/3 of the voltage U9, which is limited by the diode D3, the control circuit RS is switched on. At the terminal 1 thereof, the reference voltage Uref then appears. In addition, the voltage divider R5 /R4 becomes effective. At the terminal 3, the control amplifier receives the voltage forming the actual value, while the first bipolar transistor T1 of the blocking-oscillator type switching power supply is addressed pulsewise via the terminal 8.
Because the capacitor C6 is charged via the resistor R6, a higher voltage than Uref is present at the terminal 4 if the control circuit RS is activated. The control voltage then discharges the capacitor C6 via the terminal 4 to half the value of the reference voltage Uref, and immediately cuts off the addressing input 8 of the control circuit RS. The first driving pulse of the switching transistor T1 is thereby limited to a minimum of time. The power for switching-on the control circuit RS and for driving the transistor T1 is supplied by the capacitor C3. The voltage U9 at the capacitor C3 then drops. If the voltage U9 drops below the switching-off voltage value of the control circuit RS, the latter is then inactivated. The next positive sine half-wave would initiate the starting process again.
By switching the transistor T1, a voltage is transformed in the secondary winding II of the transformer Tr. The positive component is rectified by the diode D1, recharing of the capacitor C3 being thereby provided. The voltage U9 at the output 9 does not, therefore, drop below the minimum value required for the operation of the control circuit RS, so that the control circuit RS remains activated. The power supply continues to operate in the rhythm of the existing conditions. In operation, the voltage U9 at the supply terminal 9 of the control circuit RS has a value which meets the condition U9 >[UDs -UBE (T2)], so that the transistor T2 of the starting circuit remains cut off.
For the internal layout of the control circuit RS, the construction shown, in particular, from FIG. 3 is advisable. This construction is realized, for example, in the commercially available type TDA 4600 (Siemens AG).
The block diagram of the control circuit according to FIG. 3
shows
the power supply thereof via the terminal 9, the output stage being
supplied directly whereas all other stages are supplied via Uref. In the
starting circuit, the individual subassemblies are supplied with power
sequentially. The d-c output voltage potential of the base current gain
i.e. the voltage for the terminal 8 of the control circuit RS, and the
charging of the capacitor C2 via the terminal 7 are formed even before
the reference voltage Uref appears. Variations of the supply voltage U9
at terminal 9 and the power fluctuations at the terminal 8/terminal 7
and at the terminal 1 of the control circuit RS are leveled or smoothed
out by the voltage control. The temperature sensitivity of the control
circuit RS and, in particular, the uneven heating of the output and
input stages and input stages on the semiconductor chip containing the
control circuit in monolithically integrated form are intercepted by the
temperature compensation provided. The output values are constant in a
specific temperature range. The message for blocking the output stage,
if the supply voltage at the terminal 9 is too low, is given also by
this subassembly to a provided control logic.The outer voltage divider of the terminal 1 via the r
esistors
R5 and R4 to the control tap U forms, via terminal 3, the variable side
of the bridge for the control amplifier formed as a differential
amplifier. The fixed bridge side is formed by the reference voltage Uref
via an internal voltage divider. Similarly formed are circuit portions
serving for the detection of an overload short circuit and circuit
portions serving for the "standby" no-load detection, which can be
operated likewise via terminal 3.Within a provided trigger circuit, the driving pulse length is determined as a function of the sawtooth rise at the terminal 4, and is transmitted to the control logic. In the control logic, the commands of the trigger circuit are processed. Through the zero-crossing identification at input 2 in the control circuit RS, the control logic is enabled to start the control input only at the zero point of the frequency oscillation. If the voltages at the terminal 5 and at the terminal 9 are too low, the control logic blocks the output amplifier at the terminal 8. The output amplifier at the terminal 7 which is responsible for the base charge in the capacitor C2, is not touched thereby.
The base current gain for the transistor T1 i.e. for the first transistor in accordance with the definition of the invention, is formed by two amplifiers which mutually operate on the capacitor C2. The roof inclination of the base driving current for the transistor T1 is impressed by the collector current simulation at the terminal 4 to the amplifier at the terminal 8. The control pulse for the transistor T1 at the terminal 8 is always built up to the potential present at the terminal 7. The amplifier working into the terminal 7 ensures that each new switching pulse at the terminal 8 finds the required base level at terminal 7.
Supplementing the comments regarding FIG. 1, it should also be mentioned that the cathode of the diode D1 connected by the anode thereof to the one end of the secondary winding II of the transformer Tr is connected via a resistor R11 to the protective input 5 of the control circuit RS whereas, in the circuit according to FIG. 2, the protective input 5 of the control circuit RS is supplied via a voltage divider R8, R7 directly from the output 3', 4' of the rectifier G delivering the rectified line a-c voltage, and which obtains the voltage required for executing its function. It is evident that the first possible manner of driving the protective input 5 can be used also in the circuit according to FIG. 2, and the second possibility also in a circuit in accordance with FIG. 1.
The control circuit RS which is shown in FIG. 3 and is realized in detail by the building block TDA 4600 and which is particularly well suited in conjunction with the blocking oscillator type switching power supply according to the invention has 9 terminals 1-9, which have the following characteristics, as has been explained in essence hereinabove:
Terminal 1 delivers a reference voltage Uref which serves as the constant-current source of a voltage divider R5.R4 which supplies the required d-c voltages for the differential amplifiers provided for the functions control, overload detection, short-circuit detection and "standby"-no load detection. The dividing point of the voltage divider R5 -R4 is connected to the terminal 3 of the control circuit RS. The terminal 3 provided as the control input of RS is controlled in the manner described hereinabove as input for the actual value of the voltage to be controlled or regulated by the secondary winding III of the transformer Tr. With this input, the lengths of the control pulses for the switching transistor T1 are determined.
Via the input provided by the terminal 2 of the control circuit RS, the zero-point identification in the control circuit is addressed for detecting the zero-point o
f
the oscillations respectively applied to the terminal 2. If this
oscillation changes over to the positive part, then the addressing pulse
controlling the switching transistor T1 via the terminal 8 is released
in the control logic provided in the control circuit.A sawtooth-shaped voltage, the rise of which corresponds to the collector current of the switching transistor T1, is present at the terminal 4 and is minimally and maximally limited by two reference voltages. The sawtooth voltage serves, on the one hand as a comparator for the pulse length while, on the other hand, the slope or rise thereof is used to obtain in the base current amplification for the switching transistor T1, via the terminal 8, a base drive of this switching transistor T1 which is proportional to the collector current.
The terminal 7 of the control circuit RS as explained hereinbefore, determines the voltage potential for the addressing pulses of the transistor T2. The base of the switching transistor T1 is pulse-controlled via the terminal 8, as described hereinbefore. Terminal 9 is connected as the power supply input of the control circuit RS. If a voltage level falls below a given value, the terminal 8 is blocked. If a given positive value of the voltage level is exceeded, the control circuit is activated. The terminal 5 releases the terminal 8 only if a given voltage potential is present.
Forei
gn References:DE2417628A1 1975-10-23 363/37
DE2638225A1 1978-03-02 363/49
Other References:
Grundig Tech. Info. (Germany), vol. 28, No. 4, (1981).
IBM Technical Disclosure Bulletin, vol. 19, No. 3, pp. 978, 979, Aug. 1976.
German Periodical, "Funkschau", (1975), No. 5, pp. 40 to 44.
Inventors:
Peruth, Gunther (Munich, DE) Siemens Aktiengesellschaft (Berlin and Munich, DE)
All functions of the receiver are summed up in the left chassis the GRUNDIG CUC70KT which is the successor of the CUC720 seen HERE
BU208(A)
Silicon NPNnpn transistors,pnp transistors,transistors
Category: N
PN Transistor, TransistorMHz: <1 MHz
Amps: 5A
Volts: 1500V
HIGH VOLTAGE CAPABILITY
JEDEC TO-3 METAL CASE.
DESCRIPTION
The BU208A, BU508A and BU508AFI are
manufactured using Multiepitaxial Mesa
technology for cost-effective high performance
and use a Hollow Emitter structure to enhance
switching speeds.
APPLICATIONS:
* HORIZONTAL DEFLECTION FOR COLOUR TV With 110° or even 90° degree of deflection angle.
ABSOLUTE MAXIMUM RATINGS
Symbol Parameter Value Unit
VCES Collector-Emit ter Voltage (VBE = 0) 1500 V
VCEO Collector-Emit ter Voltage (IB = 0) 700 V
VEBO Emitter-Base Voltage (IC = 0) 10 V
IC Collector Current 8 A
ICM Collector Peak Current (tp < 5 ms) 15 A
TO - 3 TO - 218 ISOWATT218
Ptot Total Dissipation at Tc = 25 oC 150 125 50 W
Tstg Storage Temperature -65 to 175 -65 to 150 -65 to 150 oC
Tj Max. Operating Junction Temperature 175 150 150 °C
GENERAL BASIC TRANSISTOR LINE OUTPUT STAGE OPERATION:
The basic essentials of a transistor line output stage are shown in Fig. 1(a). They comprise: a line output transformer which provides the d.c. feed to the line output transistor and serves mainly to generate the high -voltage pulse from which the e.h.t. is derived, and also in practice other supplies for various sections of the receiver; the line output transistor and its parallel efficiency diode which form a bidirectional switch; a tuning capacitor which resonates with the line output transformer primary winding and the scan coils to determine the flyback time; and the scan coils, with a series capacitor which provides a d.c. block and also serves to provide slight integration of the deflection current to compensate for the scan distortion that would otherwise be present due to the use of flat screen, wide deflection angle c.r.t.s. This basic circuit is widely used in small -screen portable receivers with little elaboration - some use a pnp output transistor however, with its collector connected to chassis.
Circuit Variations:
Variations to the basic circuit commonly found include: transposition of the scan coils and the correction capacitor; connection of the line output transformer primary winding and its e.h.t. ove
rwinding
in series; connection of the deflection components to a tap on the
transformer to obtain correct matching of the components and conditions
in the stage; use of a boost diode which operates in identical manner to
the arrangement used in valve line output stages, thereby increasing
the effective supply to the stage; omission of the efficiency diode
where the stage is operated from an h.t. line, the collector -base
junction of the line output transistor then providing the efficiency
diode action without, in doing so, producing scan distortion; addition
of inductors to provide linearity and width adjustment; use of a pair of
series -connected line output transistors in some large -screen colour
chassis; and in colour sets the addition of line convergence circuitry
which is normally connected in series between the line scan coils and
chassis. These variations on the basic circuit do not alter the basic
mode of operation however.Resonance
The most important fact to appreciate about the circuit is that when the transistor and diode are cut off during the flyback period - when the beam is being rapidly returned from the right-hand side of the screen to the left-hand side the tuning capacitor together with the scan coils and the primary winding of the line output transformer form a parallel resonant circuit: the equivalent circuit is shown in Fig. 1(b). The line output transformer primary winding and the tuning capacitor as drawn in Fig. 1(a) may look like a series tuned circuit, but from the signal point of view the end of the transformer primary winding connected to the power supply is earthy, giving the equivalent arrangement shown in Fig. 1(b).
The Flyback Period:
Since the operation of the circuit depends mainly upon what happens during the line flyback period, the simplest point at which to break into the scanning cycle is at the end of the forward scan, i.e. with the
beam deflected to the right-hand side of the screen, see Fig. 2. At
this point the line output transistor is suddenly switched off by the
squarewave drive applied to its base. Prior to this action a linearly
increasing current has been flowing in the line output transformer
primary winding and the scan coils, and as a result magnetic fields have
been built up around these components. When the transistor is switched
off these fields collapse, maintaining a flow of current which rapidly
decays to zero and returns the beam to the centre of the screen. This
flow of current charges the tuning capacitor, and the voltage at A rises
to a high positive value - of the order of 1- 2k V in large -screen
sets, 200V in the case of mains/battery portable sets. The e
nergy
in the circuit is now stored in the tuning capacitor which next
discharges, reversing the flow of current in the circuit with the result
that the beam is rapidly deflected to the left-hand side of the screen -
see Fig. 3. When the tuning capacitor has discharged, the voltage at A
has fallen to zero and the circuit energy is once more stored in the
form of magnetic fields around the inductive components. One half -cycle
of oscillation has occurred, and the flyback is complete.Energy Recovery:
First Part of Forward Scan The circuit then tries to continue the cycle of oscillation, i.e. the magnetic fields again collapse, maintaining a current flow which this time would charge the tuning capacitor negatively (upper plate). When the voltage at A reaches about -0.6V however the efficiency diode becomes forward biased and switches on. This damps the circuit, preventing further oscillation, but the magnetic fields continue to collapse and in doing so produce a linearly decaying current flow which provides the first part of the forward s
can,
the beam returning towards the centre of the screen - see Fig. 4. The
diode shorts out the tuning capacitor but the scan correction capacitor
charges during this period, its right-hand plate becoming positive with
respect to its left-hand plate, i.e. point A. Completion of Forward Scan
When the current falls to zero, the diode will switch off. Shortly
before this state of affairs is reached however the transistor is
switched on. In practice this is usually about a third of the way
through the scan. The squarewave applied to its base drives it rapidly
to saturation, clamping the vol
tage
at point A at a small positive value - the collector emitter saturation
voltage of the transistor. Current now flows via the transistor and the
primary winding of the line output transformer, the scan correction
capacitor discharges, and the resultant flow of current in the line scan
coils drives the beam to the right-hand side of the screen see Fig. 5.Efficiency:
The transistor is then cut off again, to give the flyback, and the cycle of events recurs. The efficiency of the circuit is high since there is negligible resistance present. Energy is fed into the circuit in the form of the magnetic fields that build up when the output transistor is switched on. This action connects the line output transformer primary winding across the supply, and as a result a linearly increasing current flows through it. Since the width is
dependent on the supply voltage, this must be stabilised.
Harmonic Tuning:
There is another oscillatory action in the circuit during the flyback period. The considerable leakage inductance between the primary and the e.h.t. windings of the line output transformer, and the appreciable self -capacitance present, form a tuned circuit which is shocked into oscillation by the flyback pulse. Unless this oscillation is controlled, it will continue into and modulate the scan. The technique used to overcome this effect is to tune the leakage inductance and the associated capacitance to an odd harmonic of the line flyback oscillation frequency. By doing this the oscillatory actions present at the beginning of the scan cancel. Either third or fifth harmonic tuning is used. Third harmonic tuning also has the effect of increasing the amplitude of the e.h.t. pulse, and is generally used where a half -wave e.h.t. rectifier is employed. Fifth harmonic tuning results in a flat-topped e.h.t. pulse, giving improved e.h.t. regulation, and is generally used where an e.h.t. tripler is employed to produce the e.h.t. The tuning is mainly built into the line output transformer, though an external variable inductance is commonly found in colour chassis so that the tuning can be adjusted. With a following post I will go into the subject of modern TV line timebases in greater detail with other models and technology shown here at Obsolete Technology Tellye !
CHASSIS CUC70KT Simplified horizontal / line deflection circuit.
-----------------------------------------------------------------------------------------------
A horizontal deflection circuit makes a sawtoothcurrent flow through a deflection coil. The current
will have equal amounts of positive and negative
current. The horizontal switch transistor conducts
for the right hand side of the picture. The damper
diode conducts for the left side of the picture.
Current only flows through the fly back capacitor
during retrace time.
For time 1 the transistor is turned on. Current
ramps up in the yoke. The beam is moved from the
center of the picture to the right edge. Energy is
stored on the inductance of the yoke.
E=I2L/2
For time 2 the transistor is turned off. Energy
transfers from the yoke to the flyback capacitor. At
the end of time two all the energy from the yoke is
placed on the flyback capacitor. There is zero
current in the yoke and a large voltage on the
capacitor. The beam is quickly moved from the
right edge back to the middle of the picture.
During time 3 the energy on the capacitor flowsback into the yoke. The voltage on the flyback
capacitor decreases while the current in the yoke
builds until there is no voltage on the capacitor. By
the end of time 3 the yoke current is at it's
maximum amount but in the negative direction.
The beam is quickly deflected form the center to the
left edge.
Time 4 represents the left hand half of the picture.
Yoke current is negative and ramping down. The
beam moves from the left to the center of the
picture.
The current that flows when the horizontal switch is
closed is approximately:
Ipk ≅ Vcc T / Ldy
Ipk = collector current
T = 1/2 trace time
Ldy = total inductance (yoke + lin coil + size coil)
note:The lin coil inductance varies with current.
______
Tr ≅ 3.14 √ L C
The current that flows during retrace is produced by
the C and L oscillation. The retrace time is 1/2 the
oscillation frequency of the L and C.
I2L /2 ≅ V2C /2 or I2L = V2C As stated earlier the energy in the yoke moves to the
flyback capacitor during time 2.
V= the amount of the flyback pulse that is above the
supply voltage.
shores, capacitors are open and generally
semiconductors are removed. The voltage at the
point “B+” is the supply voltage. The collector
voltage of Q1 is also at the supply voltage. The
voltage across C2 is equal to the supply voltage.
When we A.C. annualize this circuit we will find
that the collector of Q1 has a voltage that ranges
from slightly negative to 1000 volts positive. The
average voltage must remain the same as the D.C.
value.
In the A.C. annualizes of the circuit, the
inductance of the yoke (DY) and the inductance of
the flyback transformer are in parallel. The
inductance of T2 is much larger than that if the
DY. This results is a total system inductance of
about 10% to 20% less than that of the DY it’s
self.
The voltage across the Q1 is a half sinusoid pulse during the flyback or retrace period and close to zero at
all other times. It is not possible or safe to observe this point on an oscilloscope without a proper high
frequency high voltage probe. Normally use a 100:1 probe suitable for 2,000V peak. The probe must have
been high frequency calibrated recently.
HORIZONTAL SIZE / E/W AMPLITUDE - CORRECTION CIRCUIT:
There are several different methods of adjusting horizontal size.
SIZE COILAdd a variable coil to the yoke current path
causes the total inductance to vary with the coils
setting.
The yoke current is related to supply voltage,
trace time and total inductance. This method
has a limited range!
The horizontal section uses a PWM to set the
horizontal size. One DAC sets the horizontal
size and another DAC sets the pincushion and
trap.
The Raster Centering (D.C. centering) is
controlled by a DAC.
On small monitors the retrace time is fixed. On
large monitors or wide frequency range monitors
two different retrace times are available. The flyback time is set by the micro computer by selecting two
different flyback capacitors. At slow frequencies the longer retrace time is selected.
Different S corrector capacitor values are selected by the micro computer. At the highest frequency the
smallest capacitor is selected.
SPLIT DIODE MODULATORThis horizontal circuit consists of two parts. D1, C1, C2 and DY are the components as described above.
D2, C3, C4 and L1 are a second “dummy” horizontal section that does not cause deflection current. By the
D.C. analyzing this circuit the voltage across C2 + C4 must equal the supply voltage (B+). Deflection
current in the DY is related to the supply voltage minus the voltage across C4. For a maximum horizontal
size the control point must be held at ground. This causes the dummy section to not operate and the DY
section will get full supply voltage. If the control point is at 1/3 supply then the DY section will be
operating at 2/3 supply.
Note: The impedance of (D1,C1,C2 and DY) and (D2,C3,C4 and L1) makes a voltage divider. If the
control point is not connected then there is some natural voltage on C4. Most split diode monitors are built
to pull power from the dummy section through L2 to ground. A single power transistor shunts from the
control point to ground. It is true that power can be supplied from some other supply through L2 to rise the
voltage on C4. For maximum range a bi-directional power amplifier can drive the control point.The most exciting feature if the split diode modulator is that the flyback pulse, as seen by the flyback
transformer, is the same size at all horizontal size settings.
HORIZONTAL SWITCH/DAMPER DIODE
On the right hand side of the screen, the H. switch transistor conducts current through the deflection yoke.
This current comes from the S correction capacitors, which have a charge equal to the effective supply
voltage. The damper diode allows current for the left hand side of the screen to flow back through the
deflection yoke to the S capacitors.
FLYBACK CAPACITOR
The flyback capacitor connects the hot side of the yoke to ground. This component determines the size and
length of the flyback pulse. ‘Tuning the flyback capacitor’ is done to match the timing of the flyback pulse
to the video blanking time of the video signal. The peak flyback voltage on the horizontal switch must be
set to less that 80% if the Vces specification. The two conditions of time and voltage can be set by three
variables (supply voltage, retrace capacitor and yoke inductance) .
S CAPACITOR
The S capacitors corrects outside versus center linearity in the horizontal scan. The voltage on the S cap
has a parabola plus the DC horizontal supply. Reducing the value of S cap increases this parabola thus
reducing the size of the outside characters and increasing the size of the center characters.
S Capacitor value: Too low: picture will be squashed towards edges.
Too high: picture will be stretched towards edges.
By simply putting a capacitor in series with each coil, the sawtooth waveform is
modified into a slightly sine-wave shape. This reduces the scanning speed near the
edges where the yoke is more sensitive. Generally the deflection angle of the electron
beam and the yoke current are closely related. The problem is the deflection angle
verses the distance of movement on the CRT screen does not have a linear effect.
DEFLECTION NPN TRANSISTOR BASE DRIVE CURRENT
T
he base drive resistor determines the amount ofbase drive. If the transistor is over driven the Vsat
looks very good, but the current fall time is poor.
If the base current is too small the current fall time is very fast. The problem is that the transistor will have many volts across C-E when closed.
The best condition is found by placing the transistor in the heaviest load condition. Adjust the base resistor for the least power consumption then increase the base drive a small amount. This will slightly over drive the base.
TUNER UNIT with TUA2000 (Siemens) + TDA5510 (Siemens) + U264B (Telefunken) + TBA120T (Telefun
ken).Video + RGB Unit (RGB Farb Baustein 29504-005.33) with TDA3562A
PAL/NTSC ONE-C
HIP DECODERDESCRIPTION:
The TDA3562A is a monolithic IC designed as
decode PAL and/or NTSC colour television standards
and it combines all functions required for the
identification and demodulation of PAL and NTSC
signals.
.CHROMINANCE SIGNALPROCESSOR
.LUMINANCE SIGNAL PROCESSING WITH
CLAMPING
.HORIZONTAL AND VERTICAL BLANKING
.LINEAR TRANSMISSION OF INSERTED
RGB SIGNALS
.LINEAR CONTRAST AND BRIGHTNESS
CONTROL ACTING ON INSERTED AND MATRIXED
SIGNALS
.AUTOMATIC CUT-OFF CONTROL
.NTSC HUE CONTROL.
THE PHILIPS TDA3562A Circuit arrangement for the control of a picture tube :1. Circuit arrangement for the control of at least one beam current in a picture tube by a picture comprising
a control loop which in one sampling interval obtains a measuring signal from the value of the beam current on the occurrence of a given reference level in the picture signal, stores a control signal derived therefrom until the next sampling interval and thereby adjusts the beam current to a value preset by a reference signal.
and a trigger circuit which suppresses auxiliary pulses used to generate the beam current after the picture tube has been started up and issues a switching signal for the purpose of closing the control loop during the sampling intervals and for releasing the control of the beam current by the picture signal after the measuring signal has exceeded the threshold value,
a change detection arrangement which delivers a change signal when the stored signal has assumed a largely constant value, and
a logic network which does not release the control of the beam current
by the picture signal outside the sampling intervals until the change
signal has also been issued after the switching signal.2. Circuit arrangement as set forth in claim 1, in which the picture signal comprises several color signals for the control of a corresponding number of beam currents for the display of a color picture in the picture tube and the control loop stores a part measuring signal or a part control signal derived therefrom for each color signal, characterized in that the change detection arrangement includes a change detector for each color signal which delivers a part change signal when the relevant stored signal has assumed a largely constant value, and the logic network does not release the control of the beam currents by the color signals outside the sampling intervals until the part change signals have been delivered by all change detectors.
3. Circuit arrangement as set forth in claim 1, including a comparator arrangement which compares the measuring signal with the reference signal and derives the control signal from this comparison, characterized in that the change detection arrangement detects a change in the control signal with respect to time and issues the change signal when the control signal has assumed a largely constant value.
4. Circuit arrangement as set forth in claims 1, 2, 3 including a control signal memory which contains at least one capacitor, characterized in that the change detection arrangement delivers the change signal when a charge-reversing current of the capacitor occuring during the starting up of the picture tube falls below a limit value.
5. Circuit arrangement as set forth in claim 2, including a comparator arrangement which compares the measuring signal with the reference signal and derives the control signal from this comparison, characterized in that the change detection arrangement detects a change in the control signal with respect to time and issues the change signal when the control signal has assumed a largely constant value.
Description:
BACKGROUND OF THE INVENTIONThe invention r
elates to a circuit arrangement for the control of at least one beam
current in a picture tube by a picture signal with a control loop which
in one sampling interval obtains a measuring signal from the value of
the beam current on the occurrence of a given reference level in the
picture signal, stores a control signal derived therefrom until the next
sampling interval and by this means adjusts the beam current to a value
preset by a reference signal, and with a trigger circuit which
suppresses auxiliary pulses used to generate the beam current after the
picture tube is turned on and issues a switching signal for the purpose
of closing the control loop during the sampling intervals and releasing
the control of the beam current by the picture signal after the
measuring signal has exceeded a threshold value. Such a circuit
arrangement has been described in Valvo Technische Information 820705
with regard to the integrated color decoder circuit PHILIPS TDA3562A and is
used in this as a so-called cut-off point control. In the known circuit
arrangement, such a cut-off point control provides automatic
compensation of the so-called cut-off point of the picture tube, i.e. it
regulates the beam current in the picture tube in such a way that for a
given reference level in the picture signal the beam current has a
constant value despite tolerances and changes with time (aging, thermal
modifications) in the picture tube and the circuit arrangement, thereby
ensuring correct picture reproduction. Such a blocking point control is particularly advantageous for the operation of a picture tube for the display of color pictures because in this case there are several beam currents for different color components of the color picture which have to be in a fixed ratio with one another. If this ratio changes, for example, as the result of manufacturing tolerances or ageing processes, distortions of the colors occur in the reproduction of the color picture. The beam currents, therefore, have to be very accurately balanced. The said cut-off point control prevents expensive adjustment and maintenance time which is otherwise necessary.
Conventional picutre tubes are constructed as cathode-ray tubes with hot cathodes which require a certain time after being turned on for the hot cathodes to heat up. Not until a final operating temperature has been reached do these hot cathodes emit the desired beam currents to the full extent, while gradually rising beam currents occur in the time interval when the hot cathodes are heating up. The instantaneous values of these beam currents depend on the instantaneous temperatures of the hot cathodes and on the accelerating voltages for the picture tube which build up simultaneously with the heating process and are undefined until the end of the heating time. After the picture tube is turned on, these values initially produce a highly distorted picture until the beam currents have attained their final value. These picture distortions after the picture tube is turned on are even further intensified by the fact that the cut-off point control is not yet adjusted to the beam currents which flow after the heating time is over.
For the purpose of suppressing distorted pictures during the heating time of the hot cathodes, the known circuit arrangement has a turn-on delay element
operating as a trigger circuit which, in essence, contains a bistable
flip-flop. When the picture tube and the circuit arrangement controlling
the beam currents flowing in it are turned on, the flip-flop is
switched into a first state in which it interrupts the supply of the
picture signal to the picture tube. Thus, during the heating time the
beam currents are suppressed, and the picture tube does not yet display
any picture. In sampling intervals which are provided subsequent to
flybacks of the cathode beam into an initial position on the changeover
from the display of one picture to the display of a subsequent picture
and even within the changeover, that is outside the display of pictures,
the picture tube is controlled for a short time in such a way that beam
currents occur when the hot cathodes are sufficiently heated up and an
accelerating voltage is resent. If these currents exceed a certain
threshold value, the flip-flop circuit switches into a second state and
releases the picture signal for the control of the beam currents and the
cut-off point control. It is found, however, that the picture displayed in the picture tube immediately after the switching over of the flip-flop is still not fault-free. Because, in fact, the beam currents are supported during the heating time of the hot cathodes, the cut-off point control cannot respond yet. This response of the cut-off point control takes place only after the beam currents are switched on, i.e. after the flip-flop is switched into the second state and therefore at a time in which the picture signal already controls the beam currents. In this way the response of the blocking point control makes its presence felt in the picture displayed.
With the known
circuit arrangement the brightness of the picture gradually increases,
during the response of the cut-off point control, from black to the
final value. This slow increase in the picture brightness after the tube is turned on is disturbing to the eyes of the viewer not only in the case of the black-and-white picture tubes with one hot cathode, but especially so in the case of colour picture tubes which usually have three hot cathodes. With a color picture tube, color purity errors can also occur in addition to the change in the picture brightness if, as a result of different speeds of response of the cut-off point control for the three beam currents, there are found to be intermittent variations from the interrelation between the beam currents required for a correct picture reproduction.
SUMMARY OF THE INVENTION
The aim of the invention is to create a circuit arrangement which suppresses the above-described disturbances of brightness and color of the displayed picture when the picture tube is being started.
The invention achieves this aim in that
a circuit arrangement of the type
mentioned in the preamble contains a change detection arrangement which
emits a change signal when the stored signal has assumed an essentially
constant value, and a logic network which does not release the control
of the beam current by the picture signal until the change signal has
also been emitted after the switching signal. In the circuit arrangement according to the invention, therefore, the display of the picture is suppressed after the picture tube is turned on until the cut-off point control has responded. If the picture signal then starts to control the beam current, a perfect picture is displayed immediately. In this way, all the disturbances of the picture which affect the viewer's pleasure are suppressed. The circuit arrangement of the invention is of simple design and can be combined on one semiconductor wafer with the existing picture signal processing circuits and also, for example, with the known circuit arrangement for cut-off point control. Such an integrated circuit arrangement not only requires very little space on the semiconductor wafer, but also needs no additional external leads. Thus the circuit arrangement of the invention can be arranged, for example, in an integrated circuit which has precisely the same external connections as known integrated circuits. This means that an integrated circuit containing the circuit arrangement of the invention can be directly incorporated in existing equipment without the need for additional measures.
In one embodiment of the said circuit arrangement, in which the picture signal contains several color signals for the control of a corresponding number of beam currents for representing a color picture in the picture tube and, for each color signal, the control loop stores a part measuring signal or a part control signal derived from it, the change detection arrangement contains a change detector for each color signal which emits a part change signal when the relevant stored signal has assumed an essentially constant value, and the logic network does not release the control of the beam currents by the color signals outside the sampling intervals until the part change signals have been emitted from all change detectors.
In principle, therefore, such a circuit arrangement has three cut-off point controls for the three beam currents controlled by the individual color signals. To reduce the cost of the circuitry, the measuring stage is common to all the cut-off point controls, as in the known circuit arrangement. All three beam currents are then measured successively by this measuring stage. In this way, a part measuring signal or a part control signal derived from it is obtained for each beam current and is stored sesparately according to which of the beam currents it belongs. Changes in the part measuring signal or part control signal are detected for each beam current by one of the change detectors each time. Each of these change detectors issues a part change signal to the logic network. The latter does not release the control of the beam currents by the picture signal outside the sampling intervals until all the part change signals indicate that the part measuring signal or the part control signal, as the case may be, remains constant. This ensures that the cut-off point controls for the beam currents of all color signals have responded when the picture appears in the picture tube.
In a further embodiment of the circuit arrangement
according to the invention with a comparator arrangement which compares
the measuring signal with the reference signal and derives the control
signal from this comparison, the change detection arrangement detects a
change in the control signal with respect to time and issues the change
signal when the control signal has assumed an essentially constant
value. In the case of the representation of a color signal the
comparator arrangement derives several part control signals, whose
changes with time are detected by the change detectors, from a
corresponding comparison of the part measuring signals with the
reference signal. In this embodiment of the circuit arrangement of the
invention, preference is given to storage of only the control signal or
the part control signals for the purpose of controlling the beam
currents. In another embodiment of the circuit arrangement of the invention which includes a control signal memory which contains at least one capacitor in which a charge or voltage corresponding to the control signal is stored, the change detection arrangement issues the change signal when a charge-reversing current of the capacitor occurring during the turning on of the picture tube has fallen below a limit value and has thus at least largely decayed. Such a detection of the steady state of the cut-off point control is independent of the actual magnitude of the control signal and therefore independent of, for example, the level of the picture tube cut-off voltage, circuit tolerances or ageing processes in the circuit arrangement or the picture tube.
Detection of whether or not the charge-reversing current exceeds the limit value is performed preferentially by a current detector which is designed with a current mirror system which is arranged in a supply line to a capacitor acting as a control signal store. A current mirror arrangement of this kind supplies a current which coincides very precisely with the charging current of the capacitor. This current is then compared, preferably in a further device contained in the change detection arrangement, with a current representing a limit value or, after conversion into a voltage, with a voltage representing the limit value. The change signal is obtained from the result of this comparison.
On the other hand, digital memories may also be used as control signal memories, especially when the picture signal is supplied as a digital signal and the blocking point control is constructed as a digital control loop. In such a case, the comparator arrangement, the change detection arrangement and the trigger circuit are also designed as digital circuits. Then, the change detection arrangement advantageously forms the difference of the signals stored in the control signal memory in two successive sampling intervals and compares this with the limit value formed by a digital value. If the difference falls short of the limit value, the change signal is issued.
BRIEF DESCRIPTION OF THE DRAWINGS
An embodiment of the invention is described in greater detail below with the aid of the drawings in which:
FIG. 1 shows a block circuit diagram of the embodiment,
FIG. 2 shows a somewhat more detailed block circuit diagram of the embodiment,
FIG. 3 shows time-dependency diagrams of some signals occurring in the circuit diagram shown in FIG. 2, and
FIG. 4 shows a somewhat moredetailed block circuit diagram of a part of the circuit diagram shown in FIG. 2.
DETAILED DESCRIPTION OF THE INVENTIONFIG. 1 shows a block circuit diagram of a circuit arrangement to which a picture signal is fed via a first input 1 of a combinatorial stage 2. From the output 3 of the combinatorial stage 2 the picture signal is fed to the picture signal input of a controllable amplifier 5 which at an output 6 issues a current controlled by the picture signal. This current is fed via a measuring stage 7 to a hot cathode 8 in a picture tube 9 and forms therein a beam current of a cathode ray by means of which a picture defined by the picture signal is displayed on a fluorescent screen of the picture tube 9.
The measuring stage 7 measures the current fed to the hot cathode 8, i.e. the the beam current in the picture tube 9, and at a measuring signal output 10, issues a measuring signal corresponding to the magnitude of this current. This is fed to a measuring signal input 11 of a comparator arrangement 12 to which a reference signal is supplied at a reference signal input 13. In a preferably periodically recurring sampling interval during the occurrence of a given reference level in the picture signal, the comparator arrangement 12 forms a control signal from the value of the measuring signal fed to the measuring signal input 11 at this time, on the one hand, and the reference signal, on the other, by means of substraction and delivers this at a control signal output 14. From there the control signal is fed to an input 15 of a control signal memory 16 and is stored in the latter. The control signal is fed via an output 17 of the control signal memory 16 to a second input 18 of combinatorial stage 2 in which it is combined with the picture signal, e.g. added to it.
The combinatorial stage 2, the controllable amplifier 5, the
measuring stage 7, the comparator arrangement 12 and the control signal
memory 16 form a control loop with which the beam current is guided
towards the reference signal in the sampling interval during the
occurrence of the reference level in the picture signal. For the
reference level, use is made in particular of a black level or a level
with small, fixed distance from the black level, i.e. a value in the
picture signal which produces a black or almost back picture area in the
displayed picture in the picture tube. In this case the control loop,
as described, forms a cut-off point control for the picture tube. If the
reference level is away from the black level, the control loop is also
designated as quasi-cut-off-point control. The circuit arrangement as shown in FIG. 1 also has a trigger circuit 19 to which the measuring signal from the measuring signal output 10 of measuring stage 7 is fed at a measuring signal input 20. When the circuit arrangement and therefore the picture tube are turned on, the trigger circuit 19 is set in a first state in which by means of a first connection 21 it blocks the comparator arrangement 12 in such a way that the latter delivers no control signal or a control signal with the value zero at its control signal output 14. This prevents the control signal memory 16 from storing undefined values for the control signal at the moment of turning on or immediately thereafter.
The circuit arrangement shown in FIG. 1 also has a logic network 22 which is connected via a second connection 23, by means of which a switching signal is supplied, with the trigger circuit 10 and via a third connection 24 with the controllable amplifier 5. Like the trigger circuit 19, the logic network 22 also finds itself controlled, when the circuit arrangement is being turned on, by the switching signal in a first stage in which by way of the third connection 24 it blocks the controllable amplifier 5 with a blocking signal in such a way that no beam currents controlled by the picture signal can yet flow in the picture tube 9. Thus the picture tube 9 is blanked; no picture is displayed yet.
When picture tube 9 is turned on, the hot cathode 8 is still cold so that no beam current can flow anyhow. The hot cathode 8 is then heated up and, after a certain time, begins gradually to emit electrons as the result of which a cathode ray and therefore a beam current can form. However, during the heating up of the hot cathode 8, and because the cut-off point control has not yet responded, this would be undefined and is therefore suppressed by the controllable amplifier 5. Only in time intervals which are provided immediately subsequent to flybacks of the cathode rays into an initial position at the changeover from the display of one image to that of a subsequent image, but even before the start of the display of the subsequent image, the controllable amplifier 5 delivers a voltage in the form of an auxiliary pulse for a short time at its output 6, and when the hot cathode 8 in the picture tube 9 is heated up sufficiently, this voltage produces a beam current. The time interval for the delivery of this voltage is selected in such a way that a cathode ray produced by its does not produce a visible image in the picture tube 9, and coincides for example with the sampling interval.
The measuring stage 7 measures the short-time cathode current produced in the manner described and, at its measuring signal output 10, delivers a corresponding measuring signal which is passed via measuring signal output 20 to the trigger circuit 19. If the measuring signal exceeds a definite preset threshold value, the trigger circuit 19 is switched into a second state in which it releases the comparator arrangement 12 via the first connection 12 and, by means of the second connection 23, uses the switching signal to also bring the logic network 22 into a second state. The comparator arrangement 12 now evaluates the measuring signal supplied to it via the measuring signal input 11, i.e. it forms the control signal as the difference between the measuring signal and the reference signal supplied via the reference signal input 13. The control signal is transferred via the control signal output 14 and the input 15 into the control signal memory 16. It is subsequently fed via the output 17 of the control signal memory 16 to the second input 18 of the combinatorial stage 2 and is there combined with the picture signal at the first input 1, e.g. is superimposed on it by addition. This superimposed picture signal is fed to the picture signal input 4 of the controllable amplifier 5 via the output 3 of the combinatorial stage 2.
During the response of the cut-off point control the control sig
nal fed to the control signal
memory 16 changes continuously. Between the control signal output 14 of
the comparator arrangement 12 and the input 15 of the control signal
memory 16 is inserted a changed detection arrangement 25 which detects
the variations of the control signal. When the cut-off point control has
responded, i.e. the control signal has assumed a constant value, the
change detection arrangement 25 delivers a change signal at an output 26
which indicates that the steady stage of the cut-off point control is
achieved and the said signal is fed to a change signal input 27 of the
logic network 22. The logic network then switches into a third state in
which via the third connection 24 it enables the controllable amplifier 5
in such a way that the beam currents are now controlled without
restriction by the picture signal. Thus a correctly represented picture
appears in the picture tube 9. A shadow-like representation of individual constituents of the circuit arrangement in FIG. 1 is used to indicate a modification by which this circuit arrangement is equipped for the representation of color pictures in the picture tube 9. For example, three color signals are fed in this case as the picture signal via the input 1 to the combinatorial stage 2. Accordingly, the input 1 is shown in triplicate, and the combinatorial stage 2 has a logic element, e.g. an adder, for example of these color signals. The controllable amplifier 5 now has three amplifier stages, one for each of the color signals, and the picture tube now contains three hot cathodes 8 instead of one so that three independent cathode rays are available for the three color signals.
However, to simplify the circuit
arrangement and to save on components, only one measuring stage 7 is
provided which measures all three beam currents successively. Also, the
comparator arrangement 12 forms part control signals from the
successively arriving part measuring signals for the individual beam
currents with the reference signal, and these part control signals are
allocated to the individual color signals and passed on to three storage
units which are contained in the control signal memory 16. From there,
the part control signals are sent via the second input 18 of the
combinatorial stage 2 to the assigned logic elements. The circuit arrangement thus forms three independently acting control loops for the cut-off point control of the individual color signals, in which case only the measuring stage 7 and to some extent at least the comparator arrangement 12 are common to these control loops.
The change detection arrangement 25 now has three change detectors each of which detects the changes with time of the part control signals relating to a color signal. Then via the output 26 each of these change detectors delivers a part change signal to the change signal input 27 of the logic network 22. These part change signals occur independently of one another when the relevent control loop has responded. The logic network 22 evaluates all three part change signals and does not switch into its third stage until all part change signals indicate a steady state of the control loops. Only then, in fact, is it ensured that all the color signals from the beam currents controlled by them are correctly reproduced in the picture tube, and thus no distortions of the displayed image, especially no color purity errors, occur. The color picture displayed then immediately has the correct brightness and color on its appearance when the picture tube is turned on.
FIG. 2 shows a
somewhat more detailed block circuit diagram of an embodiment of a
circuit arrangement equipped for the processing of a picture signal
containing three colour signals. Three color signals for the
representation of the colors red, green and blue are fed to this circuit
arrangement via three input terminals 101, 102, 103. A red color signal
is fed via the first input terminal 101 to a first adder 201, a green
colour signal is fed via the second input terminal to a second adder
202, and a blue colour signal is fed via the third input terminal 103 to
a third adder 203. From outputs 301, 302 and 303 of the adders 201,
202, 203 the color signals are fed to amplifier stages 501, 502 and 503
respectively. Each of the amplifier stages contains a switchable
amplifier 511, 512 and 513, an output amplifier 521, 522 and 523 as well
as a measuring transistor 531, 532 and 533 respectively. The emitters
of these measuring transistors 531, 532, 533 are each connected to a hot
cathode 801, 802, 803 of the picture tube 9 and deliver the cathode
currents, whereas the collectors of measuring transistors 521, 532, 533
are connected to one another and to a first terminal 701 of a measuring
resistor 702 the second terminal of which 703 is connected to earth. The
current gain of the measuring transistors 531, 532 and 533 is so great
that their collector currents coincide almost with the cathode currents.
By measuring the voltage drop produced by the cathode currents at the
measuring resistor 802 it is then possible to measure the cathode
currents and therefore the beam currents in the picture tube 9 with
great accuracy. The falling voltage at the measuring resistor 702 is fed as a measuring signal to an input 121 of a buffer amplifier 120 with a gain factor of one, at the output 122 of which the unchanged measuring signal is therefore available at low impedance. From there it is fed to a first terminal 131 of a reference voltage source 130 which is connected with its second terminal 132 to inverting inputs 111, 112 and 113 of three differential amplifiers 123, 124, 125 respectively. The differential amplifiers 123, 124, 125 also each have a non-inverting input 114, 115, and 116 respectively. These are connected to each other at a junction 117, to earth via a leakage current storage capacitor 126 and to the output 122 of the buffer amplifier 120 via decoupling resistor 118 and a leakage current sampling switch 119. In addition, the input 121 of the buffer amplifier 120 can be connected to earth via a short-circuiting switch 127.
The operation of the comparator arrangement 12 which consists mainly of the buffer amplifier 120, the reference voltage source 130 and differential amplifiers 123, 124, 125 will be explained below with the aid of the pulse diagrams in FIG. 3. FIG. 3a shows a horizontal blanking signal for a television signal which, as the picture signal, controls the beam currents in the picture tube 9. In this diagram, H represents horizontal blanking pulses which follow one another in the picture signal at the time interval of one line duration and by means of which the beam currents are switched off during line flyback between the display of the individual picture lines in the picture tube. FIG. 3b shows a vertical blanking pulse V by means of which the beam currents are switched off during the change ober from the display of one picture to the display of the next picture. FIG. 3c shows a measuring signal control pulse VH which is formed from a vertical blanking pulse lengthened by three line duration.
The short-circuiting switch 127 is now controlled in such a way that it is non-conducting only throughout the duration of the measuring signal control pulse VH and during the remaining time short-circuits the input 121 of the buffer amplifier 120 to earth. This means that a measuring signal only reaches the comparator arrangement 12 during frame change so that the parts of the picture signal which control the beam currents producing the picture in the picture tube exert no influence on comparator arrangement 12 and therefore on the blocking point control.
Throughout the duration of the measuring signal control pulse VH, the measuring signal from output 122, reduced by a reference voltage issued by the reference voltage source 130 between its first 131 and its second terminal 132, is present at the inverting inputs 111, 112, 113 of differential amplifiers 123, 124, 125. If the differential amplifiers 123, 124, 125 were not present, this difference would be fed directly as part control signals to the control signal storage capacitors 161, 162, 162. The differential amplifiers 123, 124, 125 amplify the difference and thus form the control amplifiers of the control loops.
The comparator
arrangement 12 further contains a device for compensation of the
influence of any leakage currents occurring in the picture tube 9. For
this purpose, a voltage to which the leakage current storage capacitor
126 is charged is fed to the non-inverting inputs 114, 115, 116 of the
three differential amplifiers 123, 124 and 125. The charging is
performed by the measuring signal from output 122 of the buffer
amplifier 120 via the decoupling resistor 118 and the leakage current
sampling switch 119 which is closed only within the period of the
vertical blanking pulse V, and in certain cases only during part of the
latter. Within this time the beam currents are, in fact, totally
switched off by the picture signal so that in certain cases only a
leakage current flows through the measuring resistor 702. Consequently,
throughout the duration of the vertical blanking pulse V the measuring
signal corresponds to this leakage current. Because the leakage current
also flows during the remaining time, even outside the duration of the
vertical blanking pulse the measuring signal contains a component
originating from the leakage current which therefore is also contained
in the voltage fed to the inverting inputs 111, 112, 113 of differential
amplifiers 123, 124, 125 and is subtracted out in the differential
amplifiers 123, 124, 125. The part control signal is fed from output 141 of differential amplifier 123 by the first control signal sampling switch 154 to the first terminal 151 of the first control signal storage capacitor 161 during the period of a storage pulse L1 and is stored in the said capacitor. Similarly, the part control signal from output 143 of differential amplifier 125 is fed to the third control signal storage capacitor 163 during the period of a storage pulse L2 and the part control signal from output 142 of differential amplifier 124 is fed to the second control signal storage capacitor 162 during a storage pulse L3. The storage pulses L1, L2 and L3 are illustrated in FIGS. 3d, e and f. They lie in sequence in one of the three line periods by which the measuring signal control pulse VH is longer than the vertical blan
king pulse V. These three line periods form
the sampling interval for the measuring signal or the part measuring
signals, as the case may be. During the remaining periods the outputs,
141, 152, 143 of the differential amplifiers 123, 124, 125 are isolated
from the control signal storage capacitors 161, 162, 163 so that no
interference can be transmitted from there and any distortion of the
stored part control signals caused thereby is eliminated. For the
duration of storage pulses L1, L2 and L3 the color signals at the input
terminals 101, 102, 103 are at their reference level i.e. in the present
embodiment at a level, corresponding to the blocking point or at a
fixed level with respect to it so that the control loops can adjust to
this level.The switchable amplifiers 511, 512, and 513 each receive at each input 241, 242, 243 a blanking signal BL1, BL2, BL3 respectively, the curves of which are shown in FIGS. 3g, h, i. These blanking signals interrupt the supply of the color signals during line flybacks and frame change, i.e. during the period of the measuring signal control pulse VH, and thus the beam currents in these time intervals are switched off. Naturally, the red color signal is let through during the first line period after the end of the vertical blanking pulse V, the blue color signal during the second line period after the end of the vertical blanking pulse V and the green color signal during the third line period after the end of the vertical blanking pulse V by the switchable amplifiers 511, 512, 513 respectively so that they can control the beam currents. Blanking signals BL1, BL2 and BL3 also provide for interruptions in the frame change blanking pulse, which corresponds to the measuring signal control pulse, in the corresponding time intervals. In these time intervals the beam currents are measured and part control signals are determined from the part measuring signals and stored in the control signal storage capacitors 161, 162, 163.
The circuit arrangement shown in FIG. 2 further contains a trigger circuit 19 to which a supply voltage is fed via a supply terminal 190. Via a reset input 191 a voltage is also
supplied to
the trigger circuit 19 from a third terminal 133 of the reference
voltage source 130. When the circuit arrangement is turned on, this
voltage is designed so as to be delayed with respect to the supply
voltage so that when the circuit arrangement is brought into operation
the interplay of the two voltages produces a switch-on reset signal such
that a low-value voltage pulse occurs at the reset input 191 during
turn on, which means that the trigger circuit 19 is set in its first
state. The reset input 191 can also be connected to another circuit of
any configuration which generates a switch-on reset signal when the
picture tube is turned on. The trigger circuit 19 is further connected via a second connection 23 to a logic network 22 which, when the circuit arrangement is turned on, is also set into a first state via the second connection 23. In this first state the logic network 22 delivers a blocking signal at a blocking output 240 which is fed to the three switchable amplifiers 511, 512, 513. By this means the supply of the color signals to the output amplifiers 521, 522, 523 is interrupted completely so that no beam currents can be generated by these. No picture is therefore displayed.
An insertion signal EL which extends over the three line periods by which the measuring signal control pulse VH is longer than the vertical blanking pulse V, i.e. over the sampling interval, is also fed via a line 233 to the trigger circuit 19 and the logic network 22. As long as the trigger circuit 19 is in its first state, this insertion pulse EL is issued via a control output 192 from the trigger circuit 19 and fed to the pulse generator 244. During the period of the insertion pulse EL this generator produces a voltage pulse of a definite magnitude and passes this to output amplfiiers 521, 522, 523 as an auxiliary pulse via switching diodes 245, 246, 247. By this means the beam currents are switched on for a short time so as to receive a measuring signal despite the disconnected color signals as soon as at least one of the hot cathodes 801, 802, 803 delivers a beam current.
In its first state the trigger circuit 19 also delivers a signal via a control line 211, and this signal is used to switch the outputs 141, 142, 143 of the differential amplifiers 123, 124, 125 to earth potential or practically to earth potential. This suppresses effects of voltages at the inputs 111 to 116 of the differential amplifiers 123, 124, 125, especially effects of the reference voltage source 130 which may in some cases initiate incorrect charging of the control signal storage capacitors 161, 162, 163.
When part change signals are present from all change detectors 251, 252, 253, when therefore all control loops have responded, the logic network 22 switches from its second to its third state. The blocking signal from the blocking output 240 is now completely disconnected such that the switchable amplifiers 511, 512, 513 are now switched only by the blanking signals BL1, BL2, BL3. The colour signals are then switched through to the output amplifiers 521, 522, 523 and the picture is displayed in the picture tube.
FIG. 4 shows an embodiment for a trigger circuit 19 and a logic network 22 of the circuit arrangements as shown in FIGS. 1 or 2. The trigger circuit 19 contains a flip-flop circuit formed from two NAND-gates 194, 195 to which the switch-on reset signal, by which the trigger circuit 19 is returned to its first stage, is fed via the reset input 191. All the elements of the circuit arrangement in FIG. 4 are shown in positive logic. Thus, a short-time low voltage at the reset input 191 immediately after the circuit arrangement is started up is used to set the flip-flop circuit 194, 195 in such a way that a high voltage occurs at the output of the second NAND gate 194 and a low voltage at the output of the second NAND gate 195. The low voltage at the output of the second NAND gate 195 blocks differential amplifiers 123, 124, 125 via the control line 211 in the manner described.
The insertion pulse EL is fed via the line 233 to the trigger circuit 19, is combined via an AND gate 196 with the signal from the output of the first NAND gate 194 and is delivered at the control output 192 for the purpose of controlling the pulse generator 244.
The signals from the outputs of the NAND-gates 194, 195 are fed via a first line 231 and a second line 232 of the second connection 23 as a switching signal to the logic network 22. The first line 231 is connected to reset inputs R of three part change signal memories 221, 222, 223 in the form of bistable flip-flop circuits which when the circuit arrangement is started up are reset via the first line 231 in such a way that they carry a low voltage at their outputs Q. The second line 232 of the second connection 23 leads via three AND gates 224, 225, 226 to setting inputs S of the three part change signal memories 221, 222, 223. By means of the AND gates 224, 225, 226 the signal on the second line 232 of the second connection 23 is combined each tim
e with one of the part change signals supplied via
the terminals 271, 272, 273. The signals from the outputs Q of the part
change signal memories 221, 222, 223 are combined by means of a
collecting gate 227 in the form of an NAND gate and are held ready at
its output 228. The measuring signal is fed to the trigger circuit 19 via the measuring signal input 20 and passed to a first input 197 of a threshold detector 198 to which at a second input a threshold value, in the form of a threshold voltage for example, produced by a threshold generator 199 is also supplied. When the voltage at the first input 197 of the threshold detector 198 is smaller than the voltage delivered by the threshold generator 199, the threshold detector 198 delivers a high voltage at its output 200. When, on the other hand, the voltage at the first input 197 is greater than the voltage of the threshold generator 199, the voltage at the output 200 jumps to a low value. This voltage is supplied as the setting signal of the flip-flop circuit 194, 195, reverses the latter and thereby switches the trigger circuit 19 into its second state when the voltage at the first input 197 exceeds the voltage of the threshold generator 199.
Between the output 200 and the flip-flop circuit 194, 195 in the circuit arrangem
ent
shown in FIG. 4 there is inserted an inquiry gate 181 in the form of an
OR gate to which an inquiry pulse is fed via an inquiry input 193 of
the trigger circuit 19. This ensures that the flip-flop circuit 194, 195
is switched over only at a time fixed by the inquiry pulse--in the
present case a negative voltage pulse--and not at any other times due to
disturbances. As such an inquiry pulse it is possible to use, for
example, a pulse which occurs in the second line period after the end of
the vertical blanking pulse V, i.e. one which largely corresponds to
the storage pulse L2. After the switching over of the flip-flop circuit 194, 195 corresponding to the setting of the trigger circuit 19 into the second state, appropriately modified signals are supplied via the control line 211 and the output 192 for the purpose of controlling the pulse generator 244 and the differential amplifiers 123, 124, 125. Modified voltages also appear on the lines 231, 232 of the second connection 23, and these voltages release the part change signal memories 221, 222, 223 such that they can each be set when the part change signals reach the terminals 271, 272, 273.
In certain cases, a further flip-flop circuit
234 is inserted in the lines 231, 232
to delay the signals passing along these lines; this is reset via the
first line 231 when the circuit arrangement is started up and thus it
also resets the part change signal memories 221, 222, 223. However,
after the trigger circuit 19 is switched into the second state the
further flip-flop circuit 234 is not set via the second line 232 of the
second connection 23 until a release pulse arrives via a release input
235 and another AND gate 236, for example a period of approximately the
interval of two vertical blanking pulses V after the switching of the
trigger circuit 19 into the second state. In this way it is possible to
bridge a period of time in which no defined signal values are present at
the terminals 271, 272, 273. The signal at the output 228 of the collecting gate 227 changes its state when the last of the three part change signals has also arrived and has set the last of the three part change signal memories. The signal is then combined via a gate arrangement 229 of two NAND gates and one AND gate with the insertion pulse EL of line 223 and with the signal on the second line 232 of the second connection 23 or from the output Q of the further flip-flop circuit 234 to the blocking signal delivered at the blocking output 24 which is fed to the switchable amplifiers 511, 512, 513.
The circuit arrangement described is designed in such a way that the trigger circuit 19 remains in its second state and logic network 22 remains in its third state even if charging currents reappear at the difference signal storage cpacitors 161, 162, 163 due to disturbances during the operation of the circuit arrangement. The cutoff point control then makes readjustments without the displayed picture being disturbed.
In the circuit arrangement shown in F
IG. 2, the green
color signal can also be let through during the second line period after
the end of the vertical blanking pulse V and the blue color signal
during the third line period after the end of the vertical blanking
pulse V by the switchable amplifiers 511, 512, 513 for the purpose of
controlling the beam currents. The storage pulses L2 and L3 at the
control signal sampling switches 155 and 156 and the second and third
blanking signals BL2 and BL3 at the blanking inputs 242 and 243 are then
to be interchanged. The resulting insertion signals A2 and A3 as shown
in FIGS. 3m and n are also interchanged then accordingly. In FIG. 2 a dashed line is used to indicate which components of the circuit arrangement can be combined advantageously to form an integrated circuit. The first terminals 151, 152, 153 of the difference signal storage capacitors 161, 162, 163, one terminal 128 of leakage current storage capacitor 126, three terminals 524, 525, 526 in the leads to the output amplifiers 521, 522, 523 as well as a line connection 704 between the first terminal 701 of the measuring resistor 702 and the input 121 of the buffer amplifier 120 will then form the connecting contacts of this integrated circuit
Deflection Unit (Ablenkung 110 29504-007.05) with TDA 2595 (Philips) + TDA2653A
TDA2595 Horizontal combination
GENERAL DESCRIPTION
Features
· Positive video input; capacitively coupled (source impedance < 200 W)
· Adaptive sync separator; slicing level at 50% of sync amplitude
· Internal vertical pulse separator with double slope integrator
· Output stage for vertical sync pulse or composite sync depending on the load; both are switched off at muting
· j1 phase control between horizontal sync and oscillator
· Coincidence detector j3 for automatic time-constant switching; overruled by the VCR switch
· Time-constant switch between two external time-constants or loop-gain; both controlled by the coincidence detector j3
· j1 gating pulse controlled by coincidence detector j3
· Mute circuit depending on TV transmitter identification
· j2 phase control between line flyback and oscillator; the slicing levels for j2 control and horizontal blanking can be set
separately
· Burst keying and horizontal blanking pulse generation, in combination with clamping of the vertical blanking pulse
(three-level sandcastle)
· Horizontal drive output with constant duty cycle inhibited by the protection circuit or the supply voltage sensor
· Detector for too low supply voltage
· Protection circuit for switching off the horizontal drive output continuously if the input voltage is below 4 V or higher
than 8 V
· Line flyback control causing the horizontal blanking level at the sandcastle output continuously in case of a missing
flyback pulse
· Spot-suppressor controlle
d by the line flyback control.The phase-lock loop circuit of the video display apparatus is synchronised by a horizontal synchronising input signal. The phase-lock loop circuit includes a frequency-to-voltage converter that is responsive to the synchronising input signal for generating a control voltage indicative of the frequency of the synchronising input signal. A voltage-to-current converter responsive to the control voltage generates a control current that is indicative of the frequency of the input signal. The control current varies the free running frequency of a controlled oscillator of the phase-lock loop circuit. The free running frequency of the oscillator is directly related to the frequency of the input signal. The phase of the output signal of the oscillator is controlled by a second control current that is indicative of the phase difference between the oscillator output signal and the synchronising input signal. When the frequency of the input signal is lower than a predetermined frequency, the first control current is at a level that causes the free running frequency to be above a predetermined minimum frequency.
TDA2653A Vertical deflection circuit:
DESCRIPTION
The TDA2653A is a monolithic integrated circuit for vertical deflection in large screen colour television receivers.
The circuit incorporates the following functions:
· Oscillator; switch capability for 50 Hz/60 Hz operation
· Synchronization circuit
· Blanking pulse generator with guard circuit
· Sawtooth generator with buffer stage
· Preamplifier with fed-out inputs
· Output sta
ge with thermal and short-circuit protection· Flyback generator
· Voltage stabilizers.
APPLICATION INFORMATION
The function is described against the corresponding pin number
1, 13. Oscillator
The oscillator frequency is determined by a potentiometer at pin 1 and a capacitor at pin 13.
2. Sync input/blanking output
Combination of sync input and blanking output. The oscillator has to be synchronized by a positive-going
pulse between 1 and 12 V. The integrated frequency detector delivers a switching level at pin 12.
The blanking pulse amplitude is 20 V with a load of 1 mA.
3. Sawtooth generator output
The sawtooth signal is fed via a buffer stage to pin 3. It delivers the signal which is used for linearity control,
and drive of the preamplifier. The sawtooth is applied via a shaping network to pin 11 (linearity) and via a
resistor to pin 4 (preamplifier).
4. Preamplifier input
The DC voltage is proportional to the output voltage (DC feedback). The AC voltage is proportional to the
sum of the buffered sawtooth voltage at pin 3 and the voltage, with opposite polarity, at the feedback
resistor (AC feedback).
5. Positive supply of output sta
geThis supply is obtained from the flyback generator. An electrolytic capacitor between pins 7 and 5, and a
diode between pins 5 and 9 have to be connected for proper operation of the flyback generator.
6. Output of class-B power stage
The vertical deflection coil is connected to this pin, via a series connection of a coupling capacitor and a
feedback resistor, to ground.
7. Flyback generator output
An electrolytic capacitor has to be connected between pins 7 and 5 to complete the flyback generator.
8. Negative supply (ground)
Negative supply of output stage and small signal part.
9. Positive supply
The supply voltage at this pin is used to supply the flyback generator, voltage stabilizer, blanking pulse
generator and buffer stage.
10. Reference voltage of preamplifier
External adjustment and decoupling of reference voltage of the preamplifier.
11. Sawtooth capacitor
This sawtooth capacitor has been split to realize linearity control.
12. 50 Hz/60 Hz switching level
This pin delivers a LOW voltage level for 50 Hz and a HIGH voltage level for 60 Hz. The amplitudes of the
sawtooth signals can be made equal for 50 Hz and 60 Hz with these levels.
E/W Correction unit (OST-WEST 29504-009.01)This invention relates to a deflection circuit in which the amplitude of the deflection current may be varied over a relatively wide range without changing the high voltage amplitude or the deflection retrace time. Modulation of the deflection current amplitude is desired for such purposes as east-west pincushion distortion correction, picture width adjustment, or similar applications.
General disadvantages of known east-west correction circuits are that they consume a relatively large amount of power, impose restrictions on the design of certain linearity correction circuits, or produce undesirable retrace time modulation. In addition, some east-west correction circuits require additional components serially coupled in the deflection current path. This series connection complicates the design of linearity correction circuits that require a grounded "S"-shaping capacitor for proper circuit operation. The amplitude of the deflection current may be varied or modulated over a relatively wide range without substantially affecting the high voltage amplitude or the deflection retrace time. Modulation of the deflection current amplitude is desired for such purposes as east-west pincushion distortion correction and picture width adjustment.
Tuning control Unit (Abstimmung Baustein 29504-003.34 ) Siemens based PLL Tuning Frequency Synthesizer. With SDA2030A 001 + SDA2006 + SDA2112GRUNDIG SUPER COLOR C8443 SERIE F3026 " BERLINO 26 " CHASSIS CUC70KT PLL MICROCOMPUTER Frequency synthesizer tuning system for television receivers:SDA2030A 001 + SDA2006 + SDA2112
" A method for tuning a television receiver having automatic frequency control to the carrier frequency of a selected broadcast channel with an associated channel number including generating a variable frequency signal by means of a local oscillator, generating a reference frequency signal by means of a reference oscillator, and generating a local oscillator correction signal for matching an intermediate frequency signal derived from said local oscillator signal and the carrier frequency signal with a predetermined nominal intermediate frequency signal, said method being characterized by the use of a microcomputer and comprising:
generating binary signals representing first and second digital tune words, said digital tune words representing a selected channel;
storing said first and second digital tune words in a first data memory in said microcomputer;
reading said first and second digital tune words from said first memory and generating a divided-down local oscillator frequency by the use of said first digital tune word and a divided-down reference oscillator frequency by the use of said second digital tune word;
comparing said divided-down local oscillator and reference frequencies and generating a control signal representative of the difference in frequency of said divided-down local oscillator and reference frequencies;
coupling said control signal to said local oscillator for causing it to be locked to the frequency of said received carrier signal;
mixing the local oscillator frequency signal and the carrier frequency signal to generate an intermediate frequency signal;
comparing said intermediate frequency signal with said predetermined nominal intermediate frequency signal and providing a tuning voltage to said microcomputer, said tuning voltage being indicative of the magnitude and direction of a tuning error between said intermediate frequency signal and said predetermined nominal intermediate frequency signal;
incrementally adjusting the reference oscillator frequency by means of a tuning signal provided to said reference oscillator by said microcomputer in response to said tuning voltage;
detecting when the incrementally changing, divided-down reference oscillator frequency causes the intermediate frequency signal to pass said predetermined nominal intermediate frequency signal; and
incrementally stepping the divided-down reference oscillator frequency back a predetermined number of steps following the passage of said predetermined nominal intermediate frequency signal by said intermediate frequency signal in tuning said television receiver to the selected channel.
"
A
television tuning system employs a frequency synthesizer system for
establishing the tuning of the receiver. A programmable frequency
divider counter is connected between the output of a reference
oscillator and a phase comparator to which the output of the local
oscillator in the tuner also is applied. The phase comparator output
provides a tuning voltage for controlling the tuning of the local
oscillator. A microprocessor is used to control the count of the
programmable frequency divider and initially to set a count
corresponding to the selected channel in a counter connected between the
output of the local oscillator and the phase comparator. The tuning
consists of three discrete time periods. First, a settling time to
allow channel change transients to settle; second, a short period of
forced search at a relatively rapid rate to insure proper tuning; and
third, a slower rate of step-by-step correction to accomodate for
station drift and the like during reception. This third time period is
initiated either by the passage of a fixed length of time following
the start of the forced search period or by sensing a preestablished
number of changes of state in the output of the frequency
discriminator during the forced/search period.1. A tuning system for the tuner of a television receiver capable of receiving a composite television signal and including frequency discriminator (AFT) circuit means, said system including in combination:
a reference oscillator providing a reference signal at a predetermined frequency;
a local oscillator in the tuner providing a variable output frequency in response to the application of a control signal thereto;
a programmable frequency divider means having first and second inputs coupled respectively to the output of said reference oscillator and said local oscillator for producing signals on first and second outputs having frequencies which are a programmable fraction of the frequency of the signals applied to the inputs thereto;
phase comparator means having one input coupled with the first output of said programmable frequency divider means and having another input coupled with the second output of said programmable frequency divider means for developing a control signal and applying such control signal to said local oscillator for controlling the output frequency thereof;
counter circuit means coupled with said programmable frequency divider means for initially setting said divider means to a predetermined division ratio and operating to change the programmable fraction of division thereof in accordance with changes in the count in said counter circuit means;
control circuit means coupled with the output of said frequency discriminator means and further coupled with said counter circuit means for causing said counter circuit means to count at a first rate in a predetermined direction determined by the state of the output signal from said discriminator means in the absence of a predetermined signal output from said frequency discriminator means until a predetermined maximum count is attained, thereupon resetting said counter circuit means to a count which is a predetermined amount less than said maximum predetermined count and continuing to count at said first rate in the same predetermined direction from said new count to continuously change the programmable fraction of said frequency divider means in accordance with the state of operation of said counter circuit means, said control means operating in response to said predetermined signal output from the frequency discriminator means for terminating operation of said counter circuit means; and
further means for terminating operation of said counter circuit means at said first rate and causing operation thereof at a second slower rate.
2. The combination according to claim 1 wherein said further means includes timing means initiated into operation simultaneously with the setting of said divider means to a predetermined division ratio, and after a predetermined time interval said timing means producing an output signal applied to said counter circuit means to cause operation thereof to take place at said second slower rate. 3. The combination according to claim 1 wherein said counter circuit means includes a reversible digital counter coupled with said programmable frequency divider, means and said control circuit means causes said counter circuit means to count in said predetermined direction when the output of said frequency discriminator is of a first state and to count in the opposite direction when the output of said frequency discriminator is of second state; and said further means comprises means coupled with the output of said frequency discriminator and with said counter circuit means to take place at said second slower rate in response to a predetermined number of changes of state of frequency discriminator. 4. The combination according to claim 3 further including means responsive to the selection of a new channel in said television receiver for resetting said further means to an initial condition of operation. 5. The combination according to claim 4 wherein said further means comprises a search termination counter means operative to provide an output signal applied to said counter circuit means in response to a count thereby of a predetermined number of changes of state of said frequency discriminator to cause said counter circuit means to be operated at said second slower rate.
Description:
BACKGROUND OF THE INVENTION
Both
of the above mentioned patents are directed to frequency synthesizer
tuning systems for use with television receivers to enable operation
of the receivers with minimal viewer fine tuning adjustments. By the
utilization of the frequency synthesizer tuning systems of these
patents, the fine tuning adjustment which is necessary with conventional
types of television receiver tuning systems has been substantially
eliminated. The system employed in the '953 patent permits utilization
of a frequency synthesizer tuning system which correctly tunes to a
desired television station or channel even if the transmitted signals
from that station are not precisely maintained at the proper
frequencies. The '535 patent is directed to a signal seek tuning system
adaptation of the frequency synthesizer tuning system of the '953
patent which still permits implementation of all of the desired
wide-band pull in range of the frequency synthesizer system of the '953
patent.
The systems of the foregoing patents operate effectively to correct automatically for frequency offsets in a frequency synthesizer tuning system without affecting the operation of the conventional frequency synthesizer used in the system. The systems of these patents are in widespread use commercially and permit direct selection, with automatic fine tuning adjustment, of any desired VHF channel which the viewer wishes to observe. In addition, the signal seek adaptation disclosed in the '535 patent couples all of the advantages of the frequency synthesizer tuning system of the '953 patent with the desirability of providing bidirectional signal seek operation.
While the systems disclosed in the foregoing patents operate in a highly satisfactory manner to accomplish the desired results of accurate tuning without the necessity of fine tuning adjustments, the circuitry for accomplishing the desired results is somewhat complex. It is desirable to reduce the circuit complexity and the number of signal detectors for accomplishing these results without compromising the accuracy of operation of the system.
SUMMARY OF THE INVENTION
Accordingly, it is an object of this invention to provide an improved tuning system for a television receiver.
It is an additional object of this invention to provide an improved frequency synthesizer tuning system for a television receiver.
It is another object of this invention to provide an improved frequency synthesizer tuning system for a television receiver which includes a provision for adjusting the synthesizer loop for frequency offsets in the received signal with a minimum number of signal detectors.
It
is a further object of this invention to tune the local RF
oscillator of a television receiver to the correct frequency for a
selected channel with a frequency synthesizer tuning system, and
automatically to change the reference frequency of the synthesizer
system, or adjust the count of a programmable divider that produces a
signal that divides the frequency of the local oscillator of the
tuner, if the AFT signal produced by the AFT frequency discriminator
of the receiver is outside a predetermined range corresponding to
correct tuning.
It is still another object of this invention to provide an improved frequency synthesizer tuning system for a television receiver which operates to adjust the synthesizer loop for frequency offsets in the received signal over a relatively wide pull in range in response to the output of the receiver frequency discriminator by changing the division ratio of a programmable frequency divider in the reference oscillator leg or local oscillator leg of the synthesizer loop at a first relatively high rate from an initial nominal value to a pre-established maximum in one direction, and then resetting the division ratio to a second nominal value once the maximum is reached and continuing to incrementally change the division ratio in the same direction from the second nominal value until a properly tuned condition is indicated by the output of the receiver AFT frequency discriminator, followed by control at a lower rate of operation to maintain tuning during transmitting station drifts.
In
accordance with a preferred embodiment of this invention, the
frequency synthesizer tuning system for a television receiver includes
a stable reference oscillator and a voltage controlled local
oscillator in the tuner. A programmable frequency divider is connected
between the output of the reference oscillator and one input to a
phase comparator, the other input of which is supplied by the output
of the local oscillator. The output of the phase comparator then
comprises a control signal which is supplied to the local oscillator
to control the frequency of its operation.
A counter circuit is connected to the programmable frequency divider for initially setting the divider to a predetermined division ratio upon selection of a desired channel by the viewer. The counter then operates to change the programmable fraction of the division ratio at a first relatively high rate in a direction controlled by the output from the receiver picture carrier discriminator in the absence of a predetermined signal output derived from the discriminator. A control means causes the counter circuit to count in this direction until it is determined that a station is tuned or a predetermined maximum count is attained if no station is correctly tuned, thereupon resetting the counter circuit to a count which is a predetermined amount less than the maximum predetermined count. Counting is continued in the same predetermined direction from the new lesser count to continuously change the programmable fraction of the frequency divider in accordance with the state of operation of the counter.
The
high rate operation of the counter is terminated by the control
means in response to a predetermined signal from the output of the
discriminator, indicating that a station is correctly tuned, or after a
fixed time-out interval; so that the system automatically adjusts
for frequency offsets of the received signal which otherwise would
cause the station to be mistuned if a conventional frequency
synthesizer tuning system were used. After termination of the high
rate operation of the counter, it is switched to a lower rate
operation for maintaining tuning during transmitting station drifts.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a block diagram of a television receiver employing a preferred embodiment of the invention;
FIG. 2 is a detailed block diagram of a portion of the circuit of the preferred embodiment shown in FIG. 1;
FIG. 3 is a detailed circuit diagram of a portion of a circuit shown in FIG. 1;
FIG. 4 is a flow chart of the control sequence of operation of the circuit shown in FIG. 1 and 2; and
FIG. 5 shows a waveform and time/frequency chart, respectively, useful in explaining the operation of the circuit shown in FIGS. 1, 2 and 3.
DETAILED DESCRIPTION
Referring now to the drawings, the same reference numbers are used throughout the several figures to designate the same or similar components.
FIG. 1 is a block diagram of a television receiver, which may be a black and white or color television receiver. Mo
st
of the circuitry of this receiver is conventional, and for that
reason it has not been shown in FIG. 1. Added to the conventional
television receiver circuitry of FIG. 1, however, is a frequency
synthesizer tuning system, in accordance with a preferred embodiment
of the invention, which is capable of automatically changing the
reference frequency when a frequency offset exists in the received
signal for a particular channel.
Transmitted composite television signals, either received over the air or distributed by means of a master antenna TV distribution system, are received by an antenna 10 or on antenna input terminals to the receiver. As is well known, these composite signals include picture and sound carrier components and synchronizing signal components, with the composite signal applied to an RF and tuner stage 11 of the receiver. The stage 11 includes the conventional RF amplifiers and tuner sections of the receiver, including a VHF oscillator section and a UHF oscillator section. Preferably, the UHF and VHF oscillators are voltage controlled oscillators, the freuency of operation of which are varied in response to a tuning voltage applied to them to effect the desired tuning of the receiver.
The output of the RF and tuner stages 11 is applied to the remainder of the television receiver 14, which includes the IF amplifier stages for supplying conventional picture (video) and sound IF signals to the video and sound processing stages of the receiver 14. The circuitry of the receiver 14 may be of any conventional type used to separate, amplify and otherwise process the signals for application to a cathode ray tube 16 and to a loudspeaker 17 which reproduce the picture and sound components, respectively, of the received signal.
The
receiver 14 also includes a conventional AFT or automatic fine
tuning discriminator circuit and additionally may include a synch
separator circuit for producing an output in response to the presence
of vertical synchronizatin pulses, a picture carrier detection
circuit, and an automatic gain control (AGC) amplifier. Outputs
representative of these sensor components are shown as being coupled
over a group of lead 20 to sensory circuitry 22, which in turn
couples outputs representative of the operation of these various
sensor circuits to a microprocessor unit 23 for controlling the
operation of the microprocessor unit.
The microprocessor unit 23 is utilized in the system of FIG. 1 for controlling the operation of a frequency synthesizer tuning system capable of automatic offset correction. When the viewer desires to select a new channel, he enters the desired channel number into a channel selection keyboard 25. There are a number of different keyboards which may be employed to accomplish this function, and the particular design is not important to this invention. The channel selector keyboard 25 also may include switches or keys for initiating a signal seek function in either the "up" or "down" direction.
Information represented by the selection of channel numbers on the keyboard 25 is supplied to the microprocessor unit 23 which provides output signals over a corresponding set of leads 27 to the tuners (local oscillators) 11 to effect the appropriate band switching control for the tuners 11 in accordance with the particular channel which has been selected. In addition, the keyboard 25, operating through the microprocessor unit 23, provides output signals which operate a channel number display 29 to provide an appropriate display of the selected channel number to the viewer.
The microprocessor M3870 unit 23 also processes the signals which are used to operate the channel number display 29 through a multiplexing circuit operation to decode the selected channel number into a parallel encoded signal. This signal is applied to corresponding inputs of the count-down counter or programmable frequency divider 31 to cause the division number of the divider 31 to relate to the divided down frequency of the tuner local oscillators connected to the input of the divider 31 through a prescaler divider circuit 32 to the frequency of the reference oscillator 34. Thus, the division number or division ratio of the local oscillator frequency obtained from the output of the programmable divider 31 is appropriately related to the frequency of the reference crystal oscillator 34.
The
output of the oscillator 34 also is applied through a countdown
circuit or programmable frequency divider 35. Conventional frequency
synthesizer techniques are employed; and the microprocessor unit 23
automatically compensates, through appropriate code converter
circuitry, for the non-uniform channel spacing of the television
signals. It has been found most convenient to cause the programmable
frequency divider 31 to divide by numbers corresponding directly to
the oscillator frequency of the selected channel, for example, 101,
107, 113 . . . up to 931.
In accordance with the time division multiplex operation of the microprocessor 23, the count of the programmable frequency divider 35 initially is adjusted to a fixed count by the application of appropriate output signals from the microprocessor unit 23 to a point selected to be at or near the mid-point of the operating range of the programmable frequency divider 35. Thus, the output of the divider 35 is a stable reference frequency (because the input is from the reference crystal oscillator 34) which is used to establish initially and to maintain tuning of the receiver to the selected channel.
The output of the programmable divider 35 is applied to one of two inputs of a phase comparator circuit 37. The other input to the phase comparator circuit 37 is supplied from the selected one of the VHF or UHF oscillators in the tuner stages 11 through the programmable frequency divider 31. The phase comparator circuit 37 operates in a conventional manner to supply a DC tuning control signal through a phase locked loop filter circuit 39 and over a lead 40 to the oscillators in the tuner system 11 to change and maintain their operating frequency.
With the exception of the use of the microprocessor unit 23, the operation of the system which has been described thus far is that of a relatively conventional frequency synthesizer system incorporated into a television receiver. This system is similar to the system of the '953 patent. As in the system of that patent, the system shown in FIG. 1, when the transmitted station or station received on a master antenna distribution system provides the station or channel signals at the proper frequency, operates as a relatively conventional frequency synthesizer system. If, however, there is a frequency offset in the received signal to cause the carrier of the received signal to be displaced from the frequency which it should have to some other frequency, it is possible that the system would give the appearance of mistuning to the received station. The microprocessor 23, operating in conjunction with the sensory circuitry 22, is employed in conjunction with the countdown or programmable frequency divider circuit 35 to eliminate this disadvantage and still retain the advantages of frequency synthesizer tuning.
Reference now should be made to FIG. 2 which shows details of t
he
interface between the keyboard 25, the microprocessor unit 23, and
the circuitry used in the frequency synthesizer portions of the
system. A commercially available microprocessor which has been used
for the microprocessor 23, and which forms the basis for the
diagramatic representation of the microprocessor in FIG. 2, is the
Matsushita Electronics Corporation MN1402 four-bit single-chip
microcomputer. This microcomputer has two, four-bit parallel input
ports labeled "A" and "B". In addition, three output ports, a five-bit
output port "C" and two four-bit output ports "D" and "E" are
provided. The internal configuration of the microcomputer 23 includes
an arithmetic logic unit (ALU), a read only memory (ROM) for storing
instructions and constants, and a random access memory (RAM) used for
data memory, arranged into four files, each file containing 16
four-bit words. These words are selected by X and Y registers and this
memory is used, for example, for timers, counters, etc., and also is
used to hold intermediate results. To facilitate an understanding of
the operation of the system, a portion of this memory is shown in
FIG. 2 as a clock 81 and a reversible counter 82 connected between
the "B" input port and the "D" output port. The microcomputer 23 is
programmed to permit it to operate in conjunction with the remainder
of the circuits shown in FIG. 2. The programming techniques are
standard, and the microcomputer 23 itself is a standard commercially
available circuit component.
There are several system parameters that must be selected in the operation of the system shown in FIG. 2. The selection of the nominal frequency of the two signals that feed the phase comparator circuit 37 is an example. Channel selection is provided by changing the frequency division ratio of the selector counter 31 which divides the local oscillator signal after this signal is passed through a prescaler circuit 32 and a divide-by-two divider circuit 41. The nominal frequency from the programmable frequency divider 31 (selector counter) is selected so that the local oscillator (tuner) 11 can be set exactly on frequency for all channels.
Since
the frequency divider 31 is able to divide only by integer numbers,
one distinct frequency possibility in the range of one KHz is
obtained, another in the range of two KHz, etc. A choice must be made
as to which of these values is optimum. Each value yields the
nominal frequency of all of the 82 channels by simply multiplying by
an appropriate integer for each channel. To simplify the phase locked
loop filtering problem by the filter 39, it is desirable that the
frequencies of the signals supplied to the phase comparator 37 are as
high as possible. This permits rapid acquisition of a new channel
along with a very clean DC control signal to adjust the local
oscillator. A trade-off for this, however, must be made to permit
fine tunning adjustment of the local oscillator automatically to
correctly tune in stations which are off their assigned frequency, or
to manually provide this feature, if desired. The two-speed
operation of the system in accordance with the present invention
allows a better trade-off to be made by allowing rapid acquisition
and then a slower speed for precise tuning.
A compromise solution which is utilized in the circuit of FIG. 2 is to cause the frequency division chain from the local oscillator 11 in the tuner to the phase comparator 37 to be composed of the fixed divide-by-256 prescaler 32, and a fixed divide-by-4 division, which is accomplished by the divider 41 at the input of the counter 31 and a second divider 42 at the output of the counter 31. The variable frequency divider counter 31 then is loaded by means of three latch circuits 44, 45 and 46 at an appropriate time by the time division multiplex operation of the microcomputer 23 and a number that programs the programmable frequency divider counter 31 to divide by the numerical value of the frequency of the local oscillator in MHz for the channel selected. For example, if the receiver is to be tuned to channel 2, which has a nominal local oscillator frequency of 101 MHz, the programmable frequency divider 31 is set to divide by 101. If the receiver is to be tuned to channel 83, which has a nominal local oscillator frequency of 931 MHz, the programmable frequency divider 31 is set to divide by 931. In both cases, the variable divider 31 produces a 1 MHz signal. However, because of the fixed divide-by-256 and the two fixed divide-by-two dividers in series with the programmable divider 31, an output frequency of 976.5625 Hz is supplied from the output of the divider 42 to the upper input of the phase comparator 37.
The
division ratio of the selector counter 31 is established by
appropriate output signals from the latch circuits 44, 45 and 46, as
mentioned above. The initial operation for changing, or maintaining,
the division ratio of the divider 31 is established by an entry of the
two digits of the selected channel number in the keyboard 25. The
microcomputer 23 operates as a time division multiplex system for
continuously monitoring the input ports and the output ports to control
the operation of the remainder of the system. The selection of the
two digits of the desired channel number is affected by a time
division multiplex iscanning of the outputs of the D output port of
microcomputer 23 and providing that information at the A input port.
From here the information is translated again to the D output ports to
the appropriate drivers of the channel number display circuit 29 and
to the latches 44, 45 and 46, and to a pair of similar four bit
latches 49 and 50 which control the divider ratio of the counter 35.
Although the D output ports of the microcomputer 23 are connected in common to all of these various portions of the circuit, the selection of which of the latches are enabled to respond to the particular output signals appearing on the D output ports at any given time is effected through the C and E output ports of the microcomputer 23 in a time division multiplex fashion. A decoder circuit 52, connected to the lowermost three outputs of the E output port of the microcomputer 23, is used to apply unique decoding signals at different times in the time division multiplex sequence of operation of the microcomputer 23 to the five latch circuits 44, 45, 46, 49 and 50, respectively. At any given time in the sequence, only one of these latch circuits is enabled for operation. A latch load signal is applied from the upper output (EO3) at each cycle of operation of the signals appearing on the E output port to set the latch circuit which is enabled by the output of the decoding circuit 52 with the data appearing on the other inputs to the latch circuit. This data simultaneously appears on the four outputs of the D output port of the microcomputer 23.
Thus, in rapid sequence, the latch circuits 44, 45 and 46 are set to store the division number corresponding to the selected channel entered onto the keyboard 25, and the latch circuits 49 and 50 are each operated to set the programmable divider reference counter 35 to a center or nominal count, which is always the same upon the selection of a new channel on the keyboard 25. Similarly, the two right-hand outputs of the C output port (CO6 and CO5) enter the two digits of the selected channel number in the drivers of the display circuit 29 at the proper time in the binary encoded sequence when these digits appear on the four-bit binary encoded representation of the D output port. This results in a visual display of the channel number selected.
In addition to the selection of a channel number directly by the keyboard 25, the keyboard also may include an additional switch 56, which is scanned in the time division multiplex sequence to determine if the receiver is placed in a "seek" mode of operation (when the signal seek capability is incorporated into such a receiver). Operating in conjunction with the signal seek switch 56 are a pair of "up" and "down" seek direction input switches shown with a graphic representation of the seek directions on the keyboard 25. A further provision is provided by two keys labeled "U" and "D", which are used for "manual" fine tuning of the receiver in the "up" or "down" directions depending upon which of the two keys U or D has been operated. The keyboard 25 includes one additional switch 58 which may be used to disable the automatic fine tuning (AFT) portion of the circuit by rendering the microcomputer insensitive to the signal output from the AFT circuit, in a manner described more fully subsequently.
As is apparent from the foregoing, the
microcomputer 23 provides the intelligence, decision making, and
control for the system operation. It is a complete self contained
computer. The decisions or signal inputs upon which the microcomputer
23 bases its operation include, in addition to the inputs from the
keyboard 25, inputs on sensory inputs into the B input port and into
the SNS1 and SNS0 inputs as shown in FIG. 2. These input signals are
used to provide an indication to the microcomputer 23 of the presence
or absence of a received signal; and if the presence of such a
signal is indicated, the inputs provide a further indication of the
accuracy of the tuning of the receiver to that signal. If the system
is being operated solely in a manual mode of operation (AFT switch 58
open), the microcomputer 23 disregards all of this sensory information
and tunes to the frequency allocation of the channel selected in the
manner described above. The system will stay tuned to this
condition, operating as a conventional frequency synthesizer, whether
or not a station is present in the received signal.
When the system is placed in its automatic mode of operation (similar to the mode of operation of the above mentioned '953 patent), the counter 82, integrally formed as part of the microcomputer 23, continuously adds or subtracts one number at a time from the nominal value or programmable division fraction entered into the programmable frequency divider 35 at the outset of each new channel number selection when frequency offset (mistuning) is present. The counter 82 is driven at a relatively high counting rate by clock pulses from the clock 81 during this initial or forced search mode of operation. Thus, automatic offset correction is provided for any channel which is off its assigned frequency. The offset correction automatically adjusts the frequency of the local oscillator by changing the division ratio of the signal from the reference oscillator 35 applied to the lower input of the phase comparator 37. By doing this, the output of the phase comparator 37 applied to the local oscillator 11 varies to cause the oscillator to be tuned in the proper direction to compensate for the transmitting station mistuning.
When the system is operating in its automatic mode of operation, the microcomputer 23 responds to the sensor information applied to it on its B input ports and on the S1 input port shown in FIG. 2. These inputs are obtained from the various outputs of the operational amplifiers shown connected to the corresponding input ports in the detailed circuit of FIG. 3. Depending upon whether the receiver is provided with a signal seek feature or not, one or more of the sensory inputs of the circuit of FIG. 3 are used. The s
ystem
shown in the drawings has a capability of correcting for frequency
offsets larger than 1.5 MHz on channels 2 and 7 and approximately 2
MHz on channels 6 and 13. The remainder of the channels have a range
between these two values.
If the receiver is not tuned properly, the micromputer 23 executes the localized search of the tuning range mentioned above. Since there is a necessary settling down time for the tuning of a television receiver immediately following selection of a new channel, a time interval of 250 milliseconds has been selected to prevent any localized search or offset frequency correction until the expiration of this "settling down" time period. If, at the end of this 250 millisecond time interval, a properly tuned station is present, this is indicated by the sensory outputs from the television receiver and no localized search is effected to change the division ratio or programmable divider count in the reference counter 35 for a system that also has signal seek.
A system with no signal seek capability is described later that requires less sensory input but which uses a time period where a forced search is required directly after the settling time interval.
Upon termination of the 250 millisecond settling down period, the microcomputer 23 is rendered responsive to the sensory input signals on its sensory input signal ports. In the simplest form, only the output of the frequency discriminator 60 (FIG. 3) applied to three comparators 61, 62 and 63 is used to provide the necessary tuning information to the microcomputer 23. The outputs of these comparators are applied to the B12 and B11 inputs of the microcomputer.
The
comparator 61 simply is a conventional comparator for determining
whether or not the output of the frequency discriminator is positive
or negative, as indicated in the upper waveform of FIG. 5. The
comparators 62 and 63 are each adjusted with appropriate reference
input levels to provide a narrow window centered about the center
tuning frequency (fc) of the receiver. If the tuning of the receiver,
as indicated by the output of the frequency discriminator 60, is
outside this window on either side of the central axis shown in FIG.
5, one output condition is indicated on the input terminal B11 of the
microcomputer. Only when the tuning frequency is within the tuning
window, indicative of a properly tuned receiver, is the appropriate
input applied to the microcomputer input terminal B11. This input
overrides any other input that may be present on the input terminal
B12 and is indicative of a properly tuned receiver. The input from the
frequency discriminator 60, as applied to the microcomputer on its
input port B12, is used to determine the direction of operation of the
counter 82 of the microcomputer for the localized search count
signals applied to the latch circuits 49 and 50 to change the count of
the reference programmable divider counter 35 on a step-by-step
basis.
The lower graph of FIG. 5 plots the relative frequency of the local oscillator 11 to the received signal frequency with respect to time. The various arrows are used to indicate the manner of operation of the counter 82 in the microcomputer 23 in conjunction with the reference counter 35 for adjusting for any mistuning conditions which may exist after the initial station selection has been effected in the manner described above.
If the receiver is properly tuned, the outputs from the comparators 62 and 63 of FIG. 3 which are combined together and applied to the input port B11 of the microcomputer 23, provide an indication that the tuning is within the properly tuned center frequency window. As a consequence, no further operation of the microcomputer to change any of the outputs applied to the latch circuits 49 and 50 for the duration of this condition is effected. On the other hand, if the receiver is mistuned on either side of the proper tuning frequency, the various operating characteristics shown in FIG. 5 are effected.
Assume initially that the receiver is capable of making tuning adjustments over a range of fc plus Δf to fc minus Δf, as indicated in the top waveform of FIG. 5. Three specific examples of mistuning will then be considered. Initially, assume that the local oscillator is mistuned relative to the received signal to a frequency f1 as shown in the lower graph of FIG. 5. In this condition, the outout of the frequency discriminator 60 is positive since this signal frequency lies to the lefthand side of the center or properly tuned region of operation of the discriminator. Under this condition of the operation, the input signal applied to the sensor port B12 of the microcomputer 23 is such that the microcomputer counter 82 is caused to advance in a positive direction to change the programmable division ratio or count of the reference counter 35 in a manner to force the output of the phase comparator 37 to adjust the frequency of the local oscillator until the proper tuning indicated at point B in the lower graph of FIG. 5 is reached. The time interval for accomplishing this result is measured from the upper end of the arrow representative of the frequency f1 to the point B.
Now assume that the receiver mistuning is to a frequency f2 which as shown in FIG. 5 as located on the righthand-side of the center axis fc. In this condition, the discriminator output is negative. This is reflected in the output of the comparator 61 applied to the input port B12 of the microcomputer 23. The polarity of this signal is identified by the microcomputer 23 to cause the counter 82 in it to operate in the reverse direction. As this count is applied on a step-by-step basis through the latch circuits 49 and 50 to the reference counter 35, the division ratio or count of the reference counter (divider) 35 is changed. As a result, the reference oscillator signal applied to the phase comparator 37 causes the phase comparator 37 output to drive the local oscillator frequency in a direction opposite to that considered in the first example. This is shown by the vector interconnecting the top of the arrow representative of f2 to point A on the time/frequency graph of FIG. 5.
As discussed in the general discussion above, whenever the tuning frequency reaches the narrow window on either side of fc, the outputs of the comparators 62 and 63 provide the necessary indication on the sensory input port terminal B11 to cause termination of the operation of the counter 82 in the microcomputer 23. Then the reference counter 35 remains set to the count attained just prior to the appearance of this input signal on the input port B11 of the microcomputer 23.
A third mistuning condition can exist, and ordinarily this condition results in an ambiguity which cannot be corrected simply by responding to the signal polarity at the output of the frequency discriminator. This is indicated by the mistuned condition where the difference between the local oscillator frequency f3 and the transmitter frequency is such that the signal f3 lies in the range to the right of the negative portion of the discriminator output shown in the upper waveform of FIG. 5. In this condition, the associated sound causes the discriminator output to be positive; so that the television receiver normally would attempt to tune toward the next adjacent channel and away from the properly tuned center frequency of the channel which is desired. The output of the discriminator 60 in this situation is the same as it was in th
e
first example considered for frequency f1; so that the counter 82 of
the microprocessor 23 operates to change the count in the reference
counter 35 in a manner to cause the local oscillator frequency to go
higher toward a frequency f3 +Δf, as shown in FIG. 5.
A predetermined number of counts of the counter 82 in the microcomputer 23 are necessary for the microcomputer to count through the frequency range Δf, and this range is selected to be within the pull in or operating range of the system. Once this count has been attained, the microcomputer counter 82 immediately is reset back to a count which corresponds to a frequency 2 Δf lower than the frequency attained by the maximum count. This is indicated in FIG. 5 by the frequency f3-Δf. Because the microcomputer counter 82 is limited to counting a number of counts equal to Δf, this new frequency now is on the lefthand side of the center line fc, shown in both waveforms of FIG. 5. This places the local oscillator frequency at a point such that the frequency discriminator output is the positive output shown on the lefthand-side of the upper waveform of FIG. 5. Counting continues in the same direction as previously. This time, however, it is in a proper direction to bring about correct tuning; and when the center frequency is reached, the output of the comparators 62 and 63 cause the microcomputer 23 to stop its count. The proper tuning point attained is indicated at point C on the graph of the lower part of FIG. 5.
Because the counter 82 of the microcomputer is limited to a maximum count equivalent to Δf above its initial count and thereupon is reset to a new count equivalent to 2 Δf lower than the maximum count, it is not necessary to utilize any other sensory inputs in order to properly tune the receiver over a wide pull in range (as much as plus or minus 2 MHz). Only the output of the conventional frequency discriminator 60 is used to provide the necessary sensory inputs.
The
counter 82 of the microcomputer 23 is operated by the clock 81
during the foregoing sequence of operation, immediately following the
selection of a new channel by the operation of the keyboard 25, at a
fast or high speed operation. Typically, the counter steps are 10
milliseconds per step; so that there are no initial visual effects
which can be noticed by an observer of the television screen of the
receiver being tuned. The maximum forced search period is
approximately 900 milliseconds in duration. At the end of this time
interval, a timer in the microcomputer 23 causes a signal to be
applied through the outputs of the E output port to the decoder
circuit 52 indicative of the completion of this time interval. The
decoder 52 then applies a pulse on an output lead connected to the B13
input of the B input port of the microcomputer 23. This pulse is
sensed by the microcomputer 23 and is applied to the clock 81 to
change the clock rate to a much slower rate, approximately one-third
(1/3) or one-fourth (1/4) the rate used previously during the forced
search mode of operation. This then permits the system to accomodate
station drifts which normally occur at a very slow rate during the
transmission and reception of a television signal. As a consequence,
it is possible to use more filtering in the filter 39 on the tuning
line (FIG. 1) and employ a smaller frequency window for the channel
verification sensed by the circuitry shown in FIG. 3. The result is a
more precise tuning from the receiver than is otherwise possible if
only a high speed operation of the clock 81 is utilized.
When the channel once again is changed by operation of the keys in the keyboard 25 or operation of the channel selection circuitry from a remote control unit, this new channel input is sensed by the microcomputer 23 from the signals applied to the A input port and the clock 81 is reset to its fast time or the forced search mode of operation; and the process resumes.
Instead of employing an additional decoding function in the decoder 52, a separate decoder also could be connected to the outputs of the D output ports to feed back the signal to the B13 input terminal of the B input port of the microcomputer 23. The operation of the system to change the rate or frequency of the pulses applied by the clock 81 to the counter 82 otherwise is the same as described above.
Although applicant has found that it is preferable to correct for mistuning or frequency offsets by adjusting the count or division ratio of the counter 35, such offset adjustments also could be effected by adjusting the count in the counter 31 in the local oscillator signal line. The operation in such a case is the same as described above for adjusting the count in the counter 35.
If the receiver is to be used with an automatic signal seek mode of operation, however, additional sensory inputs are necessary. These inputs operate in conjunction with the output of the frequency discriminator 60. The operation of the microcomputer 23 in controlling the count of the reference programmable frequency counter divider 35 is the same as described
above. The additional sensory inputs simply are used in conjunction
with the outputs of the comparators 62 and 63 to signal the
microcomputer 23 to assure that tuning is to a picture channel rather
than an adjacent sound channel. This is accomplished by utilizing
the output of the synchronizing signal separator 65 which is applied
to a comparator 67 to produce an output signal to the SNS1 sensory
input of the microcomputer 23 only when vertical synchronizing signal
components are present.
In addition, the output of a picture carrier detector 69 is applied to the input of a comparator 70 to produce an output to the B10 sensory input of the microcomputer 23. If the picture carrier detector 69 is producing an output indicative of the presence of a carrier, but no output is being obtained from the vertical synch separator 65 at the same time, the system is mistuned to a sound carrier and the microcomputer 23 is permitted to continue its localized search until a properly tuned station is found. Only when there is coincidence of signals from the picture carrier detector 69, the synch signal separator 65, and the automatic frequency discriminator window as determined by the comparators 62 and 63, is the microcomputer operation terminated to indicate that a properly tuned channel is present.
Further insurance of tuning the receiver only to a strong signal also can be provided by the addition of an AGC amplifier 72. This is connected to a comparator 74 coupled to the B10 input port along with the output of the picture carrier detector comparator 70. When the AGC amplifier 72 is used as a sensory input, the microcomputer operation, when the system is used in a signal seek mode, is only terminated to indicate reception of a valid signal when that signal is strong enough to produce the desired output from the comparator 74. The signal level which is acceptable is set by a potentiometer 75.
It should be noted that when the system is operated in a signal seek mode, the sensory inputs must indicate the reception of a properly tuned signal within a pre-established time period. If no signal is sensed by the various sensory input circuits operating in conjunction with one another as described above, the microcomputer 23 automatically steps to the next channel number and repeats the sequence of operation described above. This is when it is placed in its signal seek mode of operation. If signal seek is not employed, the additional sensory circuits 65, 69 and 72 are not necessary, and the inputs to the microcomputer which are provided from these sensory circuits are not utilized. The sensory signal input which is used both for a receiver without a signal seek capability of operation and for a receiver which has a signal seek mode of operation in it, is the output of the frequency discriminator 60 operating in conjunction with the comparators 61, 62 and 63 as described above.
As indicated above, the wideband method of tuning precisely to an incoming signal that is at the wrong frequency described here only needs the frequency discriminator sensory information. The method that uses the additional sensors described above is needed to make this system operate compatibly with signal seek but it is not restricted to seek operation.
For
a system that does not use signal seek operation, only the frequency
discriminator sensory input is required for proper operation. The
discriminator 60 is used for both fine tuning direction information and
to produce a frequency window to indicate the presence of a
correctly tuned station (channel verification). Initially, after a
channel change, there is a 250 millisecond settling time, the same as
the operation described above with compatible seek. After that,
however, comes a period of time where a forced localized search is
produced by the microcomputer 23. The forced search is needed to
insure that the system will correctly tune to stations that initially
may be tuned to the undesired zero voltage crossover in the right
half of the upper curve of FIG. 5. Such signals may be within the
frequency window of the discriminator 60; and if a search is not
forced, this system will not correctly tune. The compatible seek
system described previously correctly tunes the local oscillator
without a forced search, because the picture carrier detector and
vertical detector do not give an output for this situation and the
system automatically goes into its search mode of operation. However,
the non-seek system does not have a picture carrier sensor input and
must be forced to search for an initial period of time sufficient to
allow the system to tune up to its maximum frequency and then reset
(loop) back to a frequency of 2 Δf lower. Then it is tuned to the
positive left half portion of the discriminator curve (FIG. 5) and
the frequency window created by the discriminator 60 is sufficient to
insure proper tuning. If the discriminator output produced by the
desired incoming signal created an initial situation that produces
the correct tuning direction information, i.e., in the left half of
the curve of FIG. 5, or in the right half portion that gives the
correct direction and
frequency
window information, the forced search would not be needed. However,
the forced search will produce a correct tuning situation anyway. In
these cases, the tuning either is correct to begin with or correct
tuning is reached quickly. Then, even though the forced search is
active, it simply alternates up and down through the correct tuning
point because each time the receiver is tuned a little high in
frequency, it produces a negative output from the discriminator 60;
and the tuning direction signal causes the system to tune down in
frequency.
Then,
a positive discriminator output is produced, and the system tunes up
in frequency. This continues until the forced search is removed by
time-out of the microcomputer 23 (a fraction of a second). At such
time, the receiver is correctly tuned by the frequency window of the
discriminator to be very near fc. The system cannot tune to the
undesired discriminator crossover shown in the right half portion of
FIG. 5 because the polarity of the tuning direction signal always
causes it to tune away from that point.
The fast time or forced search operation of the system can be terminated in a different way other than the preestablished time-out period described above in conjunction with the operation of the circuit shown in FIG. 2. Generally, it is desirable to build into the system (or program into the system by means of software) such a maximum time-out period to effect the operation which has been described above to terminate the search and cause the clock 81 thereafter to operate in a low speed mode of operation. Termination also can be accomplished by sensing the number of changes in the direction sensor input applied to the B12 terminal of the B input port to cause the search to be terminated when this direction changes three times (or more). By doing this, any flicker that might be observed on the screen of the television receiver is minimized, since the forced search still takes place at the high rate of application of clock pulses from the clock 81 to the counter 82 in the same manner described above.
Termination of the search, however, also may be effected by means of a search terminate counter 78 (FIG. 3), which is advanced by pulses applied to it each time the output of the comparator 61 changes its sign (indicative of a change in direction for the counter 82) as applied to it through the B12 input port, as described earlier. After three of these changes, or some other number if desired, an output pulse is obtained from the search terminate counter 78 and is applied to the SNS0 input of the microcomputer 23. This causes the operation of the clock 81 to be switched to its low speed mode of operation to terminate the fast or "forced search" mode of operation. The next time a new channel number is entered on the keyboard 25, a reset pulse is applied to the search terminate counter 78 to reset it to its original or zero count, thereby readying it for another sequence of operation. It is apparent that the search terminate counter 78 may not always be operated to terminate the count, since the time-out interval which is sensed by the decode circuit 52 and applied to the B13 input port of the microcomputer 23 may occur before there are three changes of direction of the search. In any event, the next time a new channel number is entered into the keyboard 25, the search terminate counter 78 is reset; so that it is irrelevant whether this counter reaches a full count or not to effect the termination of the forced search operation of the system.
FIG. 4 shows the control sequence of the system which is stored in the ROM (Read Only Memory) of the microcomputer 23. The microcomputer 23 operates by always running through the flow sequence, via loops L1, L2 and L3. Loop L1 corresponds to a new channel selection by two digit number entry. Loop L2 corresponds to channel number increment or decrement by an up or down key operation, respectively, or by seek operation. Loop L3 corresponds to fine tuning, either manual or automatic. To obtain exact timing for system control, the microcomputer 23 receives a standard timing pulse from the output of the reference counter 35 divided in a divide-by-five counter 80 and applied to the A13 input port of the microcomputer 23. The control functions which are programmed into the microcomputer 23, as indicated in the flow chart of FIG. 4, are outlined in the following paragraphs.
Channel Number Correction: An invalid two digit channel number entry (0, 1, 84, 99) is corrected. When the operation of the receiver is in the signal seek mode, the next channel up from 83 is channel 2, and the next lower channel from channel 2 is 83.
PLL Control I: For a
given channel number, a corresponding binary code for the PLL selector
counter 31 is derived as described previously. For UHF channels, the
local oscillator frequency separation between two adjacent channels
is 6 MHz and the code for PLL is generated by the microcomputer 23
through means of a simple calculation. This code then is transferred
from the microcomputer 23 to the latches 44, 45 and 46 as described
previously.
PLL Control II: This routine of the microcomputer 23 is used to transfer the fine tuning data to the latches 49 and 50 which control the count of the reference counter 35 in the PLL circuit.
Channel Number Display: The channel number is transferred from the microcomputer 23 to the driver latches of the display driver circuit 29.
Key Input Detection: The keyboard is arranged as the matrix circuit shown in FIG. 2. ROM programming for scanning and acknowledging a keyboard entry only after successive indications provides protection against false entry due to contact bounce. The four data output lines of the D output port of the microcomputer 23 are used to transfer data to the phase lock loop section of the circuit and to the display circuit 29, as well as for scanning the keyboard matrix circuit.
Time Count: The microcomputer 23 receives a basic timing pulse of approximately 200 Hz from the output of the divider 80 and performs various controls for each timing pulse. By way of example, sensing for the vertical synch input (when the system is used with a signal seek capability) on the input port SNS1 takes place every 2.5 milliseconds. Automatic seek timing is selected to be 133 milliseconds for UHF channels. All of these timing pulses are derived from the basic synchronization timing pulse applied to the microcomputer on the A13 input port from the output of the divider 80. Various other timing values used in the microcomputer to properly time multiplex sequence the operation are derived from this basic timing pulse.
Sensor Input Detection: As described previously, the output of the comparators shown in FIG. 3 reflect the status of the tuning of the television receiver. If no signal seek mode of operation is used, only the frequency discriminator or AFT discriminator 60 is necessary. When a system is being used in a signal seek mode, a proper television signal receipt is indicated by the presence of a vertical synch signal at the output of the synch signal separator 65 and corresponding outputs are applied to the input leads B10 and B11 (high level input signals) indicative of tuning to the "correct tuned" frequency discriminator window and reception of a picture carrier. As stated previously, the signal present on the B12 input lead is used to determine the direction of tuning when the receiver is operated in its automatic mode.
Mode Detection: The status of the seek and automatic/manual (A/M) switches are detected. If the A/M switch (not shown) is in its automatic position, automatic seek and offset correction are active. If only the seek switch is on, only seek is performed. If the A/M switch is in manual, manual fine tuning (MFT) is active.
Automatic Mode: If the TV receiver is not properly tuned for VHF channels in automatic, the local oscillator frequency is shifted automatically toward proper tuning. The fine tuning data is generated in the microcomputer 23 and is transferred to the latches 49 and 50 for the reference counter 35 in the PLL circuit.
Manual Fine Tuning (MFT) Control: The local oscillator frequency is shifted by pushing the fine tuning up (U) or down (D) pushbutton or switch. This MFT control can be applied to VHF channels as well as to UHF channels.
Channel Up/Down: When a channel up (upward pointing arrow) or down (downward pointing arrow) key closure in the keyboard 25 is detected, or upon a direct access to an unused channel, this routine is activated and the system will advance to the next channel in the selected direction.
The
foregoing embodiment of the invention which has been described above
and which is illustrated in the drawings is to be considered
illustrative of the invention, which is not limited to the specific
embodiment selected for this purpose. For example, hard-wired logic
could be used to achieve the various circuit operations which are
accomplished by the microcomputer 23 in conjunction with the other
portions of the system. The relative ease of programming and debugging
the microcomputer 23, however, make it much simpler to implement the
system operation with the microcomputer than with hard-wired logic.
With respect to the sensor circuit inputs to the system, an added
degree of operating assurance can be provided by the addition of a
sound carrier sensor in addition to the picture carrier sensor shown
in FIG. 3. If this feature is desired, the output of the comparator
for the sound carrier is combined with the outputs of the comparators
70 and 74 at the input terminal B10 of the B input port of the
microcomputer 23. Because of the manner of the circut operation which
has been described previously, however, the addition of a sound
carrier detector to the system is not considered necessary, even for a
system operating in the signal seek mode of operation. This is in
contrast to conventional television receivers having a signal seek
operation, in which detection of the sound carrier generally is a
necessity to insure that mistuning of the receiver to an adjacent
sound carrier does not take place.
Both
of the above mentioned patents are directed to frequency synthesizer
tuning systems for use with television receivers to enable operation
of the receivers with minimal viewer fine tuning adjustments. By the
utilization of the frequency synthesizer tuning systems of these
patents, the fine tuning adjustment which is necessary with conventional
types of television receiver tuning systems has been substantially
eliminated. The system employed in the '953 patent permits utilization
of a frequency synthesizer tuning system which correctly tunes to a
desired television station or channel even if the transmitted signals
from that station are not precisely maintained at the proper
frequencies. The '535 patent is directed to a signal seek tuning system
adaptation of the frequency synthesizer tuning system of the '953
patent which still permits implementation of all of the desired
wide-band pull in range of the frequency synthesizer system of the '953
patent.The systems of the foregoing patents operate effectively to correct automatically for frequency offsets in a frequency synthesizer tuning system without affecting the operation of the conventional frequency synthesizer used in the system. The systems of these patents are in widespread use commercially and permit direct selection, with automatic fine tuning adjustment, of any desired VHF channel which the viewer wishes to observe. In addition, the signal seek adaptation disclosed in the '535 patent couples all of the advantages of the frequency synthesizer tuning system of the '953 patent with the desirability of providing bidirectional signal seek operation.
While the systems disclosed in the foregoing patents operate in a highly satisfactory manner to accomplish the desired results of accurate tuning without the necessity of fine tuning adjustments, the circuitry for accomplishing the desired results is somewhat complex. It is desirable to reduce the circuit complexity and the number of signal detectors for accomplishing these results without compromising the accuracy of operation of the system.
SUMMARY OF THE INVENTION
Accordingly, it is an object of this invention to provide an improved tuning system for a television receiver.
It is an additional object of this invention to provide an improved frequency synthesizer tuning system for a television receiver.
It is another object of this invention to provide an improved frequency synthesizer tuning system for a television receiver which includes a provision for adjusting the synthesizer loop for frequency offsets in the received signal with a minimum number of signal detectors.
It is still another object of this invention to provide an improved frequency synthesizer tuning system for a television receiver which operates to adjust the synthesizer loop for frequency offsets in the received signal over a relatively wide pull in range in response to the output of the receiver frequency discriminator by changing the division ratio of a programmable frequency divider in the reference oscillator leg or local oscillator leg of the synthesizer loop at a first relatively high rate from an initial nominal value to a pre-established maximum in one direction, and then resetting the division ratio to a second nominal value once the maximum is reached and continuing to incrementally change the division ratio in the same direction from the second nominal value until a properly tuned condition is indicated by the output of the receiver AFT frequency discriminator, followed by control at a lower rate of operation to maintain tuning during transmitting station drifts.
In
accordance with a preferred embodiment of this invention, the
frequency synthesizer tuning system for a television receiver includes
a stable reference oscillator and a voltage controlled local
oscillator in the tuner. A programmable frequency divider is connected
between the output of the reference oscillator and one input to a
phase comparator, the other input of which is supplied by the output
of the local oscillator. The output of the phase comparator then
comprises a control signal which is supplied to the local oscillator
to control the frequency of its operation.A counter circuit is connected to the programmable frequency divider for initially setting the divider to a predetermined division ratio upon selection of a desired channel by the viewer. The counter then operates to change the programmable fraction of the division ratio at a first relatively high rate in a direction controlled by the output from the receiver picture carrier discriminator in the absence of a predetermined signal output derived from the discriminator. A control means causes the counter circuit to count in this direction until it is determined that a station is tuned or a predetermined maximum count is attained if no station is correctly tuned, thereupon resetting the counter circuit to a count which is a predetermined amount less than the maximum predetermined count. Counting is continued in the same predetermined direction from the new lesser count to continuously change the programmable fraction of the frequency divider in accordance with the state of operation of the counter.
matically adjusts
for frequency offsets of the received signal which otherwise would
cause the station to be mistuned if a conventional frequency
synthesizer tuning system were used. After termination of the high
rate operation of the counter, it is switched to a lower rate
operation for maintaining tuning during transmitting station drifts.BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a block diagram of a television receiver employing a preferred embodiment of the invention;
FIG. 2 is a detailed block diagram of a portion of the circuit of the preferred embodiment shown in FIG. 1;
FIG. 3 is a detailed circuit diagram of a portion of a circuit shown in FIG. 1;
FIG. 4 is a flow chart of the control sequence of operation of the circuit shown in FIG. 1 and 2; and
FIG. 5 shows a waveform and time/frequency chart, respectively, useful in explaining the operation of the circuit shown in FIGS. 1, 2 and 3.
DETAILED DESCRIPTION
Referring now to the drawings, the same reference numbers are used throughout the several figures to designate the same or similar components.
FIG. 1 is a block diagram of a television receiver, which may be a black and white or color television receiver. Mo
st
of the circuitry of this receiver is conventional, and for that
reason it has not been shown in FIG. 1. Added to the conventional
television receiver circuitry of FIG. 1, however, is a frequency
synthesizer tuning system, in accordance with a preferred embodiment
of the invention, which is capable of automatically changing the
reference frequency when a frequency offset exists in the received
signal for a particular channel.Transmitted composite television signals, either received over the air or distributed by means of a master antenna TV distribution system, are received by an antenna 10 or on antenna input terminals to the receiver. As is well known, these composite signals include picture and sound carrier components and synchronizing signal components, with the composite signal applied to an RF and tuner stage 11 of the receiver. The stage 11 includes the conventional RF amplifiers and tuner sections of the receiver, including a VHF oscillator section and a UHF oscillator section. Preferably, the UHF and VHF oscillators are voltage controlled oscillators, the freuency of operation of which are varied in response to a tuning voltage applied to them to effect the desired tuning of the receiver.
The output of the RF and tuner stages 11 is applied to the remainder of the television receiver 14, which includes the IF amplifier stages for supplying conventional picture (video) and sound IF signals to the video and sound processing stages of the receiver 14. The circuitry of the receiver 14 may be of any conventional type used to separate, amplify and otherwise process the signals for application to a cathode ray tube 16 and to a loudspeaker 17 which reproduce the picture and sound components, respectively, of the received signal.
The
receiver 14 also includes a conventional AFT or automatic fine
tuning discriminator circuit and additionally may include a synch
separator circuit for producing an output in response to the presence
of vertical synchronizatin pulses, a picture carrier detection
circuit, and an automatic gain control (AGC) amplifier. Outputs
representative of these sensor components are shown as being coupled
over a group of lead 20 to sensory circuitry 22, which in turn
couples outputs representative of the operation of these various
sensor circuits to a microprocessor unit 23 for controlling the
operation of the microprocessor unit.The microprocessor unit 23 is utilized in the system of FIG. 1 for controlling the operation of a frequency synthesizer tuning system capable of automatic offset correction. When the viewer desires to select a new channel, he enters the desired channel number into a channel selection keyboard 25. There are a number of different keyboards which may be employed to accomplish this function, and the particular design is not important to this invention. The channel selector keyboard 25 also may include switches or keys for initiating a signal seek function in either the "up" or "down" direction.
Information represented by the selection of channel numbers on the keyboard 25 is supplied to the microprocessor unit 23 which provides output signals over a corresponding set of leads 27 to the tuners (local oscillators) 11 to effect the appropriate band switching control for the tuners 11 in accordance with the particular channel which has been selected. In addition, the keyboard 25, operating through the microprocessor unit 23, provides output signals which operate a channel number display 29 to provide an appropriate display of the selected channel number to the viewer.
The microprocessor M3870 unit 23 also processes the signals which are used to operate the channel number display 29 through a multiplexing circuit operation to decode the selected channel number into a parallel encoded signal. This signal is applied to corresponding inputs of the count-down counter or programmable frequency divider 31 to cause the division number of the divider 31 to relate to the divided down frequency of the tuner local oscillators connected to the input of the divider 31 through a prescaler divider circuit 32 to the frequency of the reference oscillator 34. Thus, the division number or division ratio of the local oscillator frequency obtained from the output of the programmable divider 31 is appropriately related to the frequency of the reference crystal oscillator 34.
In accordance with the time division multiplex operation of the microprocessor 23, the count of the programmable frequency divider 35 initially is adjusted to a fixed count by the application of appropriate output signals from the microprocessor unit 23 to a point selected to be at or near the mid-point of the operating range of the programmable frequency divider 35. Thus, the output of the divider 35 is a stable reference frequency (because the input is from the reference crystal oscillator 34) which is used to establish initially and to maintain tuning of the receiver to the selected channel.
The output of the programmable divider 35 is applied to one of two inputs of a phase comparator circuit 37. The other input to the phase comparator circuit 37 is supplied from the selected one of the VHF or UHF oscillators in the tuner stages 11 through the programmable frequency divider 31. The phase comparator circuit 37 operates in a conventional manner to supply a DC tuning control signal through a phase locked loop filter circuit 39 and over a lead 40 to the oscillators in the tuner system 11 to change and maintain their operating frequency.
With the exception of the use of the microprocessor unit 23, the operation of the system which has been described thus far is that of a relatively conventional frequency synthesizer system incorporated into a television receiver. This system is similar to the system of the '953 patent. As in the system of that patent, the system shown in FIG. 1, when the transmitted station or station received on a master antenna distribution system provides the station or channel signals at the proper frequency, operates as a relatively conventional frequency synthesizer system. If, however, there is a frequency offset in the received signal to cause the carrier of the received signal to be displaced from the frequency which it should have to some other frequency, it is possible that the system would give the appearance of mistuning to the received station. The microprocessor 23, operating in conjunction with the sensory circuitry 22, is employed in conjunction with the countdown or programmable frequency divider circuit 35 to eliminate this disadvantage and still retain the advantages of frequency synthesizer tuning.
Reference now should be made to FIG. 2 which shows details of t
he
interface between the keyboard 25, the microprocessor unit 23, and
the circuitry used in the frequency synthesizer portions of the
system. A commercially available microprocessor which has been used
for the microprocessor 23, and which forms the basis for the
diagramatic representation of the microprocessor in FIG. 2, is the
Matsushita Electronics Corporation MN1402 four-bit single-chip
microcomputer. This microcomputer has two, four-bit parallel input
ports labeled "A" and "B". In addition, three output ports, a five-bit
output port "C" and two four-bit output ports "D" and "E" are
provided. The internal configuration of the microcomputer 23 includes
an arithmetic logic unit (ALU), a read only memory (ROM) for storing
instructions and constants, and a random access memory (RAM) used for
data memory, arranged into four files, each file containing 16
four-bit words. These words are selected by X and Y registers and this
memory is used, for example, for timers, counters, etc., and also is
used to hold intermediate results. To facilitate an understanding of
the operation of the system, a portion of this memory is shown in
FIG. 2 as a clock 81 and a reversible counter 82 connected between
the "B" input port and the "D" output port. The microcomputer 23 is
programmed to permit it to operate in conjunction with the remainder
of the circuits shown in FIG. 2. The programming techniques are
standard, and the microcomputer 23 itself is a standard commercially
available circuit component.There are several system parameters that must be selected in the operation of the system shown in FIG. 2. The selection of the nominal frequency of the two signals that feed the phase comparator circuit 37 is an example. Channel selection is provided by changing the frequency division ratio of the selector counter 31 which divides the local oscillator signal after this signal is passed through a prescaler circuit 32 and a divide-by-two divider circuit 41. The nominal frequency from the programmable frequency divider 31 (selector counter) is selected so that the local oscillator (tuner) 11 can be set exactly on frequency for all channels.
Since
the frequency divider 31 is able to divide only by integer numbers,
one distinct frequency possibility in the range of one KHz is
obtained, another in the range of two KHz, etc. A choice must be made
as to which of these values is optimum. Each value yields the
nominal frequency of all of the 82 channels by simply multiplying by
an appropriate integer for each channel. To simplify the phase locked
loop filtering problem by the filter 39, it is desirable that the
frequencies of the signals supplied to the phase comparator 37 are as
high as possible. This permits rapid acquisition of a new channel
along with a very clean DC control signal to adjust the local
oscillator. A trade-off for this, however, must be made to permit
fine tunning adjustment of the local oscillator automatically to
correctly tune in stations which are off their assigned frequency, or
to manually provide this feature, if desired. The two-speed
operation of the system in accordance with the present invention
allows a better trade-off to be made by allowing rapid acquisition
and then a slower speed for precise tuning.A compromise solution which is utilized in the circuit of FIG. 2 is to cause the frequency division chain from the local oscillator 11 in the tuner to the phase comparator 37 to be composed of the fixed divide-by-256 prescaler 32, and a fixed divide-by-4 division, which is accomplished by the divider 41 at the input of the counter 31 and a second divider 42 at the output of the counter 31. The variable frequency divider counter 31 then is loaded by means of three latch circuits 44, 45 and 46 at an appropriate time by the time division multiplex operation of the microcomputer 23 and a number that programs the programmable frequency divider counter 31 to divide by the numerical value of the frequency of the local oscillator in MHz for the channel selected. For example, if the receiver is to be tuned to channel 2, which has a nominal local oscillator frequency of 101 MHz, the programmable frequency divider 31 is set to divide by 101. If the receiver is to be tuned to channel 83, which has a nominal local oscillator frequency of 931 MHz, the programmable frequency divider 31 is set to divide by 931. In both cases, the variable divider 31 produces a 1 MHz signal. However, because of the fixed divide-by-256 and the two fixed divide-by-two dividers in series with the programmable divider 31, an output frequency of 976.5625 Hz is supplied from the output of the divider 42 to the upper input of the phase comparator 37.
Although the D output ports of the microcomputer 23 are connected in common to all of these various portions of the circuit, the selection of which of the latches are enabled to respond to the particular output signals appearing on the D output ports at any given time is effected through the C and E output ports of the microcomputer 23 in a time division multiplex fashion. A decoder circuit 52, connected to the lowermost three outputs of the E output port of the microcomputer 23, is used to apply unique decoding signals at different times in the time division multiplex sequence of operation of the microcomputer 23 to the five latch circuits 44, 45, 46, 49 and 50, respectively. At any given time in the sequence, only one of these latch circuits is enabled for operation. A latch load signal is applied from the upper output (EO3) at each cycle of operation of the signals appearing on the E output port to set the latch circuit which is enabled by the output of the decoding circuit 52 with the data appearing on the other inputs to the latch circuit. This data simultaneously appears on the four outputs of the D output port of the microcomputer 23.
Thus, in rapid sequence, the latch circuits 44, 45 and 46 are set to store the division number corresponding to the selected channel entered onto the keyboard 25, and the latch circuits 49 and 50 are each operated to set the programmable divider reference counter 35 to a center or nominal count, which is always the same upon the selection of a new channel on the keyboard 25. Similarly, the two right-hand outputs of the C output port (CO6 and CO5) enter the two digits of the selected channel number in the drivers of the display circuit 29 at the proper time in the binary encoded sequence when these digits appear on the four-bit binary encoded representation of the D output port. This results in a visual display of the channel number selected.
In addition to the selection of a channel number directly by the keyboard 25, the keyboard also may include an additional switch 56, which is scanned in the time division multiplex sequence to determine if the receiver is placed in a "seek" mode of operation (when the signal seek capability is incorporated into such a receiver). Operating in conjunction with the signal seek switch 56 are a pair of "up" and "down" seek direction input switches shown with a graphic representation of the seek directions on the keyboard 25. A further provision is provided by two keys labeled "U" and "D", which are used for "manual" fine tuning of the receiver in the "up" or "down" directions depending upon which of the two keys U or D has been operated. The keyboard 25 includes one additional switch 58 which may be used to disable the automatic fine tuning (AFT) portion of the circuit by rendering the microcomputer insensitive to the signal output from the AFT circuit, in a manner described more fully subsequently.
As is apparent from the foregoing, the
microcomputer 23 provides the intelligence, decision making, and
control for the system operation. It is a complete self contained
computer. The decisions or signal inputs upon which the microcomputer
23 bases its operation include, in addition to the inputs from the
keyboard 25, inputs on sensory inputs into the B input port and into
the SNS1 and SNS0 inputs as shown in FIG. 2. These input signals are
used to provide an indication to the microcomputer 23 of the presence
or absence of a received signal; and if the presence of such a
signal is indicated, the inputs provide a further indication of the
accuracy of the tuning of the receiver to that signal. If the system
is being operated solely in a manual mode of operation (AFT switch 58
open), the microcomputer 23 disregards all of this sensory information
and tunes to the frequency allocation of the channel selected in the
manner described above. The system will stay tuned to this
condition, operating as a conventional frequency synthesizer, whether
or not a station is present in the received signal.When the system is placed in its automatic mode of operation (similar to the mode of operation of the above mentioned '953 patent), the counter 82, integrally formed as part of the microcomputer 23, continuously adds or subtracts one number at a time from the nominal value or programmable division fraction entered into the programmable frequency divider 35 at the outset of each new channel number selection when frequency offset (mistuning) is present. The counter 82 is driven at a relatively high counting rate by clock pulses from the clock 81 during this initial or forced search mode of operation. Thus, automatic offset correction is provided for any channel which is off its assigned frequency. The offset correction automatically adjusts the frequency of the local oscillator by changing the division ratio of the signal from the reference oscillator 35 applied to the lower input of the phase comparator 37. By doing this, the output of the phase comparator 37 applied to the local oscillator 11 varies to cause the oscillator to be tuned in the proper direction to compensate for the transmitting station mistuning.
When the system is operating in its automatic mode of operation, the microcomputer 23 responds to the sensor information applied to it on its B input ports and on the S1 input port shown in FIG. 2. These inputs are obtained from the various outputs of the operational amplifiers shown connected to the corresponding input ports in the detailed circuit of FIG. 3. Depending upon whether the receiver is provided with a signal seek feature or not, one or more of the sensory inputs of the circuit of FIG. 3 are used. The s
ystem
shown in the drawings has a capability of correcting for frequency
offsets larger than 1.5 MHz on channels 2 and 7 and approximately 2
MHz on channels 6 and 13. The remainder of the channels have a range
between these two values.If the receiver is not tuned properly, the micromputer 23 executes the localized search of the tuning range mentioned above. Since there is a necessary settling down time for the tuning of a television receiver immediately following selection of a new channel, a time interval of 250 milliseconds has been selected to prevent any localized search or offset frequency correction until the expiration of this "settling down" time period. If, at the end of this 250 millisecond time interval, a properly tuned station is present, this is indicated by the sensory outputs from the television receiver and no localized search is effected to change the division ratio or programmable divider count in the reference counter 35 for a system that also has signal seek.
A system with no signal seek capability is described later that requires less sensory input but which uses a time period where a forced search is required directly after the settling time interval.
Upon termination of the 250 millisecond settling down period, the microcomputer 23 is rendered responsive to the sensory input signals on its sensory input signal ports. In the simplest form, only the output of the frequency discriminator 60 (FIG. 3) applied to three comparators 61, 62 and 63 is used to provide the necessary tuning information to the microcomputer 23. The outputs of these comparators are applied to the B12 and B11 inputs of the microcomputer.
The
comparator 61 simply is a conventional comparator for determining
whether or not the output of the frequency discriminator is positive
or negative, as indicated in the upper waveform of FIG. 5. The
comparators 62 and 63 are each adjusted with appropriate reference
input levels to provide a narrow window centered about the center
tuning frequency (fc) of the receiver. If the tuning of the receiver,
as indicated by the output of the frequency discriminator 60, is
outside this window on either side of the central axis shown in FIG.
5, one output condition is indicated on the input terminal B11 of the
microcomputer. Only when the tuning frequency is within the tuning
window, indicative of a properly tuned receiver, is the appropriate
input applied to the microcomputer input terminal B11. This input
overrides any other input that may be present on the input terminal
B12 and is indicative of a properly tuned receiver. The input from the
frequency discriminator 60, as applied to the microcomputer on its
input port B12, is used to determine the direction of operation of the
counter 82 of the microcomputer for the localized search count
signals applied to the latch circuits 49 and 50 to change the count of
the reference programmable divider counter 35 on a step-by-step
basis.The lower graph of FIG. 5 plots the relative frequency of the local oscillator 11 to the received signal frequency with respect to time. The various arrows are used to indicate the manner of operation of the counter 82 in the microcomputer 23 in conjunction with the reference counter 35 for adjusting for any mistuning conditions which may exist after the initial station selection has been effected in the manner described above.
If the receiver is properly tuned, the outputs from the comparators 62 and 63 of FIG. 3 which are combined together and applied to the input port B11 of the microcomputer 23, provide an indication that the tuning is within the properly tuned center frequency window. As a consequence, no further operation of the microcomputer to change any of the outputs applied to the latch circuits 49 and 50 for the duration of this condition is effected. On the other hand, if the receiver is mistuned on either side of the proper tuning frequency, the various operating characteristics shown in FIG. 5 are effected.
Assume initially that the receiver is capable of making tuning adjustments over a range of fc plus Δf to fc minus Δf, as indicated in the top waveform of FIG. 5. Three specific examples of mistuning will then be considered. Initially, assume that the local oscillator is mistuned relative to the received signal to a frequency f1 as shown in the lower graph of FIG. 5. In this condition, the outout of the frequency discriminator 60 is positive since this signal frequency lies to the lefthand side of the center or properly tuned region of operation of the discriminator. Under this condition of the operation, the input signal applied to the sensor port B12 of the microcomputer 23 is such that the microcomputer counter 82 is caused to advance in a positive direction to change the programmable division ratio or count of the reference counter 35 in a manner to force the output of the phase comparator 37 to adjust the frequency of the local oscillator until the proper tuning indicated at point B in the lower graph of FIG. 5 is reached. The time interval for accomplishing this result is measured from the upper end of the arrow representative of the frequency f1 to the point B.
Now assume that the receiver mistuning is to a frequency f2 which as shown in FIG. 5 as located on the righthand-side of the center axis fc. In this condition, the discriminator output is negative. This is reflected in the output of the comparator 61 applied to the input port B12 of the microcomputer 23. The polarity of this signal is identified by the microcomputer 23 to cause the counter 82 in it to operate in the reverse direction. As this count is applied on a step-by-step basis through the latch circuits 49 and 50 to the reference counter 35, the division ratio or count of the reference counter (divider) 35 is changed. As a result, the reference oscillator signal applied to the phase comparator 37 causes the phase comparator 37 output to drive the local oscillator frequency in a direction opposite to that considered in the first example. This is shown by the vector interconnecting the top of the arrow representative of f2 to point A on the time/frequency graph of FIG. 5.
As discussed in the general discussion above, whenever the tuning frequency reaches the narrow window on either side of fc, the outputs of the comparators 62 and 63 provide the necessary indication on the sensory input port terminal B11 to cause termination of the operation of the counter 82 in the microcomputer 23. Then the reference counter 35 remains set to the count attained just prior to the appearance of this input signal on the input port B11 of the microcomputer 23.
A third mistuning condition can exist, and ordinarily this condition results in an ambiguity which cannot be corrected simply by responding to the signal polarity at the output of the frequency discriminator. This is indicated by the mistuned condition where the difference between the local oscillator frequency f3 and the transmitter frequency is such that the signal f3 lies in the range to the right of the negative portion of the discriminator output shown in the upper waveform of FIG. 5. In this condition, the associated sound causes the discriminator output to be positive; so that the television receiver normally would attempt to tune toward the next adjacent channel and away from the properly tuned center frequency of the channel which is desired. The output of the discriminator 60 in this situation is the same as it was in th
e
first example considered for frequency f1; so that the counter 82 of
the microprocessor 23 operates to change the count in the reference
counter 35 in a manner to cause the local oscillator frequency to go
higher toward a frequency f3 +Δf, as shown in FIG. 5.A predetermined number of counts of the counter 82 in the microcomputer 23 are necessary for the microcomputer to count through the frequency range Δf, and this range is selected to be within the pull in or operating range of the system. Once this count has been attained, the microcomputer counter 82 immediately is reset back to a count which corresponds to a frequency 2 Δf lower than the frequency attained by the maximum count. This is indicated in FIG. 5 by the frequency f3-Δf. Because the microcomputer counter 82 is limited to counting a number of counts equal to Δf, this new frequency now is on the lefthand side of the center line fc, shown in both waveforms of FIG. 5. This places the local oscillator frequency at a point such that the frequency discriminator output is the positive output shown on the lefthand-side of the upper waveform of FIG. 5. Counting continues in the same direction as previously. This time, however, it is in a proper direction to bring about correct tuning; and when the center frequency is reached, the output of the comparators 62 and 63 cause the microcomputer 23 to stop its count. The proper tuning point attained is indicated at point C on the graph of the lower part of FIG. 5.
Because the counter 82 of the microcomputer is limited to a maximum count equivalent to Δf above its initial count and thereupon is reset to a new count equivalent to 2 Δf lower than the maximum count, it is not necessary to utilize any other sensory inputs in order to properly tune the receiver over a wide pull in range (as much as plus or minus 2 MHz). Only the output of the conventional frequency discriminator 60 is used to provide the necessary sensory inputs.
The
counter 82 of the microcomputer 23 is operated by the clock 81
during the foregoing sequence of operation, immediately following the
selection of a new channel by the operation of the keyboard 25, at a
fast or high speed operation. Typically, the counter steps are 10
milliseconds per step; so that there are no initial visual effects
which can be noticed by an observer of the television screen of the
receiver being tuned. The maximum forced search period is
approximately 900 milliseconds in duration. At the end of this time
interval, a timer in the microcomputer 23 causes a signal to be
applied through the outputs of the E output port to the decoder
circuit 52 indicative of the completion of this time interval. The
decoder 52 then applies a pulse on an output lead connected to the B13
input of the B input port of the microcomputer 23. This pulse is
sensed by the microcomputer 23 and is applied to the clock 81 to
change the clock rate to a much slower rate, approximately one-third
(1/3) or one-fourth (1/4) the rate used previously during the forced
search mode of operation. This then permits the system to accomodate
station drifts which normally occur at a very slow rate during the
transmission and reception of a television signal. As a consequence,
it is possible to use more filtering in the filter 39 on the tuning
line (FIG. 1) and employ a smaller frequency window for the channel
verification sensed by the circuitry shown in FIG. 3. The result is a
more precise tuning from the receiver than is otherwise possible if
only a high speed operation of the clock 81 is utilized.When the channel once again is changed by operation of the keys in the keyboard 25 or operation of the channel selection circuitry from a remote control unit, this new channel input is sensed by the microcomputer 23 from the signals applied to the A input port and the clock 81 is reset to its fast time or the forced search mode of operation; and the process resumes.
Instead of employing an additional decoding function in the decoder 52, a separate decoder also could be connected to the outputs of the D output ports to feed back the signal to the B13 input terminal of the B input port of the microcomputer 23. The operation of the system to change the rate or frequency of the pulses applied by the clock 81 to the counter 82 otherwise is the same as described above.
Although applicant has found that it is preferable to correct for mistuning or frequency offsets by adjusting the count or division ratio of the counter 35, such offset adjustments also could be effected by adjusting the count in the counter 31 in the local oscillator signal line. The operation in such a case is the same as described above for adjusting the count in the counter 35.
If the receiver is to be used with an automatic signal seek mode of operation, however, additional sensory inputs are necessary. These inputs operate in conjunction with the output of the frequency discriminator 60. The operation of the microcomputer 23 in controlling the count of the reference programma
ble frequency counter divider 35 is the same as described
above. The additional sensory inputs simply are used in conjunction
with the outputs of the comparators 62 and 63 to signal the
microcomputer 23 to assure that tuning is to a picture channel rather
than an adjacent sound channel. This is accomplished by utilizing
the output of the synchronizing signal separator 65 which is applied
to a comparator 67 to produce an output signal to the SNS1 sensory
input of the microcomputer 23 only when vertical synchronizing signal
components are present.In addition, the output of a picture carrier detector 69 is applied to the input of a comparator 70 to produce an output to the B10 sensory input of the microcomputer 23. If the picture carrier detector 69 is producing an output indicative of the presence of a carrier, but no output is being obtained from the vertical synch separator 65 at the same time, the system is mistuned to a sound carrier and the microcomputer 23 is permitted to continue its localized search until a properly tuned station is found. Only when there is coincidence of signals from the picture carrier detector 69, the synch signal separator 65, and the automatic frequency discriminator window as determined by the comparators 62 and 63, is the microcomputer operation terminated to indicate that a properly tuned channel is present.
Further insurance of tuning the receiver only to a strong signal also can be provided by the addition of an AGC amplifier 72. This is connected to a comparator 74 coupled to the B10 input port along with the output of the picture carrier detector comparator 70. When the AGC amplifier 72 is used as a sensory input, the microcomputer operation, when the system is used in a signal seek mode, is only terminated to indicate reception of a valid signal when that signal is strong enough to produce the desired output from the comparator 74. The signal level which is acceptable is set by a potentiometer 75.
It should be noted that when the system is operated in a signal seek mode, the sensory inputs must indicate the reception of a properly tuned signal within a pre-established time period. If no signal is sensed by the various sensory input circuits operating in conjunction with one another as described above, the microcomputer 23 automatically steps to the next channel number and repeats the sequence of operation described above. This is when it is placed in its signal seek mode of operation. If signal seek is not employed, the additional sensory circuits 65, 69 and 72 are not necessary, and the inputs to the microcomputer which are provided from these sensory circuits are not utilized. The sensory signal input which is used both for a receiver without a signal seek capability of operation and for a receiver which has a signal seek mode of operation in it, is the output of the frequency discriminator 60 operating in conjunction with the comparators 61, 62 and 63 as described above.
As indicated above, the wideband method of tuning precisely to an incoming signal that is at the wrong frequency described here only needs the frequency discriminator sensory information. The method that uses the additional sensors described above is needed to make this system operate compatibly with signal seek but it is not restricted to seek operation.
Then,
a positive discriminator output is produced, and the system tunes up
in frequency. This continues until the forced search is removed by
time-out of the microcomputer 23 (a fraction of a second). At such
time, the receiver is correctly tuned by the frequency window of the
discriminator to be very near fc. The system cannot tune to the
undesired discriminator crossover shown in the right half portion of
FIG. 5 because the polarity of the tuning direction signal always
causes it to tune away from that point.The fast time or forced search operation of the system can be terminated in a different way other than the preestablished time-out period described above in conjunction with the operation of the circuit shown in FIG. 2. Generally, it is desirable to build into the system (or program into the system by means of software) such a maximum time-out period to effect the operation which has been described above to terminate the search and cause the clock 81 thereafter to operate in a low speed mode of operation. Termination also can be accomplished by sensing the number of changes in the direction sensor input applied to the B12 terminal of the B input port to cause the search to be terminated when this direction changes three times (or more). By doing this, any flicker that might be observed on the screen of the television receiver is minimized, since the forced search still takes place at the high rate of application of clock pulses from the clock 81 to the counter 82 in the same manner described above.
Termination of the search, however, also may be effected by means of a search terminate counter 78 (FIG. 3), which is advanced by pulses applied to it each time the output of the comparator 61 changes its sign (indicative of a change in direction for the counter 82) as applied to it through the B12 input port, as described earlier. After three of these changes, or some other number if desired, an output pulse is obtained from the search terminate counter 78 and is applied to the SNS0 input of the microcomputer 23. This causes the operation of the clock 81 to be switched to its low speed mode of operation to terminate the fast or "forced search" mode of operation. The next time a new channel number is entered on the keyboard 25, a reset pulse is applied to the search terminate counter 78 to reset it to its original or zero count, thereby readying it for another sequence of operation. It is apparent that the search terminate counter 78 may not always be operated to terminate the count, since the time-out interval which is sensed by the decode circuit 52 and applied to the B13 input port of the microcomputer 23 may occur before there are three changes of direction of the search. In any event, the next time a new channel number is entered into the keyboard 25, the search terminate counter 78 is reset; so that it is irrelevant whether this counter reaches a full count or not to effect the termination of the forced search operation of the system.
FIG. 4 shows the control sequence of the system which is stored in the ROM (Read Only Memory) of the microcomputer 23. The microcomputer 23 operates by always running through the flow sequence, via loops L1, L2 and L3. Loop L1 corresponds to a new channel selection by two digit number entry. Loop L2 corresponds to channel number increment or decrement by an up or down key operation, respectively, or by seek operation. Loop L3 corresponds to fine tuning, either manual or automatic. To obtain exact timing for system control, the microcomputer 23 receives a standard timing pulse from the output of the reference counter 35 divided in a divide-by-five counter 80 and applied to the A13 input port of the microcomputer 23. The control functions which are programmed into the microcomputer 23, as indicated in the flow chart of FIG. 4, are outlined in the following paragraphs.
Channel Number Correction: An invalid two digit channel number entry (0, 1, 84, 99) is corrected. When the operation of the receiver is in the signal seek mode, the next channel up from 83 is channel 2, and the next lower channel from channel 2 is 83.
PLL Control I: For a
given channel number, a corresponding binary code for the PLL selector
counter 31 is derived as described previously. For UHF channels, the
local oscillator frequency separation between two adjacent channels
is 6 MHz and the code for PLL is generated by the microcomputer 23
through means of a simple calculation. This code then is transferred
from the microcomputer 23 to the latches 44, 45 and 46 as described
previously.PLL Control II: This routine of the microcomputer 23 is used to transfer the fine tuning data to the latches 49 and 50 which control the count of the reference counter 35 in the PLL circuit.
Channel Number Display: The channel number is transferred from the microcomputer 23 to the driver latches of the display driver circuit 29.
Key Input Detection: The keyboard is arranged as the matrix circuit shown in FIG. 2. ROM programming for scanning and acknowledging a keyboard entry only after successive indications provides protection against false entry due to contact bounce. The four data output lines of the D output port of the microcomputer 23 are used to transfer data to the phase lock loop section of the circuit and to the display circuit 29, as well as for scanning the keyboard matrix circuit.
Time Count: The microcomputer 23 receives a basic timing pulse of approximately 200 Hz from the output of the divider 80 and performs various controls for each timing pulse. By way of example, sensing for the vertical synch input (when the system is used with a signal seek capability) on the input port SNS1 takes place every 2.5 milliseconds. Automatic seek timing is selected to be 133 milliseconds for UHF channels. All of these timing pulses are derived from the basic synchronization timing pulse applied to the microcomputer on the A13 input port from the output of the divider 80. Various other timing values used in the microcomputer to properly time multiplex sequence the operation are derived from this basic timing pulse.
Sensor Input Detection: As described previously, the output of the comparators shown in FIG. 3 reflect the status of the tuning of the television receiver. If no signal seek mode of operation is used, only the frequency discriminator or AFT discriminator 60 is necessary. When a system is being used in a signal seek mode, a proper television signal receipt is indicated by the presence of a vertical synch signal at the output of the synch signal separator 65 and corresponding outputs are applied to the input leads B10 and B11 (high level input signals) indicative of tuning to the "correct tuned" frequency discriminator window and reception of a picture carrier. As stated previously, the signal present on the B12 input lead is used to determine the direction of tuning when the receiver is operated in its automatic mode.
Mode Detection: The status of the seek and automatic/manual (A/M) switches are detected. If the A/M switch (not shown) is in its automatic position, automatic seek and offset correction are active. If only the seek switch is on, only seek is performed. If the A/M switch is in manual, manual fine tuning (MFT) is active.
Automatic Mode: If the TV receiver is not properly tuned for VHF channels in automatic, the local oscillator frequency is shifted automatically toward proper tuning. The fine tuning data is generated in the microcomputer 23 and is transferred to the latches 49 and 50 for the reference counter 35 in the PLL circuit.
Manual Fine Tuning (MFT) Control: The local oscillator frequency is shifted by pushing the fine tuning up (U) or down (D) pushbutton or switch. This MFT control can be applied to VHF channels as well as to UHF channels.
Channel Up/Down: When a channel up (upward pointing arrow) or down (downward pointing arrow) key closure in the keyboard 25 is detected, or upon a direct access to an unused channel, this routine is activated and the system will advance to the next channel in the selected direction.
A Cockcroft-Walton cascade circuit comprises an input voltage source and a pumping and storage circuit with a series array of capacitors with pumping and storage portions of the circuit being interconnected by silicon rectifiers, constructed and arranged so that at least the capacitor nearest the voltage source, and preferably one or more of the next adjacent capacitors in the series array, have lower tendency to internally discharge than the capacitors in the array more remote from the voltage source.
1. An improved voltage multiplying circuit comprising,
2. An improved voltage multiplying circuit in accordance with claim 1 wherein said first pumping capacitor is a self-healing impregnated capacitor which is impregnated with a high voltage impregnant.
3. An improved voltage multiplying circuit in accordance with claim 1 wherein said first pumping capacitor comprises a foil capacitor.
Description:
BACKGROUND OF THE INVENTION
The invention relates in general to Cockcroft-Walton cascade circuits for voltage multiplication and more particularly to such circuits with a pumping circuit and a storage circuit composed of capacitors connected in series, said pumping circuits and storage circuit being linked with one another by a rectifier circuit whose rectifiers are preferably silicon rectifiers, especially for a switching arrangement sensitive to internal discharges of capacitors, and more especially a switching arrangement containing transistors, and especially an image tube switching arrangement.
Voltage multiplication cascades composed of capacitors and rectifiers are used to produce high D.C. voltages from sinusoidal or pulsed alternating voltages. All known voltage multiplication cascades and voltage multipliers are designed to be capacitance-symmetrical, i.e., all capacitors used have the same capacitance. If U for example is the maximum value of an applied alternating voltage, the input capacitor connected directly to the alternating voltage source is charged to a D.C. voltage with a value U, while all other capacitors are charged to the value of 2U. Therefore, a total voltage can be obtained from the series-connected capacitors of a capacitor array.
In voltage multipliers, internal resistance is highly significant. In order to obtain high load currents on the D.C. side, the emphasis in the prior art has been on constructing voltage multipliers with internal resistances that are as low as possible.
Internal resistance of voltage multipliers can be reduced by increasing the capacitances of the individual capacitors by equal amounts. However, the critical significance of size of the assembly in the practical application of a voltage multiplier, limits the extent to which capacitance of the individual capacitors can be increased as a practical matter.
In television sets, especially color television sets, voltage multiplication cascades are required whose internal resistance is generally 400 to 500 kOhms. Thus far, it has been possible to achieve this low internal resistance with small dimensions only by using silicon diodes as rectifiers and metallized film capacitors as the capacitors.
When silicon rectifiers are used to achieve low internal resistance, their low forward resistance produces high peak currents and therefore leads to problems involving the pulse resistance of the capacitors. Metallized film capacitors are used because of space requirements, i.e., in order to ensure that the assembly will have the smallest possible dimensions, and also for cost reasons. These film capacitors have a self-healing effect, in which the damage caused to the capacitor by partial evaporation of the metal coating around the point of puncture (pinhole), which develops as a result of internal spark-overs, is cured again. This selfhealing effect is highly desirable as far as the capacitors themselves are concerned, but is not without its disadvantages as far as the other cirucit components are concerned, especially the silicon rectifiers, the image tubes, and the components which conduct the image tube voltage.
It is therefore an important object of the invention to improve voltage multiplication cascades of the type described above.
It is a further object of the invention to keep the size of the entire assembly small and the internal resistance low.
It is a further object of the invention to increase pulse resistance of the entire circuit.
It is a further object of the invention to avoid the above-described disadvantageous effects on adjacent elements.
It is a further object of the invention to achieve multiples of the foregoing objects and preferably all of them consistent with each other.
SUMMARY OF THE INVENTION
In accordance with the invention, the foregoing objects are met by making at least one of the capacitors in the pumping circuit, preferably including the one which is adjacent to the input voltage source, one which is less prone to internal discharges than any of the individual capacitors in the storage circuit.
The Cockcroft-Walton cascade circuit is not provided with identical capacitors. Instead, the individual capacitors are arranged according to their loads and designed in such a way that a higher pulse resistance is attained only in certain capacitors. It can be shown that the load produced by the voltage in all the capacitors in the multiplication circuit is approximately the same. But the pulse currents of the capacitors as well as their forward flow angles are different. In particular, the capacitors of the pumping circuit are subjected to very high loads in a pulsed mode. In the voltage multiplication cascade according to the invention, these capacitors are arranged so that they exhibit fewer internal discharges than the capacitors in the storage circuit.
The external dimensions of the entire assembly would be unacceptably large if one constructed the entire switching arrangement using such capacitors.
The voltage multiplication cascade according to the invention also makes it possible to construct a reliably operating
arrangement which has no tendency toward spark-overs, consistent with satisfactory internal resistance of the voltage multiplication cascade and small dimensions of the entire assembly. This avoids the above cited disadvantages with respect to the particularly sensitive components in the rest of the circuit and makes it possible to design voltage multiplication cascades with silicon rectifiers, which are characterized by long lifetimes. Hence, a voltage multiplication cascade has been developed particularly for image tube circuits in television sets, especially color television sets, and this cascade satisfies the highest requirements in addition to having an average lifetime which in every case is greater than that of the television set.
A further aspect of the invention is that at least one of the capacitors that are less prone to internal discharges is a capacitor which is impregnated with a high-voltage impregnating substance, especially a high-voltage oil such as polybutene or silicone oil, or mixtures thereof. In contrast to capacitors made of metallized film which have not been impregnated, this allows the discharge frequency due to internal discharges or spark-overs to be reduced by a factor of 10 to 100.
According to a further important aspect of the invention, at least one of the capacitors that are less prone to internal discharges is either a foil capacitor or a self-healing capacitor. In addition, the capacitor in the pumping circuit which is adjacent to the voltage source input can be a foil capacitor which has been impregnated in the manner described above, while the next capacitor in the pumping circuit is a self-healing capacitor impregnated in the same fashion.
Other objects, features and advantages of the invention will be apparent from the following detailed description of preferred embodiments, taken in connection with the accompanying drawing, the single FIGURE of which:
BRIEF DESCRIPTION OF THE DRAWING
is a schematic diagram of a circuit made according to a preferred embodiment of the invention.
DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS
The voltage multiplier comprises capacitors C1 to C5 and rectifiers D1 to D5 connected in a cascade. An alternating voltage source UE is connected to terminals 1 and 2, said voltage source supplying for example a pulsed alternating voltage. Capacitors C1 and C2 form the pumping circuit while capacitors C3, C4 and C5 form the storage circuit.
In the steady state, capacitor C1 is charged to the maximum value of the alternating voltage UE as are the other capacitors C2 to C5. The desired high D.C. voltage UA is picked off at terminals 3 and 4, said D.C. voltage being composed of the D.C. voltages from capacitors C3 to C5. Terminal 3 and terminal 2 are connected to one pole of the alternating voltage source UE feeding the circuit, which can be at ground potential. In the circuit described here, a D.C. voltage UA can be picked off whose voltage value is approximately 3 times the maximum value of the pulsed alternating voltage UE. By using more than five capacitors, a correspondingly higher D.C. voltage can be obtained.
The individual capacitors are discharged by disconnecting D.C. voltage UA. However, they are constantly being recharged by the electrical energy supplied by the alternating voltage source UE, so that the voltage multiplier can be continuously charged on the output side.
According to the invention, in this preferred embodiment, capacitor C1 and/or C2 in the pumping circuit are designed so that they have a lower tendency toward internal discharges than any of the individual capacitors C3, C4 and C5 in the storage circuit.
It is evident that those skilled in the art, once given the benefit of the foregoing disclosure, may now make numerous other uses and modifications of, and departures from the specific embodiments described herein without departing from the inventive concepts. Consequently, the invention is to be construed as embracing each and every novel feature and novel combination of features present in, or possessed by, the apparatus and techniques herein disclosed and limited solely by the scope and spirit of the appended claims.
Inventors:Petrick, Paul (Landshut, DT)
Schwedler, Hans-peter (Landshut, DT)
Holzer, Alfred (Schonbrunn, DT)
ERNST ROEDERSTEIN SPEZIALFABRIK
US Patent References:
3714528 ELECTRICAL CAPACITOR WITH FILM-PAPER DIELECTRIC 1973-01-30 Vail
3699410 SELF-HEALING ELECTRICAL CONDENSER 1972-10-17 Maylandt
3463992 ELECTRICAL CAPACITOR SYSTEMS HAVING LONG-TERM STORAGE CHARACTERISTICS 1969-08-26 Solberg
3457478 WOUND FILM CAPACITORS 1969-07-22 Lehrer
3363156 Capacitor with a polyolefin dielectric 1968-01-09 Cox
2213199 Voltage multiplier 1940-09-03 Bouwers et al.
Grundig CUC70 BLANK RASTER WITH FLYBACK LINES C514 150pF IN LINE MODULE AFFECTING SANDCASTLE PULSE
Grundig CUC70 Dead - blows Chopper tr BU208A D661 s/c; R661 0R15 o/c
Grundig CUC70 FIELD C2791 2757 IF IC FAULTY C2758 SIGNALS C2217 C2221
Grundig CUC70 Field collapse - pic cut off D2758 to pin 5 of field chip leaky
Grundig CUC70 Foldover if ajusting for correct height. C2757 ( 1 , 000uf )
Grundig CUC70 int dead dry joints chopper trfmr
Grundig CUC70 int E-W correction dry joints deflection panel where it plugs into main pcb.
Grundig CUC70 Int goes to st/by C642 100µ 10v
Grundig CUC70 Line ringing C622 600µ 63v
Grundig CUC70 LINE TEARING C501 1000uF 25v
Grundig CUC70 NO EHT NO LINE DRIVE NO 12v AS D689 OC
Grundig CUC70 No frame. Frame IC but also check C2758 ( 100uf ) & safety res o/c.
Grundig CUC70 No go - fuse blown. R661 ( 0.15ohm ) secondary side of chopper PSU o/c , TDA4600-2 , Fuse.
Grundig CUC70 No linear frame - kink at bottom on sides. C2791 ( 2 , 200uf or 3 , 300uf )
Grundig CUC70 No picture - 12v B+ rail missing. D689 ( 1N4001 ) o/c.
Grundig CUC70 No signals dry joints IC681 reg on main pcb
Grundig CUC70 No sound R366 ( 82R ) o/c , C366 ( 1 , 000uf ) s/c.
Grundig CUC70 off with a bang s/c mains rect block bew 2.5A fuse
Grundig CUC70 Poor picture with crinkly verticals from cold Check C2217 ( 10u f , 16V ) & C2221 ( 4.7u f , 25V ) in the tin IF module.
Grundig CUC70 Poor sync - sideways scrolling Txt C2216 1µ; C2221 4µ7
Grundig CUC70 Pulling - esp BBC1 & VCR chan - Txt scrolls sideways Replaced 2 x 1µ & 47µ caps in IF/Tuner unit near TDA4442
Grundig CUC70 Ragged verts Line driver supply smoother 1000µ low cap
Grundig CUC70 Reduced frame - with kink at bottom on sides. C2791 ( 2 , 200uf or 3 , 300uf )
Grundig CUC70 SNOW NO PICTURE R2127 180R IN IF CAN
Grundig CUC70 Snow on picture & no sound. DJ IC681.
Grundig CUC70 Sound uncontrollable - full vol - may be int R2250 4k7 preset sound pot int
Grundig CUC70 Tuning - no tuning. C2020 in tuner & T2304 in tuning module.
Grundig CUC70KT BENT VERTICALS WORSE FROM COLD C2731 100u 25v ON LINE/FIELD PANEL
Grundig CUC70KT Dead. D661 ( SKE4G2/06 ) o/c can use SKE4F2/02
Grundig CUC70KT East west - poor. R571 o/c.
Grundig CUC70KT Line - cogging. C662 ( 600u f , 63v )
Grundig CUC70KT No Picture - sound Ok - static on screen. Frame PCB R2756 ( 7R 1/4watt ) o/c if so IC TDA2653A
Grundig CUC70KT Tripping - starts up then trips. D471 s/c can use a BY299.
Grundig CUC70 Dead - blows Chopper tr BU208A D661 s/c; R661 0R15 o/c
Grundig CUC70 FIELD C2791 2757 IF IC FAULTY C2758 SIGNALS C2217 C2221
Grundig CUC70 Field collapse - pic cut off D2758 to pin 5 of field chip leaky
Grundig CUC70 Foldover if ajusting for correct height. C2757 ( 1 , 000uf )
Grundig CUC70 int dead dry joints chopper trfmr
Grundig CUC70 int E-W correction dry joints deflection panel where it plugs into main pcb.
Grundig CUC70 Int goes to st/by C642 100µ 10v
Grundig CUC70 Line ringing C622 600µ 63v
Grundig CUC70 LINE TEARING C501 1000uF 25v
Grundig CUC70 NO EHT NO LINE DRIVE NO 12v AS D689 OC
Grundig CUC70 No frame. Frame IC but also check C2758 ( 100uf ) & safety res o/c.
Grundig CUC70 No go - fuse blown. R661 ( 0.15ohm ) secondary side of chopper PSU o/c , TDA4600-2 , Fuse.
Grundig CUC70 No linear frame - kink at bottom on sides. C2791 ( 2 , 200uf or 3 , 300uf )
Grundig CUC70 No picture - 12v B+ rail missing. D689 ( 1N4001 ) o/c.
Grundig CUC70 No signals dry joints IC681 reg on main pcb
Grundig CUC70 No sound R366 ( 82R ) o/c , C366 ( 1 , 000uf ) s/c.
Grundig CUC70 off with a bang s/c mains rect block bew 2.5A fuse
Grundig CUC70 Poor picture with crinkly verticals from cold Check C2217 ( 10u f , 16V ) & C2221 ( 4.7u f , 25V ) in the tin IF module.
Grundig CUC70 Poor sync - sideways scrolling Txt C2216 1µ; C2221 4µ7
Grundig CUC70 Pulling - esp BBC1 & VCR chan - Txt scrolls sideways Replaced 2 x 1µ & 47µ caps in IF/Tuner unit near TDA4442
Grundig CUC70 Ragged verts Line driver supply smoother 1000µ low cap
Grundig CUC70 Reduced frame - with kink at bottom on sides. C2791 ( 2 , 200uf or 3 , 300uf )
Grundig CUC70 SNOW NO PICTURE R2127 180R IN IF CAN
Grundig CUC70 Snow on picture & no sound. DJ IC681.
Grundig CUC70 Sound uncontrollable - full vol - may be int R2250 4k7 preset sound pot int
Grundig CUC70 Tuning - no tuning. C2020 in tuner & T2304 in tuning module.
Grundig CUC70KT BENT VERTICALS WORSE FROM COLD C2731 100u 25v ON LINE/FIELD PANEL
Grundig CUC70KT Dead. D661 ( SKE4G2/06 ) o/c can use SKE4F2/02
Grundig CUC70KT East west - poor. R571 o/c.
Grundig CUC70KT Line - cogging. C662 ( 600u f , 63v )
Grundig CUC70KT No Picture - sound Ok - static on screen. Frame PCB R2756 ( 7R 1/4watt ) o/c if so IC TDA2653A
Grundig CUC70KT Tripping - starts up then trips. D471 s/c can use a BY299.
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