The RGB Amplifier in the chassis PHILIPS K12 is operating on the G1 instead on katode in conventional schemes.
These are realized with HYBRID IC'S technology and they're in ceramic substrate.
PHILIPS CHASSIS K12 (20AX) CRT TUBE RGB AMPLIFIER DETAIL RGB HYBRID AMPLIFIER 4822 212 20307 3122 128 58455
Hybrid RGB Amplifiers on CRT Socket.
Driving directly: G1 blue G1 red G1 green.Hybrid IC Technology on Ceramic substrate.
On a ceramic substrate, spiral-type inductors of a single layer wiring of a metal thin film are provided and respectively connected to a wiring pattern formed on another face of the substrate via through holes. A semiconductor chip is flip-chip mounted on the substrate in a face-down manner. On the face of the semiconductor chip, capacitors composed of a highly dielectric material, resistors formed by an ion implantation method or a thin-film forming method, and FETs are provided, respectively. Interconnection between the substrate and an external circuit board is achieved employing terminals formed at end faces of the substrate. The terminals have a concave shape with respect to the end face of the substrate. Thus, there is no need to use a package, and miniaturization and reduction in cost of a high-performance hybrid IC is achieved.
1. A hybrid IC comprising:
a substrate including a front face, a back face opposite the front face, and side faces interposed between the front face and the back face which define an outer perimeter of the substrate;
at least one inductor formed on at least one of the front face and the back face of the substrate;
a semiconductor chip mounted on the front face of the substrate by flip-chip bonding;
at least one terminal formed in a predetermined portion of the side faces of the substrate,
wherein the semiconductor chip comprises a plurality of circuit elements provided therein, at least one of the plurality of circuit elements being an MIM capacitor having a metal-insulation film-metal (MIM) structure, the insulation film being composed of a highly dielectric material.
2. A hybrid IC according to claim 1 further comprising at least one matching circuit for matching an input signal to the circuit elements provided inside the semiconductor chip, the matching circuit comprising at least one inductor.
3. A hybrid IC according to claim 2, wherein a wiring pattern is formed of a single metal layer on both the front and back faces of the substrate, the wiring patterns on the respective front and back faces of the substrate being interconnected with each other via through holes, and the at least one inductor comprised in the matching circuit is formed in the wiring pattern on one of the respective front and back faces of the substrate.
4. A hybrid IC according to claim 2, wherein the matching circuit is constituted only by inductors and comprises at least one serial inductor and at least one parallel inductor.
5. A hybrid IC according to claim 4, wherein the parallel inductor comprised in the matching circuit is a spiral-type inductor, outermost wiring of the spiral-type inductor being grounded.
6. A hybrid IC according to claim 2, wherein the inductors comprised in the matching circuit are a spiral-type inductor or a meander-type inductor.
7. A hybrid IC according to claim 2, wherein the matching circuit comprises an inductor and a capacitor, the capacitor being formed inside the semiconductor chip.
8. A hybrid IC according to claim 7, wherein the inductor comprised in the matching circuit is a spiral-type inductor or a meander-type inductor.
9. A hybrid IC according to claim 1, wherein the at least one terminal includes at least an RF terminal functioning as an input terminal for an RF signal, an LO terminal functioning as an input terminal for an LO signal, an IF terminal functioning as an output terminal for an IF signal, a ground terminal, and a supply terminal.
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HERE BELOW A DETAILED DESCRIPTION OF THE PHILIPS CHASSIS K12 CRT DRIVING TECHNOLOGY:
PHILIPS CHASSIS K12 Automatic gray scale control circuit for a color television receiver
The present invention relates to a novel automatic gray scale control circuit for a color television receiver. The circuit senses the cut-off voltage of each gun during the blanking interval, and uses a voltage equal to the cut-off voltage to energize the driver and bias the gun during the video field. The effect is to standardize the emission of each of the three guns against variation in gun cut-off voltage and to produce improved gray scale accuracy at the lowest emission levels. Since the gray scale adjustment is optimized at the lowest emission levels, where the eye is most intolerant to error in hue, one may avoid the need for manual adjustment of the cut-off point, and in cases where the gain does not vary widely from gun to gun, avoid the need for separate gain adjustment. Thus, the circuit may be used either to simplify or eliminate the color set up process at the factory when the receiver is manufactured. It may also reduce or avoid the need for readjustment after periods of use.
CHASSIS K12 CRT Beam current control apparatus:
Introducing beam current control:
1. A television picture display device for displaying pictures derived from video signals, said display device comprising a picture display tube having a cathode; an emitter-follower device coupled to said cathode for producing a beam current for driving said picture display tube; a beam current reference level control circuit coupled to said emitter-follower device for controlling the black level of the beam current; a blacker-than-black current compensation circuit incorporated in said beam current reference level control circuit for compensating for a leakage current in said cathode caused by a blacker-than-black level in the video signal; and a blacker-than-black current conduction circuit coupled to said cathode to enable compensation of said leakage current when said beam current is blanked.
2. A television picture display device as claimed in claim 1, wherein the blacker-than-black current conduction circuit comprises a direct current source coupled to the cathode of the picture display tube for keeping said emitter-follower device conductive at the occurrence of a blacker-than-black current in the cathode circuit of the picture display tube thereby allowing for the compensation of said blacker-than-black signal.
3. A television picture display device as claimed in claim 2, wherein the direct current source comprises a first and second resistor serially connected to a supply voltage said first resistor being a.c. coupled to the base of the emitter-follower device.
5. A television picture display device as claimed in claim 4, wherein said difference-forming circuit comprises a current mirror circuit having an input and an output, the input of which is coupled to the collector of the second emitter-follower device and the output to the collector of the first-mentioned emitter-follower device.
6. A television picture display device as claimed in claims 2, 3, 4, 5 or 1, which further comprises a heater, a wehnelt electrode connection of the picture display tube and a leakage current conducting circuit, said heater and said wehnelt electrode connection being coupled to the collector of the first-mentioned emitter-follower device through a said leakage current conducting circuit.
Description:
BACKGROUND OF THE INVENTION Dutch Patent Application No. 7604463 discloses a television picture display device of the above-defined type. In this device a control of the black level, serving as the reference level, of the beam current, to a constant value takes inter alia place by means of a voltage produced across the collector resistor of the emitter-follower. Although in principle this control should furnish a very constant black level this appears not to be the case.
SUMMARY OF THE INVENTION
It is an object of the invention to improve the constancy of the controlled reference level of the beam current.
A television picture display de
vice of the above-defined type according to the invention is therefore characterized in that the beam current reference level control circuit comprises a blacker-than-black current compensation circuit by which the influence of the blacker-than-black current on the beam current reference level control circuit is compensated for while a blacker-than-black current conductive circuit is coupled to the cathode of the picture display tube so that the beam current reference level control circuit can also process a blacker-than-black current of the picture display tube occurring in case of a blanked beam current. DESCRIPTION OF THE DRAWINGS
The invention will now be further explained with reference to the drawing.
In the drawing
FIG. 1 shows a circuit of a picture display device according to the invention in which the emitter-follower can be kept conductive by means of a direct current and
FIG. 2 shows a further circuit of a picture display device according to the invention in which the emitter-follower current can be taken over by another circuit and passed to a control circuit.
DESCRIPTION OF THE PREFERRED EMBODIMENT
In FIG. 1 a video signal is applied to an input of an adder circuit 3. A control signal originating from an output 7 of a beam current reference level control circuit 9 appears at a further input 5 of the adder circuit 3. This control signal is added to the video signal and the sum signal is applied to the base of an emitter-follower 13 via an amplifier 11. The emitter of the emitter-follower 13 drives a cathode of a picture display tube 14. The collector circuit of the emitter-follower 13 comprises a resistor 15, an end of which is connected to an input 17 of the beam current reference level control circuit 9. The beam current supplied by the emitter-follower 13 flows through this resistor 15.
The input 17 is, at the same time, the input of an amplifier 19, an output 21 of which is connected to an input 25 of an amplifier 27 via a switch 23. A capacitor 29 is also connected to the input 25. The switch 23 closes periodically under the influence of a signal derived from an output 31 of a switching signal generator 33 during the occurrence of the reference level, for example the black level, in the video signal.
Consequently a voltage, which is a measure of the cathode current of the picture display tube 14 during the occurrence of the reference level, is produced across the capacitor 29. This cathode current is composed of a beam current and a leakage current. To compensate for the influence of this leakage current, a switch 35, which is also connected to the output 21 of the amplifier 19, is periodically closed during the occurrence of a blacker-than-black level in the video signal, under the influence of a signal originating from an output 37 of the switching signal generator 33.
The switching signal generator 33 is synchronized with the video signal by a synchronisation signal, applied to an input 39 thereof, obtained from the video signal.
A capacitor 41, connected to the output of the switch 35, now has a voltage there across which is a measure of the leakage current of the cathode of the picture display tube 14. This voltage is supplied to an input 43 of the amplifier 27. The amplifier 27 amplifies the difference of the voltages across the capacitors 29 and 41 so that a control signal is produced at the output 7 thereof which is a measure of the beam current of the picture display tube during the occurrence of the reference level in the video signal. This control signal counteracts changes in the beam current reference level which corresponds to the video signal reference level.
To enable also a compensation of this direct current in the the blacker-than-black compensation circuit, constituted by the switch 35, the capacitor 41 and the difference formation in the amplifier 27, the value of this direct current must not be dependent on the video signal. Therefore this video signal is applied, via a capacitor 49, to the junction of the resistors 45, 47 so that the voltage across the resistor 45 becomes independent of the video signal.
The variation in the leakage current in the cathode circuit of the picture display tube 14 can be many times larger than that in the beam current, which condition occurs if the reference level in the video signal is present. The variation in the voltage difference across the capacitors 28 and 41 would then be many times smaller than the variation in the voltage across one of the capacitors 28, 41. This might adversely affect the accuracy of the control system. To prevent this, two resistors 53 and 55 are provided which pass the leakage current from the wehnelt circuit and the heater circuit to the resistor 15 and, consequently, compensate the cathode leakage current to a large extent.
It will be obvious that the direct current supplied to the cathode of the picture display tube may be obtained, if so desired, by means of a transistor connected as a current source. This transistor should then be suitable for a rather high voltage because the voltage at the cathode of the picture display tube may change very much.
In FIG. 2 elements corresponding to elements of the circuit of FIG. 1 have been given the same reference numerals as in FIG. 1. The blacker-than-black current compensation circuit of FIG. 2 is arranged somewhat differently then in FIG. 1. The capacitor 41 is now arranged in series with the output 21 of the amplifier 19 and the two switches 23 and 35, and the input 43 of the amplifier 27 is connected to ground. The switch 35 now operates as a clamping switch which ensures that the blacker-than-black level is connected to ground and the voltage across the capacitor 29 becomes a measure of the beam current occurring at the reference level in the video signal.
A further difference relative to the circuit of FIG. 1 is that the collector circuit of the second emitter-follower 51 includes a circuit which acts as blacker-than-black current conduction circuit and supplies any current flowing to the cathode to the input 17 of the beam current reference level control circuit 9 so that also these currents can be measured. The direct current supply circuit (45, 47) at the cathode of the picture display tube can then be dispensed with.
The collector current of the second emitter-follower 51 is supplied to the resistor 15 via two current mirror circuits. A first current mirror circuit is constituted by a series arrangement of a resistor 57 and a diode 59 in parallel with the series arrangement of the base-emitter path of a transistor 61 and a resistor 63 to a positive supply voltage. The collector current of the transistor 61 is supplied to a second current mirror circuit having a transistor 65 and a diode 67, which is in parallel with the base-emitter path of the transistor 65. The collector of the transistor 61 is connected to the resistor 15 through which the difference in the collector currents of the two emitter-followers 13, 51 now flows. Independent of the direction of the cathode currents of the picture display tube 14, a voltage, which is a measure of that cathode current, is now produced across the resistor 15.
Instead of determining the difference current by means of current mirror circuits in the described manner, it is alternatively possible, if so desired, to use other difference-determining circuits.
If so desired also the measuring data for a beam current limiting control of the resistor 15 can be obtained in the described circuits.
The amplifier 19 may comprise
a circuit which limits the amplitude of the video signal outside the instants in which measuring takes place. This may be a circuit operated by an auxiliary signal or a self-switching circuit, for example a diode limiter circuit. If the picture display tube is a color display tube having several electron guns, the emitter-followers for each of the guns may have the resistor 15 in common and a sequential measurement may take place at a reference level sequentially occurring in the different video signals, so that only a portion of the control circuits is not common.
It will be obvious that the choice of the measuring instants and the associated occurrence of the reference levels and blacker-than-black levels are not important for the essence of the invention and may be chosen in a suitable manner.
The blacker-than-black current compensation circuits 9 of the above-described embodiments are interchangeable.
The amplifier 27 may comprise a level reference circuit so that the value of the beam current is determined which is associated with the corresponding reference level in the video signal.
PHILIPS 22C850 /88Z " Tiepolo" CHASSIS K12 (20AX) System for stabilizing cathode ray tube operation:
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The invention pertains to a video display apparatus which includes a cathode ray tube having at least one electron gun and means fo
r deriving a source of excitatory voltage signal representative of picture information. In accordance with the invention there is provided a system for stabilizing the display intensity attributable to the electron gun comprising means coupling the source of voltage signal to the gun for generating a stabilized current in the gun, the level of current in the gun being substantially independent of variations in the operating characteristics of the gun. In the preferred embodiment of the invention a test signal is periodically applied at a predetermined level to the gun. A sampler samples the beam current of the gun during the test signal and generates a correction signal in accordance with the sample value. The correction signal is applied to the electron gun so as to stabilize its reference current level.
1. In a television display apparatus which includes a cathode ray tube having at least one electron gun and means for applying a television video signal to said electron gun; a system for stabilizing the display intensity attributable to said electron gun, comprising:
means for generating first and second test signals during the vertical blanking intervals of said television video signal;
means for applying said first and second test signals to said electron gun during first and second portions, respectively, of said vertical blanking intervals;
means for sampling the beam current of said electron gun during said first and second portions and for generating first and second correction signals, respectively, in accordance with the sampled values; and
means for applying said first and second correction signals to said electron gun so as to stabilize the reference current level of said electron gun.
2. The system as defined by claim 1 wherein said first and second correction signals are applied to grids of said electron gun.
3. The system as defined by claim 2 wherein means are provided for maintaining the correction signals during the intervals between successive test signals.
4. The system as defined by claim 1 wherein means are provided for maintaining the correction signals during the intervals between the successive test signals.
5. The system as defined by claim 1 wherein said first and second test signals are at white and black level viedo, respectively.
6. The system as defined by claim 5 wherein said first correction signal is applied as a multiplying factor to said television video signal.
7. The system as defined by claim 6 wherein the corrected television video signal is applied to the control grid of said electron gun.
8. The system as defined by claim 6 wherein the corrected television video signal is applied to the cathode of said electron gun.
9. The system as defined by claim 6 wherein said second correction signal is applied as a DC reference level to the corrected television video signal.
10. The system as defined by claim 1 wherein said means for sampling the beam current comprises a resistor in series with the cathode of said gun.
11. In a video display apparatus which includes a cathode ray tube having a plurality of electron guns and means for applying a plurality of television video signals representative of color picture information to said electron guns; a system for balancing the color screen temperature of said cathode ray tube, comprising:
means for generating first and second test signals during the vertical blanking intervals of said television video signal;
means for applying said first and second test signals to said electron guns during first and second portions, respectively, of said vertical blanking intervals;
means for sampling the beam current of each electron gun during said first and second portions and for generating first and second correction signals, respectively, for each electron gun in accordance with the sampled values; and
means for applying said first and second correction signals to their respective electron guns so as to balance the screen color temperature of said cathode ray tube.
12. The system as defined by claim 11 wherein said first and second correction signals are applied to grids of said electron guns.
13. The system as defined by claim 12 wherein means are provided for maintaining the correction signals during the intervals between successive test signals.
14. The system as defined by claim 11 wherein means are provided for maintaining the correction signals during the intervals between successive test signals.
15. The system as defined by claim 11 wherein said first and second test signals are at white and black level video, respectively.
16. The system as defined by claim 15 wherein said first correction signal is applied as a multiplying factor to each of said plurality of television video signals.
17. The system as defined by claim 16 wherein the corrected plurality of television video signals are applied to the control grids of their respective electron guns.
18. The system as defined by claim 16 wherein the corrected plurality of television video signals are applied to the cathodes of their respective electron guns.
19. The system as defined by claim 16 wherein said second correction signals are applied as DC reference levels to the corrected television video signals.
20. The system as defined by claim 11 wherein said means for sampling the beam current comprises a plurality of resistors in series with the cathode of said electron guns.
Description:
BACKGROUND OF THE INVENTION
This invention relates to improvements in video display apparatus and, more particularly, to a system for stabilizing the display intensity or "color temperature" of a cathode ray tube. The subject matter of this invention is related to subject matter disclosed in copending U.S. application Ser. No. 572,128 of C. W. Smith and R. H. McMann, filed of even date herewith and assigned to the same assignee.
Conventional television display systems employing kinescope cathode ray tubes are subject to performance degradation resulting from instabilities in the operating characteristics of the kinescope or the circuits which drive or bias the kinescope. Prior techniques have been developed which serve to stabilize the signals driving a kinescope. For example, the drive voltages applied to the cathodes of a color kinescope can be stabilized using a feedback scheme; e.g., circuitry which periodically senses the drive voltage at input "black" and "white" levels of operation and corrects for deviations from standard reference voltages by gain adjustment. DC voltages applied to the kinescope can also be stabilized by using precise voltage regulation techniques.
There remains, however, the recognized problem of kinescope electron gun drift which manifests itself as a drift in screen color temperature in a three gun color kinescope. As the electron guns age, their generated beam current per unit of applied voltage (which can be considered a transconductance function) varies, the variations being generally non-uniform in the three different guns. This is a cause of noticeable and undesirable drifts in the display screen color.
The major sources of drift are: aging or long term variations caused by a gradual decrease in cathode activity, not necessarily constant or uniform for each cathode; and cathode operating temperature. The relatively long term variations in emission are caused by filament voltage changes and heat build-up in the gun area, generally a function of how many hours a display tube has been operating. Dynamic heating of each gun depends on the ratio of gun currents drawn to provide the colored picture being instantaneously presented. For example, a long persisting mostly red field causes red gun current almost exclusively, thereby causing an unbalanced heating of the red cathode, which changes its emission characteristics to a different degree than the other cathodes, this change remaining until relative cooling occurs.
Cathode thermal current, I th , is represented by the Dushman equation: I th = SA 0 T 2 e - b s /T amperes
where S and A 0 are constants and b 0 = Dushman constant ≉ 11,600° for an oxide coated cathode.
The derivative of the natural logarithm of this equation gives the change in emission with respect to temperature change: dI th /I th = 2 + (b 0 /t ) (dT/T)
the temperature of the CRT cathode is approximately 1,160° K, which yields (dI th /I th) = 12dT/T
typical ambient temperature variations, such as in a display monitor, are about 40° C, so that the net change of gun current is of the order of 12 . 40/1,160 ≉ 40%
Therefore, a 1° C change in cathode temperature yields about a 1% change in gun current, if the gun is fixed bias and not near cut-off.
It is an object of the present invention to provide a stabilizing system which overcomes the problems set forth.
SUMMARY OF THE INVENTION
The invention pertains to a video display apparatus which includes a cathode ray tube having at least one electron gun and means for deriving a source of excitatory voltage signal representative of picture information. In accordance with the invention there is provided a system for stabilizing the display intensity attributable to the electron gun comprising means coupling the source of voltage signal to the gun for generating a stabilized current in the gun, the level of current in the gun being substantially independent of variations in the operating characteristics of the gun.
In the preferred embodiment of the invention a test signal is periodically applied at a predetermined level to the electron gun. A sampler samples the beam current of the gun during the test signal and generates a correction signal in accordance with the sample value. The correction signal is applied to the gun so as to stabilize its reference current level.
Further features and advantages of the invention will become more readily apparent from the following detailed description when taken in conjunction with the accompanying drawings.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a simplified diagram of a color television display kinescope;
FIGS. 2A, 2B and 2C are block diagrams of embodiments of the invention which utilize periodically applied test signals;
FIG. 3 is a schematic representation of an embodiment of the invention which employs a "constant current" technique;
FIG. 4 is a schematic representation of another embodiment of the invention employing a differential amplifier; and
FIG. 5 is a schematic representation of another embodiment of the invention employing direct cathode temperature sensing and heater control.
DESCRIPTION OF THE PREFERRED EMBODIMENTS
Referring to FIG. 1, there is shown a simplified diagram of a color television display cathode ray tube or kinescope 10 as driven by excitatory video voltage signals designated R, G and B, these signals having typical ranges of about 100 volts peak-to-peak. The kinescope 10 has three electron guns, each including a cathode and associated grids. For clarity of illustration, only one of the three guns, designated by reference numeral 11, is represented in some detail, but it will be appreciated that two other complete electron guns (indicated in the Figure by only the two dashed cathodes coupled to the G and B inputs) are normally provided and are substantially identical to the gun 11. Hereinafter, and in the description of the embodiments of the invention, the circuitry associated with only one electron gun in a given kinescope will be described for illustrative clarity, but it will be understood that if the kinescope has two or more guns, similar circuitry can be employed in conjunction with the remaining guns.
The electron gun 11 comprises a cathode 21 and first, second and third grids, 22, 23, and 24, which are sometimes designated as "grid 1," "grid 2," and "grid 3," or as the "control electrode," the "accelerating electrode," and the "focusing electrode," respectively. Generated electrons impinge on an anode 25 near display screen 26 which is coated with an electron-sensitive phosphor as is conventional in the art. Typical voltages applied to the cathode, grid 1, grid 2, grid 3 and the anode are about 235 volts, 150 volts, 700 volts, 5000 volts and 25,000 volts, respectively. In alternate modes of operation, the excitatory voltage input signal may be applied to a control grid with the remaining grid and cathode voltages being set at appropriate values.
Referring to FIG. 2A, there is shown an embodiment of the invention which comprises a system for stabilizing the display intensity or "color temperature" of the electron guns in a kinescope 30. An excitatory voltage signal at input terminal 31, which may be the R, B or G signal in a color system or the luminance signal in a black and white system, is coupled through an adder 41 and DC restorer circuit 42 to grid 1 of the kinescope 30. Cathode 43 is coupled through a resistor R 1 to ground reference. A bias voltage is applied to grid 2 via a voltage amplifier 44 which receives a signal on a line 58A which determines the level of the bias voltage applied to grid 2. Suitable focus and anode voltages are applied to grid 3 and the anode from sources not shown.
The vertical and horizontal synchronizing signals of the composite television signal, available in the television receiver, are applied to a line counter 51 which is adapted to count horizontal scanlines of the television field and to be reset to zero at the end of each television field. The counter generates a first output on a line 51A during the scanlines 15-17 of each television field and a signal on line 51B during lines 18-20 of each television field, all of the lines 15-20 occuring during the vertical blanking period. The signal on line 51A enables a gate 55 and also enables a sample-and-hold circuit 56. The signal on line 51B enables a gate 57 and a sample-and-hold circuit 58. The gates 55 and 57 respectively receive voltages at reference "black level" and "white level." The outputs of gates 55 and 57 are coupled over lines 55A and 57A, respectively, to inputs of the adder 41.
Operation of the system of FIG. 2A is as follows: During lines 15-17 of the vertical blanking interval the gate 55 is enabled so that black level voltage is coupled through adder 41 and circuit 42 to grid 1. With this voltage applied to grid 1 the cathode current should ideally have a certain nominal value that does not vary with the tube life or cathode temperature but, as indicated above in the Background, this is not generally the case in actual practice. The actual cathode current is sampled across resistor R 1 , and a voltage representative of this current is coupled to the sample-and-hold circuit 56 which is enabled to sample the voltage across resistor R 1 during the lines 15-17. The circuit 56 holds the sampled voltage through the subsequent video field and couples the held voltage to circuit 42 via line 56A, this voltage serving to adjust the DC reference level of the output of circuit 42. In this manner, the voltage on line 56A controls the bias level at grid 1 so as to correct for any variations in the cathode current at nominal black level. Thus, for example, if at some point in operation the cathode current for a black level input voltage is lower than its nominal value, the voltage drop across sampling resistor R 1 will also be low. This will decrease the output of sample-and-hold circuit 56 fed to circuit 42 which, in turn, will cause the bias level at grid 1 to decrease (typically, to a less negative value with respect to the cathode). A lesser negative bias level on the control grid 1 will, in turn, cause a proportionate increase in the electron current flowing from cathode 43; the desired result.
Similarly, during lines 18-20 of the vertical blanking period white level voltage is applied via adder 41 and amplifier 42 to grid 1, and during this time the cathode current is sampled by circuit 58 which is enabled to sample by the signal on line 51B. During the remainder of the television field, the bias voltage applied to grid 2, via voltage amplifier 33, is a function of the voltage which had been sampled by circuit 58. For example, in an instance where the cathode current sensed at a white level voltage input is lower than the nominal value, the resultant low voltage sampled by circuit 58 will cause the grid 2 accelerating voltage to decrease. This causes the sampled voltage at black level to appear too negative (when next sampled during the succeeding vertical blanking interval) which, in turn, results in a decrease in grid bias by the black level circuit causing the desired increase in beam current over prior conditions, as previously described.
The embodiment of FIG. 2B is similar to that of FIG. 2A except that the output of sample-and-hold circuit 58 (which is a measure of the sampled white level current) is coupled to an analog multiplier circuit 59, which is in series with DC restorer circuit 52. In this embodiment, corrections resulting from both the white level and black level measurements are achieved via grid 1, with operation otherwise being substantially as described above.
In the embodiment of FIG. 2C the electron gun is driven by application of the video signal to the cathode 43 via a complementary emitter-follower 120 which comprises NPN transistor 121 and PNP transistor 122. (The system to the left of blocks 42 and 56 is the same as in FIG. 2B). The transistor emitters are coupled to the cathode 43 of kinescope 30 and the transistor bases receive the video signal from DC restorer circuit 42. The collector of transistor 121 is coupled to a suitable bias voltage, e.g., 150 volts, and the collector of transistor 122 is coupled to ground reference potential through sampling resistor R 1 .
In operation, during the lines 15-20 the test signals are applied via circuit 42 and cathode 43 is driven while the cathode current is sampled by resistor R 1 , a typical value for which is 1K ohm. Transistor 122 is "on" during the white level test signal (output of circuit 42 about 25 volts) and the black level test signal (output of circuit 42 about 125 volts), and the gun current-representative voltages sampled across resistor R 1 are coupled to the appropriate sample-and-hold circuits as previously described. During the active portion of the television field the analog multiplier 59 and DC restorer circuit 42 apply appropriate corrections, with transistor 122 normally "on." During rapid lighter-to-darker transitions of the video signal the transistor 121 turns momentarily "on" and the tube capacitance and stray capacitance (collectively represented by C in the Figure can be thought of as charging. Diode D 1 prevents inordinate voltage drops across R 1 during the active picture area when R 1 is not used for sampling.
In the embodiment of FIG. 3 the video voltage signal at terminal 61 is coupled to cathode 71 of a kinescope 75 by the parallel combination of capacitor 62 and amplifier 63 in a series with resistor R 2 . Amplifier 63 comprises transistors 64 and 65 and has a voltage gain of about 5 and an output capability of about 500 volts. The resistor R 2 is selected to be substantially greater than the input impedance of the cathode 71 and preferably has a resistance at least five times higher than the cathode resistance. Since the effective cathode resistance is the inverse of the gun transconductance (about 8.6 micromhos), a suitable value for R 2 is of the order of 600K ohms. Accordingly, the amplifier 63 in conjunction with resistor R 2 operates as a so-called "constant current" source, which effectively transforms the voltage signal at terminal signal to a current source input to the cathode 71, this current source input being relatively insensitive to variations in the kinescope characteristics. Since normal wiring capacitance and electron gun interelectrode capacitance render high frequency response impractical i
n a high impedance amplifier drive, the higher frequency portions of the video signal are shunted across the amplifier by capacitor 62 which may have a typical value of about 0.05 microfarads. The higher frequency signals arrive at substantially the same relative level as the low frequencies, thereby preserving their relationship. This is because the lower frequency signals are amplified by a factor of 5 and then undergo a one-fifth loss by virtue of the voltage divider action of resistor R 2 and the cathode impedance.
FIG. 4 shows a further embodiment wherein the video voltage signal at an input terminal 81 is applied to one input of a differential amplifier comprising transistors 82, 83 and 84. The output stage 84 drives the grid 1 electrode of kinescope 90 through series peaking inductor L 1 and shunt peaking inductor L 2 . The cathode 91 of kinescope 90 is coupled to ground reference potential through resistor R 3 which is used to continuously monitor the cathode current, the line 89 coupling a voltage representative of the cathode current to the other input of the differential amplifier; viz., the base of transistor 83.
In operation, the voltage developed across resistor R 3 is proportional to the cathode current. This voltage, for a stable transconductance, should be in a stable relationship with respect to input voltage at terminal 81, and R 3 is selected empirically at a value, typically about 2K ohms, which generates a sample voltage nominally equal to the input voltage at terminal 81. When a deviation exists between the inputs to transistors 82 and 83, the output of the differential amplifier adjusts up or down to correct for the difference, thereby adjusting control of the drive to grid 1 and correcting for drifts in the kinescope transconductance.
A characteristic of the circuit of FIG. 4 is that it linearizes the electron gun transfer function which normally is non-linear, the non-linear function conventionally being known as "gamma" of the kinescope. Television video signals are conventionally precorrected for the gamma of the kinescope. In a color kinescope the gamma may be different for each gun, making it difficult to match the effective light output attributable to each gun over the grey scale; a problem known as "tracking" in the prior art. The present invention allows use of an inverse gamma circuit (which eliminates the precorrection in the conventional television signal) and the linearized gun transfer functions reduce tracking problems.
The invention has been described with reference to particular embodiments, but it will be understood that variations within the spirit and scope of the invention will occur to those skilled in the art. For example, the circuits of the "constant current" generator of FIG. 3 or the differential amplifier of FIG. 4 may be of other suitable forms. Also, in the embodiment of FIG. 2, sampling could be achieved during any suitable blanking or active period. The beam could be deflected off the tube face during sampling time to avoid displaying the trace during this time. Finally, stabilization of cathode temperatures could be achieved directly, such as by providing heater/thermistor stabilization circuits for each cathode. A suitable circuit is shown in FIG. 5 wherein a negative temperature coefficient thermistor 101 is attached to the cathode metal. V 0 is a precision voltage source providing a voltage typically in the range 5-12 volts and R 0 is selected as being substantially equal to the resistance of the thermistor at nominal cathode temperature. If the cathode becomes unduly hot, the resistance of thermistor 101 will decrease which, in turn, causes the voltage at terminal 103 to decrease. This results in a decreased output of operational amplifier 102, so that the cathode heater drive is reduced, as desired. Insufficient cathode temperature can be seen to cause the opposite effect.
This invention relates to improvements in video display apparatus and, more particularly, to a system for stabilizing the display intensity or "color temperature" of a cathode ray tube. The subject matter of this invention is related to subject matter disclosed in copending U.S. application Ser. No. 572,128 of C. W. Smith and R. H. McMann, filed of even date herewith and assigned to the same assignee. Conventional television display systems employing kinescope cathode ray tubes are subject to performance degradation resulting from instabilities in the operating characteristics of the kinescope or the circuits which drive or bias the kinescope. Prior techniques have been developed which serve to stabilize the signals driving a kinescope. For example, the drive voltages applied to the cathodes of a color kinescope can be stabilized using a feedback scheme; e.g., circuitry which periodically senses the drive voltage at input "black" and "white" levels of operation and corrects for deviations from standard reference voltages by gain adjustment. DC voltages applied to the kinescope can also be stabilized by using precise voltage regulation techniques.
There remains, however, the recognized problem of kinescope electron gun drift which manifests itself as a drift in screen color temperature in a three gun color kinescope. As the electron guns age, their generated beam current per unit of applied voltage (which can be considered a transconductance function) varies, the variations being generally non-uniform in the three different guns. This is a cause of noticeable and undesirable drifts in the display screen color.
The major sources of drift are: aging or long term variations caused by a gradual decrease in cathode activity, not necessarily constant or uniform for each cathode; and cathode operating temperature. The relatively long term variations in emission are caused by filament voltage changes and heat build-up in the gun area, generally a function of how many hours a display tube has been operating. Dynamic heating of each gun depends on the ratio of gun currents drawn to provide the colored picture being instantaneously presented. For example, a long persisting mostly red field causes red gun current almost exclusively, thereby causing an unbalanced heating of the red cathode, which changes its emission characteristics to a different degree than the other cathodes, this change remaining until relative cooling occurs.
Cathode thermal current, I th , is represented by the Dushman equation: I th = SA 0 T 2 e - b s /T amperes
where S and A 0 are constants and b 0 = Dushman constant ≉ 11,600° for an oxide coated cathode.
The derivative of the natural logarithm of this equation gives the change in emission with respect to temperature change: dI th /I th = 2 + (b 0 /t ) (dT/T)
the temperature of the CRT cathode is approximately 1,160° K, which yields (dI th /I th) = 12dT/T
typical ambient temperature variations, such as in a display monitor, are about 40° C, so that the net change of gun current is of the order of 12 . 40/1,160 ≉ 40%
Therefore, a 1° C change in cathode temperature yields about a 1% change in gun current, if the gun is fixed bias and not near cut-off.
It is an object of the present invention to provide a stabilizing system which overcomes the problems set forth.
SUMMARY OF THE INVENTION
The invention pertains to a video display apparatus which includes a cathode ray tube having at least one electron gun and means for deriving a source of excitatory voltage signal representative of picture information. In accordance with the invention there is provided a system for stabilizing the display intensity attributable to the electron gun comprising means coupling the source of voltage signal to the gun for generating a stabilized current in the gun, the level of current in the gun being substantially independent of variations in the operating characteristics of the gun.
In the preferred embodiment of the invention a test signal is periodically applied at a predetermined level to the electron gun. A sampler samples the beam current of the gun during the test signal and generates a correction signal in accordance with the sample value. The correction signal is applied to the gun so as to stabilize its reference current level.
Further features and advantages of the invention will become more readily apparent from the following detailed description when taken in conjunction with the accompanying drawings.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a simplified diagram of a color television display kinescope;
FIGS. 2A, 2B and 2C are block diagrams of embodiments of the invention which utilize periodically applied test signals;
FIG. 3 is a schematic representation of an embodiment of the invention which employs a "constant current" technique;
FIG. 4 is a schematic representation of another embodiment of the invention employing a differential amplifier; and
FIG. 5 is a schematic representation of another embodiment of the invention employing direct cathode temperature sensing and heater control.
DESCRIPTION OF THE PREFERRED EMBODIMENTS
Referring to FIG. 1, there is shown a simplified diagram of a color television display cathode ray tube or kinescope 10 as driven by excitatory video voltage signals designated R, G and B, these signals having typical ranges of about 100 volts peak-to-peak. The kinescope 10 has three electron guns, each including a cathode and associated grids. For clarity of illustration, only one of the three guns, designated by reference numeral 11, is represented in some detail, but it will be appreciated that two other complete electron guns (indicated in the Figure by only the two dashed cathodes coupled to the G and B inputs) are normally provided and are substantially identical to the gun 11. Hereinafter, and in the description of the embodiments of the invention, the circuitry associated with only one electron gun in a given kinescope will be described for illustrative clarity, but it will be understood that if the kinescope has two or more guns, similar circuitry can be employed in conjunction with the remaining guns. The electron gun 11 comprises a cathode 21 and first, second and third grids, 22, 23, and 24, which are sometimes designated as "grid 1," "grid 2," and "grid 3," or as the "control electrode," the "accelerating electrode," and the "focusing electrode," respectively. Generated electrons impinge on an anode 25 near display screen 26 which is coated with an electron-sensitive phosphor as is conventional in the art. Typical voltages applied to the cathode, grid 1, grid 2, grid 3 and the anode are about 235 volts, 150 volts, 700 volts, 5000 volts and 25,000 volts, respectively. In alternate modes of operation, the excitatory voltage input signal may be applied to a control grid with the remaining grid and cathode voltages being set at appropriate values.
Referring to FIG. 2A, there is shown an embodiment of the invention which comprises a system for stabilizing the display intensity or "color temperature" of the electron guns in a kinescope 30. An excitatory voltage signal at input terminal 31, which may be the R, B or G signal in a color system or the luminance signal in a black and white system, is coupled through an adder 41 and DC restorer circuit 42 to grid 1 of the kinescope 30. Cathode 43 is coupled through a resistor R 1 to ground reference. A bias voltage is applied to grid 2 via a voltage amplifier 44 which receives a signal on a line 58A which determines the level of the bias voltage applied to grid 2. Suitable focus and anode voltages are applied to grid 3 and the anode from sources not shown. The vertical and horizontal synchronizing signals of the composite television signal, available in the television receiver, are applied to a line counter 51 which is adapted to count horizontal scanlines of the television field and to be reset to zero at the end of each television field. The counter generates a first output on a line 51A during the scanlines 15-17 of each television field and a signal on line 51B during lines 18-20 of each television field, all of the lines 15-20 occuring during the vertical blanking period. The signal on line 51A enables a gate 55 and also enables a sample-and-hold circuit 56. The signal on line 51B enables a gate 57 and a sample-and-hold circuit 58. The gates 55 and 57 respectively receive voltages at reference "black level" and "white level." The outputs of gates 55 and 57 are coupled over lines 55A and 57A, respectively, to inputs of the adder 41.
Operation of the system of FIG. 2A is as follows: During lines 15-17 of the vertical blanking interval the gate 55 is enabled so that black level voltage is coupled through adder 41 and circuit 42 to grid 1. With this voltage applied to grid 1 the cathode current should ideally have a certain nominal value that does not vary with the tube life or cathode temperature but, as indicated above in the Background, this is not generally the case in actual practice. The actual cathode current is sampled across resistor R 1 , and a voltage representative of this current is coupled to the sample-and-hold circuit 56 which is enabled to sample the voltage across resistor R 1 during the lines 15-17. The circuit 56 holds the sampled voltage through the subsequent video field and couples the held voltage to circuit 42 via line 56A, this voltage serving to adjust the DC reference level of the output of circuit 42. In this manner, the voltage on line 56A controls the bias level at grid 1 so as to correct for any variations in the cathode current at nominal black level. Thus, for example, if at some point in operation the cathode current for a black level input voltage is lower than its nominal value, the voltage drop across sampling resistor R 1 will also be low. This will decrease the output of sample-and-hold circuit 56 fed to circuit 42 which, in turn, will cause the bias level at grid 1 to decrease (typically, to a less negative value with respect to the cathode). A lesser negative bias level on the control grid 1 will, in turn, cause a proportionate increase in the electron current flowing from cathode 43; the desired result.
Similarly, during lines 18-20 of the vertical blanking period white level voltage is applied via adder 41 and amplifier 42 to grid 1, and during this time the cathode current is sampled by circuit 58 which is enabled to sample by the signal on line 51B. During the remainder of the television field, the bias voltage applied to grid 2, via voltage amplifier 33, is a function of the voltage which had been sampled by circuit 58. For example, in an instance where the cathode current sensed at a white level voltage input is lower than the nominal value, the resultant low voltage sampled by circuit 58 will cause the grid 2 accelerating voltage to decrease. This causes the sampled voltage at black level to appear too negative (when next sampled during the succeeding vertical blanking interval) which, in turn, results in a decrease in grid bias by the black level circuit causing the desired increase in beam current over prior conditions, as previously described.
The embodiment of FIG. 2B is similar to that of FIG. 2A except that the output of sample-and-hold circuit 58 (which is a measure of the sampled white level current) is coupled to an analog multiplier circuit 59, which is in series with DC restorer circuit 52. In this embodiment, corrections resulting from both the white level and black level measurements are achieved via grid 1, with operation otherwise being substantially as described above.
In the embodiment of FIG. 2C the electron gun is driven by application of the video signal to the cathode 43 via a complementary emitter-follower 120 which comprises NPN transistor 121 and PNP transistor 122. (The system to the left of blocks 42 and 56 is the same as in FIG. 2B). The transistor emitters are coupled to the cathode 43 of kinescope 30 and the transistor bases receive the video signal from DC restorer circuit 42. The collector of transistor 121 is coupled to a suitable bias voltage, e.g., 150 volts, and the collector of transistor 122 is coupled to ground reference potential through sampling resistor R 1 . In operation, during the lines 15-20 the test signals are applied via circuit 42 and cathode 43 is driven while the cathode current is sampled by resistor R 1 , a typical value for which is 1K ohm. Transistor 122 is "on" during the white level test signal (output of circuit 42 about 25 volts) and the black level test signal (output of circuit 42 about 125 volts), and the gun current-representative voltages sampled across resistor R 1 are coupled to the appropriate sample-and-hold circuits as previously described. During the active portion of the television field the analog multiplier 59 and DC restorer circuit 42 apply appropriate corrections, with transistor 122 normally "on." During rapid lighter-to-darker transitions of the video signal the transistor 121 turns momentarily "on" and the tube capacitance and stray capacitance (collectively represented by C in the Figure can be thought of as charging. Diode D 1 prevents inordinate voltage drops across R 1 during the active picture area when R 1 is not used for sampling.
In the embodiment of FIG. 3 the video voltage signal at terminal 61 is coupled to cathode 71 of a kinescope 75 by the parallel combination of capacitor 62 and amplifier 63 in a series with resistor R 2 . Amplifier 63 comprises transistors 64 and 65 and has a voltage gain of about 5 and an output capability of about 500 volts. The resistor R 2 is selected to be substantially greater than the input impedance of the cathode 71 and preferably has a resistance at least five times higher than the cathode resistance. Since the effective cathode resistance is the inverse of the gun transconductance (about 8.6 micromhos), a suitable value for R 2 is of the order of 600K ohms. Accordingly, the amplifier 63 in conjunction with resistor R 2 operates as a so-called "constant current" source, which effectively transforms the voltage signal at terminal signal to a current source input to the cathode 71, this current source input being relatively insensitive to variations in the kinescope characteristics. Since normal wiring capacitance and electron gun interelectrode capacitance render high frequency response impractical i
n a high impedance amplifier drive, the higher frequency portions of the video signal are shunted across the amplifier by capacitor 62 which may have a typical value of about 0.05 microfarads. The higher frequency signals arrive at substantially the same relative level as the low frequencies, thereby preserving their relationship. This is because the lower frequency signals are amplified by a factor of 5 and then undergo a one-fifth loss by virtue of the voltage divider action of resistor R 2 and the cathode impedance. 
FIG. 4 shows a further embodiment wherein the video voltage signal at an input terminal 81 is applied to one input of a differential amplifier comprising transistors 82, 83 and 84. The output stage 84 drives the grid 1 electrode of kinescope 90 through series peaking inductor L 1 and shunt peaking inductor L 2 . The cathode 91 of kinescope 90 is coupled to ground reference potential through resistor R 3 which is used to continuously monitor the cathode current, the line 89 coupling a voltage representative of the cathode current to the other input of the differential amplifier; viz., the base of transistor 83. In operation, the voltage developed across resistor R 3 is proportional to the cathode current. This voltage, for a stable transconductance, should be in a stable relationship with respect to input voltage at terminal 81, and R 3 is selected empirically at a value, typically about 2K ohms, which generates a sample voltage nominally equal to the input voltage at terminal 81. When a deviation exists between the inputs to transistors 82 and 83, the output of the differential amplifier adjusts up or down to correct for the difference, thereby adjusting control of the drive to grid 1 and correcting for drifts in the kinescope transconductance.
A characteristic of the circuit of FIG. 4 is that it linearizes the electron gun transfer function which normally is non-linear, the non-linear function conventionally being known as "gamma" of the kinescope. Television video signals are conventionally precorrected for the gamma of the kinescope. In a color kinescope the gamma may be different for each gun, making it difficult to match the effective light output attributable to each gun over the grey scale; a problem known as "tracking" in the prior art. The present invention allows use of an inverse gamma circuit (which eliminates the precorrection in the conventional television signal) and the linearized gun transfer functions reduce tracking problems.
PHILIPS 22C850 /88Z " Tiepolo" CHASSIS K12 Television display apparatus including a beam current clamping control circuit:
A beam current clamping control circuit in which in case of a too large deviation of the desired value of the field frequency measured beam current an accelerated correction of this deviation is effected.
1. Television display apparatus including a beam current clamping control circuit having a beam current measuring circuit operable by a pulse generator and coupled to a television display tube for measuring, during at least part of a line scan time of a field blanking time a beam current reference level to be corrected, a level insertion circuit for inserting the reference level during the measuring time into a video signal to be applied to the television display tube and a level correction circuit coupled to an output of the measuring circuit and to a control electrode of the television display tube, the measuring circuit including a threshold circuit and a storage circuit, characterized in that the threshold circuit is a circuit which applies a signal to an output thereof when a too large beam current occurs, said output being coupled to an operation signal input of a circuit means for extending the charge correction time of the storage circuit per field period.
2. Television display apparatus as claimed in claim 1, characterized in that the circuit extending the charge correction time is a circuit means for increasing the measuring frequency, whereby a greater number of measurements per unit time is effected.
3. Television display apparatus as claimed in claim 1, characterized in that the circuit extending the charge correction time is a change-over switch a first input of which is coupled to an input of the measuring circuit, a second input is coupled to a beam current independent charging circuit coupled to a supply source and an output is coupled to the storage circuit.
4. Television display apparatus as claimed in claim 1, characterized in that the pulse generator includes a counting circuit having a feedback which can be switched on by the threshold circuit.
5. A beam current clamping control circuit for a television display tube comprising a beam current measuring circuit means for measuring during at least part of a line scan time of a field blanking time a beam current reference level to be corrected, a pulse generator coupled to said measuring circuit, a level insertion circuit means coupled to said pulse generator for inserting the reference level during the measuring time into a video signal to be applied to the television display tube, a level correction circuit coupled to an output of the measuring circuit, the measuring circuit including a threshold circuit means for coupling to said tube for applying a signal to an output thereof when a beam current above a selected value occurs, a storage circuit, and a circuit means for extending the charge correction time of the storage circuit per field period having an input coupled to said threshold circuit output and an output coupled to said storage circuit.
Description:
The invention relates to television display apparatus including a beam current clamping control circuit comprising a beam current measuring circuit coupled to a television display tube and operable by a pulse generator for measuring during at least part of a line scan time of a field blanking period a beam current reference level to be corrected, a level insertion circuit for inserting during the measuring time the reference level into a video signal to be applied to the television display tube, and a level correction circuit coupled to an output of the measuring circuit and to a control electrode of the television display tube, the measuring circuit including a threshold circuit and a storage circuit.
Television display apparatus of this type is known from U.S. Patent No. 3,562,409 in which the threshold circuit ensures that in case of a too large deviation of the beam current relative to the desired value the drive of the storage circuit is limited.
It is an object of the invention to provide improved television display apparatus of the kind described in the preamble which is characterized in that the threshold circuit is a circuit which in case of occurrence of a too large beam current applies a signal to an output thereof which output is coupled to an operation signal input of a circuit extending the charge corrections time of the storage circuit per field period.
Due to the limiting circuit in the circuit arrangement described in the abovementioned U.S. Patent Specification, in combination with the short measuring time available it could occur in the known circuit arrangement that, for example, after switching on the apparatus a very brightly lit picture occurred for several seconds. The storage circuit must include a capacitor having a fairly large capacitance the charge of which must be corrected in the known circuit with a current limited by the threshold circuit during the measuring time.
As a result of the step according to the invention the threshold circuit does not serve to limit the current intensity during the correction of the charge but to activate a circuit in case of a too large deviation of the nominal beam current, which circuit accelerates a feedback of the beam current to the nominal value by realising an extended charge correction time for the storage circuit. Consequently a feedback of the beam current to a level which is not troublesome can be obtained in a very quick manner without the current intensity in circuit elements correcting the charge of the storage circuit becoming too large.
The extended charge correction time of the storage circuit can be obtained by applying a correction charge from a current source which is independent of the measuring value under the influence of the threshold circuit during a time which is considerably longer than the measuring time, or by increasing the measuring frequency so that a current source which is dependent on the measuring value can recharge the storage circuit more frequently per unit of time.
The invention will now be described with reference to the drawing and some embodiments.
In the drawing:
FIG. 1 illustrates by way of a block-schematic diagram a television display apparatus according to the invention.
FIG. 2 illustrates by way of a block-schematic diagram a further elaboration of a television display apparatus according to the invention,
FIG. 3 likewise illustrates by way of a block-schematic diagram a further embodiment of a television display apparatus according to the invention,
FIG. 4 illustrates partly by way of a block-schematic diagram and partly by way of a principle circuit diagram a colour television display apparatus according to the invention including a sequential measuring circuit and
FIG. 5 shows a number of wave forms as may occur in the circuit of FIG. 4.
In FIG. 1 a video signal is applied to an input 1 of a level insertion circuit 3 which signal is provided with a reference level every time during some line times of a field some time after a field flyback with the aid of a signal obtained from an output 7 of a pulse generator 9 and applied to a further input 5. The video signal with the reference level is passed from an output 11 of the level insertion circuit 3 to an input 13 of a beam current level correction circuit 15. A level correction quantity obtained from an output 19 of a measuring circuit 21 is applied to a further input 17 of the beam current level correction circuit 15. A video signal having a reference level and an added level correction quantity is obtained from an output 23 and is applied to an input 25 of an amplifier 27 an output 29 of which is connected to the wehnelt electrode of a television display tube 31. A cathode of the television display tube 31 is connected to an input 33 of the measuring circuit 21.
The measuring circuit 21 includes a gating circuit 35 connected to the input 33 which applies a quantity dependent on the beam current to an output 37 during the occurrence of the reference level. The gating circuit 35 is arranged with the aid of a gating signal applied to a gating signal input 39 and originating from an output 41 of the pulse generator 9.
The quantity dependent on the beam current produced during the occurrence of the reference level is applied from the output 37 of the gating circuit 35 to an input 43 of a detection circuit 45 and to an input 47 of a threshold circuit 49.
An output 51 of the detection circuit 45 is connected in the normal operating condition to a storage capacitor 53 which constitutes a storage circuit with the circuit arrangement connected thereto and which has a time constant which is long relative to the field period. The beam current level correction quantity occurring at the output 19 of the measuring circuit 21 is then obtained from the storage capacitor 53 which quantity readjusts the beam current to a constant value.
The threshold circuit 49 has an output 55 which is connected to an operation signal input 57 of a circuit 59 extending the charge correction time of the storage circuit. The circuit 59 extending the charge correction time includes a second storage capacitor 60 which can retain a voltage optionally provided by the threshold circuit for some time and a changeover switch 61 an operation signal input 63 of which is connected to the operation signal input 57 of the circuit 59 extending the recharge time. The changeover switch 61 has an input 67 which is connected to the output 51 of the detection circuit 45 and an input 69 which is connected through a resistor 71 to a positive supply voltage. An output 73 of the change-over switch is connected to the storage capacitor 53 and the output 19 of the measuring circuit 21.
In the normal operating condition the change-over switch 61 occupies the position shown and the above-described beam current control occurs.
When the beam current is very large during the measuring time, which means that the voltage at the storage capacitor 53 is very low, threshold circuit 49 applies a voltage to its output 55 which sets the change-over switch 61 to the position not shown and, dependent on the discharge time of the second storage capacitor 60, retains it in that position for some time, for example, several hundred line times. The resistor 71 then constitutes a current source independent of the measuring value which increases the voltage across the storage capacitor 53. During the next measurement the threshold circuit 49 determines whether this increase has been sufficient or not so that the change-over switch 61 can remain in the position shown or must be reset once more for some time.
The pulse generator 9 has an input 75 to which field frequency pulses derived from field synchronising or field flyback pulses are applied and an input 77 to which pulses derived from line synchronizing or line flyback pulses are applied. The input 75 is connected to an input 79 of a shift register 81 which in this case has four sections but of which the number of sections can of course be chosen arbitrarily. A line frequency pulse originating from the input 77 serves as a clock pulse for the shift register 81 into which, for example, at the end of the field flyback time a one is shifted which is shifted one section further at every subsequently occurring clock pulse. The last two sections of the shift register have outputs 83 and 85 which are connected to inputs 87 and 89, respectively, of a gating circuit 91 which has a further input 93 connected to the input 77. The gating circuit 91 supplies signals to two outputs 95 and 97 which signals are applied to the outputs 7 and 41 of the pulse generator 9 and which have the functions described hereinbefore.
In FIG. 2 corresponding components have the same reference numerals as those in FIG. 1. For the description of this Figure reference is made to FIG. 1.
The circuit arrangement of FIG. 2 differs from that of FIG. 1 in that the second storage capacitor 60 of the circuit 59 extending the recharge time of the first storage capacitor 53 is replaced by a trigger circuit 99 a set input of which is the input 57 which is connected to the output 55 of the threshold circuit and a reset input 101 of which is connected to the input 75 of the pulse generator 9 to which the field frequency pulses are applied. An output 103 of the trigger circuit is connected to the input 63 of the change-over switch 61 and to an operation signal input 105 of a second change-over switch 107 in the pulse generator 9. An input 109 of the second change-over switch 107 is connected to an output 111 of the shift register 81 and an input 113 is connected to the input 75 for field frequency pulses from the pulse generator 9. An output 115 of the change-over switch 107 is connected to the input 79 of the shift register 81.
In the position shown of the change-over switches 61, 107 the operation of the circuit arrangement is the same as in the case of FIG. 1. When a too high beam current is detected by the threshold circuit 49, the trigger circuit 99 will be set and the change-over switches 61 and 107 will be brought to the position not shown. The storage capacitor 53 will then be charged through the resistor 71 likewise as in the case of FIG. 1. The change-over switch 107 connects the output 111 of the shift register 81 to its input 79 so that always a one circulates in the shift register 81 and a continuous measurement of the beam current is effected until the threshold circuit 49 no longer detects a too high beam current and due to the output voltage dropping out or due to the next field frequency pulse at the reset input 101 the trigger circuit 99 is reset and the normal state is restored.
In FIG. 3 corresponding components have the same reference numerals as those in FIGS. 1 and 2 and reference is made to the description associated with the relevent Figures.
The circuit arrangement differs from that of FIG. 2 by the absence of the first change-over switch 61. Consequently, when a too high beam current occurs the normal control loop is maintained and only due to the second change-over switch 107 the shift register 81 is enabled so that a strongly increased measuring frequency and an accelerated feedback occurs through the normal control loop to the normal state. The recharge time of the storage capacitor 53 is in this case extended because per field period a larger number of measurements and associated recharges of this capacitor is effected.
FIG. 4 shows a colour television receiver including a circuit arrangement according to the invention in which the principle as described with reference to FIG. 2 is used. The circuit arrangement has a colour television display apparatus having a sequential beam current measuring circuit whose principle is described in prior application Ser. No. 402,159, filed Oct. 1, 1973, now abandoned.
When a colour television signal is applied to an input 201 of a high frequency-intermediate frequency and detection section 203, a luminance signal Y is produced at an output 205, a chrominance signal Chr is produced at an output 207 and a synchronizing signal S is produced at an output 209.
The synchronizing signal S is applied to an input 211 of a time base generator 213 connected to the output 209. Deflection currents for a television display tube 221 are obtained at two outputs 215 and 217 and an EHT for the supply of the display tube 221 is obtained at an output 223.
An output 227 of the time base generator 213 applies line flyback pulses to an input 229 and an output 231 applies field flyback pulses to an input 233 of a pulse generator 235. These pulses are shown in FIG. 5 by the waveforms 529 and 533, respectively.
Furthermore FIG. 5 shows a number of waveforms 537, 509, 514, 544, 548, 520, 539, 541, 543, 547, 549, 551, 553, 656, 555, 565 and 563 and a number of instants t 1 , t 1 +T 1 , t 3 to t 10 , t 10a , and t 10 + T 2 , t 10 + 2T 2 , T 11 , t 11 + T 1 , t 11 + 2 T 1 , and t 12 which are important for explaining the operation of the circuit arrangement. The waveforms are not shown to scale.
The pulse generator 235 has a number of outputs 237, 239, 241, 243, 244, 245, 247, 249, 251, 253, 255 and an input 256 at which the waveforms 537, 539, 541, 543, 545, 547, 549, 551, 553, 555 and 556, respectively, are present.
The output 255 of the pulse generator 235 is connected to an input 257 of a level insertion circuit 259 an input 261 of which is connected to the output 205 of the section 201 and receives the luminance signal Y therefrom. The level insertion circuit 259 has an output 263 at which the modified luminance signal denoted by the waveform 563 of FIG. 5 is produced which includes a reference level 521 during three line times t 5 -t 6 , t 7 -t 8 , t 9 -t 10 at the commencement of the field, which level is inserted with the aid of the wave-form 555 applied to the input 257.
The waveform 565 is then produced at a further output 265 of the level insertion circuit 259, which waveform is applied to an input 267 of a chrominance signal amplifier 269 for suppressing the chrominance signal applied to an input 271 connected to the output 207 of the section 201.
The output 263 of the level insertion circuit 259 is connected to inputs 285, 287 and 289 of suppression and level correction circuits 291, 293 and 295, respectively, inputs 297, 299 and 301 of which are connected to outputs 247, 249 and 251 of the pulse generator 235 and inputs 303, 305 and 307 of which are connected to outputs 309, 311 and 313, respectively, of a demodulator and matrix circuit 315 to whose input 317 a chrominance signal originating from an output 319 of the chrominance signal amplifier 219 is applied.
The demodulator and matrix circuit applies a (B-Y), (G-Y) and (R-Y) colour difference signal to its outputs 309, 311 and 313, respectively. These colour difference signals are combined in the suppression and level correction circuits 291, 293 and 295 with the modified luminance signal applied to the inputs 285, 287 and 289 to form colour signals R, G and B, while as a result of the signals with the waveforms 547, 549, 551 at the inputs 297, 299 and 301 the reference level 521 originating from the luminance signal is suppressed in a special sequence. In this example a sequence is maintained in which only a reference level is left in the R-signal from t 5 to t 6 , in the G signal from t 7 to t 8 and in the B signal from t 9 to t 10 .
The suppression and level correction circuits 291, 293 and 295 furthermore have outputs 339, 341, 343 connected to the wehnelt electrode of the red, blue and green guns of the display tube 221 and inputs 345, 347 and 349 connected to the output 237 of the pulse generator 235 for receiving a suppression signal of the waveform 537 which prevent a signal supply to the display tube 221 during the time base flyback times. Inputs 351, 353 and 355 are furthermore connected to outputs 357, 359 and 361, respectively, of a measuring circuit 362 which outputs are connected to storage capacitors 363, 365 and 367, respectively, whose other ends are connected to earth. The storage capacitors 363, 365 and 367 are connected to switches 377, 378 and 379 respectively, operation signal inputs 383, 384 and 385 of which are connected to outputs 239, 241 and 243, respectively, of the pulse generator 235.
The switch 377 conducts from t 5 to t 6 , the switch 378 conducts from t 7 to t 8 and the switch 379 conducts from t 9 to t 10 . Each of these switches introduces a level applied to an input 381 thereof into the relevant storage capacitors 363, 365, 367.
The cathodes of the television display tube 221 are interconnected and are connected to an input 389 of the measuring circuit 362. Sequentially by each cathode voltages is generated across a resistor 393 connected to the input 389 which resistor is shunted by a diode 394 and is connected at its other end to a voltage of + 130 Volt during the said periods. This generated voltage produces a corresponding voltage across the storage capacitors 363, 365 and 367 which is retained during the next field flyback time and is passed on for beam current control to the inputs 351, 353 and 355 of the suppression and level correction circuits 291, 293 and 295. The two cathodes which are not measured then do not convey any beam current.
An input 391 of the measuring circuit is connected to the output 245 and an input 392 is connected to the output 253 of the pulse generator 235.
The screen grids of the guns of the television display tube 221 are each connected to an adjusting point of potentiometers 395, 396 and 397 between + 130 Volt and a higher voltage ++.
A signal occurring at the cathodes of the television display tube 221 produces a voltage across the resistor 393 which voltage is limited by the diode 394 blocked during the measuring periods. This voltage is applied to the base of a pnp transistor 401 whose emitter is connected to the other end of the resistor 393. The collector of the transistor 401 is connected to earth through a resistor 403, connected to earth through a capacitor 405 for high frequencies and has a negative feedback to the base through a resistor 407.
A signal amplified by the transistor 401 is applied through a series arrangement of a capacitor 409, a resistor 410 and a resistor 411 to the base of an npn transistor 413. The collector of an npn transistor 415 serving as a clamping switch is connected between the resistors 410 and 411. The emitter is connected to a clamping voltage V kl and the base is connected to the input 391 of the measuring circuit 362.
As a result of the waveform applied to the input 391, which waveform is the inverse form of waveform 544, the transistor 415 is cut off during the occurrence of the signal produced by the reference level 521. During the preceding line periods the capacitor 511 has reached a constant charge condition (clamped) through the then conducting transistor 415 because both the luminance signal and the chrominance signal are suppressed and the cathodes of the display tube 221 do not convey any current.
During the period t 4 -t 10 signals caused by the reference level 521 occur successively during the periods t 5 -t 6 , t 7 - t 8 and t 9 - t 10 at the successive cathodes of the display tube 221 which signals are applied in an amplified manner to the base of the transistor 413 and which furthermore appear at the emitter of an npn transistor 412 connected to the base of the transistor 413.
When the signals have an amplitude which is not too large they are passed on by the transmitter 413 through its emitter and a resistor 423 to the base of an npn transistor 425.
When the signals at the base of the transistor 413 have a too large amplitude the emitter of the transistor 412 will become more negative than its base connected to a threshold voltage V dr and this transistor 412 will start conducting and hence operate a set input of a trigger circuit 414 connected to its collector which then provides a positive voltage for its output 416. This positive voltage (waveform 556) is applied to an output 418 which is connected to the input 256 of the pulse generator 235 and which causes a variation to a free-running state which will be described hereinafter, while this voltage also controls the base of an npn transistor 422 through a resistor 420 and the emitter of this transistor is connected to earth while the collector is connected to the base of the transistor 413. The base of this transistor 413 then is no longer controlled because the transistor 422 acts as a short circuit for its input signal. The trigger circuit 414 furthermore has a reset input 424 to which a field flyback pulse (waveform 533) originating from the output 231 of the time base generator 213 is applied so that the trigger circuit 414 is reset every time at the commencement of the field flyback.
In the state in which the beam current is not too large the transistors 412 and 422 do not conduct and the signal is passed on through the transistor 413 to the base of the transistor 425 whose emitter is connected to the collector of an npn transistor 427 whose emitter is connected to earth and whose base is connected through a resistor 428 to the input 392 of the measuring circuit. The collector of the transistor 425 is connected through a resistor 430 to the base of an npn transistor 433 whose emitter is connected through a resistor 435 to the input 381 of the sequence switches 377, 378 and 379.
The collector of the transistor 425 is connected through a resistor 432 to the emitter of an npn transistor 434 arranged as an emitter follower whose base is connected through a resistor 436 to an input 438 of the measuring circuit 362 which is connected to the output 244 of the pulse generator 235 at which the waveform 544 occurs. This transistor 434 conducts during the measuring period and furthermore every time during a number of line times in the field scan time when the pulse generator 235 has acquired a free-running state (waveform 544) due to a too high beam current with the aid of the trigger circuit 414.
The transistor 427 conducts due to the waveform 553 during the measuring periods so that in the periods t 5 -t 6 , t 7 -t 8 , t 9 -t 10 the emitter of the transistor 425 is connected to earth through the transistor 427 and the transistor 425 acts as an amplifier and passes on a signal to the base of the transistor 433 and recharges through its emitter the storage capacitors 363, 365 and 367 during the corresponding periods t 5 -t 6 , t 7 -t 8 and t 9 -t 10 , respectively.
In the state of the circuit arrangement where a too large beam current has been detected and where the trigger circuit 414 is set, the transistor 425 does not conduct due to the short circuit of the base of the transistor 413 so that then the storage capacitors 363, 365 and 367 are recharged by the emitter follower 434, 433 every time during the periods when the waveform 544 is positive. This is effected a large number of times during field scan periods following the set of the trigger circuit 414 so that the storage capacitors are charged quickly to a higher voltage to reduce the beam current through the control loops as quickly as possible again.
The operation of a control loop will now be described. This control loop not only includes the television dislay tube 221 and the measuring circuit 362 but also the suppression and level correction circuit 295.
A modified luminance signal Y m according to the waveform 563 is applied to the input 289 in which signal the reference level 521 is present during the periods t 5 -t 6 , t 7 -t 8 and t 9 -t 10 and which level is suppressed from t 1 to t 5 . The input 289 is connected to the base of an npn transistor 601 whose emitter is connected to earth through a resistor 603 and whose collector is connected through a resistor 605 to the emitter of an npn transistor 607. The collector of the transistor 607 is connected to a positive voltage. The base is connected to the input 307 and receives a red colour difference signal -(R-Y) which is suppressed from t 1 to t 10a .
Furthermore the emitter of the transistor 601 is connected to the emitter of an npn transistor 609 whose collector is connected to a positive voltage and whose base is connected to the input 301.
A voltage having the waveform 547 is applied to the input 301 so that the transistor 609 conducts during the periods t 7 -t 8 and t 9 -t 10 and thus cuts off the transistor 601 so that also the transistor 607 does not convey current and the signal at the collector of the transistor 601 is suppressed.
When the transistor 609 does not conduct a signal is generated at the collector of the transistor 601 which signal is a combination of the -(R-Y) signal supplied through the emitter follower 607 and the -Y m signal amplified through the transistor 601. This combination is a -R signal in which a level corresponding to the level 521 is present during the period t 5 -t 6 and which is suppressed in the line periods t 1 -t 5 and t 6 -t 10a . This signal is applied to the base of an npn transistor 611 whose collector is connected to a positive voltage and whose emitter is connected through a resistor 613 to the collector of an npn transistor 615 whose emitter is connected to earth through a resistor 617 and whose base is connected to a reference voltage of 1.4 V.
The emitter of the transistor 615 is furthermore connected to the emitters of two npn transistors 619 and 621 whose collectors are connected to a positive supply voltage.
A level correction voltage originating from the capacitor 367 and applied to the input 355 is transferred to the base of the transistor 619 through an npn transistor 623 arranged as an emitter follower while a suppression signal having the waveform 537 applied to the input 349 is passed on to the base of the transistor 621. The latter signal ensures the common suppression of the beam current during the line and field flyback periods. Due to the fact that the otherwise cut-off transistor 621 is blocked during these flyback periods, the transistor 615 is cut off and no signal is passed to the collector of this transistor because then the emitter circuit of the transistor 611 is interrupted.
The level correction voltage applied to the input 355 is then applied through the emitters of the transistors 623 and 619 to the emitter of the transistor 615 and influences the direct current through the resistor 613 provided by the transistor 615 and hence of the direct current level of the -R signal passed through the emitter of the transistor 611 to the collector of the transistor 615.
The collector of the transistor 615 is connected to the base of an npn transistor 625 whose collector is connected to a positive voltage and whose emitter is connected through an adjustable resistor 629 to the base of an npn transistor 631. The emitter of the transistor 631 is connected to earth and the collector is connected to the emitter of an npn transistor 633 whose collector is connected through a resistor 635 to a positive supply voltage of +130 V and whose base is adjusted to a bias voltage of +5 V. The base of the transistor 631 receives a negative feedback voltage from the collector of the transistor 633 through a potential divider 637, 639 to a negative voltage.
A red colour signal amplified by the transistors 631 and 633 is obtained from the collector of the transistor 633 and is applied through the output 343 to the wehnelt electrode of the red gun of the display tube 221. This signal includes the correction level originating from the capacitor 363 with which the beam current in the red gun is adjusted to a desired value.
The amplification of the circuit arrangement is adjustable with the resistor 629 so as to perform for example, a white point correction. Due to the control loop such an adjustment has substantially no influence on the beam current which is produced by the reference level 521 so that the black level and hence the colour of dark picture parts does not change due to the adjustment.
The modified luminance signal Y m for the input 289 is obtained in the level insertion circuit 259.
A luminance signal is applied to the input 261 which signal is applied to an input 701 of an amplifier 703. Furthermore the amplifier 703 has an input 707 to which an adjustable direct voltage is applied with the aid of a potentiometer 709 which serves for luminance adjustment and an input 771 to which a suppression signal is applied as is shown by the waveform 555. The black level of the picture 520 in the waveform 563 is adjustable with the aid of the potentiometer 709 relative to the level occurring during the suppression periods. The waveform 563 is the luminance signal which occurs at the output 263 across an emitter resistor 713 of an npn transistor 715 arranged as an emitter follower whose base is connected to an output 705 of the amplifier 703.
A signal having the waveform 555 is applied to the input 257 of the level insertion circuit through a capacitor 717, which signal is passed at one end through a resistor 721 to the base of an npn transistor 723 and at the other end through a potential divider 725, 727 to the base of an npn transistor 729.
During the most positive parts of the waveform 555 a low voltage is produced across a collector resistor 731 of the transistor 723. The level of the signal during the periods t 5 -t 10 lies below the cut-off point of the transistor 727 so that this is not found back in the collector signal 565 of this transistor. As a result of the potential divider 725, 727 connected to the positive voltage the transistor 729 only reacts to the most negative parts of the signal 555 and the transistor 729 is cut off during the periods t 5 -t 6 , t 7 -t 8 and t 9 -t 10 so that then positive going square-wave voltages occur at the collector of this transistor.
The collector of the transistor 729 is connected through a resistor 733 to a potential divider including a resistor 735 and a series arrangement of a diode 736 and a resistor 737. The voltage at the wiper of this potential divider has a constant value of + 2.2 V which also occurs at the collector of the transistor 729 during the said periods. This voltage is passed on through a resistor 738 to the base of an npn transistor 739 whose collector is connected to a positive supply voltage and whose emitter is connected to the emitter of the transistor 715. During the periods t 5 -t 6 , t 7 -t 8 and t 9 -t 10 the emitter of the transistor 739 is brought to a voltage of +1.5 V due to its base voltage so that the transistor 715 is cut off and the reference level 521 is produced at the output 263. The rest of the period the transistor 739 is cut off and a signal is applied to the output 263 through the transistor 715.
In the waveform 563 arrows show that the black level 520 of the signal is adjustable relative to the constant reference level 521 which as a result of the said control corresponds to a constant beam current valve in the guns of the display tube 221.
The pulses for the various parts of the circuit arrangement are supplied by the pulse generator 235.
This generator has four trigger circuits 801, 803, 805 and 807. A signal which relative to the signal 533 at the input is delayed over a time T 1 is applied to a set input 809 of the trigger 801 from the input 233 through a delay circuit including a resistor 811 and a capacitor 813. This signal is shown in the waveform 509. Before this pulse appears the triggers 801, 803, 805 and 807 are assumed to be in the reset condition. At the instant t 1 + T 1 the trigger 801 is brought to the set condition. A signal represented by the waveform 514 then appears at an output 814 thereof which signal is applied to four AND gate circuits 817, 819, 821 and 823.
Furthermore an inverted set signal from the trigger 801 is applied to the gate 817 as well as a line frequency pulse signal originating from the input 229.
In addition a signal represented by the waveform 544 originating from an output 816 of the trigger 803 and a signal originating from an AND gate 817 are applied to the gate 819.
The gate 821 furthermore receives a signal having a waveform 518 from an output 818 of the trigger 805 and a signal originating from an AND gate 829.
Furthermore the gate 823 receives a signal having a waveform 520 from an output 820 of the trigger 807 and an inverted line frequency pulse signal originating from the input 229. An output of this gate 823 is connected through a delay circuit including a resistor 831 and a capacitor 833 to an AND gate 835 which also receives a line frequency pulse signal from the input 229. The output of this gate 835 is connected to the reset inputs of all four triggers. The gates 827 and 829 receive an inverted signal from the input 229 and an output signal from the triggers 805 and 807, respectively.
Furthermore a switchable feedback is present between the output 814 of the trigger 801 and the set input thereof, which feedback is only switched on when the beam current in the display tube 221 is too high. This feedback starts from the output 814 through a resistor 837 which is connected to the base of an npn transistor 839 whose emitter is connected to earth and whose collector is connected through a resistor 841 to the input 256 and is connected to the output of the trigger circuit 414, the base of a transistor 843 connected to the collector of the transistor 839 having its collector connected to a positive supply voltage and being connected through a resistor 845 connected to the emitter of this transistor 843 to the set input 809 of the trigger 801. Together with the capacitor 813 the resistor 845 constitutes a delay circuit having a time delay T 2 . The feedback is not present when the beam current is so low that the trigger circuit 414 is not set; the voltage at the collector of the transistor 839 is then low and the transistor 843 is cut off. This condition is assumed to be present at the commencement of the time axis in FIG. 5.
When the voltage at the input 256 becomes high due to the trigger 414 coming in the set condition the feedback will be switched on. The transistor 839 will then act as an inverter.
When at the instant t 1 + T 1 the trigger 801 is set, the trigger 803 is blocked at the same time by the inverted set signal from the trigger 801 applied to the gate 817. The triggers 803, 805 and 807 then remain in the reset condition. After the end of the set pulse from the trigger 801, i.e. after the instant t 3 = t 2 +T 1 the gate circuit 817 becomes conducting at the instant t 4 due to the next line frequency pulse and brings the trigger 803 in the set condition. The gate 819 then becomes conducting after the end of the line pulse under the influence of the gate 827 and applies a pulse through a delay network 845, 847 to an input of an AND gate 849 to which also the line frequency signals of the input 229 are applied. At the next line pulse at the instant t 6 the trigger 805 is set so that the gate 821 provides a pulse which is delayed by a network 851, 853 and is applied to an input of an AND gate 855 to which furthermore the line frequency pulse signal from the input 229 is applied. This gate 855 becomes conducting at the next line pulse at the instant t 8 under the influence of the gate 829 and brings the trigger 807 to the set condition.
A pulse is obtained from the output of the gate 823 which begins after termination of the line pulse and which is applied delayed through the network 831, 833 to the gate 835 which then passes the next line pulse at the instant t 10 so that the triggers 801, 803, 805 and 807 are reset.
When meanwhile the trigger circuit 414 has not reached the set condition during the time between t 5 and t 10 in which the beam currents in the display tube are measured, the triggers 801, 803, 805 and 807 remain in the reset condition until the next field pulse.
When the trigger circuit 414 changes to the set condition due to a too high beam current for example between t 5 and t 6 , the described feedback from the output 814 of the trigger 801 is provided at its set input 809 and a positive pulse will appear again at the instant t 10 + T 2 at the input 809 which pulse will bring the trigger 809 to the set condition again so that at an instant t 10 + 2T 2 the trigger 803 is set whereafter the triggers repeat the above-described conditions every time.
At the next field pulse at the instant t 11 , however, the trigger 414 is reset so that the feedback in the pulse generator 235 is interrupted and a set cycle of the triggers 801, 803, 805 and 807 already started is terminated at the instant t 12 when the gate 835 applies a line pulse to the reset input of the trigger 801 so that voltage at the output 814 remains equally low and all triggers are reset. After the line pulse drops out at its reset input the trigger 801 is brought immediately to the set condition so that at the next line pulse which lies after the instant which is T 1 later than the trailing edge of the field pulse 533 the trigger 803 is set again and three line times later everything is reset and remains reset when the beam current has become sufficiently low due to the repeated recharge of the capacitors 363, 365, 367 in the measuring circuit 362.
A number of gate circuits supplying the output signals from the pulse generator 235 are coupled to the outputs of the trigger circuits.
The output 239 is connected to an output of an AND gate 857 which has an input connected to the output 816 of the trigger 803, an inverted input which is connected to the output 818 of the trigger 805 and an inverted input which is connected to the input 229 of the pulse generator. The output of this gate then applies the waveform 539.
The output 241 is connected to an output of an AND gate 859 and input of which is connected to the output 818 of the trigger 805, an inverted input is connected to the output 820 of the trigger 807 and an inverted input is connected to the input 229 of the pulse generator 235. The AND gate 859 provides an output signal having the waveform 541.
The output 243 is connected to an output of an AND gate 861 an input of which is connected to the output 820 of the trigger 807 and an inverted input is connected to the input 229 of the pulse generator 235. The AND gate 861 provides an output signal having the waveform 543.
The output 244 is connected to the output 816 of the trigger 803 and the output 245 is connected to an inverted output of this trigger.
The output 247 is connected to an output of an OR gate 863, the output 249 is connected to an output of an OR gate 865 and the output 251 is connected to an output of an OR gate 867. The inputs of the gate 863 are connected to the outputs of the gates 861 and 859, the inputs of the gate 865 are connected to the outputs of the gates 861 and 857 and the inputs of the gate 867 are connected to the outputs of the gates 857 and 859. The gates 863, 865 and 867 supply the signals 547, 549 and 551, respectively.
The output 253 is connected to an output of an AND gate 869 an input of which is connected to the output 816 of the trigger 803 and an inverted input is connected to the input 219 of the pulse generator 235. The waveform 553 is supplied by this gate.
The output 237 is connected to an output of an OR gate 871 whose inputs are connected to the inputs 229 and 233 of the pulse generator. The waveform 537 is provided by the gate 871.
The output 255 is connected to an output of a gate and superimposition circuit 873 an input of which is connected to an inverted output of the trigger 801, an inverted input is connected to the output of the gate 871 and an inverted input is connected to the output of the gate 869. The gate and superimposition circuit 873 provides the waveform 555.
Since the latter waveform is applied through the capacitor 717 to the input 257 of the level insertion circuit 259, an unsatisfactory amplitude selection will take place when the trigger 801, 803, 805 and 807 run free for a long period so that a charge of the capacitors 363, 365, 367 in the measuring circuit 362 is effected independently of the beam current when it has exceeded a given maximum value as described in the foregoing. The level shift due to this capacitor 717 and the results thereof are not important for the understanding of the invention and are therefore not shown in the waveforms.
Although the control voltage in the described embodiments is applied to a wehnelt electrode of the display tube it may alternatively be applied to a different control electrode.
The insertion of the reference level may of course alternatively be effected in a different manner such as, for example, by clamping the video signal on the control signal.
For switching over the described functions in case of a too large beam current value a schmitt trigger or a monostable multivibrator may be used if desired instead of a bistable trigger circuit having a set and reset, input, or an output signal from the threshold circuit may be used.
The type of measuring circuit is not important for using the step according to the invention. Instead of a sequence measuring circuit it is alternatively possible to use a simultaneous measuring circuit in which all beam currents are simultaneously measured in the case of a multigun display tube. Also the input circuit of the measuring circuit may of course be adapted as desired.
If desired other counting circuits may be used instead of a shift register.
Television display apparatus of this type is known from U.S. Patent No. 3,562,409 in which the threshold circuit ensures that in case of a too large deviation of the beam current relative to the desired value the drive of the storage circuit is limited.
It is an object of the invention to provide improved television display apparatus of the kind described in the preamble which is characterized in that the threshold circuit is a circuit which in case of occurrence of a too large beam current applies a signal to an output thereof which output is coupled to an operation signal input of a circuit extending the charge corrections time of the storage circuit per field period.
Due to the limiting circuit in the circuit arrangement described in the abovementioned U.S. Patent Specification, in combination with the short measuring time available it could occur in the known circuit arrangement that, for example, after switching on the apparatus a very brightly lit picture occurred for several seconds. The storage circuit must include a capacitor having a fairly large capacitance the charge of which must be corrected in the known circuit with a current limited by the threshold circuit during the measuring time.
As a result of the step according to the invention the threshold circuit does not serve to limit the current intensity during the correction of the charge but to activate a circuit in case of a too large deviation of the nominal beam current, which circuit accelerates a feedback of the beam current to the nominal value by realising an extended charge correction time for the storage circuit. Consequently a feedback of the beam current to a level which is not troublesome can be obtained in a very quick manner without the current intensity in circuit elements correcting the charge of the storage circuit becoming too large.
The extended charge correction time of the storage circuit can be obtained by applying a correction charge from a current source which is independent of the measuring value under the influence of the threshold circuit during a time which is considerably longer than the measuring time, or by increasing the measuring frequency so that a current source which is dependent on the measuring value can recharge the storage circuit more frequently per unit of time.
The invention will now be described with reference to the drawing and some embodiments.
In the drawing:
FIG. 1 illustrates by way of a block-schematic diagram a television display apparatus according to the invention.
FIG. 2 illustrates by way of a block-schematic diagram a further elaboration of a television display apparatus according to the invention,
FIG. 3 likewise illustrates by way of a block-schematic diagram a further embodiment of a television display apparatus according to the invention,
FIG. 4 illustrates partly by way of a block-schematic diagram and partly by way of a principle circuit diagram a colour television display apparatus according to the invention including a sequential measuring circuit and
FIG. 5 shows a number of wave forms as may occur in the circuit of FIG. 4.
In FIG. 1 a video signal is applied to an input 1 of a level insertion circuit 3 which signal is provided with a reference level every time during some line times of a field some time after a field flyback with the aid of a signal obtained from an output 7 of a pulse generator 9 and applied to a further input 5. The video signal with the reference level is passed from an output 11 of the level insertion circuit 3 to an input 13 of a beam current level correction circuit 15. A level correction quantity obtained from an output 19 of a measuring circuit 21 is applied to a further input 17 of the beam current level correction circuit 15. A video signal having a reference level and an added level correction quantity is obtained from an output 23 and is applied to an input 25 of an amplifier 27 an output 29 of which is connected to the wehnelt electrode of a television display tube 31. A cathode of the television display tube 31 is connected to an input 33 of the measuring circuit 21.The measuring circuit 21 includes a gating circuit 35 connected to the input 33 which applies a quantity dependent on the beam current to an output 37 during the occurrence of the reference level. The gating circuit 35 is arranged with the aid of a gating signal applied to a gating signal input 39 and originating from an output 41 of the pulse generator 9.
The quantity dependent on the beam current produced during the occurrence of the reference level is applied from the output 37 of the gating circuit 35 to an input 43 of a detection circuit 45 and to an input 47 of a threshold circuit 49.An output 51 of the detection circuit 45 is connected in the normal operating condition to a storage capacitor 53 which constitutes a storage circuit with the circuit arrangement connected thereto and which has a time constant which is long relative to the field period. The beam current level correction quantity occurring at the output 19 of the measuring circuit 21 is then obtained from the storage capacitor 53 which quantity readjusts the beam current to a constant value.
The threshold circuit 49 has an output 55 which is connected to an operation signal input 57 of a circuit 59 extending the charge correction time of the storage circuit. The circuit 59 extending the charge correction time includes a second storage capacitor 60 which can retain a voltage optionally provided by the threshold circuit for some time and a changeover switch 61 an operation signal input 63 of which is connected to the operation signal input 57 of the circuit 59 extending the recharge time. The changeover switch 61 has an input 67 which is connected to the output 51 of the detection circuit 45 and an input 69 which is connected through a resistor 71 to a positive supply voltage. An output 73 of the change-over switch is connected to the storage capacitor 53 and the output 19 of the measuring circuit 21.
In the normal operating condition the change-over switch 61 occupies the position shown and the above-described beam current control occurs.
When the beam current is very large during the measuring time, which means that the voltage at the storage capacitor 53 is very low, threshold circuit 49 applies a voltage to its output 55 which sets the change-over switch 61 to the position not shown and, dependent on the discharge time of the second storage capacitor 60, retains it in that position for some time, for example, several hundred line times. The resistor 71 then constitutes a current source independent of the measuring value which increases the voltage across the storage capacitor 53. During the next measurement the threshold circuit 49 determines whether this increase has been sufficient or not so that the change-over switch 61 can remain in the position shown or must be reset once more for some time.
The pulse generator 9 has an input 75 to which field frequency pulses derived from field synchronising or field flyback pulses are applied and an input 77 to which pulses derived from line synchronizing or line flyback pulses are applied. The input 75 is connected to an input 79 of a shift register 81 which in this case has four sections but of which the number of sections can of course be chosen arbitrarily. A line frequency pulse originating from the input 77 serves as a clock pulse for the shift register 81 into which, for example, at the end of the field flyback time a one is shifted which is shifted one section further at every subsequently occurring clock pulse. The last two sections of the shift register have outputs 83 and 85 which are connected to inputs 87 and 89, respectively, of a gating circuit 91 which has a further input 93 connected to the input 77. The gating circuit 91 supplies signals to two outputs 95 and 97 which signals are applied to the outputs 7 and 41 of the pulse generator 9 and which have the functions described hereinbefore.
In FIG. 2 corresponding components have the same reference numerals as those in FIG. 1. For the description of this Figure reference is made to FIG. 1.The circuit arrangement of FIG. 2 differs from that of FIG. 1 in that the second storage capacitor 60 of the circuit 59 extending the recharge time of the first storage capacitor 53 is replaced by a trigger circuit 99 a set input of which is the input 57 which is connected to the output 55 of the threshold circuit and a reset input 101 of which is connected to the input 75 of the pulse generator 9 to which the field frequency pulses are applied. An output 103 of the trigger circuit is connected to the input 63 of the change-over switch 61 and to an operation signal input 105 of a second change-over switch 107 in the pulse generator 9. An input 109 of the second change-over switch 107 is connected to an output 111 of the shift register 81 and an input 113 is connected to the input 75 for field frequency pulses from the pulse generator 9. An output 115 of the change-over switch 107 is connected to the input 79 of the shift register 81.
In the position shown of the change-over switches 61, 107 the operation of the circuit arrangement is the same as in the case of FIG. 1. When a too high beam current is detected by the threshold circuit 49, the trigger circuit 99 will be set and the change-over switches 61 and 107 will be brought to the position not shown. The storage capacitor 53 will then be charged through the resistor 71 likewise as in the case of FIG. 1. The change-over switch 107 connects the output 111 of the shift register 81 to its input 79 so that always a one circulates in the shift register 81 and a continuous measurement of the beam current is effected until the threshold circuit 49 no longer detects a too high beam current and due to the output voltage dropping out or due to the next field frequency pulse at the reset input 101 the trigger circuit 99 is reset and the normal state is restored.
In FIG. 3 corresponding components have the same reference numerals as those in FIGS. 1 and 2 and reference is made to the description associated with the relevent Figures.
The circuit arrangement differs from that of FIG. 2 by the absence of the first change-over switch 61. Consequently, when a too high beam current occurs the normal control loop is maintained and only due to the second change-over switch 107 the shift register 81 is enabled so that a strongly increased measuring frequency and an accelerated feedback occurs through the normal control loop to the normal state. The recharge time of the storage capacitor 53 is in this case extended because per field period a larger number of measurements and associated recharges of this capacitor is effected.FIG. 4 shows a colour television receiver including a circuit arrangement according to the invention in which the principle as described with reference to FIG. 2 is used. The circuit arrangement has a colour television display apparatus having a sequential beam current measuring circuit whose principle is described in prior application Ser. No. 402,159, filed Oct. 1, 1973, now abandoned.
When a colour television signal is applied to an input 201 of a high frequency-intermediate frequency and detection section 203, a luminance signal Y is produced at an output 205, a chrominance signal Chr is produced at an output 207 and a synchronizing signal S is produced at an output 209.
The synchronizing signal S is applied to an input 211 of a time base generator 213 connected to the output 209. Deflection currents for a television display tube 221 are obtained at two outputs 215 and 217 and an EHT for the supply of the display tube 221 is obtained at an output 223.
An output 227 of the time base generator 213 applies line flyback pulses to an input 229 and an output 231 applies field flyback pulses to an input 233 of a pulse generator 235. These pulses are shown in FIG. 5 by the waveforms 529 and 533, respectively.
Furthermore FIG. 5 shows a number of waveforms 537, 509, 514, 544, 548, 520, 539, 541, 543, 547, 549, 551, 553, 656, 555, 565 and 563 and a number of instants t 1 , t 1 +T 1 , t 3 to t 10 , t 10a , and t 10 + T 2 , t 10 + 2T 2 , T 11 , t 11 + T 1 , t 11 + 2 T 1 , and t 12 which are important for explaining the operation of the circuit arrangement. The waveforms are not shown to scale.The pulse generator 235 has a number of outputs 237, 239, 241, 243, 244, 245, 247, 249, 251, 253, 255 and an input 256 at which the waveforms 537, 539, 541, 543, 545, 547, 549, 551, 553, 555 and 556, respectively, are present.
The output 255 of the pulse generator 235 is connected to an input 257 of a level insertion circuit 259 an input 261 of which is connected to the output 205 of the section 201 and receives the luminance signal Y therefrom. The level insertion circuit 259 has an output 263 at which the modified luminance signal denoted by the waveform 563 of FIG. 5 is produced which includes a reference level 521 during three line times t 5 -t 6 , t 7 -t 8 , t 9 -t 10 at the commencement of the field, which level is inserted with the aid of the wave-form 555 applied to the input 257.
The waveform 565 is then produced at a further output 265 of the level insertion circuit 259, which waveform is applied to an input 267 of a chrominance signal amplifier 269 for suppressing the chrominance signal applied to an input 271 connected to the output 207 of the section 201.The output 263 of the level insertion circuit 259 is connected to inputs 285, 287 and 289 of suppression and level correction circuits 291, 293 and 295, respectively, inputs 297, 299 and 301 of which are connected to outputs 247, 249 and 251 of the pulse generator 235 and inputs 303, 305 and 307 of which are connected to outputs 309, 311 and 313, respectively, of a demodulator and matrix circuit 315 to whose input 317 a chrominance signal originating from an output 319 of the chrominance signal amplifier 219 is applied.
The demodulator and matrix circuit applies a (B-Y), (G-Y) and (R-Y) colour difference signal to its outputs 309, 311 and 313, respectively. These colour difference signals are combined in the suppression and level correction circuits 291, 293 and 295 with the modified luminance signal applied to the inputs 285, 287 and 289 to form colour signals R, G and B, while as a result of the signals with the waveforms 547, 549, 551 at the inputs 297, 299 and 301 the reference level 521 originating from the luminance signal is suppressed in a special sequence. In this example a sequence is maintained in which only a reference level is left in the R-signal from t 5 to t 6 , in the G signal from t 7 to t 8 and in the B signal from t 9 to t 10 .
The suppression and level correction circuits 291, 293 and 295 furthermore have outputs 339, 341, 343 connected to the wehnelt electrode of the red, blue and green guns of the display tube 221 and inputs 345, 347 and 349 connected to the output 237 of the pulse generator 235 for receiving a suppression signal of the waveform 537 whi
ch prevent a signal supply to the display tube 221 during the time base flyback times. Inputs 351, 353 and 355 are furthermore connected to outputs 357, 359 and 361, respectively, of a measuring circuit 362 which outputs are connected to storage capacitors 363, 365 and 367, respectively, whose other ends are connected to earth. The storage capacitors 363, 365 and 367 are connected to switches 377, 378 and 379 respectively, operation signal inputs 383, 384 and 385 of which are connected to outputs 239, 241 and 243, respectively, of the pulse generator 235.The switch 377 conducts from t 5 to t 6 , the switch 378 conducts from t 7 to t 8 and the switch 379 conducts from t 9 to t 10 . Each of these switches introduces a level applied to an input 381 thereof into the relevant storage capacitors 363, 365, 367.
The cathodes of the television display tube 221 are interconnected and are connected to an input 389 of the measuring circuit 362. Sequentially by each cathode voltages is generated across a resistor 393 connected to the input 389 which resistor is shunted by a diode 394 and is connected at its other end to a voltage of + 130 Volt during the said periods. This generated voltage produces a corresponding voltage across the storage capacitors 363, 365 and 367 which is retained during the next field flyback time and is passed on for beam current control to the inputs 351, 353 and 355 of the suppression and level correction circuits 291, 293 and 295. The two cathodes which are not measured then do not convey any beam current.
An input 391 of the measuring circuit is connected to the output 245 and an input 392 is connected to the output 253 of the pulse generator 235.
The screen grids of the guns of the television display tube 221 are each connected to an adjusting point of potentiometers 395, 396 and 397 between + 130 Volt and a higher voltage ++.
A signal occurring at the cathodes of the television display tube 221 produces a voltage across the resistor 393 which voltage is limited by the diode 394 blocked during the measuring periods. This voltage is applied to the base of a pnp transistor 401 whose emitter is connected to the other end of the resistor 393. The collector of the transistor 401 is connected to earth through a resistor 403, connected to earth through a capacitor 405 for high frequencies and has a negative feedback to the base through a resistor 407.
A signal amplified by the transistor 401 is applied through a series arrangement of a capacitor 409, a resistor 410 and a resistor 411 to the base of an npn transistor 413. The collector of an npn transistor 415 serving as a clamping switch is connected between the resistors 410 and 411. The emitter is connected to a clamping voltage V kl and the base is connected to the input 391 of the measuring circuit 362.
As a result of the waveform applied to the input 391, which waveform is the inverse form of waveform 544, the transistor 415 is cut off during the occurrence of the signal produced by the reference level 521. During the preceding line periods the capacitor 511 has reached a constant charge condition (clamped) through the then conducting transistor 415 because both the luminance signal and the chrominance signal are suppressed and the cathodes of the display tube 221 do not convey any current.
During the period t 4 -t 10 signals caused by the reference level 521 occur successively during the periods t 5 -t 6 , t 7 - t 8 and t 9 - t 10 at the successive cathodes of the display tube 221 which signals are applied in an amplified manner to the base of the transistor 413 and which furthermore appear at the emitter of an npn transistor 412 connected to the base of the transistor 413.
When the signals have an amplitude which is not too large they are passed on by the transmitter 413 through its emitter and a resistor 423 to the base of an npn transistor 425.
When the signals at the base of the transistor 413 have a too large amplitude the emitter of the transistor 412 will become more negative than its base connected to a threshold voltage V dr and this transistor 412 will start conducting and hence operate a set input of a trigger circuit 414 connected to its collector which then provides a positive voltage for its output 416. This positive voltage (waveform 556) is applied to an output 418 which is connected to the input 256 of the pulse generator 235 and which causes a variation to a free-running state which will be described hereinafter, while this voltage also controls the base of an npn transistor 422 through a resistor 420 and the emitter of this transistor is connected to earth while the collector is connected to the base of the transistor 413. The base of this transistor 413 then is no longer controlled because the transistor 422 acts as a short circuit for its input signal. The trigger circuit 414 furthermore has a reset input 424 to which a field flyback pulse (waveform 533) originating from the output 231 of the time base generator 213 is applied so that the trigger circuit 414 is reset every time at the commencement of the field flyback.
In the state in which the beam current is not too large the transistors 412 and 422 do not conduct and the signal is passed on through the transistor 413 to the base of the transistor 425 whose emitter is connected to the collector of an npn transistor 427 whose emitter is connected to earth and whose base is connected through a resistor 428 to the input 392 of the measuring circuit. The collector of the transistor 425 is connected through a resistor 430 to the base of an npn transistor 433 whose emitter is connected through a resistor 435 to the input 381 of the sequence switches 377, 378 and 379.
The collector of the transistor 425 is connected through a resistor 432 to the emitter of an npn transistor 434 arranged as an emitter follower whose base is connected through a resistor 436 to an input 438 of the measuring circuit 362 which is connected to the output 244 of the pulse generator 235 at which the waveform 544 occurs. This transistor 434 conducts during the measuring period and furthermore every time during a number of line times in the field scan time when the pulse generator 235 has acquired a free-running state (waveform 544) due to a too high beam current with the aid of the trigger circuit 414.
The transistor 427 conducts due to the waveform 553 during the measuring periods so that in the periods t 5 -t 6 , t 7 -t 8 , t 9 -t 10 the emitter of the transistor 425 is connected to earth through the transistor 427 and the transistor 425 acts as an amplifier and passes on a signal to the base of the transistor 433 and recharges through its emitter the storage capacitors 363, 365 and 367 during the corresponding periods t 5 -t 6 , t 7 -t 8 and t 9 -t 10 , respectively.
In the state of the circuit arrangement where a too large beam current has been detected and where the trigger circuit 414 is set, the transistor 425 does not conduct due to the short circuit of the base of the transistor 413 so that then the storage capacitors 363, 365 and 367 are recharged by the emitter follower 434, 433 every time during the periods when the waveform 544 is positive. This is effected a large number of times during field scan periods following the set of the trigger circuit 414 so that the storage capacitors are charged quickly to a higher voltage to reduce the beam current through the control loops as quickly as possible again.
The operation of a control loop will now be described. This control loop not only includes the television dislay tube 221 and the measuring circuit 362 but also the suppression and level correction circuit 295.
A modified luminance signal Y m according to the waveform 563 is applied to the input 289 in which signal the reference level 521 is present during the periods t 5 -t 6 , t 7 -t 8 and t 9 -t 10 and which level is suppressed from t 1 to t 5 . The input 289 is connected to the base of an npn transistor 601 whose emitter is connected to earth through a resistor 603 and whose collector is connected through a resistor 605 to the emitter of an npn transistor 607. The collector of the transistor 607 is connected to a positive voltage. The base is connected to the input 307 and receives a red colour difference signal -(R-Y) which is suppressed from t 1 to t 10a .
Furthermore the emitter of the transistor 601 is connected to the emitter of an npn transistor 609 whose collector is connected to a positive voltage and whose base is connected to the input 301.
A voltage having the waveform 547 is applied to the input 301 so that the transistor 609 conducts during the periods t 7 -t 8 and t 9 -t 10 and thus cuts off the transistor 601 so that also the transistor 607 does not convey current and the signal at the collector of the transistor 601 is suppressed.
When the transistor 609 does not conduct a signal is generated at the collector of the transistor 601 which signal is a combination of the -(R-Y) signal supplied through the emitter follower 607 and the -Y m signal amplified through the transistor 601. This combination is a -R signal in which a level corresponding to the level 521 is present during the period t 5 -t 6 and which is suppressed in the line periods t 1 -t 5 and t 6 -t 10a . This signal is applied to the base of an npn transistor 611 whose collector is connected to a positive voltage and whose emitter is connected through a resistor 613 to the collector of an npn transistor 615 whose emitter is connected to earth through a resistor 617 and whose base is connected to a reference voltage of 1.4 V.
The emitter of the transistor 615 is furthermore connected to the emitters of two npn transistors 619 and 621 whose collectors are connected to a positive supply voltage.
A level correction voltage originating from the capacitor 367 and applied to the input 355 is transferred to the base of the transistor 619 through an npn transistor 623 arranged as an emitter follower while a suppression signal having the waveform 537 applied to the input 349 is passed on to the base of the transistor 621. The latter signal ensures the common suppression of the beam current during the line and field flyback periods. Due to the fact that the otherwise cut-off transistor 621 is blocked during these flyback periods, the transistor 615 is cut off and no signal is passed to the collector of this transistor because then the emitter circuit of the transistor 611 is interrupted.
The level correction voltage applied to the input 355 is then applied through the emitters of the transistors 623 and 619 to the emitter of the transistor 615 and influences the direct current through the resistor 613 provided by the transistor 615 and hence of the direct current level of the -R signal passed through the emitter of the transistor 611 to the collector of the transistor 615.
The collector of the transistor 615 is connected to the base of an npn transistor 625 whose collector is connected to a positive voltage and whose emitter is connected through an adjustable resistor 629 to the base of an npn transistor 631. The emitter of the transistor 631 is connected to earth and the collector is connected to the emitter of an npn transistor 633 whose collector is connected through a resistor 635 to a positive supply voltage of +130 V and whose base is adjusted to a bias voltage of +5 V. The base of the transistor 631 receives a negative feedback voltage from the collector of the transistor 633 through a potential divider 637, 639 to a negative voltage.
A red colour signal amplified by the transistors 631 and 633 is obtained from the collector of the transistor 633 and is applied through the output 343 to the wehnelt electrode of the red gun of the display tube 221. This signal includes the correction level originating from the capacitor 363 with which the beam current in the red gun is adjusted to a desired value.
The amplification of the circuit arrangement is adjustable with the resistor 629 so as to perform for example, a white point correction. Due to the control loop such an adjustment has substantially no influence on the beam current which is produced by the reference level 521 so that the black level and hence the colour of dark picture parts does not change due to the adjustment.
The modified luminance signal Y m for the input 289 is obtained in the level insertion circuit 259.
A luminance signal is applied to the input 261 which signal is applied to an input 701 of an amplifier 703. Furthermore the amplifier 703 has an input 707 to which an adjustable direct voltage is applied with the aid of a potentiometer 709 which serves for luminance adjustment and an input 771 to which a suppression signal is applied as is shown by the waveform 555. The black level of the picture 520 in the waveform 563 is adjustable with the aid of the potentiometer 709 relative to the level occurring during the suppression periods. The waveform 563 is the luminance signal which occurs at the output 263 across an emitter resistor 713 of an npn transistor 715 arranged as an emitter follower whose base is connected to an output 705 of the amplifier 703.
A signal having the waveform 555 is applied to the input 257 of the level insertion circuit through a capacitor 717, which signal is passed at one end through a resistor 721 to the base of an npn transistor 723 and at the other end through a potential divider 725, 727 to the base of an npn transistor 729.
During the most positive parts of the waveform 555 a low voltage is produced across a collector resistor 731 of the transistor 723. The level of the signal during the periods t 5 -t 10 lies below the cut-off point of the transistor 727 so that this is not found back in the collector signal 565 of this transistor. As a result of the potential divider 725, 727 connected to the positive voltage the transistor 729 only reacts to the most negative parts of the signal 555 and the transistor 729 is cut off during the periods t 5 -t 6 , t 7 -t 8 and t 9 -t 10 so that then positive going square-wave voltages occur at the collector of this transistor.
The collector of the transistor 729 is connected through a resistor 733 to a potential divider including a resistor 735 and a series arrangement of a diode 736 and a resistor 737. The voltage at the wiper of this potential divider has a constant value of + 2.2 V which also occurs at the collector of the transistor 729 during the said periods. This voltage is passed on through a resistor 738 to the base of an npn transistor 739 whose collector is connected to a positive supply voltage and whose emitter is connected to the emitter of the transistor 715. During the periods t 5 -t 6 , t 7 -t 8 and t 9 -t 10 the emitter of the transistor 739 is brought to a voltage of +1.5 V due to its base voltage so that the transistor 715 is cut off and the reference level 521 is produced at the output 263. The rest of the period the transistor 739 is cut off and a signal is applied to the output 263 through the transistor 715.
In the waveform 563 arrows show that the black level 520 of the signal is adjustable relative to the constant reference level 521 which as a result of the said control corresponds to a constant beam current valve in the guns of the display tube 221.
The pulses for the various parts of the circuit arrangement are supplied by the pulse generator 235.
This generator has four trigger circuits 801, 803, 805 and 807. A signal which relative to the signal 533 at the input is delayed over a time T 1 is applied to a set input 809 of the trigger 801 from the input 233 through a delay circuit including a resistor 811 and a capacitor 813. This signal is shown in the waveform 509. Before this pulse appears the triggers 801, 803, 805 and 807 are assumed to be in the reset condition. At the instant t 1 + T 1 the trigger 801 is brought to the set condition. A signal represented by the waveform 514 then appears at an output 814 thereof which signal is applied to four AND gate circuits 817, 819, 821 and 823.
Furthermore an inverted set signal from the trigger 801 is applied to the gate 817 as well as a line frequency pulse signal originating from the input 229.
In addition a signal represented by the waveform 544 originating from an output 816 of the trigger 803 and a signal originating from an AND gate 817 are applied to the gate 819.
The gate 821 furthermore receives a signal having a waveform 518 from an output 818 of the trigger 805 and a signal originating from an AND gate 829.
Furthermore the gate 823 receives a signal having a waveform 520 from an output 820 of the trigger 807 and an inverted line frequency pulse signal originating from the input 229. An output of this gate 823 is connected through a delay circuit including a resistor 831 and a capacitor 833 to an AND gate 835 which also receives a line frequency pulse signal from the input 229. The output of this gate 835 is connected to the reset inputs of all four triggers. The gates 827 and 829 receive an inverted signal from the input 229 and an output signal from the triggers 805 and 807, respectively.
Furthermore a switchable feedback is present between the output 814 of the trigger 801 and the set input thereof, which feedback is only switched on when the beam current in the display tube 221 is too high. This feedback starts from the output 814 through a resistor 837 which is connected to the base of an npn transistor 839 whose emitter is connected to earth and whose collector is connected through a resistor 841 to the input 256 and is connected to the output of the trigger circuit 414, the base of a transistor 843 connected to the collector of the transistor 839 having its collector connected to a positive supply voltage and being connected through a resistor 845 connected to the emitter of this transistor 843 to the set input 809 of the trigger 801. Together with the capacitor 813 the resistor 845 constitutes a delay circuit having a time delay T 2 . The feedback is not present when the beam current is so low that the trigger circuit 414 is not set; the voltage at the collector of the transistor 839 is then low and the transistor 843 is cut off. This condition is assumed to be present at the commencement of the time axis in FIG. 5.
When the voltage at the input 256 becomes high due to the trigger 414 coming in the set condition the feedback will be switched on. The transistor 839 will then act as an inverter.
When at the instant t 1 + T 1 the trigger 801 is set, the trigger 803 is blocked at the same time by the inverted set signal from the trigger 801 applied to the gate 817. The triggers 803, 805 and 807 then remain in the reset condition. After the end of the set pulse from the trigger 801, i.e. after the instant t 3 = t 2 +T 1 the gate circuit 817 becomes conducting at the instant t 4 due to the next line frequency pulse and brings the trigger 803 in the set condition. The gate 819 then becomes conducting after the end of the line pulse under the influence of the gate 827 and applies a pulse through a delay network 845, 847 to an input of an AND gate 849 to which also the line frequency signals of the input 229 are applied. At the next line pulse at the instant t 6 the trigger 805 is set so that the gate 821 provides a pulse which is delayed by a network 851, 853 and is applied to an input of an AND gate 855 to which furthermore the line frequency pulse signal from the input 229 is applied. This gate 855 becomes conducting at the next line pulse at the instant t 8 under the influence of the gate 829 and brings the trigger 807 to the set condition.
A pulse is obtained from the output of the gate 823 which begins after termination of the line pulse and which is applied delayed through the network 831, 833 to the gate 835 which then passes the next line pulse at the instant t 10 so that the triggers 801, 803, 805 and 807 are reset.
When meanwhile the trigger circuit 414 has not reached the set condition during the time between t 5 and t 10 in which the beam currents in the display tube are measured, the triggers 801, 803, 805 and 807 remain in the reset condition until the next field pulse.
When the trigger circuit 414 changes to the set condition due to a too high beam current for example between t 5 and t 6 , the described feedback from the output 814 of the trigger 801 is provided at its set input 809 and a positive pulse will appear again at the instant t 10 + T 2 at the input 809 which pulse will bring the trigger 809 to the set condition again so that at an instant t 10 + 2T 2 the trigger 803 is set whereafter the triggers repeat the above-described conditions every time.
At the next field pulse at the instant t 11 , however, the trigger 414 is reset so that the feedback in the pulse generator 235 is interrupted and a set cycle of the triggers 801, 803, 805 and 807 already started is terminated at the instant t 12 when the gate 835 applies a line pulse to the reset input of the trigger 801 so that voltage at the output 814 remains equally low and all triggers are reset. After the line pulse drops out at its reset input the trigger 801 is brought immediately to the set condition so that at the next line pulse which lies after the instant which is T 1 later than the trailing edge of the field pulse 533 the trigger 803 is set again and three line times later everything is reset and remains reset when the beam current has become sufficiently low due to the repeated recharge of the capacitors 363, 365, 367 in the measuring circuit 362.
A number of gate circuits supplying the output signals from the pulse generator 235 are coupled to the outputs of the trigger circuits.
The output 239 is connected to an output of an AND gate 857 which has an input connected to the output 816 of the trigger 803, an inverted input which is connected to the output 818 of the trigger 805 and an inverted input which is connected to the input 229 of the pulse generator. The output of this gate then applies the waveform 539.
The output 241 is connected to an output of an AND gate 859 and input of which is connected to the output 818 of the trigger 805, an inverted input is connected to the output 820 of the trigger 807 and an inverted input is connected to the input 229 of the pulse generator 235. The AND gate 859 provides an output signal having the waveform 541.
The output 243 is connected to an output of an AND gate 861 an input of which is connected to the output 820 of the trigger 807 and an inverted input is connected to the input 229 of the pulse generator 235. The AND gate 861 provides an output signal having the waveform 543.
The output 244 is connected to the output 816 of the trigger 803 and the output 245 is connected to an inverted output of this trigger.
The output 247 is connected to an output of an OR gate 863, the output 249 is connected to an output of an OR gate 865 and the output 251 is connected to an output of an OR gate 867. The inputs of the gate 863 are connected to the outputs of the gates 861 and 859, the inputs of the gate 865 are connected to the outputs of the gates 861 and 857 and the inputs of the gate 867 are connected to the outputs of the gates 857 and 859. The gates 863, 865 and 867 supply the signals 547, 549 and 551, respectively.
The output 253 is connected to an output of an AND gate 869 an input of which is connected to the output 816 of the trigger 803 and an inverted input is connected to the input 219 of the pulse generator 235. The waveform 553 is supplied by this gate.
The output 237 is connected to an output of an OR gate 871 whose inputs are connected to the inputs 229 and 233 of the pulse generator. The waveform 537 is provided by the gate 871.
The output 255 is connected to an output of a gate and superimposition circuit 873 an input of which is connected to an inverted output of the trigger 801, an inverted input is connected to the output of the gate 871 and an inverted input is connected to the output of the gate 869. The gate and superimposition circuit 873 provides the waveform 555.
Since the latter waveform is applied through the capacitor 717 to the input 257 of the level insertion circuit 259, an unsatisfactory amplitude selection will take place when the trigger 801, 803, 805 and 807 run free for a long period so that a charge of the capacitors 363, 365, 367 in the measuring circuit 362 is effected independently of the beam current when it has exceeded a given maximum value as described in the foregoing. The level shift due to this capacitor 717 and the results thereof are not important for the understanding of the invention and are therefore not shown in the waveforms.
Although the control voltage in the described embodiments is applied to a wehnelt electrode of the display tube it may alternatively be applied to a different control electrode.
The insertion of the reference level may of course alternatively be effected in a different manner such as, for example, by clamping the video signal on the control signal.
For switching over the described functions in case of a too large beam current value a schmitt trigger or a monostable multivibrator may be used if desired instead of a bistable trigger circuit having a set and reset, input, or an output signal from the threshold circuit may be used.
The type of measuring circuit is not important for using the step according to the invention. Instead of a sequence measuring circuit it is alternatively possible to use a simultaneous measuring circuit in which all beam currents are simultaneously measured in the case of a multigun display tube. Also the input circuit of the measuring circuit may of course be adapted as desired.
If desired other counting circuits may be used instead of a shift register.
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