MIVAR 16C3V CHASSIS TV2962/2 + TV2965/2 INTERNAL VIEW.

This CHASSIS TV2962/2 + TV2965/2 is the first MIVAR color Television chassis with complete separation from mains, and it's a completely new design particularly in deflection and power stages.

Previous models have had a Line Mains direct connected chassis type.

It's the first MIVAR chassis featuring the AV SCART Socket for external AV sources.

It's an unusual chassis because of the use of an isolation mains transformer in power supply part on bottom cabinet side.



The PLL SYNTHESIZER TUNING UNIT RT18A-99  uses a uController M3870 combined with a M206B1 all made by SGS.
SGS is Società Generale Semiconduttori - Aquila Tubi E Semiconduttori (SGS-ATES, "Semiconductor General Society - Tubes and Semiconductors Aquila"), later SGS Microelettronica, a former Italian company now merged into STMicroelectronics
SGS Microelettronica and Thomson Semiconducteurs were both long-established semiconductor companies. SGS Microelettronica originated in 1972 from a previous merger of two companies:

• ATES (Aquila Tubi e Semiconduttori), a vacuum tube and semiconductor maker headquartered in the Abruzzese city of l'Aquila, who in 1961 changed its name into Azienda Tecnica ed Elettronica del Sud and relocated its manufacturing plant in the outskirts of the Sicilian city of Catania
• Società Generale Semiconduttori (founded in 1957 by Adriano Olivetti).

CIRCUITS DESCRIPTIONS:
Switching regulator power supply device combined with the horizontal deflection circuit of a television receiver which it supplies:
MIVAR CHASSIS TV2962/2 + TV2965/2 POWER SUPPLY WITH THOMSON TEA2026 STEP UP SMPS EXPLANATION THEORY.1 - COLOR TV SCANNING AND POWER SUPPLY PROCESSOR THOMSON TEA2026

DEFLECTION :
.CERAMIC 500kHz RESONATOR FREQUENCY REFERENCE
.NO LINE AND FRAME OSCILLATOR ADJUSTMENT
.DUAL PLL FOR LINE DEFLECTION
.HIGH PERFORMANCE SYNCHRONIZATION
.SUPER SANDCASTLE OUTPUT
.VIDEO IDENTIFICATION CIRCUIT
.AUTOMATIC 50/60Hz STANDARD IDENTIFICATION
.EXCELLENT INTERLACING CONTROL
.SPECIALPATENTED FRAME SYNCHRO DEVICE
FOR VCR OPERATION
.FRAME SAW-TOOTH GENERATOR
.FRAME PHASE MODULATOR FOR THYRISTOR,SMPS CONTROL
.ERROR AMPLIFIER AND PHASE MODULATOR
.SYNCHRONIZATION WITH HORIZONTAL DEFLECTION
.SECURITY CIRCUIT AND START UP PROCESSOR

GENERAL DESCRIPTION
the TEA2028 combines 3
major functionsof a TV set as follows :
- Horizontal (line) and vertical (frame) time base
generation for spot deviation. The video signal is
used for the synchronization of both time bases.
- On-chip switching power supply controller synchronized
on line frequency.
This integrated circuit has been implemented in
bipolar I2L technology, and various functions are
digitally processed. In fact, resorting to logic functions
has the advantage of working with pure and
accurate signals while full benefit is drawn from
high integration of logic gates (approx. 110 gates
per mm2).
The main objective is to drive all functions using an
accurate time base generated by a master 500kHz
oscillator.
Also, horizontal and vertical time bases, are obtained
by binary division of reference frequency.
This has the advantage of eliminating the 2 adjustments
which were necessary in former devices.

- MAIN FUNCTIONS
- Detection and extraction of line and frame synchronization
pulses from the composite video
signal.
- Horizontal scanning control and synchronization
by two phase-locked loop devices.
- Video identification.
- 50 or 60Hz standardrecognition for vertical scanning.
- Generation of a self-synchronized frame sawtooth
for 50/60Hz standards.
- Line time constant switching for VCR operation
through an input labeled ”VCR” (Video Cassette
Recorder).
- Control and regulation of a primary-connected
switching power supply by on-chip controller device
combining :
• an error amplifier
• a pulse width modulator synchronized on line
frequency
• a start-up and protection system
- Overall TV set protection input
- Frame blanking and super sandcastle output signals
- Frame blanking safety input for CRT protection in
case of vertical stage failure.

FUNCTIONAL DESCRIPTION
Majority of the on-chip analog functions were computer
simulated and results such as temperature
variation, technological characteristic dispersion
and stability, have led to the enhancement and
implementation of actually employed structures.A
parallel in-depth study of the device implemented
in form of integrated sub-sections is provided to
analyze the overall performance in a TV set.


TEA2026 Scanning control circuit for a television receiver, with gradual startup

This invention concerns television scanning circuits.
It involves a scanning startup circuit comprising a capacitor, and means of gradual charging or discharging, to produce a priority voltage transmitted at startup to replace a control voltage, regulating a chopped power supply circuit which is to be started up progressively. At the end of a certain period of time, the regulation control voltage takes over from the original voltage which, at startup, is at a level such that it prevents over-consumption of current in power components. The circuit is also protected against voltage surges, being halted and restarted automatically and gradually in the event of such a surge.

1. In a television receiver comprising a switched mode power supply and a scanning control circuit, said power supply circuit comprising a power switching element (T2) and a pulse-width modulator (24, 26) having an input for receiving a modulating voltage and an output for providing pulses of modulated width, wherein said scanning control circuit comprises, in view of ensuring gradual start-up of scanning:
supply terminals for receiving a low voltage supply (Vcc),
a capacitor (C3) having a first plate and a second plate which are connected to said supply terminals in such a way that upon starting up of the receiver, said first plate will follow the rising potential of one supply terminal,
a voltage limiter (38) to limit the voltage on said capacitor first plate to a predetermined value Vz,
an analogic voltage transmission device (32) having a first input connected to said capacitor first plate, a second input for receiving a control voltage which may vary between predetermined limits, and an output which is connected to the input of said pulse width modulator, said voltage transmission device being designed so that its output will transmit either the voltage on its first input or the voltage on its second input, depending on the relative magnitude of these voltages, the transmitted voltage being the one which corresponds to the narrower pulse width at the output of the pulse width modulator,
the value of Vz being such that it corresponds to a pulse width narrower than the minimum pulse width that may be produced by the modulator when said modulator receives the said control voltage from the transmission device,
means (34) for progressively altering the electrical charge of said capacitor so as to vary the potential of its first plate in a sense corresponding to an increase of the pulse width defined by this voltage.
2. A circuit as claimed in claim 1 wherein said second plate of the capacitor is directly connected to said one supply terminal. 3. A circuit as claimed in claim 1, wherein said means for progressively altering the charge of said capacitor comprises a current source controlled so as to supply current to the capacitor only when receiving fly-back pulses indicating that horizontal scanning is operating. 4. A circuit as claimed in claim 1 further comprising a threshold comparator (62) having one input receiving said low voltage supply and another input connected to a voltage reference, and an output for supplying to the pulse width modulator a disabling signal whenever said low voltage supply falls below a predetermined value (VS2). 5. A circuit as claimed in claim 3, wherein a second means for gradually altering the electrical charge of said capacitor is provided, said second means comprising a current source connected to the capacitor in such a way that it will change the voltage on the first plate in a sense corresponding to a decrease of the pulse width defined by this voltage.

Description:

This invention concerns television receivers, and more specifically circuits to control the sweep of the light spot produced on the television screen. In particular, it relates to the way in which such circuits start up when the receiver is switched on.
A television receiver usually contains a horizontal scan transformer, surrounded by several closely interconnected circuits:
stabilized supply circuit, providing a regulated DC voltage of about 100 volts, to supply the transformer;
horizontal scan control circuit supplying periodical signals to the base of a transistor mounted in series with the primary transformer winding, with a horizontal deflection coil connected to this transistor;
vertical deflection control circuit using a secondary winding of the horizontal scan transformer as source of supply, to produce a periodical voltage gradient for vertical scanning;
very high voltage circuit using a secondary winding of the transformer to create a high potential in the cathode-ray tube, for the purpose of producing and accelerating the electron beam.
In one embodiment, described in French patent application No. 81 08 337 of Apr. 27, 1981 on the present applicant's behalf, these different circuits are closely interconnected and precise provision has to be made for their startup and stoppage, taking account of their reciprocal interactions.
FIG. 1 shows the general layout of the horizontal and vertical scanning circuit, with a regulated chopped power supply circuit. This invention applies specifically to such a circuit. Further details are to be found in the aforementioned patent application, but the general structure will be described here, in order to define the purposes of this invention.
The horizontal scan transformer TL is provided with a primary winding EP, connected on one side to the collector of a transistor T1, the emitter of which is earthed, and on the other side to the output of a stabilized chopped power supply circuit, which is itself powered by the AC mains current.
A horizontal scanning control circuit 10 supplies the base of transistor T1 with periodical signals, synchronized with a synchronization signal SL, extracted by a synchronization extractor (not shown here) from the video signal reaching the receiver.
The transistor collector T1 is connected to a horizontal deflection coil 12, which produces a periodical deflection of the electronic beam of the tube at every changeover of transistor T1.
Transformer TL contains one or more secondary windings, e.g. one winding to produce a very high voltage THT, and another to activate a vertical deflection circuit. A line return signal RL, a negative pulse produced whenever the beam returns as a result of sudden blocking of transistor T1, is taken from one of these windings. This negative pulse RL is used for the horizontal scanning circuit 10, since scanning is synchronized with the syncronization signal SL by shifting the transistor T1 control signals until SL and RL are synchronized. The same pulse RL is also used to operate the chopped power supply circuit.
The vertical deflection circuit 14 is connected to a vertical deflection coil 16, and to a secondary winding of the transformer TL, and is also connected to the horizontal scanning control circuit, to ensure the necessary correspondence between line scanning and frame scanning.
The chopped power supply ciruit, comprises, following a transformer 18 powered by the mains, a rectifier bridge 20 followed by a filter capacitor C1 and inductance L1, mounted in series with the collector of a transistor T2, the emitter of which is connected to earth.
The collector of this transistor T2 is connected to the anode of a diode 22, the cathode of which is connected to a capacitor C2. The supply circuit output voltage reaches the terminals of this capacitor C2. The supply circuit functions by means of high-frequency switching (at the horizontal scan frequency) of transistor T2, by means of periodical signals (line scanning period 64 microseconds).
Capacitor C2 forms an energy accumulator, which discharges into the utilization circuit while transistor T2 is conducting, and which recharges through inductance L1 when transistor T2 is blocked; the width of the signals is automatically regulated, so that the charge lost every time transistor T2 becomes conducting is exactly counterbalanced by the charge regained during each blocking.
Regulation is obtained by taking a regulation control voltage at the supply circuit output, and comparing it with a stabilized reference voltage, the difference between them being amplified in a differential amplifier 30, and compared in a comparator 24 with a periodical sweep voltage, DSC, supplied at the line scan frequency by a generator 26, which to this effect receives signals at this frequency from the horizontal scanning circuit 10.
The comparator thereby supplies variable-width signals at this frequency, and these are amplified in an amplifier 28, and delivered to the base of transistor T2.
Finally, since transistor T2 has to block a high current every period, the strongly negative line return pulse RL from the secondary winding of transformer TL is delivered to its base. As already stated, RL is synchronized with the horizontal scanning signals from circuit 10, and therefore with returns of the sweep voltage DSC, so that a negative pulse encouraging blocking of transistor T2 appears immediately after delivery of a blocking signal from comparator 24.
The electronic control circuits are supplied with low-voltage DC current Vcc, obtainable either by rectifying a fraction of the mains AC current, or by taking a fraction of the DC voltage from the stabilized chopped power supply circuit, or even by a combination of both methods.
In conclusion, close interconnections exist among the chopped power supply circuit (operating at the horizontal scanning frequency), horizontal scanning circuit, vertical scanning circuit, and even very-high-voltage circuits. This invention is intended to perform the following functions relevant to this situation.
First, it is preferable for vertical scanning to function whenever horizontal scanning is in operation (something that involves the appearance of a very high voltage); otherwise, a motionless bright horizontal line is created on the screen, and this will ultimately burn out the photosensitive layer of the television tube.
Second, on startup of the chopped power supply circuit, when the receiver is first switched on, the regulation system will tend to make the comparator 24 generate signals of maximum length, until the supply circuit has reached its nominal output voltage. Current in transistor T2 increases in a linear way (because of inductance L1) during the period of the signal, resulting in far too high a consumption of current in the transistor during supply circuit startup, and serious overheating of the transistor, which has to cut out this high current (several amps) at high voltage (several hundred volts). In any case, the duration of these signals needs to be limited, for normal functioning of the supply circuit (e.g. by ensuring that the regulation control voltage cannot fall below a certain threshold). However, such limitation, although adequate during normal operation, does not eliminate the risk of destruction of transistor T2 during startup.
Apart from this startup problem, there is also the matter of stoppage of scanning circuits. The stabilized power supply circuit has to be switched off before horizontal scanning stops, since absence of scanning results in abnormal functioning of the power supply (removal of the sweep voltage and of the line return signal allowing transistor T2 to be properly blocked). When the supply circuit is off, scanning can be stopped, but horizontal and vertical scans have to be switched off simultaneously, and this should in fact not be done too quickly after the removal of power, since a very high voltage remains for some time, producing an electron beam which has to sweep the whole screen, so as not to risk burning a central point on it.
Finally, in exceptional cases, such as occurrence of overvoltage at a critical point, emergency switchoff of the supply circuit and scanning circuits must be possible, with automatic restarting, unless such voltage surges recur too quickly.
This invention aims to overcome such difficulties by means of a scan control circuit with a chopped power supply circuit, comprising a controlled switch (transistor T2) and a comparator or phase modulator 24, to produce control siqnals for this switch, the comparator being capable of receiving a variable regulation control voltage at one input and a sweep voltage at a second input, to produce signals of variable width, a startup circuit comprising a capacitor, means of having the potential of a first plate of the capacitor follow quickly the potential of a low voltage source Vcc, whenever this low voltage rises above zero to a value Vz, a priority transmission device, one input of which is connected to the first capacitor plate, while another input receives the regulation control voltage, and an output connected to the first comparator input, in order to ensure priority of delivery to this comparator input of whichever of the two voltages will produce narrower signals to cause the controlled switch to conduct, the value of Vz being such that when it is reached and the chopped power supply circuit is functioning, it is given priority for delivery to the comparator, to generate signals much shorter than the maximum possible length, and means of loading or unloading the capacitor gradually, in such a way that the potential of the first plate is gradually altered until it no longer has priority.
On startup, regardless of the regulation control voltage, signals making transistor T2 conduct are very short, since they widen as the capacitor discharges, until the supply circuit regulation control voltage takes over and issues signals, in accordance with normal functioning of the chopped power supply circuit.
Furthermore, control of horizontal and vertical scanning is inhibited if the low voltage is below a threshold VS1, and control of the chopped power supply circuit is inhibited until the lower voltage reaches a threshold VS2, above VS1.
The priority transmission device may, for example, comprise a pair of diodes or transistors.
Other features and benefits of the invention will emerge from the following detailed description, with reference to the accompanying figures:
FIG. 1, already described, showing the general layout of a television receiver scanning circuit, particularly suitable for use with this invention;
FIG. 2, showing the startup circuit for the invention;
FIG. 3, showing a chart of voltages during receiver startup and stoppage;
FIG. 4, showing a chart of voltages in the event of voltage surges;
FIG. 5, showing a detail of the circuit.
FIG. 2 shows a capacitor C3, the first plate of which is connected to a terminal of a circuit 31, which is itself connected to an input E1 of a priority transmission device 32, possessing another input E2, which receives a control voltage to regulate the chopped power supply circuit in FIG. 1 (voltage taken from the divider bridge at the supply circuit output and amplified by amplifier 30). The priority transmission device output S is connected to the first input of comparator 24, the other input of which receives the sweep voltage DSC.
The priority transmission device 32 delivers the higher of the two voltages entering inputs E1 and E2 at its output S. To this effect, the device may, for example, comprise two transistors T3 and T4, mounted as follower emitters. Both emitters are connected to the output S, while the base of T3 is connected to input E1 and the base of T4 to input E2.
Transistor T4 can also in fact form part of the output stage of amplifier 30. The device may also comprise two diodes, instead of the base/emitter connections.
The second plate of capacitor C3 is connected to a low-voltage source, preferably the voltage Vcc used to supply all scan control circuits in the receiver, and which may be 12 volts, for example.
This voltage Vcc is obtained by rectifying a low-voltage AC current, which may be taken from the transformer 18 in FIG. 1. In one recommended embodiment (not shown here), arrangements may be made so that, when the scanning circuit has been started up, a rectified, filtered voltage, from the line transformer, takes over from the rectified filtered voltage from transformer 18, to provide the voltage Vcc.
The first plate of capacitor C3 is also connected to a charging device (current source 34) and discharging device (current source 36), to charge and discharge the capacitor fairly slowly (a few tens or even hundreds of milliseconds). The plate is also connected to a voltage limiter (Zener diode 38 and possibly additional diodes, to adjust the limiter output voltage to 6 or 7 volts). Current source 34 functions only in the presence of the line return signal RL.
Finally, the first plate of capacitor C3 is connected to the input of a threshold comparator 40, which delivers a positive logic signal when the voltage at the plate exceeds a threshold VS4 of about 6 volts.
The comparator output is connected to one input of an AND gate 42, another input of which receives the output signal from a decoder 44 at the output from a counter 46, in such a way that the AND gate theoretically opens when threshold VS4 is exceeded, except when a predetermined content of counter 46 is reached.
The output from the AND gate 42 is connected to one input of an OR gate 48, the other input of which receives a resetting signal RAZ, for resetting input R of a bistable flipflop 50, the output of which controls incrementation of a counter 46. This counter can also be cleared by the RAZ signal from a logic circuit (not shown here), producing a logic level resetting the flipflop and counter to 0, either when Vcc falls below a low threshold VS5, of about 4 volts (i.e. when the receiver is switched on again after complete stoppage); or at the end of a given period of time.
The flipflop 50 receives a switching control signal from an AND gate 52, one input of which receives the line return signal RL from the horizontal scanning circuit, while another input receives the output signal from a threshold comparator 54, which delivers a high logic level whenever any voltage surge appears at a safety terminal 55 connected to a critical point in the receiver.
One input terminal of comparator 54 accordingly receives the voltage from this critical point, while the other input is raised to a reference potential VS3.
When this threshold VS3 is exceeded, flipflop 50 is switched over at the next line return signal RL (and therefore on condition that horizontal scanning is taking place). This switchover increments the counter and stops the horizontal and vertical scanning circuits. Here, the output of flipflop 50 is connected through an OR gate 56 to the horizontal scanning control circuit 10 and vertical scanning control circuit 14, in order to inhibit functioning of these two circuits. OR gate 56 also receives a logic signal from a threshold comparator 60, which also delivers a scanning control circuit inhibiting signal whenever supply voltage Vcc falls below a threshold VS1 of about 6 volts, significantly higher than the threshold VS5 below which the clearing signal RAZ is issued.
Finally, a last threshold comparator 62 compares Vcc with a threshold VS2 of about 9 volts, higher than VS1, and transmits, through an AND gate 64 receiving the line return signal RL and an OR gate 66, a chopped power supply circuit inhibiting signal, for example a signal inhibiting comparator 24 which establishes signals of variable width, at the first line return after voltage Vcc has fallen below threshold VS2.
Another input of OR gate 66 is connected to the output from OR gate 56.
All threshold comparators comparing Vcc with a threshold have been shown with one input at Vcc and the other at the required threshold. In practice, it is preferable to compare a voltage KVcc with a single threshold voltage common to all comparators, the coefficient K varying from one comparator to another, controled by divider bridges. The common threshold voltage may be a reference voltage of 1.26 volts, a band-gap reference that is easily generated, and which offers very good temperature stability.
The description below covers functioning of the various parts of the circuit when the receiver is switched on, switched off, and also when repeated voltage surges occur at a critical point in the receiver.
FIG. 3 shows a time chart of voltages when the receiver is switched on and switched off.
When it is switched on (at time 0), voltage Vcc quickly reaches its nominal value of 12 volts (upper graph in FIG. 3).
The capacitor C3 is initially discharged, and its first plate (connected to E1) is at a potential Ve which initially follows the potential of the other plate, i.e. Vcc, up to a value Vz of about 7 volts, depending on the Zener diode limiter 38.
Before reaching Vz, voltage Vcc passes through a phase during which it is below a threshold VS5 of about 4 volts, causing resetting to 0 of flipflop 50 and counter 46. Vcc then passes through threshold VS1 of about 6 volts, causing, by means of comparator 60, simultaneous startup of horizontal and vertical scanning (which can function with a voltage Vcc of about 6 volts).
The appearance of horizontal scanning generates a periodical signal RL at the level of the horizontal scan transformer, thereby enabling the current source 34 (the average amplitude of which is greater than that of current source 36) to function, so that capacitor C3 begins to charge, reducing potential Ve, with a time constant of a few tens of milliseconds. The decrease in Ve is approximately linear overall, but in reality it takes place in a stepped fashion, each line return signal RL causing a small drop in the capacitor load. Meanwhile, almost as soon as scanning starts, voltage Vcc has passed through threshold VS2 of about 9 volts, enabling the chopped power supply circuit to function.
Ve is therefore initially about 7 volts, i.e. much higher than the regulation control voltage delivered to the chopped power supply circuit. This control voltage tends to be at its minimum level, to produce, by comparison with the sweep voltage DSC, very wide signals. In fact, voltage Ve is transmitted with priority through transistor T3, so that narrow signals are produced. The width of the signals increases as Ve falls; the chopped power supply circuit gradually reaches normal operating conditions, and the regulation control voltage rises. The system is thus in a phase of gradual startup of the power supply circuit.
After a certain period, voltage Ve has dropped to a level below the regulation control voltage, which has risen with the supply circuit output voltage. The regulation control voltage thereupon takes over from voltage Ve in the priority transmission device 32, initiating a phase of normal regulation of the power supply circuit.
Because the chopped power supply circuit cannot begin to function until Vcc is above VS2 (9 V), whereas scanning is already in operation from VS1 (6 V), the line return signal RL is still present, as well as the sweep voltage DSC, to ensure proper functioning of the supply circuit.
When the receiver is switched off, Vcc falls gradually, for lack of mains supply. When it drops below VS2 (9 V), functioning of the chopped power supply circuit is inhibited at the next line return signal RL (comparator 62 and AND gate 64). The power supply is therefore switched off in the blocked state of transistor T2 in FIG. 1.
Vcc then falls below VS1 (6 V), stopping scanning. Meanwhile, the supply circuit output voltage has dropped, so that the very high voltage has itself fallen; the electron beam is therefore no longer so intense when scanning stops, and there is no danger of an intense motionless beam burning any point on the screen.
Occurrence of voltage surges at a critical point on the circuit is illustrated in the voltage chart in FIG. 4, showing potential Vcc from the time of switching on the receiver, and potential Ve from the time of switching on and after the appearance of three successive voltage surges, resulting, for instance, from an ionizing discharge in the cathode-ray tube at the start of its lifespan.
At switch-on (time 0), potential Vcc rises from 0 to 12 volts. Potential Ve follows it, stabilizing at Vz (about 7 volts), after which it decreases slowly, to establish the gradual startup phase of the chopped power supply circuit as already explained in connection with FIG. 3.
A voltage surge is assumed to occur at time t ₁ , causing threshold VS3 (which can be regulated as necessary, to suit the voltage surges to be detected), to be exceeded at the input to comparator 54, which therefore delivers a signal switching over flipflop 50 at the first line return signal RL after the threshold has been exceeded. Counter 46, initially reset to 0, because Vcc has passed through a level below 4 volts during switching-on of the receiver, is incremented by a unit.
Simultaneously, the flipflop output delivers a signal through OR gate 56, to inhibit horizontal and vertical scanning, and through OR gate 66, to inhibit the chopped power supply circuit.
Since there is no further horizontal scanning, current source 34 can no longer keep capacitor C3 charged, and current source 36 begins to discharge it slowly, raising potential Ve.
When Ve reaches threshold VS4, comparator 40 delivers a logic signal which resets flipflop 50, simultaneously causing restarting of scanning, and the chopped power supply circuit since Vcc has remained at 12 volts in the meantime.
Horizontal scanning causes reappearance of the periodical line return signal RL, and current source 34 can once again charge capacitor C3, causing potential Ve to fall gradually, initiating a phase of gradual startup of the supply circuit.
Further surges can occur, for example at times t ₂ and t ₃ . The same procedure prevails, unless the content of counter 46 is high enough for the decoder 44 to deliver a signal blocking AND gate 42, therefore blocking resetting of flipflop 50. If the decoder is programmed for example, to deliver a blocking signal when the contents of the counter indicate the third successive voltage surge without resetting to 0 (it should be remembered that resetting can occur either when the receiver is switched off, or after a certain period), the voltage surge occuring at time t ₃ will again cause simultaneously stoppage of scanning and of the chopped power supply circuit at the first line signal RL after appearance of the voltage surge, and therefore raising potential Ve. But decoder 44 will prevent resetting of the flipflop and therefore startup of scanning circuits and supply circuit Ve will stabilize at the level Vz determined by the limiter. No fresh startup can occur until the receiver has been switched off and on again.
The number of surges and the period for which they can occur without causing definite stoppage of scanning circuits can be selected as required.
The clock governing the period within which n succeeding surges (where n is 3, for example) must not occur can be formed of a monostable flipflop (not shown here), tripped by the first status-change in the counter, or of any other timing device. The counter and flipflop can also be cleared periodically, every 20 milliseconds, by resetting to 0 of the vertical scan.
FIG. 5 shows a constructional detail illustrating a further improvement. The circuit in FIG. 1 remains unchanged, but only the capacitor C3 and current sources 34 and 36 are shown here.
These current sources supply current of the same amplitude, but they are activated during different periodical time intervals. The time interval for control of source 34 lasts 4 microseconds on each period (provided that the line return signal RL is present); this time interval is established by a logic circuit, and is located within the duration of the RL signal. The interval controlling source 36 lasts 2 microseconds for each horizontal scanning period, and can be generated from a 500 kHz clock and a logic circuit allowing a single clock impulse to pass for each horizontal scanning period.
In the presence of RL, the average value of currents delivered means that capacitor C3 charges; in the absence of the signal, it discharges.
Provision can also be made for the lower level of voltage Ve to be limited, by a voltage from a divider bridge supplied by voltage Vcc. A diode providing a direct polarization link between the divider bridge and the first plate of capacitor C3 prevents Ve from falling below a certain level. This obviously involves absolute limitation of the width of signals which the chopped power supply circuit comparator 24 can supply.
If the scanning circuit is an integrated circuit (except for power components), capacitor C3 is kept outside this integrated circuit, as well as the divider bridge and diode, so that maximum signal width can be adjusted at will.
In the description above, the first plate of capacitor C3 has to follow potential Vcc quickly, when the receiver is switched on. For this purpose, the second plate is connected to the low-voltage source Vcc. This plate could also be earthed, and a fairly intense additional current source included, activated only when Vcc is below a threshold of 8 to 10 volts. Such a source would be parallel to source 36, and would cease to function once Vcc reached its nominal level of 12 volts.

MIVAR  16C3V  CHASSIS TV2962/2 +  TV2965/2
2 - Step-up switching regulator power supply device comprising, connected between the poles of a rectifier circuit supplied by an isolating voltage step-down transformer and loaded by a first filter capacitor, and inductance and the collector-emitter path of a first switching transistor of NPN type, a first diode whose anode is connected to the junction of the inductance and to the collector of said transistor and whose cathode is connected to a second filter and storage capacitor supplying a voltage at its output which supplies a horizontal deflection circuit of a television receiver.
This horizontal deflection circuit which comprises in cascade a horizontal oscillator, a driver stage and an output stage, forms an integral part of the circuit controlling said first transistor and determines the repetition period of the switching, because it is started under an initial voltage slightly less than the unregulated input voltage of the device.
The switching transistor is being turned off in synchronism with the turning off of the trace switch transistor by using flyback pulses of negative polarity to bias the base thereof.

1. A power supply device with switching regulation and boosting of its DC output voltage, combined with a horizontal deflection circuit of a television receiver, supplied thereby and which comprises in cascade a horizontal oscillator, a driver stage and an output stage including a trace switch transistor and a line transformer, this device comprising an inductance and the collector-emitter path of a switching transistor connected in series between the poles of a DC input voltage source, a rectifying diode connected by its anode to the junction between the inductance and the collector of said switching transistor and by its cathode to one of the terminals of a filtering and storage capacitor whose other terminal is connected to the emitter of said transistor, so as to apply across its terminals an initial DC voltage slightly lower than said input voltage, when said switching transistor is turned off, and a regulated DC output voltage with a level higher than said input voltage, when said transistor is recurrently, alternately turned on and off, the level of said output voltage depending on the duty cycle of said switching transistor states, and a control circuit feeding the base of said switching transistor and including a regulator stage comparing an adjustable fraction of said output voltage to a fixed reference voltage and supplying a regulating current or voltage proportional to the difference between said compared voltages, a pulse-width modulator triggered by means of a recurrent signal and supplying a rectangular signal whose duty cycle varies as a function of said regulating current or voltage, another driver stage receiving the rectangular signal and controlling said switching transistor, the regulation and boosting of said output voltage being controlled by the initially independent starting up of the entire horizontal deflection circuit when supplied by said initial voltage from said power supply device as soon as a DC input voltage is applied thereto and which then delivers recurrent trigger pulses to said pulse-width modulator, one of the supply inputs of said other driver stage receiving directly a first voltage waveform whose positive alternations comprise constant-voltage plateau and whose negative alternations comprise negative-going horizontal flyback pulses provided by a first secondary winding of said line transformer, so as to control the turning off of said switching transistor substantially simultaneously with that of the trace switch transistor.

2. A power supply device as claimed in claim 1, wherein said other driver circuit comprises a third transistor whose emitter is connected to the base of said switching transistor and which is of the same type as the latter, whose collector is connected, through said supply input, to said first secondary winding of said line transformer to receive therefrom said first waveform and whose base is coupled to the output of said pulse-width modulator.

3. A power supply device as claimed in claim 2, wherein the collector of said third transistor is connected, through a resistor to the supply input and its emitter is connected, furthermore, to that of the switching transistor through another resistor so that the negative-going flyback pulses, applied to the collector of said third transistor, control the symmetric (reverse) saturation thereof so as to reversely bias the base-emitter junction of said switching transistor.

4. A power supply device as claimed in claim 2, wherein the collector of said third transistor is connected to said power supply input through a fourth diode conducting in the normal direction of its collector-emitter path, and wherein its emitter is further connected, on the one hand, through a resistor, to the emitter of the switching transistor and, on the other hand, through another resistor and a fifth diode conducting in the reverse direction to that of the base-emitter junction of the switching transistor, so as to transmit to the base thereof negative-going flyback pulses through a voltage divider formed by said two resistors in series.

5. A power supply device as claimed in claim 1, wherein said other driver circuit comprises a third transistor whose emitter is connected to the base of said switching transistor, whose collector is connected to that of this latter so as to form a so-called Darlington circuit and whose base coupled, moreover, to said pulse-width modulator is further connected, through a resistor and a diode in series, to said first secondary winding of said line transformer so as to control the simultaneous turn off of both transistors of said Darlington circuit by simultaneously reversely biasing their respective base-emitter junctions, connected in series, by means of negative-going flyback pulses.

6. A power supply device as claimed in any one of the preceding claims, wherein said pulse-width modulator, supplied at its input with a voltage waveform whose positive alternations comprise positive-going flyback pulses and whose negative alternations comprise constant negative-voltage plateaux, comprises a passive circuit which forms a simple integrator during positive alternations because one of its resistors is shunted by a diode and which is a cascaded double integrator during negative alternations of this waveform so as to deliver during the trace periods of the scan a linearly decreasing negative current which, added to the positive regulating current, supplies the base of a fourth comparator transistor, so that the turning off of this latter through equality of the negative and positive currents supplied to this base controls the beginnings of the saturation of said switching transistor in such a manner that the duration of this saturation varies inversely with variation of said output voltage.

7. A power supply device as claimed in claim 6, wherein said comparator transistor is biased, furthermore, at its base by means of a resistor which connects it to the positive pole of said input voltage source, so that it remains saturated in the absence of flyback pulses supplied by said horizontal deflection circuit so as to maintain the switching transistor in a cut off state.

8. A power supply device as claimed in any one of the preceding claims, wherein said control circuit, except for the regulator stage which is supplied by said output voltage, is supplied by said input voltage.

9. A power supply device as claimed in any one of the preceding claims 1 to 6, wherein said DC supply voltage of said control circuit, with the exception of one of the inputs of said regulator stage receiving said output voltage, is supplied by a secondary winding of said line transformer, through a rectifier circuit including a diode and a filtering capacitor.

Description:

BACKGROUND OF THE INVENTION
The present invention relates to a switching voltage regulator power supply device combined with the horizontal deflection circuit of a television receiver which it supplies with DC voltage. It relates, more particularly, to DC voltage supply devices of the type which boost or increase the voltage supplied at the output of the device in relation to the level of a DC voltage applied to its input and which regulate this level by recurrent switching of this input voltage, this switching being synchronous with the (horizontal) line frequency of the television receiver supplied by this device.
Switched step-up or boost voltage regulator devices of this type are known, particularly from the publications U.S. Pat. Nos. 3,571,697 (or 3,736,496) and they are related to switched mode power supply devices or DC-DC converters of the so-called unisolated flyback type, in which the collector-emitter path of a bipolar switching transistor is connected in series with a commutating inductance between the terminals of a DC source supplying an input voltage and a rectifying diode is connected between the junction of the inductance with the transistor and one of the plates of a filtering or storage capacitor (in parallel with the load), so that the current stored in the inductance during the conducting period of the transistor is used for charging the capacitor (and supplying the load) through the diode during its consecutive cut-off period. The use of a switched-mode power supply device of this type in television receivers for supplying, particularly, the horizontal deflection circuit thereof has been described, for example, in two articles by VAN SCHAIK entitled respectively "AN INTRODUCTION TO SWITCHED-MODE POWER SUPPLIES IN TV RECEIVERS" and "CONTROL CIRCUITS FOR SMPS IN TV RECEIVERS," appearing respectively on pages 93 to 108 of No. 3, Vol. 34, of September 1976 and on pages 162 to 180 of No. 4 of this same volume, of December 1976, in the English language Dutch review "ELECTRONIC APPLICATIONS BULLETIN" of PHILIPS', or on pages 181 to 195 of No. 135 of July 1977 and on pages 210 to 226 of No. 136 of October 1977 of the British review "MULLARD TECHNICAL COMMUNICATIONS." Since none of the switched-mode power supply devices described in these articles, isolated or not from the mains, whether they use a forward or a flyback converter, supplies at its output a DC voltage for supplying the horizontal deflection circuit before the switching transistor has been turned on (saturated or conducting) one or more times, the control circuit of this transistor must comprise an independent relaxation oscillator and must be supplied by the same DC input voltage (rectified and smoothed voltage of the AC mains) as the switching circuit comprising the inductance and the transistor in series. Synchronization of the switching with the horizontal deflection can only occur subsequently, when the horizontal oscillator and/or the horizontal deflection circuit as a whole have begun to operate, as soon as the supply voltage supplied thereto by the device which operates independently on starting up, has become sufficient. This synchronization of the switching with the horizontal deflection, advantageous for reducing or eliminating the interferences visible on the screen which are caused by high-frequency energy radiation due to abrupt transitions of power switching, particularly when the switching transistor is being cutt off, is generally carried out by means of a signal comprising flyback or retrace pulses, taken at the terminals of an auxiliary secondary winding of the line tranformer whose primary winding is generally connected between the output of the switched-mode power supply device and one of the terminals of the trace switch which is provided in the output stage. It is also possible to use for this purpose the signal provided by the horizontal oscillator (see, for example, the publication FR-A-2 040 217).
In a switched-mode supply for a television receiver described in the publication FR-A-2 261 670, the circuit for controlling the switching transistor of a forward-type converter, supplied with the rectified and smoothed voltage of the mains, comprises a bistable trigger circuit of flip-flop one of whose outputs is coupled back to one of its trigger inputs through a regulating circuit comprising a sawtooth voltage generator and a voltage comparator providing transitions which control the setting of the flip-flop, when the sawtooth voltage reaches the level of a voltage proportional to the amplitude of the flyback pulse. The other one of the two complementary outputs of this flip-flop is coupled back to its other trigger input through a so-called starting loop comprising an ascending voltage wave-form which approaches asymptotically a predetermined voltage level smaller than a predetermined fraction of the nominal level which the amplitude of the flyback pulse must reach in normal operation, and a voltage comparator providing transitions which control the recurrent resetting of the flip-flop to its initial state until the flyback pulse has reached or exceeded a threshold amplitude slightly below its nominal amplitude. When this threshold amplitude has been exceeded, resetting of the flip-flop is controlled by the flyback pulses themselves, negative-going in the present case, which supplant the starting pulses. Such an arrangement is equivalent to an astable multivibrator during the starting period, which later becomes a monostable one and triggered by the flyback pulses and whose quasi-stable state has a variable duration, depending on the amplitude of these pulses so as to obtain regulation thereof by the duty cycle. The pulse which controls the closing of the switch (saturation of the switching transistor) begins here with the leading edge of the flyback pulse and its duration or length is modulated as a function of the current drawn by the load and of the variation of the rectified and smoothed voltage, so that its end controlling the opening of the supply switch (cutting off the transistor) occurs during the trace portion of the horizontal deflection. Thus it can be seen that this switched-mode supply, like most of the known ones, effects regulation of its output voltage by varying the duty cycle as a reverse function of the level thereof.
Since the high-frequency radiation is precisely at its most intense during abrupt transitions of current in the switching inductance and of the voltage accross its terminals, the appearance of one or more vertical lines (light or dark according to the sense of the modulation of the carrier wave by the video signal) may be observed, contrasting with the normal contents of the picture, whose location on the screen depends on the duration of the pulse controlling the switching transistor. The effect of this radiation becomes particularly troublesome when the input signal of the radio-frequency stages or tuner is small, particularly when the selected channel is situated in the lower part of the VHF band, for the automatic gain-control device of the receiver acts on the gain of the high-frequency and/or intermediate-frequency input stages, so that the sensitivity (amplification) of the receiver is then maximum and this also as concerns the spurious radiated signals.
SUMMARY OF THE INVENTION
The present invention, on the one hand, avoids or at least appreciably reduces the interferences visible on the screen by controlling the cutting off of the switching transistor in synchronism with the leading edge or the flyback pulse and, on the other hand, the starting of the horizontal deflection circuit by means of a simple circuit without any special oscillator, and provides efficient protection of the switching transistor which remains cut off when the horizontal deflection circuit is not operating. This is made possible by using a step-up switching regulator supply device of the type described in the publication U.S. Pat. No. 3,571,697 and whose control circuit includes, in accordance with the invention, the horizontal deflection circuit, which it supplies.
The object of the present invention is a power supply device with boosting and regulation of its output voltage by switching, combined with a horizontal sweep circuit of a television receiver, which it supplies and which comprises a horizontal oscillator, a driver stage and an output stage including a line transformer, this device comprising an inductance and the collector-emitter path of a switching transistor connected in series between the poles of a DC input voltage source, a rectifiying diode connected by its anode to the junction between the inductance and the collector of the transistor and by its cathode to one of the terminals of a filtering capacitor whose other terminal is connected to the emitter of the transistor so as to supply between its terminals an initial output voltage, slightly lower than the input voltage, when the transistor is cut off permanently, and a regulated DC output voltage with a level higher than the input voltage, when the transistor is recurrently alternately turned on and off, the level of this output voltage depending on the duty cycle of the respective states of this transistor, and a control circuit for driving the base of the transistor and including a regulator stage comparing an adjustable fraction of the output voltage to a fixed reference voltage and supplying a regulating current or voltage proportional to the difference between these compared voltages, to a pulse-width modulator triggered by means a recurrent signal and supplying a rectangular signal whose duty cycle varies as a function of this regulating current or voltage, and another driver stage receiving the rectangular signal and controlling the switching transistor.
In accordance with the invention, the horizontal deflection forming an integral part of the circuit controlling the switching transistor, determines therefor, from the start, the repetition period of the rectangular signal controlling it, and one of the supply inputs of the other driver stage receives directly a first voltage waveform whose positive alternations, comprise DC voltage plateaux and whose negative alternations comprise negative-going flyback pulses supplied by a first secondary winding of the line transformer, so as to control the cut-off the switching transistor substantially simultaneously with that of the trace switch transistor.
DESCRIPTION OF THE DRAWINGS
The invention will be better understood and other of its objects, characteristics, features and advantages will become clear from the following description and the accompanying drawings which refer thereto, given solely by way of example, in which:
FIG. 1 is partly a block diagram and partly a schematic diagram of a power supply device combined with the horizontal deflection circuit in accordance with the invention;
FIG. 2 shows waveforms of two voltages and of a current at different points of the circuit of FIG. 1;
FIG. 3 is a block diagram of the circuit for controlling the switching transistor;
FIGS. 4 and 5 are schematic diagrams of two different embodiments of the driver circuit 20 forming the output stage of the control circuit of FIG. 3;
FIG. 6 is the block diagram of one embodiment of the pulse-width modulator 10 of the circuit of FIG. 3;
FIG. 7 shows three voltage waveforms at different points of the circuit of FIG. 6;
FIG. 8 is a schematic diagram of one embodiment of the pulse-width modulator 10 of the circuit of FIG. 3, using discrete components;
FIG. 9 shows a current waveform and two voltage waveforms at different points of the circuit of FIG. 8;
FIG. 10 is a schematic diagram of a conventional embodiment of a regulator stage 30 adapted to supply the modulation input of the modulator of FIG. 8; and
FIGS. 11 and 12 are partial respective schematic diagrams of two embodiments of a power supply device in accordance with the invention.
DESCRIPTION OF THE PREFERRED EMBODIMENTS
FIG. 1 shows the schematic diagram of the power stages of the power supply device and of the horizontal deflection circuit of the television receiver, which it supplies and in block diagram form the respective circuits which control them.
The DC input voltage VE which is not regulated is supplied by a rectifier bridge R with four diodes, supplied at its input by the secondary winding of an insulating step-down transformer TS, whose primary winding is supplied by the AC mains. The output terminals of rectifier bridge R are connected respectively to the terminals of a first filtering capacitor C1 across which this input voltage VE is taken.
The positive pole P of this source of the input voltage VE is connected to one of the terminals of an energy-storage inductance L, whereas its negative pole N is connected to ground G of the receiver, which is isolated from the mains. The other terminal of inductance L is connected, on the one hand, to the collector of a first NPN bipolar switching transistor T1, whose emitter is connected to ground G and, on the other hand, to the anode of a first diode D1 whose cathode is connected to the positive terminal of a second filtering and storage capacitor C2. With the negative terminal of this second capacitor C2 connected to ground G, the output voltage VS which supplies the load is taken between its terminals.
Such a supply device BS provides both step-up or boost and regulation of its output voltage level, because the first switching transistor T1 and the first diode D1 thereof are connected so as to conduct respectively currents flowing through inductance L in the same direction, it supplies at its output formed by the terminals of the second capacitor C2, an initial DC voltage VSI as soon as the primary winding of the insulating transformer TS is connected to the mains. This initial voltage VSI which is equal to the input voltage VE less the forward voltage drop VD1 across the first diode D1, is then supplied to the load until the control circuit SC is started up, whose output 6 is connected to the base of the first transistor T1 so as to cause it to be alternately turned on and off.
When the first transistor T1 is turned on by positively biasing its base-emitter junction, its collector-emitter path connects the junction of the inductance L with the anode of the first diode D1 to ground G. Diode D1 being then reversely biased, it ceases to conduct and the inductance L connected by the first transistor T1 between the positive P and negative N poles of the source supplying the unregulated DC input voltage VE, then conducts a linearly increasing current IL so as to store the energy which increases with the square of the conduction duration of the first transistor T1, until this latter is cut off. At the instant when the first transistor T1 is cut off after the control circuit SC has brought its base-emitter voltage to zero or below, the voltage at the terminals of inductance L is reversed so that, at its junction with the collector of transistor T1 and the anode of diode D1, there appears a voltage VM greater than the input voltage VE, which results in the forward biasing of diode D1. Consequently, from the instant when transistor T1 is cut off, diode D1 conducts a linearly decreasing current until the energy stored in the form of a current IL in the inductance L, which charges the second capacitor C2 to an output voltage VS greater than the input voltage VE, disappears. The regulation of the level of the output voltage VS is here effected in a conventional way, by varying the duty cycle, i.e. the radio (quotient) between the duration of the conducting period of transistor T1 and the sum of the respective durations of two of its successive conducting and cut off periods, as a function of the desired output voltage VS (determined by comparison to a stable reference voltage).
According to the invention, a supply device BS of the above-described type is combined with the horizontal deflection circuit SH of a television receiver, which it supplies, so that this latter forms an integral part of its control circuit SC and for determining the repetition period of its operation and so that the above-mentioned regulation by varying the duty cycle maintains a stable peak-to-peak amplitude of the sawtooth scanning current and/or the very high voltage for biasing the electrodes (anode, focusing electrode and accelerating grid) of the cathode-ray tube, which are obtained by rectifying the horizontal flyback pulses supplied by a step-up secondary winding (not shown) of the line transformer TL.
The horizontal deflection circuit SH which comprises in cascade the horizontal oscillator OH whose known phase control circuit with respect to the horizontal sync signal separated from the composite video signal has not been shown here, the driver stage HD controlled by the horizontal oscillator OH and controlling the output stage OS of the horizontal deflection, is as a whole supplied by the above-described regulated power supply device BS. In fact, the positive supply input AL of the horizontal deflection circuit SH is connected by means of a fuse FS to the junction of the cathode of the first diode D1 with the positive terminal of the second capacitor C2, which forms the positive output terminal SP of the regulated power supply device BS. This supply input AL is connected directly to that of the driver circuit HD and, preferably, through a conventional Zener diode or series ballast transistor voltage regulator VR, to that of the horizontal oscillator OH, which are moreover connected to the isolated ground G.
The supply input AL of the horizontal deflection circuit SH is furthermore connected to one of the primary winding terminals B1 of the line transformer TL, whose other terminal AB is connected in parallel to the collector of another switching transistor TH, of NPN type, called trace switch transistor, to the cathode of a second so-called shunt recovery diode DR, to one of the terminals of another capacitor CR, called line-retrace capacitor, and to one of the plates of an additional capacitor CS, called trace capacitor, which supplies the horizontal deflection coils LH one terminal of which is connected to its other terminal during the trace periods of the scanning. The emitter of the scanning transistor TH, the anode of the "shunt" recovery diode DR, the other terminal of the retrace capacitor CR and the other terminal of the horizontal deflection coils LH are all connected to ground G. This assembly of components thus connected forms the output stage OS whose operation is well-known and does not form part of the invention.
As was mentioned above, as soon as the primary winding of the step-down isolating transformer TS is connected to the mains, rectifier R supplies the first filtering capacitor C1 so as to provide between its terminals P and N a unregulated low DC voltage VE. With the first transistor T1 then turned off, this input voltage is applied through the inductance L and the first diode D1 to the second capacitor C2 so as to obtain between the terminal SP and ground G an initial output voltage VSI substantially equal to VE-VD1, which is approximately equal to 60 percent of the regulated output voltage VS. This initial output voltage VSI (equal to about 0.6 VS) is sufficient to cause the generation of autonomous oscillations by the horizontal oscillator OH. This latter supplies at its output, connected to the input of driver circuit HD, pulses at an independent frequency close to the line frequency. In response to these pulses, driver circuit HD, also supplied by device BS, provides at the base of the trace switch transistor TH pulses controlling its periodical cut off at this independent frequency and its consecutive turning on after a period greater than the duration of the flyback period, so that the recovery diode DR may take the current from the deflector LH during substantially the first half of the trace portion of the scan. During flyback or retrace, with both transistor TH and diode DR cut off, the energy stored in the form of currents respectively in the inductances of deflector LH and of the primary winding B1 of the line transformer TL which are then, from the AC current point of view, connected in parallel, flow in an oscillating manner through the retrace capacitor CR which forms therewith a parallel resonant circuit whose resonance period determines the duration of the flyback period.
There then appears periodically between point AB and ground G a voltage pulse VTH having substantially a sinusoidal half-wave form, which is shown in Diagram A of FIG. 2. The average value of this voltage VTH being then equal to VSI, at start-up, and to VS, during established operation. The line transformer TL comprises, in addition to a very-high-voltage winding and other windings for supplying rectifying circuits, not shown, two secondary windings B2, B3 respectively supplying across their terminals, voltage waveforms comprising flyback pulses with zero average values and with respectively negative and positive polarities.
This means that the first secondary winding B2 supplies a voltage waveform -VTL which, between two successive flyback pulses, comprises a positive plateau whose level is equal to the average value of these pulses and which is used, in accordance with the invention, to control the turn off of the first transistor T1 so that the interferences which would otherwise be visible only occur during the line-blanking periods comprising the line-retrace periods. The second secondary winding B3 then supplies a voltage waveform +VTL which is the reverse of or complementary to the preceding one -VTL.
One of the terminals of each of these secondary windings B2, B3 is connected to ground G, whereas their other terminals are respectively connected to two inputs 2 and 1 of the control circuit SC. A third input 3 of this latter is connected to the SP output of the supply device BS and a fourth input 4 is connected to the positive pole P of the input voltage source VE. A fifth terminal 5 of the control circuit SC is connected to ground G (or negative pole N) and its output 6 is connected to the base of the first transistor T1. This control circuit SC causes, following the start up of the horizontal deflection circuit SH, a first saturation of the first transistor T1 at a time determined by a pulse-width modulator operating by conventional comparison of a sawtooth voltage waveform the elaboration of which is controlled by a first flyback pulse, with a regulating voltage, depending on the output voltage VS. During this saturation period of transistor T1 which extends as far as the leading edge of the next flyback pulse, energy is stored in inductance L.
From the instant when transistor T1 is turned off, diode D1 transfers this stored energy to the second capacitor C2, at the terminals of which it causes an increase of the voltage VS with respect to its initial value VSI, until the current in diode D1 is canceled out, when it becomes reverse biased.
The collector-emitter voltage waveforms VTH of the trace switch transistor TH and VCE of the switching transistor T1 in established operation have been shown respectively by the diagrams A and B of FIG. 2. Diagram C of FIG. 2 shows the corresponding waveform of the current IL flowing through the inductance L.
When the base of the first transistor T1 receives from the output 6 of the control circuit SC a rectangular signal which turns it on at time instant t1, its collector-emitter voltage VCE (Diagram B) becomes close to zero (V _CEsat ) and a linearly increasing current IL (Diagram C) flows through inductance L from time t1 until time t2 when transistor T1 is again turned off, which is controlled by the leading edge of the flyback pulse VTH (Diagram A). With the collector current of transistor T1 canceled at the end of the storage time of the excess minority carriers in the base, the voltage across the terminals of the inductance L inverses its polarity so as to be added to the input voltage VE, so that the collector-emitter voltage VCE (Diagram B) then reaches a level VM greater than VS (as well as VE), so as to apply forward bias to the first diode D1, which then conducts the current IL through the inductance L. This current IL, from time instant t2 when it reaches its maximum value IM, becomes linearly decreasing and it flows through the first diode D1 in the passing direction in order to recharge the second capacitor C2 and supply, in particular, the horizontal deflection circuit SH.
When the current IL passing through the first diode D1 is canceled out at time t3, the collector-emitter voltage VCE of the first transistor T1 becomes equal to the unregulated input voltage VE until the next turn on of the transistor T1, and the first diode D1 remains reversely biased until the time when this latter is cut off again.
From the above it can be easily seen that the principal advantage of this combined device resides in the fact that a single oscillator OH belonging to the horizontal deflection circuit SH is sufficient for controlling the two power switching transistors TH and T1.
Furthermore, a possible overload in the circuitry of the television receiver, such for example as a short-circuit of the trace switch transistor TH, results in overloading the diode D and the inductance L. The first transistor T1 which is consequently cut off is not subjected to this overload and is therefore protected. In order to protect the rest of the television receiver as well as inductance L and the first diode D1, a fuse FS may be connected in series in the supply line from the second capacitor C2. This fuse FS may also be inserted between pole P and inductance L.
It is moreover known that it is difficult to construct switched supplies for obtaining correct operation when it is not fully charged (for supplying, for example, a ready-state remote-control receiver). In the present case, the problem does not come up since, when the supply is in operation, there is always a minimum load formed by the horizontal deflection circuit. When this circuit is not operating, the supply circuit BS does not operate either, but it supplies an output voltage VSI of a value less than the nominal voltage VS which cannot cause damage and which may, for example, supply a ready-state receiver for television receivers having a remote control.
Finally, the control circuit SC allows transistor T1 to be cut off at the beginning of each flyback period, when the blanking circuit has extinguished the spot (s) on the cathode-ray tube. Thus, the spurious signals radiated into the receiver input circuits will cause no visible effect on the screen of the cathode-ray tube.
FIG. 3 shows in block diagram form the control circuit SC of FIG. 1.
This control circuit SC comprises a pulse-width modulator stage 10 a first input 11 of which, connected to input 1, receives flyback pulses of positive polarity +VTL from the second secondary winding B3 of the line transformer TL (see FIG. 1 and a second input 12 of which receives a so-called regulating voltage or current whose level is proportional to the difference between the actual output voltage VS and a constant reference value, delivered by the output 32 of a regulating circuit or stage 30 whose input 31 is connected through input 3 to the positive output pole SP of the supply device BS supplying the regulated voltage VS. The variation of the regulating current or voltage causes the variation of the time instant when the instantaneous amplitude of a sawtooth voltage waveform, either with substantially constant slope and amplitude, reaches the level of this regulating voltage, or with a slope variable depending of the regulating current (which is added to the current for linearly charging a capacitor), reaches the predetermined level of a fixed reference (threshold) voltage, with respect to the beginning or the end of the sawtooth waveform. Thus a two-level rectangular signal with constant periodicity is generated, whose duty cycle varies as a function of the regulating current or voltage. If it is arranged, which is possible, for a reduction of the output voltage VS with respect to its nominal value defined by the reference voltage, to cause an increase in the duty cycle and for an increase in VS to have the opposite effect, regulation of this output voltage VS is provided, which tends to be stabilized to this nominal value.
The output 14 of modulator 10 supplies a first input 21 of the driver stage 20 of the first switching transistor T1, a second input 22 of which receives the flyback pulses of negative polarity -VTL, coming from the first secondary winding B2 of the line transformer TL.
FIGS. 4 and 5 illustrate two different embodiments of the driver stage 20 of FIG. 3, providing efficient turn off of the first transistor T1.
In FIG. 4, the driver stage 20A comprises a third supply input 23 which connected to the positive pole (P) of the source of the (unregulated) input voltage VE and to one of the terminals of a first resistor R1 (1.8 kiloohms) whose other terminal is connected in parallel to the anodes of two diodes D2 and D3 (of type 1N4148). The second of these diodes D3 has its cathode connected to the base of a third NPN transistor T2 and to one of the terminals of a second resistor R2 (220 ohms). The emitter of the third transistor T2 is connected to the other terminal of the second resistor R2 and to the output 24 of stage 20A, which is connected through the output 6 of the control circuit SC to the base of the first transistor T1. The collector of the second transistor T2 is connected through a third resistor R3 (10 ohms) to the second input 22 of stage 20A receiving the signal -VTL which comprises the negative-going flyback pulses and, between them, plateaux of a constant positive level (zero average value). The base of the first transistor T1 is coupled to its emitter and to ground G, through a fourth resistor R4 (100 ohms). The third transistor T2 is thus mounted as a common collector (emitter-follower) stage.
When the output 14 of modulator 10 (FIG. 3) which is connected to the input 21 of stage 20A supplies a low state (level), i.e. a voltage close to zero, the thus positively biased diode D2 becomes conducting so that its anode will be at a voltage of a few tenths of a volt (0.7+V _CEsat ) which is less than the voltage required for making the three series PN junctions orientated in the same direction conductive, the first of which is formed by the third diode D3, the second is the base-emitter junction of a third transistor T2 and the third that of the first transistor T1, which will thus remain turned off. When, on the other hand, output 14 supplies a high state or forms an open circuit (the output stage of modulator 10 being formed by an open-collector transistor), diode D2 is cut off by its reverse bias and the voltage VE applied to the input 23 causes a current to flow through the first resistor R1, the diode D3 and the respective base-emitter junctions of transistors T2 and T1 connected in series. Under these circumstances and if, at the same time, the voltage waveform -VTL applied to the collector of transistor T3 presents its constant positive level portion, coinciding with the trace periods of the horizontal scan, transistors T2 and T1 become simultaneously saturated with the effect previously described insofar as the supply device BS of FIG. 1 is concerned. On the other hand, when the voltage waveform -VTL applied to the collector of the third transistor T2 becomes negative, during flyback periods, the current then flows between terminals 23 and 22 of driver stage 20 A, through resistor R1, diode D3, the base-collector junction of the third transistor T2 and resistor R3. The third transistor T2 then operates along its symmetrical saturation characteristics, i.e. it is inverted so that its collector becomes emitter and vice versa. It then conducts a current in the reverse direction between ground and the input 22 (negative) through the resistor R4 across the terminals of which it causes, after removal of the excess minority carriers from the base of the first transistor T1 through the third transistor T2, a voltage drop biasing said base negatively with respect to the emitter. This negative voltage applied to the base of reversely saturated transistor T3 allows a considerable reduction in the storage time and a rapid turnoff of the first transistor T1. Since the sawtooth generator of the pulse-width modulator 10 described above is controlled by positive-going flyback pulses, the rectangular signal applied by its output 14 (FIG. 14) to input 21 of stage 20A undergoes, during the flyback period following the turn off of the first transistor T1, a transition from its high state to its low state which causes diode D2 to conduct and, consequently, the third transistor T2 (reversed) to be cut off before the waveform -VTL becomes positive again and rebiases this transistor T2 the right way round.
FIG. 5 shows the schematic diagram of another embodiment of the driver circuit 20 of FIG. 3, designated by 20B, which has only been modified with respect to circuit 20A of FIG. 4 insofar as the collector circuit of the third transistor T2 and the base circuit of the first transistor T1 are concerned.
This modification is more particularly intented for the case where the negative peak amplitude of the voltage waveform -VTL applied to the base of the first transistor T1 through resistor R3 and the emitter-collector path of the reversely saturated third transistor T2, exceeds the reverse (Zener) avalanche-effect breakdown voltage of one of the base-emitter or base-collector junctions of the first transistor T1. This may occur when the first secondary winding B2 of the line transformer TL is also used for other functions in the television receiver.
To prevent the third transistor T2 from being reversely saturated (symmetrically), the circuit 20B comprises a fourth diode D4 inserted between the input 22 receiving the voltage waveform -VTL and the collector thereof, in series with the resistor R3 and connected to conduct in the same direction as its collector-emitter path. The input 22 is more over connected to the cathode of a fifth diode D5 (1N4148) whose anode is connected through a circuit formed by a fifth resistor R5 (330 ohms) and a third capacitor C3 (1nF) connected in parallel, to the base of the first transistor T1.
Diode D5 isolates the base of transistor T1 from the input 22, when the waveform -VTL is positive, and connects them together through a resistive voltage divider formed by resistors R5 and R4 in series, when it becomes negative. Capacitor C3 accelerates the turn-off by favoring the transmission to the base of T1 of abrupt transitions of the negative flybacd pulses.
FIG. 6 is a diagram, partly in block form, of a possible embodiment of the pulse-width modulator 10 of the control circuit SC of FIG. 3. Diagrams D, E and F of FIG. 7 show the voltage waveforms applied respectively to the input 11 (+VTL) and supplied by the output SI (VI) of the sawtooth generator GD and by the output 14 (VP) of circuit 10A.
Modulator 10A of FIG. 5 comprises a sawtooth generator GD formed by a conventional integrator circuit comprising a first amplifier A1 (integrated operational amplifier, for example), an integrating resistor R1 inserted in series between the input 11 receiving the voltage waveform +VTL illustrated by Diagram D of FIG. 7 and supplied by the second secondary winding B3 of the line transformer TL, and the input (inverting) of amplifier A1, as well as an interating capacitor CI connected between this input and the output SI of amplifier A1 (capacitive feedback). In response to this waveform +VTL, the output of amplifier A1 forming the output SI of sawtooth generator GD, supplies a voltage waveform VI illustrated by the diagram E of FIG. 7 which comprises, during the period between time instants t0 and t2 corresponding to the trace period TA of the scan, a voltage decreasing linearly between a maximum value (positive) and a minimum value (negative), and during the flyback intervals preceding time instant t0 and succeding to time instant t2, an increasing voltage of substantially semi-cosinusoidal shape.
Voltage VI is applied to one of the inputs (-) of an analog voltage comparator which may be formed by means of a second differential-type amplifier A2 (integrated operational amplifier), whose other input (+) connected to the input 12 of modulator 10A, receives the regulating voltage VR supplied by the regulator stage (30 of FIG. 3). This regulating voltage VR, which is obtained by comparing the output voltage VS of the supply device BS of the circuit of FIG. 1 with a reference voltage (VZ supplied by a Zener diode, for example), is a DC voltage undergoing slow variations, shown in Diagram E of FIG. 7 by a dash-dot line.
When the waveform VI applied to the inverting input (-) of comparator A2 is greater than the regulating voltage VR, which is the case during the period between time instants t0 and t1, its output connected to the output 14 of modulator 10A provides a low state. When, on the other hand, it (VI) reaches or becomes less than VR, which occurs from the time instant t1, the output 14 of modulator 10A provides a high state (which causes saturation of the first transistor T1). This high state continues until time instant t4 subsequent to the time instant t2 of the beginning of the following flyback pulse whose leading edge controls the turn-off of the first transistor T1, when the waveform VI becomes greater than the regulating voltage VR. Thus there is obtained at the output 14 of modulator 10A a rectangular signal VP shown in Diagram F of FIG. 7, formed successively of a low-level (zero or negative) beginning during the first half of the flyback period TR and ending at time instant t1, and a high level going from time instant t1 to time instant t4. Time instant t1 of the positive transition of signal VP, which determines the beginning of conduction of the first transistor T1 is then situated during the trace period of the scan TA and its position with respect to the beginning t0 or to the end t2 thereof varies as a function of the regulating voltage VR. When the regulating voltage VR is negative (as on the Diagram E of FIG. 7), a predetermined fraction of the output voltage VS is greater than the reference voltage, the duration of the high level state (t2-t1) is less than half of the trace period of the scan T1. In the opposite case, this duration (t2-t1) is greater than TA/2. The modification of this duration (t2-t1) and thus of the duty cycle is carried out in the reverse direction of the variation of the output voltage VS so as to stabilize it at a previously adjusted level, with respect to this reference voltage. The waveform -VTL may also be applied to the input 11 of modulator 10A. In this case, the input of comparator A2 must also be inverted.
To obtain suitable operating limits, while taking into consideration particularly the value of inductance L, the duty cycle or the durations (t2-t1) must vary between 0, the case where the input voltage VE is equal to the nominal output voltage VS, and about two-thirds, the case where the maximum power is supplied for a minimum voltage at the input.
The ratio between the residual alternating voltage (hum) at the output and the alternating voltage at the input must also allow an image to be obtained which is not perturbed for the eye. A value less than or equal to a hundredth for this ratio gives satisfactory results.
FIG. 8 shows the simplified diagram of a practical embodiment (by means of discrete components) of the pulse-width modulator 10 of FIG. 3. Different waveforms of a current I1 and input +VTL and output VP voltages are respectively illustrated by the Diagrams H, J and K of FIG. 9.
The input 11 of modulator 10B of FIG. 3 receives the voltage waveform +VTL which may be suppled either directly by the second secondary winding B3 of line transformer TL, or through a coupling capacitor whose one terminal is connected to the collector of the trace switch transistor TH (see FIG. 1). This input 11 supplies a passive shaping circuit, supplying negative-going (decreasing) sawtooth waveforms during the trace periods of scan T1. This passive circuit comprises a fourth coupling capacitor C4 (0.1μ) one terminal of which is connected to the input 11 and the other of which is connected to one of the terminals of a sixth resistor R6 (10 Kohms). The other terminal of this resistor R6 is connected to one of the terminals of a seventh resistor R7 (5.6 Kohms), to one of the terminals of a fifth capacitor C5 (5.6 nF) and to the anode of a sixth diode D6. The other terminal of capacitor C5 is connected to ground G. The cathode of the sixth diode D6 and the other terminal of resistor R7 are both connected to one of the terminals of an eighth resistor R8 (33 kohms), to that of a ninth resistor R9 (470 ohms), to that of a sixth capacitor C6 (4.7 nF) and to the regulation input 12 of modulator 10B, which is connected to the output 32 of the regulator stage 30 (see FIG. 3). The other terminal of capacitor C6 is connected to ground. The other terminal of resistor R8 is connected to the supply input 13 of modulator 10B receiving the input voltage VE. The other terminal of the ninth resistor R9 is connected to the base of a fourth NPN transistor T3, which forms the voltage comparator stage, whose emitter is connected to ground and whose collector (open), which forms the output 14 of modulator 10 B, is connected to the input 21 of the driver stage 20A (of FIG. 4) or 20B (of FIG. 5), formed by the cathode of the second diode D2. The value of capacitor C6 has been chosen so as to limit the maximum negative voltage applied to the base-emitter junction of transistor T3 to a value less than its reverse avalanche breakdown voltage. When the input voltage waveform +VTL is positive, as during the major portion of the flyback periods TR, diode D6 short-circuits resistor R7 and we have then a simple passive RC integrator formed by resistor R6 in series and two capacitors C5 and C6 in parallel, whose output is connected to the base of transistor T3 through resistor R9. Transistor T3 becomes conducting when its base current IB formed by the sum of currents I1 and I2 becomes positive. The current I1 shown by an arrow in FIG. 8 and on the Diagram H of FIG. 9, results from the application of the +VTL waveform of Diagram J to the above-mentionned simple integrator, during its positive alternation, and to the cascaded double integrator R6, C5, R7, C6 during its negative plateau going from t0 to t2. During this negative voltage plateau of the +VTL signal, the current I1 becomes negative and linearly decreasing. When the instantaneous negative amplitude of current I1 becomes equal to the positive current I2 shown by another arrow in FIG. 8 and by means of a reversed constant level (-I2) shown by a broken line in diagram H of FIG. 7, which occurs at time t1, the base current of transistor T3 is cancelled out and this latter is cut off. Since the current I2 is due for a large part to the regulating current IR supplied by the output of the regulator stage (30 in FIG. 3) and proportional to the error voltage, the duration of the cut-off state (t4-t1) of transistor T3 and, consequently, that (t2-t1) of the saturated state of the first transistor T1 (as well as the duty cycle) will vary reversely to the variation of this current IR. The current IE shown by an arrow in FIG. 8, which flows through the high-value resistor R8 from the input voltage source VE and which is one of the components with IR of current I2, forms a small current for maintaining transistor T3 saturated in the absence of flyback pulses and thus of horizontal deflection. The fact that resistor R8 is supplied by the unregulated input voltage VE allows another parameter to be added for acting on the duty cycle of transistor T3 as a function thereof. Diagram K of FIG. 9 illustrates the rectangular signal VP obtained at the output 14 of the modulator 10B of FIG. 8.
FIG. 10 is a schematic diagram of a conventional regulator stage 30 of the control circuit of FIG. 3. It is formed essentially by a well-known circuit called differential amplifier having two inputs, the first of which receives an adjustable fraction of the voltage to be stabilized, formed, in the present case, by the output voltage VS of the power supply device (BS, FIG. 1) and the second input of which receives a stable reference voltage which is generally generated within this stage (as in most known ballast or switched-mode voltage regulator).
The reference voltage VZ is here produced by means of a Zener diode D7 (of the BZX83C type having a stabilized Zener voltage of 7.5 V) whose cathode is connected to the input 31 receiving the output voltage VS of the device BS (FIG. 1) and whose anode is connected through an eleventh resistor R11 (10 Kohms) to ground G. The second input of the differential amplifier used here is formed by the emitter of a fifth PNP transistor T4 which is connected to the anode of the Zener diode D7. The voltage (VS-VZ) biasing this emitter is then fixed with respect to the output voltage VS. The first input of the differential amplifier is here formed by the base of transistor T4 which is biased by a voltage-divider circuit, formed from a fifteenth resistor R15 (4.7 Kohms), a potentiometer R16 (5 Kohms) and a fourteenth resistor R14 (22 Kohms) connected in series between the input terminal 31 and ground G. The base of transistor T4, connected to the slider of potentiometer R16 receives then a previously adjusted fraction of the output voltage VS supplying the horizontal deflection circuit (SH), so that it forms a constant current generator supplying a current proportional to its emitter-base voltage which is equal to the difference (error voltage) between the reference voltage VZ and the selected fraction of the output voltage VS supplied by potentiometer R16. The collector of the fourth transistor T4, connected by a tenth resistor R10 (2.2 Kohms) to the output 32, supplies then the regulating current IR to the regulating input (12, FIGS. 3 and 8) of the pulse-width modulator (10 or 10B, FIGS. 3 and 8).
It will be noted here that a feedback circuit comprising a twelfth resistor R12 (5.6 Kohms) and a seventh capacitor C7 (4.7 nF) in series connects the collector of transistor 14 to its base.
The difference between the voltage respectively provided by the potentiometer R16 and the Zener diode D7 causes more or less heavy conduction of transistor T4 which delivers the current IR.
In short, when the output voltage VS increases, the voltage (VS-VZ) at the emitter of transistor T4 increases more than that applied to its base and current IR increases. The value of I1 at which transistor T3 is cut off increases then in absolute value and this transistor T3 is turned off later, which reduces the conducting period of transistor T1. The peak current in inductance L then diminishes, which causes a reduction of the output voltage VS which comes back to its nominal value, taking into account the residual error required for controlled operation.
FIG. 11 shows the complete simplified diagram of a power supply device BS of FIG. 1 whose control circuit SCA is respectively formed by the driver circuit 20A of FIG. 4, by the modulator 10B of FIG. 8 and the regulator stage 30 of FIG. 10, except for a few variations.
The variations concern a damping resistor R17 of 1 kiloohm shunting the inductance L, resistor R8 and resistor R10 which are both connected directly to the base of transistor T3 instead of being connected to the cathode of diode D6, resistor R11 which has been omitted and a resistor R13 which shunts the slider of potentiometer R16 to ground. These details of construction have no influence at all on the operation of the circuit such as it has been described above, but simply allow easier adjustment.
Another embodiment is shown in FIG. 12. It allows more especially a television set to be supplied with power in which the horizontal deflection circuit operates from a higher DC voltage VS, of about 100 volts for example, itself obtained from an initial output voltage VSI of about 60 volts. The operation of the circuit is fundamentally the same as that of FIG. 11 and only the differences will be described below. The components playing the same role in both diagrams bear the same references. The values may however be different but their dimensioning is within the scope of a man skilled in the art. The voltage VS delivered by the power supply is used principally in the horizontal deflection circuit which is the component consuming most power in the television set. The power supply circuit components receiving permanently a voltage when the horizontal deflection circuit is not operating, but when the mains is connected, are solely those indispensable for activating the power supply, i.e. the first switching transistor T1 and the circuit for measuring the output voltage in the regulator stage 300.
To simplify the driver stage 100, instead of the single switching transistor T1, an integrated Darlington circuit T10 is used of the BU 807 type, for example. Therefore, the gain is sufficient to omit a discrete driver transistor T2 and to connect the cathode of diode D3 directly to the base input of T10. The negative -VTH pulses, coming from an intermediate tapping on coil B2 of the line output transformer, are applied directly to the base of T10 through resistor R3 which is connected in series with a diode D9 whose cathode is connected to this intermediate tapping.
Instead of the input voltage VE, the power supply input 4 of the control circuit SCB is fed by a voltage obtained by rectifying the positive half-waves (plateaux) of the -VTL voltage supplied by the first secondary winding B2, by means of a diode D8 and a capacitor C8. Thus considerably lower voltage may be obtained than that supplying the horizontal deflection circuit, of the order of 13 volts, for example. A voltage of this value allows video amplification circuits as well as other circuits of the television set to be supplied while providing for these latter a very great reliability. This voltage is applied through resistor R1 to the anodes of diodes D2 and D3 and through resistor R8 to the base of the transistor T3 of modulator 10B.
The regulator stage 300 here comprises two PNP transistors T4 and T5 connected differentially. For that, their emitters receive the voltage rectified by D8 through a resistor R18 of 1.5 kiloohms. The collector of transistor T5 is connected to ground through a resistor R20 of 3.9 kiloohms and the collector of transistor T4, which supplies the regulating current IR, is connected to the cathode of diode D6 through a resistor R10 of 4.7 kiloohms.
The reference voltage (6.2 volts) is supplied by a Zener diode D7 whose anode is connected to ground, and cathode to a resistor R19 (6.8 kiloohms) which receives the voltage rectified by D8. This reference voltage is applied to the base of transistor 14. A capacitor C9 (49 microfarads) shunts diode D7 so as to cause the reference voltage to rise gradually when the apparatus is switched on, which allows a gradual rise of the output voltage VS to be obtained.
A potentiometer R16 of 10 kiloohms connected between two stopper resistors R15 (68 kiloohms) and R14 (5.6 kiloohms) receives the voltage VS through the resistor R15 and is connected to ground through resistor R14. The sliding contact of potentiometer R16 allows a fraction of the voltage VS to be applied to the base T5. A resistor R13 (47 kiloohms) also connects this base to the common point between R15 and R16.
An anti-oscillation capacitor C10 (15 nanofarads) connects the base of the collector of transistor T5.
Thus the regulating current IR supplied by resistor R10 is directly dependent on the difference between the output voltage VS, applied to the horizontal deflection circuit, and the reference voltage determined by the Zener diode D7. The power supply BS thus stabilizes this voltage VS and at the same time the rectified voltage supplied by diode D8.
To stop this power supply, as well as that of FIG. 11 moreover, it is sufficient to stop by means of a remote control receiver, for example, the operation of the horizontal oscillator.
In this case, the input voltage VE is still present, but is considerably smaller than voltage VS. For the power supply of FIG. 12, this reduced voltage is only applied to the Darlington transistor T10 and a fraction thereof to the base of transistor T5 of the regulator stage 300. Thus the life expectation of the other components of the device BS is increased. Since the voltage supplied by diode D8 is itself regulated, it may be used for supplying a major portion of the television set, except for the horizontal deflection circuit supplied by voltage VS and the remote control receiver which must be capable of operating permanently (also in the ready state) so as to detect the turn-on control signal. The protection which was mentioned earlier on is then extended to the greatest part of the components of the television set.
It will be noted here that the three stages 10, 20 and 30 of control circuit SC (see FIGS. 1 and 3) may be formed by means of circuits different from those described and shown and which are known per se, and that it is sufficient to have a secondary winding B2 (in addition to the very-high-voltage winding) of the line transformer TL, supplying negative line-flyback pulses which may be used for generating a decreasing or increasing sawtooth voltage waveform as well as for controlling the cutting off of the first switching transistor T1.

TDA3562A (Philips) PAL/NTSC ONE-CHIP DECODER
DESCRIPTION


The TDA3562A is a monolithic IC designed as
decode PAL and/or NTSC colour television standards
and it combines all functions required for the
identification and demodulation of PAL and NTSC
signals.


.CHROMINANCE SIGNAL PROCESSOR

.LUMINANCE SIGNAL PROCESSING WITH
CLAMPING

.HORIZONTAL AND VERTICAL BLANKING
.LINEAR TRANSMISSION OF INSERTED
RGB SIGNALS
.LINEAR CONTRAST AND BRIGHTNESS
CONTROL ACTING ON INSERTED AND MATRIXED
SIGNALS
.AUTOMATIC CUT-OFF CONTROL
.NTSC HUE CONTROL

FEATURES

· A black-current stabilizer which
controls the black-currents of the
three electron-guns to a level low
enough to omit the black-level
adjustment

· Contrast control of inserted RGB
signals

· No black-level disturbance when
non-synchronized external RGB
signals are available on the inputs
· NTSC capability with hue control.


APPLICATIONS
· Teletext/broadcast antiope

· Channel number display.


GENERAL DESCRIPTION



It follows that the
external switches and filters which
are required for the TDA3562A are
not required for the TDA3566A.
There is no difference between the
amplitudes of the colour output
signals in the PAL or NTSC mode.

· The clamp capacitor at pins 10, 20
and 21 in the black-level
stabilization loop can be reduced to
100 nF provided the stability of the
loop is maintained. Loop stability
depends on complete application.
The clamp capacitors receive a
pre-bias voltage to avoid coloured
background during switch-on.

· The crystal oscillator circuit has
been changed to prevent parasitic
oscillations on the third overtone of
the crystal. Consequently the
optimum tuning capacitance must
be reduced to 10 pF.

· The hue control has been improved
(linear)

THE PHILIPS TDA3562A Circuit arrangement for the control of a picture tube :

 1. Circuit arrangement for the control of at least one beam current in a picture tube by a picture comprising
a control loop which in one sampling interval obtains a measuring signal from the value of the beam current on the occurrence of a given reference level in the picture signal, stores a control signal derived therefrom until the next sampling interval and thereby adjusts the beam current to a value preset by a reference signal.
and a trigger circuit which suppresses auxiliary pulses used to generate the beam current after the picture tube has been started up and issues a switching signal for the purpose of closing the control loop during the sampling intervals and for releasing the control of the beam current by the picture signal after the measuring signal has exceeded the threshold value,
a change detection arrangement which delivers a change signal when the stored signal has assumed a largely constant value, and
a logic network which does not release the control of the beam current by the picture signal outside the sampling intervals until the change signal has also been issued after the switching signal.


2. Circuit arrangement as set forth in claim 1, in which the picture signal comprises several color signals for the control of a corresponding number of beam currents for the display of a color picture in the picture tube and the control loop stores a part measuring signal or a part control signal derived therefrom for each color signal, characterized in that the change detection arrangement includes a change detector for each color signal which delivers a part change signal when the relevant stored signal has assumed a largely constant value, and the logic network does not release the control of the beam currents by the color signals outside the sampling intervals until the part change signals have been delivered by all change detectors.

3. Circuit arrangement as set forth in claim 1, including a comparator arrangement which compares the measuring signal with the reference signal and derives the control signal from this comparison, characterized in that the change detection arrangement detects a change in the control signal with respect to time and issues the change signal when the control signal has assumed a largely constant value.

4. Circuit arrangement as set forth in claims 1, 2, 3 including a control signal memory which contains at least one capacitor, characterized in that the change detection arrangement delivers the change signal when a charge-reversing current of the capacitor occuring during the starting up of the picture tube falls below a limit value.

5. Circuit arrangement as set forth in claim 2, including a comparator arrangement which compares the measuring signal with the reference signal and derives the control signal from this comparison, characterized in that the change detection arrangement detects a change in the control signal with respect to time and issues the change signal when the control signal has assumed a largely constant value.

Description:

BACKGROUND OF THE INVENTION

The invention relates to a circuit arrangement for the control of at least one beam current in a picture tube by a picture signal with a control loop which in one sampling interval obtains a measuring signal from the value of the beam current on the occurrence of a given reference level in the picture signal, stores a control signal derived therefrom until the next sampling interval and by this means adjusts the beam current to a value preset by a reference signal, and with a trigger circuit which suppresses auxiliary pulses used to generate the beam current after the picture tube is turned on and issues a switching signal for the purpose of closing the control loop during the sampling intervals and releasing the control of the beam current by the picture signal after the measuring signal has exceeded a threshold value.
Such a circuit arrangement has been described in Valvo Technische Information 820705 with regard to the integrated color decoder circuit PHILIPS TDA3562A and is used in this as a so-called cut-off point control. In the known circuit arrangement, such a cut-off point control provides automatic compensation of the so-called cut-off point of the picture tube, i.e. it regulates the beam current in the picture tube in such a way that for a given reference level in the picture signal the beam current has a constant value despite tolerances and changes with time (aging, thermal modifications) in the picture tube and the circuit arrangement, thereby ensuring correct picture reproduction.
Such a blocking point control is particularly advantageous for the operation of a picture tube for the display of color pictures because in this case there are several beam currents for different color components of the color picture which have to be in a fixed ratio with one another. If this ratio changes, for example, as the result of manufacturing tolerances or ageing processes, distortions of the colors occur in the reproduction of the color picture. The beam currents, therefore, have to be very accurately balanced. The said cut-off point control prevents expensive adjustment and maintenance time which is otherwise necessary.
Conventional picutre tubes are constructed as cathode-ray tubes with hot cathodes which require a certain time after being turned on for the hot cathodes to heat up. Not until a final operating temperature has been reached do these hot cathodes emit the desired beam currents to the full extent, while gradually rising beam currents occur in the time interval when the hot cathodes are heating up. The instantaneous values of these beam currents depend on the instantaneous temperatures of the hot cathodes and on the accelerating voltages for the picture tube which build up simultaneously with the heating process and are undefined until the end of the heating time. After the picture tube is turned on, these values initially produce a highly distorted picture until the beam currents have attained their final value. These picture distortions after the picture tube is turned on are even further intensified by the fact that the cut-off point control is not yet adjusted to the beam currents which flow after the heating time is over.
For the purpose of suppressing distorted pictures during the heating time of the hot cathodes, the known circuit arrangement has a turn-on delay element operating as a trigger circuit which, in essence, contains a bistable flip-flop. When the picture tube and the circuit arrangement controlling the beam currents flowing in it are turned on, the flip-flop is switched into a first state in which it interrupts the supply of the picture signal to the picture tube. Thus, during the heating time the beam currents are suppressed, and the picture tube does not yet display any picture. In sampling intervals which are provided subsequent to flybacks of the cathode beam into an initial position on the changeover from the display of one picture to the display of a subsequent picture and even within the changeover, that is outside the display of pictures, the picture tube is controlled for a short time in such a way that beam currents occur when the hot cathodes are sufficiently heated up and an accelerating voltage is resent. If these currents exceed a certain threshold value, the flip-flop circuit switches into a second state and releases the picture signal for the control of the beam currents and the cut-off point control.
It is found, however, that the picture displayed in the picture tube immediately after the switching over of the flip-flop is still not fault-free. Because, in fact, the beam currents are supported during the heating time of the hot cathodes, the cut-off point control cannot respond yet. This response of the cut-off point control takes place only after the beam currents are switched on, i.e. after the flip-flop is switched into the second state and therefore at a time in which the picture signal already controls the beam currents. In this way the response of the blocking point control makes its presence felt in the picture displayed.
With the known circuit arrangement the brightness of the picture gradually increases, during the response of the cut-off point control, from black to the final value.
This slow increase in the picture brightness after the tube is turned on is disturbing to the eyes of the viewer not only in the case of the black-and-white picture tubes with one hot cathode, but especially so in the case of colour picture tubes which usually have three hot cathodes. With a color picture tube, color purity errors can also occur in addition to the change in the picture brightness if, as a result of different speeds of response of the cut-off point control for the three beam currents, there are found to be intermittent variations from the interrelation between the beam currents required for a correct picture reproduction.

SUMMARY OF THE INVENTION
The aim of the invention is to create a circuit arrangement which suppresses the above-described disturbances of brightness and color of the displayed picture when the picture tube is being started.
The invention achieves this aim in that a circuit arrangement of the type mentioned in the preamble contains a change detection arrangement which emits a change signal when the stored signal has assumed an essentially constant value, and a logic network which does not release the control of the beam current by the picture signal until the change signal has also been emitted after the switching signal.
In the circuit arrangement according to the invention, therefore, the display of the picture is suppressed after the picture tube is turned on until the cut-off point control has responded. If the picture signal then starts to control the beam current, a perfect picture is displayed immediately. In this way, all the disturbances of the picture which affect the viewer's pleasure are suppressed. The circuit arrangement of the invention is of simple design and can be combined on one semiconductor wafer with the existing picture signal processing circuits and also, for example, with the known circuit arrangement for cut-off point control. Such an integrated circuit arrangement not only requires very little space on the semiconductor wafer, but also needs no additional external leads. Thus the circuit arrangement of the invention can be arranged, for example, in an integrated circuit which has precisely the same external connections as known integrated circuits. This means that an integrated circuit containing the circuit arrangement of the invention can be directly incorporated in existing equipment without the need for additional measures.
In one embodiment of the said circuit arrangement, in which the picture signal contains several color signals for the control of a corresponding number of beam currents for representing a color picture in the picture tube and, for each color signal, the control loop stores a part measuring signal or a part control signal derived from it, the change detection arrangement contains a change detector for each color signal which emits a part change signal when the relevant stored signal has assumed an essentially constant value, and the logic network does not release the control of the beam currents by the color signals outside the sampling intervals until the part change signals have been emitted from all change detectors.
In principle, therefore, such a circuit arrangement has three cut-off point controls for the three beam currents controlled by the individual color signals. To reduce the cost of the circuitry, the measuring stage is common to all the cut-off point controls, as in the known circuit arrangement. All three beam currents are then measured successively by this measuring stage. In this way, a part measuring signal or a part control signal derived from it is obtained for each beam current and is stored sesparately according to which of the beam currents it belongs. Changes in the part measuring signal or part control signal are detected for each beam current by one of the change detectors each time. Each of these change detectors issues a part change signal to the logic network. The latter does not release the control of the beam currents by the picture signal outside the sampling intervals until all the part change signals indicate that the part measuring signal or the part control signal, as the case may be, remains constant. This ensures that the cut-off point controls for the beam currents of all color signals have responded when the picture appears in the picture tube.
In a further embodiment of the circuit arrangement according to the invention with a comparator arrangement which compares the measuring signal with the reference signal and derives the control signal from this comparison, the change detection arrangement detects a change in the control signal with respect to time and issues the change signal when the control signal has assumed an essentially constant value. In the case of the representation of a color signal the comparator arrangement derives several part control signals, whose changes with time are detected by the change detectors, from a corresponding comparison of the part measuring signals with the reference signal. In this embodiment of the circuit arrangement of the invention, preference is given to storage of only the control signal or the part control signals for the purpose of controlling the beam currents.
In another embodiment of the circuit arrangement of the invention which includes a control signal memory which contains at least one capacitor in which a charge or voltage corresponding to the control signal is stored, the change detection arrangement issues the change signal when a charge-reversing current of the capacitor occurring during the turning on of the picture tube has fallen below a limit value and has thus at least largely decayed. Such a detection of the steady state of the cut-off point control is independent of the actual magnitude of the control signal and therefore independent of, for example, the level of the picture tube cut-off voltage, circuit tolerances or ageing processes in the circuit arrangement or the picture tube.

Detection of whether or not the charge-reversing current exceeds the limit value is performed preferentially by a current detector which is designed with a current mirror system which is arranged in a supply line to a capacitor acting as a control signal store. A current mirror arrangement of this kind supplies a current which coincides very precisely with the charging current of the capacitor. This current is then compared, preferably in a further device contained in the change detection arrangement, with a current representing a limit value or, after conversion into a voltage, with a voltage representing the limit value. The change signal is obtained from the result of this comparison.
On the other hand, digital memories may also be used as control signal memories, especially when the picture signal is supplied as a digital signal and the blocking point control is constructed as a digital control loop. In such a case, the comparator arrangement, the change detection arrangement and the trigger circuit are also designed as digital circuits. Then, the change detection arrangement advantageously forms the difference of the signals stored in the control signal memory in two successive sampling intervals and compares this with the limit value formed by a digital value. If the difference falls short of the limit value, the change signal is issued.


BRIEF DESCRIPTION OF THE DRAWINGS

An embodiment of the invention is described in greater detail below with the aid of the drawings in which:

FIG. 1 shows a block circuit diagram of the embodiment,
FIG. 2 shows a somewhat more detailed block circuit diagram of the embodiment,
FIG. 3 shows time-dependency diagrams of some signals occurring in the circuit diagram shown in FIG. 2, and
FIG. 4 shows a somewhat moredetailed block circuit diagram of a part of the circuit diagram shown in FIG. 2.

DETAILED DESCRIPTION OF THE INVENTION
FIG. 1 shows a block circuit diagram of a circuit arrangement to which a picture signal is fed via a first input 1 of a combinatorial stage 2. From the output 3 of the combinatorial stage 2 the picture signal is fed to the picture signal input of a controllable amplifier 5 which at an output 6 issues a current controlled by the picture signal. This current is fed via a measuring stage 7 to a hot cathode 8 in a picture tube 9 and forms therein a beam current of a cathode ray by means of which a picture defined by the picture signal is displayed on a fluorescent screen of the picture tube 9.
The measuring stage 7 measures the current fed to the hot cathode 8, i.e. the the beam current in the picture tube 9, and at a measuring signal output 10, issues a measuring signal corresponding to the magnitude of this current. This is fed to a measuring signal input 11 of a comparator arrangement 12 to which a reference signal is supplied at a reference signal input 13. In a preferably periodically recurring sampling interval during the occurrence of a given reference level in the picture signal, the comparator arrangement 12 forms a control signal from the value of the measuring signal fed to the measuring signal input 11 at this time, on the one hand, and the reference signal, on the other, by means of substraction and delivers this at a control signal output 14. From there the control signal is fed to an input 15 of a control signal memory 16 and is stored in the latter. The control signal is fed via an output 17 of the control signal memory 16 to a second input 18 of combinatorial stage 2 in which it is combined with the picture signal, e.g. added to it.
The combinatorial stage 2, the controllable amplifier 5, the measuring stage 7, the comparator arrangement 12 and the control signal memory 16 form a control loop with which the beam current is guided towards the reference signal in the sampling interval during the occurrence of the reference level in the picture signal. For the reference level, use is made in particular of a black level or a level with small, fixed distance from the black level, i.e. a value in the picture signal which produces a black or almost back picture area in the displayed picture in the picture tube. In this case the control loop, as described, forms a cut-off point control for the picture tube. If the reference level is away from the black level, the control loop is also designated as quasi-cut-off-point control.
The circuit arrangement as shown in FIG. 1 also has a trigger circuit 19 to which the measuring signal from the measuring signal output 10 of measuring stage 7 is fed at a measuring signal input 20. When the circuit arrangement and therefore the picture tube are turned on, the trigger circuit 19 is set in a first state in which by means of a first connection 21 it blocks the comparator arrangement 12 in such a way that the latter delivers no control signal or a control signal with the value zero at its control signal output 14. This prevents the control signal memory 16 from storing undefined values for the control signal at the moment of turning on or immediately thereafter.

The circuit arrangement shown in FIG. 1 also has a logic network 22 which is connected via a second connection 23, by means of which a switching signal is supplied, with the trigger circuit 10 and via a third connection 24 with the controllable amplifier 5. Like the trigger circuit 19, the logic network 22 also finds itself controlled, when the circuit arrangement is being turned on, by the switching signal in a first stage in which by way of the third connection 24 it blocks the controllable amplifier 5 with a blocking signal in such a way that no beam currents controlled by the picture signal can yet flow in the picture tube 9. Thus the picture tube 9 is blanked; no picture is displayed yet.
When picture tube 9 is turned on, the hot cathode 8 is still cold so that no beam current can flow anyhow. The hot cathode 8 is then heated up and, after a certain time, begins gradually to emit electrons as the result of which a cathode ray and therefore a beam current can form. However, during the heating up of the hot cathode 8, and because the cut-off point control has not yet responded, this would be undefined and is therefore suppressed by the controllable amplifier 5. Only in time intervals which are provided immediately subsequent to flybacks of the cathode rays into an initial position at the changeover from the display of one image to that of a subsequent image, but even before the start of the display of the subsequent image, the controllable amplifier 5 delivers a voltage in the form of an auxiliary pulse for a short time at its output 6, and when the hot cathode 8 in the picture tube 9 is heated up sufficiently, this voltage produces a beam current. The time interval for the delivery of this voltage is selected in such a way that a cathode ray produced by its does not produce a visible image in the picture tube 9, and coincides for example with the sampling interval.

The measuring stage 7 measures the short-time cathode current produced in the manner described and, at its measuring signal output 10, delivers a corresponding measuring signal which is passed via measuring signal output 20 to the trigger circuit 19. If the measuring signal exceeds a definite preset threshold value, the trigger circuit 19 is switched into a second state in which it releases the comparator arrangement 12 via the first connection 12 and, by means of the second connection 23, uses the switching signal to also bring the logic network 22 into a second state. The comparator arrangement 12 now evaluates the measuring signal supplied to it via the measuring signal input 11, i.e. it forms the control signal as the difference between the measuring signal and the reference signal supplied via the reference signal input 13. The control signal is transferred via the control signal output 14 and the input 15 into the control signal memory 16. It is subsequently fed via the output 17 of the control signal memory 16 to the second input 18 of the combinatorial stage 2 and is there combined with the picture signal at the first input 1, e.g. is superimposed on it by addition. This superimposed picture signal is fed to the picture signal input 4 of the controllable amplifier 5 via the output 3 of the combinatorial stage 2.
In the second state of the logic network 22 the controllable amplifier 5 is switched via the third connection 24 by the blocking signal in such a way that the picture signal controls the beam currents only during the sampling intervals and that, for the rest, no image appears yet in the picture tube. The cut-off point control now gebins to respond, i.e. the value of the control signal is changed by the control loop comprising the combinatorial stage 2, the controllable amplifier 5, the measuring stage 7, the comparator arrangement 12 and the control signal memory 16 until such time as the beam current in the picture tube 9 at the blocking point or at a fixed level with respect to it is adjusted to a value preset by the reference signal. For this purpose the sampling interval, in which the picture signal controls the beam current via the controllable amplifier 5 is selected in such a way that within it the picture signal just assumes a value corresponding to the cut-off point or to a fixed level with respect to it.

During the response of the cut-off point control the control signal fed to the control signal memory 16 changes continuously. Between the control signal output 14 of the comparator arrangement 12 and the input 15 of the control signal memory 16 is inserted a changed detection arrangement 25 which detects the variations of the control signal. When the cut-off point control has responded, i.e. the control signal has assumed a constant value, the change detection arrangement 25 delivers a change signal at an output 26 which indicates that the steady stage of the cut-off point control is achieved and the said signal is fed to a change signal input 27 of the logic network 22. The logic network then switches into a third state in which via the third connection 24 it enables the controllable amplifier 5 in such a way that the beam currents are now controlled without restriction by the picture signal. Thus a correctly represented picture appears in the picture tube 9.
A shadow-like representation of individual constituents of the circuit arrangement in FIG. 1 is used to indicate a modification by which this circuit arrangement is equipped for the representation of color pictures in the picture tube 9. For example, three color signals are fed in this case as the picture signal via the input 1 to the combinatorial stage 2. Accordingly, the input 1 is shown in triplicate, and the combinatorial stage 2 has a logic element, e.g. an adder, for example of these color signals. The controllable amplifier 5 now has three amplifier stages, one for each of the color signals, and the picture tube now contains three hot cathodes 8 instead of one so that three independent cathode rays are available for the three color signals.
However, to simplify the circuit arrangement and to save on components, only one measuring stage 7 is provided which measures all three beam currents successively. Also, the comparator arrangement 12 forms part control signals from the successively arriving part measuring signals for the individual beam currents with the reference signal, and these part control signals are allocated to the individual color signals and passed on to three storage units which are contained in the control signal memory 16. From there, the part control signals are sent via the second input 18 of the combinatorial stage 2 to the assigned logic elements.
The circuit arrangement thus forms three independently acting control loops for the cut-off point control of the individual color signals, in which case only the measuring stage 7 and to some extent at least the comparator arrangement 12 are common to these control loops.
The change detection arrangement 25 now has three change detectors each of which detects the changes with time of the part control signals relating to a color signal. Then via the output 26 each of these change detectors delivers a part change signal to the change signal input 27 of the logic network 22. These part change signals occur independently of one another when the relevent control loop has responded. The logic network 22 evaluates all three part change signals and does not switch into its third stage until all part change signals indicate a steady state of the control loops. Only then, in fact, is it ensured that all the color signals from the beam currents controlled by them are correctly reproduced in the picture tube, and thus no distortions of the displayed image, especially no color purity errors, occur. The color picture displayed then immediately has the correct brightness and color on its appearance when the picture tube is turned on.

FIG. 2 shows a somewhat more detailed block circuit diagram of an embodiment of a circuit arrangement equipped for the processing of a picture signal containing three colour signals. Three color signals for the representation of the colors red, green and blue are fed to this circuit arrangement via three input terminals 101, 102, 103. A red color signal is fed via the first input terminal 101 to a first adder 201, a green colour signal is fed via the second input terminal to a second adder 202, and a blue colour signal is fed via the third input terminal 103 to a third adder 203. From outputs 301, 302 and 303 of the adders 201, 202, 203 the color signals are fed to amplifier stages 501, 502 and 503 respectively. Each of the amplifier stages contains a switchable amplifier 511, 512 and 513, an output amplifier 521, 522 and 523 as well as a measuring transistor 531, 532 and 533 respectively. The emitters of these measuring transistors 531, 532, 533 are each connected to a hot cathode 801, 802, 803 of the picture tube 9 and deliver the cathode currents, whereas the collectors of measuring transistors 521, 532, 533 are connected to one another and to a first terminal 701 of a measuring resistor 702 the second terminal of which 703 is connected to earth. The current gain of the measuring transistors 531, 532 and 533 is so great that their collector currents coincide almost with the cathode currents. By measuring the voltage drop produced by the cathode currents at the measuring resistor 802 it is then possible to measure the cathode currents and therefore the beam currents in the picture tube 9 with great accuracy.
The falling voltage at the measuring resistor 702 is fed as a measuring signal to an input 121 of a buffer amplifier 120 with a gain factor of one, at the output 122 of which the unchanged measuring signal is therefore available at low impedance. From there it is fed to a first terminal 131 of a reference voltage source 130 which is connected with its second terminal 132 to inverting inputs 111, 112 and 113 of three differential amplifiers 123, 124, 125 respectively. The differential amplifiers 123, 124, 125 also each have a non-inverting input 114, 115, and 116 respectively. These are connected to each other at a junction 117, to earth via a leakage current storage capacitor 126 and to the output 122 of the buffer amplifier 120 via decoupling resistor 118 and a leakage current sampling switch 119. In addition, the input 121 of the buffer amplifier 120 can be connected to earth via a short-circuiting switch 127.

From outputs 141, 142, and 143 respectively of the differential amplifiers 123, 124 and 125, part control signals relating to the individual color signals are fed in the form of electrical voltages (or, in some cases, charge-reversing currents) via control signal sampling switches 154, 155 and 156, in the one instance, to first terminals 151, 152 and 153 respectively of control signal storage capacitors 161, 162, 163 which form the storage units of the control signal memory 16 and store inside them charges corresponding to these voltages (or formed by the charge-reversing currents). In the other instance, the part control signals are fed to second inputs 181, 182 and 183 of the first, second or third adders 201, 202, 203 respectively and are added therein to the color signals from the first, second or third input terminals 101, 102 or 103 respectively.

The operation of the comparator arrangement 12 which consists mainly of the buffer amplifier 120, the reference voltage source 130 and differential amplifiers 123, 124, 125 will be explained below with the aid of the pulse diagrams in FIG. 3. FIG. 3a shows a horizontal blanking signal for a television signal which, as the picture signal, controls the beam currents in the picture tube 9. In this diagram, H represents horizontal blanking pulses which follow one another in the picture signal at the time interval of one line duration and by means of which the beam currents are switched off during line flyback between the display of the individual picture lines in the picture tube. FIG. 3b shows a vertical blanking pulse V by means of which the beam currents are switched off during the change ober from the display of one picture to the display of the next picture. FIG. 3c shows a measuring signal control pulse VH which is formed from a vertical blanking pulse lengthened by three line duration.
The short-circuiting switch 127 is now controlled in such a way that it is non-conducting only throughout the duration of the measuring signal control pulse VH and during the remaining time short-circuits the input 121 of the buffer amplifier 120 to earth. This means that a measuring signal only reaches the comparator arrangement 12 during frame change so that the parts of the picture signal which control the beam currents producing the picture in the picture tube exert no influence on comparator arrangement 12 and therefore on the blocking point control.

Throughout the duration of the measuring signal control pulse VH, the measuring signal from output 122, reduced by a reference voltage issued by the reference voltage source 130 between its first 131 and its second terminal 132, is present at the inverting inputs 111, 112, 113 of differential amplifiers 123, 124, 125. If the differential amplifiers 123, 124, 125 were not present, this difference would be fed directly as part control signals to the control signal storage capacitors 161, 162, 162. The differential amplifiers 123, 124, 125 amplify the difference and thus form the control amplifiers of the control loops.
The comparator arrangement 12 further contains a device for compensation of the influence of any leakage currents occurring in the picture tube 9. For this purpose, a voltage to which the leakage current storage capacitor 126 is charged is fed to the non-inverting inputs 114, 115, 116 of the three differential amplifiers 123, 124 and 125. The charging is performed by the measuring signal from output 122 of the buffer amplifier 120 via the decoupling resistor 118 and the leakage current sampling switch 119 which is closed only within the period of the vertical blanking pulse V, and in certain cases only during part of the latter. Within this time the beam currents are, in fact, totally switched off by the picture signal so that in certain cases only a leakage current flows through the measuring resistor 702. Consequently, throughout the duration of the vertical blanking pulse V the measuring signal corresponds to this leakage current. Because the leakage current also flows during the remaining time, even outside the duration of the vertical blanking pulse the measuring signal contains a component originating from the leakage current which therefore is also contained in the voltage fed to the inverting inputs 111, 112, 113 of differential amplifiers 123, 124, 125 and is subtracted out in the differential amplifiers 123, 124, 125.
The part control signal is fed from output 141 of differential amplifier 123 by the first control signal sampling switch 154 to the first terminal 151 of the first control signal storage capacitor 161 during the period of a storage pulse L1 and is stored in the said capacitor. Similarly, the part control signal from output 143 of differential amplifier 125 is fed to the third control signal storage capacitor 163 during the period of a storage pulse L2 and the part control signal from output 142 of differential amplifier 124 is fed to the second control signal storage capacitor 162 during a storage pulse L3. The storage pulses L1, L2 and L3 are illustrated in FIGS. 3d, e and f. They lie in sequence in one of the three line periods by which the measuring signal control pulse VH is longer than the vertical blanking pulse V. These three line periods form the sampling interval for the measuring signal or the part measuring signals, as the case may be. During the remaining periods the outputs, 141, 152, 143 of the differential amplifiers 123, 124, 125 are isolated from the control signal storage capacitors 161, 162, 163 so that no interference can be transmitted from there and any distortion of the stored part control signals caused thereby is eliminated. For the duration of storage pulses L1, L2 and L3 the color signals at the input terminals 101, 102, 103 are at their reference level i.e. in the present embodiment at a level, corresponding to the blocking point or at a fixed level with respect to it so that the control loops can adjust to this level.

The switchable amplifiers 511, 512, and 513 each receive at each input 241, 242, 243 a blanking signal BL1, BL2, BL3 respectively, the curves of which are shown in FIGS. 3g, h, i. These blanking signals interrupt the supply of the color signals during line flybacks and frame change, i.e. during the period of the measuring signal control pulse VH, and thus the beam currents in these time intervals are switched off. Naturally, the red color signal is let through during the first line period after the end of the vertical blanking pulse V, the blue color signal during the second line period after the end of the vertical blanking pulse V and the green color signal during the third line period after the end of the vertical blanking pulse V by the switchable amplifiers 511, 512, 513 respectively so that they can control the beam currents. Blanking signals BL1, BL2 and BL3 also provide for interruptions in the frame change blanking pulse, which corresponds to the measuring signal control pulse, in the corresponding time intervals. In these time intervals the beam currents are measured and part control signals are determined from the part measuring signals and stored in the control signal storage capacitors 161, 162, 163.

The circuit arrangement shown in FIG. 2 further contains a trigger circuit 19 to which a supply voltage is fed via a supply terminal 190. Via a reset input 191 a voltage is also supplied to the trigger circuit 19 from a third terminal 133 of the reference voltage source 130. When the circuit arrangement is turned on, this voltage is designed so as to be delayed with respect to the supply voltage so that when the circuit arrangement is brought into operation the interplay of the two voltages produces a switch-on reset signal such that a low-value voltage pulse occurs at the reset input 191 during turn on, which means that the trigger circuit 19 is set in its first state. The reset input 191 can also be connected to another circuit of any configuration which generates a switch-on reset signal when the picture tube is turned on.
The trigger circuit 19 is further connected via a second connection 23 to a logic network 22 which, when the circuit arrangement is turned on, is also set into a first state via the second connection 23. In this first state the logic network 22 delivers a blocking signal at a blocking output 240 which is fed to the three switchable amplifiers 511, 512, 513. By this means the supply of the color signals to the output amplifiers 521, 522, 523 is interrupted completely so that no beam currents can be generated by these. No picture is therefore displayed.

An insertion signal EL which extends over the three line periods by which the measuring signal control pulse VH is longer than the vertical blanking pulse V, i.e. over the sampling interval, is also fed via a line 233 to the trigger circuit 19 and the logic network 22. As long as the trigger circuit 19 is in its first state, this insertion pulse EL is issued via a control output 192 from the trigger circuit 19 and fed to the pulse generator 244. During the period of the insertion pulse EL this generator produces a voltage pulse of a definite magnitude and passes this to output amplfiiers 521, 522, 523 as an auxiliary pulse via switching diodes 245, 246, 247. By this means the beam currents are switched on for a short time so as to receive a measuring signal despite the disconnected color signals as soon as at least one of the hot cathodes 801, 802, 803 delivers a beam current.
In its first state the trigger circuit 19 also delivers a signal via a control line 211, and this signal is used to switch the outputs 141, 142, 143 of the differential amplifiers 123, 124, 125 to earth potential or practically to earth potential. This suppresses effects of voltages at the inputs 111 to 116 of the differential amplifiers 123, 124, 125, especially effects of the reference voltage source 130 which may in some cases initiate incorrect charging of the control signal storage capacitors 161, 162, 163.
The measuring signal produced by means of the pulse generator 244 at the input 121 of the buffer amplifier 120 is also fed to the trigger circuit 19 via a measuring signal input 20. If it exceeds a preset threshold value, the trigger circuit 19 switched into its second state. The logic network 22 is then also switched into its second state via the second connection 23. The differential amplifiers 123, 124, 125, too, are triggered by the signal along the control line 211 into issuing a control signal defined by the difference in the voltages at its inputs 111 to 116. The pulse generator 244 is blocked by the control output 192. The blocking signal issued from the blocking output 240 of the logic network 22 now turns on the switchable amplifiers 511, 512, 513 in the time intervals defined by the storage pulses L1, L2, L3 in such a way that in these time intervals the color signals can produce beam currents to form a measuring signal by which the control loops respond. However, the display of the picture is still suppressed. The control signal storage capacitors 161, 162, 163 are charged up in this process. In the leads to the first terminals 151, 152, 153 there are change detectors 251, 252, 253 which detect the changes of the charging currents of the control signal storage capacitors 161, 162, 163 and at their outputs 261, 262, 263 in each case deliver a part change signal when the charging current of the control signal storage capacitor in question has decayed and thus the relevant control loop has responded. The part change signals are fed to three terminals 271, 272, 273 of the change signal input 27 of the logic network 22.
When part change signals are present from all change detectors 251, 252, 253, when therefore all control loops have responded, the logic network 22 switches from its second to its third state. The blocking signal from the blocking output 240 is now completely disconnected such that the switchable amplifiers 511, 512, 513 are now switched only by the blanking signals BL1, BL2, BL3. The colour signals are then switched through to the output amplifiers 521, 522, 523 and the picture is displayed in the picture tube.

FIG. 4 shows an embodiment for a trigger circuit 19 and a logic network 22 of the circuit arrangements as shown in FIGS. 1 or 2. The trigger circuit 19 contains a flip-flop circuit formed from two NAND-gates 194, 195 to which the switch-on reset signal, by which the trigger circuit 19 is returned to its first stage, is fed via the reset input 191. All the elements of the circuit arrangement in FIG. 4 are shown in positive logic. Thus, a short-time low voltage at the reset input 191 immediately after the circuit arrangement is started up is used to set the flip-flop circuit 194, 195 in such a way that a high voltage occurs at the output of the second NAND gate 194 and a low voltage at the output of the second NAND gate 195. The low voltage at the output of the second NAND gate 195 blocks differential amplifiers 123, 124, 125 via the control line 211 in the manner described.
The insertion pulse EL is fed via the line 233 to the trigger circuit 19, is combined via an AND gate 196 with the signal from the output of the first NAND gate 194 and is delivered at the control output 192 for the purpose of controlling the pulse generator 244.
The signals from the outputs of the NAND-gates 194, 195 are fed via a first line 231 and a second line 232 of the second connection 23 as a switching signal to the logic network 22. The first line 231 is connected to reset inputs R of three part change signal memories 221, 222, 223 in the form of bistable flip-flop circuits which when the circuit arrangement is started up are reset via the first line 231 in such a way that they carry a low voltage at their outputs Q. The second line 232 of the second connection 23 leads via three AND gates 224, 225, 226 to setting inputs S of the three part change signal memories 221, 222, 223. By means of the AND gates 224, 225, 226 the signal on the second line 232 of the second connection 23 is combined each time with one of the part change signals supplied via the terminals 271, 272, 273. The signals from the outputs Q of the part change signal memories 221, 222, 223 are combined by means of a collecting gate 227 in the form of an NAND gate and are held ready at its output 228.
The measuring signal is fed to the trigger circuit 19 via the measuring signal input 20 and passed to a first input 197 of a threshold detector 198 to which at a second input a threshold value, in the form of a threshold voltage for example, produced by a threshold generator 199 is also supplied. When the voltage at the first input 197 of the threshold detector 198 is smaller than the voltage delivered by the threshold generator 199, the threshold detector 198 delivers a high voltage at its output 200. When, on the other hand, the voltage at the first input 197 is greater than the voltage of the threshold generator 199, the voltage at the output 200 jumps to a low value. This voltage is supplied as the setting signal of the flip-flop circuit 194, 195, reverses the latter and thereby switches the trigger circuit 19 into its second state when the voltage at the first input 197 exceeds the voltage of the threshold generator 199.
Between the output 200 and the flip-flop circuit 194, 195 in the circuit arrangement shown in FIG. 4 there is inserted an inquiry gate 181 in the form of an OR gate to which an inquiry pulse is fed via an inquiry input 193 of the trigger circuit 19. This ensures that the flip-flop circuit 194, 195 is switched over only at a time fixed by the inquiry pulse--in the present case a negative voltage pulse--and not at any other times due to disturbances. As such an inquiry pulse it is possible to use, for example, a pulse which occurs in the second line period after the end of the vertical blanking pulse V, i.e. one which largely corresponds to the storage pulse L2.
After the switching over of the flip-flop circuit 194, 195 corresponding to the setting of the trigger circuit 19 into the second state, appropriately modified signals are supplied via the control line 211 and the output 192 for the purpose of controlling the pulse generator 244 and the differential amplifiers 123, 124, 125. Modified voltages also appear on the lines 231, 232 of the second connection 23, and these voltages release the part change signal memories 221, 222, 223 such that they can each be set when the part change signals reach the terminals 271, 272, 273.
In certain cases, a further flip-flop circuit 234 is inserted in the lines 231, 232 to delay the signals passing along these lines; this is reset via the first line 231 when the circuit arrangement is started up and thus it also resets the part change signal memories 221, 222, 223. However, after the trigger circuit 19 is switched into the second state the further flip-flop circuit 234 is not set via the second line 232 of the second connection 23 until a release pulse arrives via a release input 235 and another AND gate 236, for example a period of approximately the interval of two vertical blanking pulses V after the switching of the trigger circuit 19 into the second state. In this way it is possible to bridge a period of time in which no defined signal values are present at the terminals 271, 272, 273.
The signal at the output 228 of the collecting gate 227 changes its state when the last of the three part change signals has also arrived and has set the last of the three part change signal memories. The signal is then combined via a gate arrangement 229 of two NAND gates and one AND gate with the insertion pulse EL of line 223 and with the signal on the second line 232 of the second connection 23 or from the output Q of the further flip-flop circuit 234 to the blocking signal delivered at the blocking output 24 which is fed to the switchable amplifiers 511, 512, 513.

FIGS. 31, m, n show the combinations of the blocking signal with the blanking signals BL1, BL2, and BL3 at the blanking inputs 241, 242, 243 of the switchable amplifiers 511, 512, 513 in the form of logic AND operations. The dot-dash lines show resulting insertion signals A1, A2, A3 formed by these operations after the starting up of the circuit arrangement and before the occurrence of a beam current, i.e. in the first state of the logic network 22. Here the resulting insertion signals A1, A2, A3 are constant at low level. The dash curves show the resulting insertion signals A1, A2, A3 after the appearance of a beam current and before the steady state of the cut-off point control is reached, i.e. in the second state of the logic network 22, while the continuous curves represent the resulting insertion signals A1, A2, A3 in the steady state of the cut-off point control, i.e. in the third state of logic network 22. The dash curves have similar shapes to storage pulses L1, L2, L3, whereas the continuous curves correspond in shape to the inverses of the blanking signals BL1, BL2, BL3. In this case a high level of the resulting insertion signals A1, A2 or A3 means that the switchable amplifier 511, 512 or 513 feeds the colour signal to the relevant output amplifier 521, 522 or 523 respectively, whereas a low level in the resulting insertion signal A1, A2 or A3 means that the relevant switchable amplifier 511, 512 or 513 is blocked for the color signal.
The circuit arrangement described is designed in such a way that the trigger circuit 19 remains in its second state and logic network 22 remains in its third state even if charging currents reappear at the difference signal storage cpacitors 161, 162, 163 due to disturbances during the operation of the circuit arrangement. The cutoff point control then makes readjustments without the displayed picture being disturbed.
In the circuit arrangement shown in FIG. 2, the green color signal can also be let through during the second line period after the end of the vertical blanking pulse V and the blue color signal during the third line period after the end of the vertical blanking pulse V by the switchable amplifiers 511, 512, 513 for the purpose of controlling the beam currents. The storage pulses L2 and L3 at the control signal sampling switches 155 and 156 and the second and third blanking signals BL2 and BL3 at the blanking inputs 242 and 243 are then to be interchanged. The resulting insertion signals A2 and A3 as shown in FIGS. 3m and n are also interchanged then accordingly.
In FIG. 2 a dashed line is used to indicate which components of the circuit arrangement can be combined advantageously to form an integrated circuit. The first terminals 151, 152, 153 of the difference signal storage capacitors 161, 162, 163, one terminal 128 of leakage current storage capacitor 126, three terminals 524, 525, 526 in the leads to the output amplifiers 521, 522, 523 as well as a line connection 704 between the first terminal 701 of the measuring resistor 702 and the input 121 of the buffer amplifier 120 will then form the connecting contacts of this integrated circuit

TDA8341 Television IF amplifier and demodulator

DESCRIPTION
The TDA8340;Q and TDA8341;Q are integrated IF
amplifier and demodulator circuits for colour or black/white
television receivers, the TDA8340;Q is for application with
n-p-n tuners and the TDA8341;Q for p-n-p tuners.
The TDA8340;Q and TDA8341;Q are pin-compatible
successors with improved performance to types
TDA2540/2541;Q and TDA3540/3541;Q.
Features
· Full range gain-controlled wide-band IF amplifier
· Linear synchronous demodulator with excellent
intermodulation performance
· White spot inverter
· Wide-band video amplifier with noise protection
· AFC circuit with AFC on/off switching and
sample-and-hold function
· Low impedance AFC output
· AGC circuit with noise gating
· Tuner AGC output for n-p-n tuners (TDA8340) or p-n-p
tuners (TDA8341)
· External video switch for switching-off the video output
· Reduced sensitivity for high sound carriers
· Integrated filter to limit second harmonic IF signals
· Wide supply voltage range
· Requires few external components

- Audio IC TDA8190.


DESCRIPTION
The TDA8190is a completeTV soundchannel with
DC tone and volume controls plus separate VCR
input and output connections. Mounted in a Powerdip
16 + 2 + 2 package, the device delivers an
output power of 4W into 16W (d = 10%, Vs = 24V)
or 1.5W into 8W (d = 10%, Vs = 12V). Included in
the TDA8190 are : IF amplifier limiter, active lowpass
filter, AF pre-amplifier and power amplifier,
turn-off muting, mute circuit and thermal protection.
High output, high sensitivity, excellentAM rejection
and low distortion make the device suitable for use
in TVs of almost every type. Further, no screening
is necessary because the device is free of radiation
problems.







MIVAR  16C3V  CHASSIS TV2962/2 +  TV2965/2   PLL MICROCOMPUTER Frequency synthesizer tuning system for television receivers:MIVAR RT18A-99

" A method for tuning a television receiver having automatic frequency control to the carrier frequency of a selected broadcast channel with an associated channel number including generating a variable frequency signal by means of a local oscillator, generating a reference frequency signal by means of a reference oscillator, and generating a local oscillator correction signal for matching an intermediate frequency signal derived from said local oscillator signal and the carrier frequency signal with a predetermined nominal intermediate frequency signal, said method being characterized by the use of a microcomputer and comprising:
generating binary signals representing first and second digital tune words, said digital tune words representing a selected channel;
storing said first and second digital tune words in a first data memory in said microcomputer;
reading said first and second digital tune words from said first memory and generating a divided-down local oscillator frequency by the use of said first digital tune word and a divided-down reference oscillator frequency by the use of said second digital tune word;
comparing said divided-down local oscillator and reference frequencies and generating a control signal representative of the difference in frequency of said divided-down local oscillator and reference frequencies;
coupling said control signal to said local oscillator for causing it to be locked to the frequency of said received carrier signal;
mixing the local oscillator frequency signal and the carrier frequency signal to generate an intermediate frequency signal;
comparing said intermediate frequency signal with said predetermined nominal intermediate frequency signal and providing a tuning voltage to said microcomputer, said tuning voltage being indicative of the magnitude and direction of a tuning error between said intermediate frequency signal and said predetermined nominal intermediate frequency signal;
incrementally adjusting the reference oscillator frequency by means of a tuning signal provided to said reference oscillator by said microcomputer in response to said tuning voltage;
detecting when the incrementally changing, divided-down reference oscillator frequency causes the intermediate frequency signal to pass said predetermined nominal intermediate frequency signal; and
incrementally stepping the divided-down reference oscillator frequency back a predetermined number of steps following the passage of said predetermined nominal intermediate frequency signal by said intermediate frequency signal in tuning said television receiver to the selected channel.
"

A television tuning system employs a frequency synthesizer system for establishing the tuning of the receiver. A programmable frequency divider counter is connected between the output of a reference oscillator and a phase comparator to which the output of the local oscillator in the tuner also is applied. The phase comparator output provides a tuning voltage for controlling the tuning of the local oscillator. A microprocessor is used to control the count of the programmable frequency divider and initially to set a count corresponding to the selected channel in a counter connected between the output of the local oscillator and the phase comparator. The tuning consists of three discrete time periods. First, a settling time to allow channel change transients to settle; second, a short period of forced search at a relatively rapid rate to insure proper tuning; and third, a slower rate of step-by-step correction to accomodate for station drift and the like during reception. This third time period is initiated either by the passage of a fixed length of time following the start of the forced search period or by sensing a preestablished number of changes of state in the output of the frequency discriminator during the forced/search period.


1. A tuning system for the tuner of a television receiver capable of receiving a composite television signal and including frequency discriminator (AFT) circuit means, said system including in combination:
a reference oscillator providing a reference signal at a predetermined frequency;
a local oscillator in the tuner providing a variable output frequency in response to the application of a control signal thereto;
a programmable frequency divider means having first and second inputs coupled respectively to the output of said reference oscillator and said local oscillator for producing signals on first and second outputs having frequencies which are a programmable fraction of the frequency of the signals applied to the inputs thereto;
phase comparator means having one input coupled with the first output of said programmable frequency divider means and having another input coupled with the second output of said programmable frequency divider means for developing a control signal and applying such control signal to said local oscillator for controlling the output frequency thereof;
counter circuit means coupled with said programmable frequency divider means for initially setting said divider means to a predetermined division ratio and operating to change the programmable fraction of division thereof in accordance with changes in the count in said counter circuit means;
control circuit means coupled with the output of said frequency discriminator means and further coupled with said counter circuit means for causing said counter circuit means to count at a first rate in a predetermined direction determined by the state of the output signal from said discriminator means in the absence of a predetermined signal output from said frequency discriminator means until a predetermined maximum count is attained, thereupon resetting said counter circuit means to a count which is a predetermined amount less than said maximum predetermined count and continuing to count at said first rate in the same predetermined direction from said new count to continuously change the programmable fraction of said frequency divider means in accordance with the state of operation of said counter circuit means, said control means operating in response to said predetermined signal output from the frequency discriminator means for terminating operation of said counter circuit means; and
further means for terminating operation of said counter circuit means at said first rate and causing operation thereof at a second slower rate.
2. The combination according to claim 1 wherein said further means includes timing means initiated into operation simultaneously with the setting of said divider means to a predetermined division ratio, and after a predetermined time interval said timing means producing an output signal applied to said counter circuit means to cause operation thereof to take place at said second slower rate. 3. The combination according to claim 1 wherein said counter circuit means includes a reversible digital counter coupled with said programmable frequency divider, means and said control circuit means causes said counter circuit means to count in said predetermined direction when the output of said frequency discriminator is of a first state and to count in the opposite direction when the output of said frequency discriminator is of second state; and said further means comprises means coupled with the output of said frequency discriminator and with said counter circuit means to take place at said second slower rate in response to a predetermined number of changes of state of frequency discriminator. 4. The combination according to claim 3 further including means responsive to the selection of a new channel in said television receiver for resetting said further means to an initial condition of operation. 5. The combination according to claim 4 wherein said further means comprises a search termination counter means operative to provide an output signal applied to said counter circuit means in response to a count thereby of a predetermined number of changes of state of said frequency discriminator to cause said counter circuit means to be operated at said second slower rate.

Description:

BACKGROUND OF THE INVENTION
Both of the above mentioned patents are directed to frequency synthesizer tuning systems for use with television receivers to enable operation of the receivers with minimal viewer fine tuning adjustments. By the utilization of the frequency synthesizer tuning systems of these patents, the fine tuning adjustment which is necessary with conventional types of television receiver tuning systems has been substantially eliminated. The system employed in the '953 patent permits utilization of a frequency synthesizer tuning system which correctly tunes to a desired television station or channel even if the transmitted signals from that station are not precisely maintained at the proper frequencies. The '535 patent is directed to a signal seek tuning system adaptation of the frequency synthesizer tuning system of the '953 patent which still permits implementation of all of the desired wide-band pull in range of the frequency synthesizer system of the '953 patent.
The systems of the foregoing patents operate effectively to correct automatically for frequency offsets in a frequency synthesizer tuning system without affecting the operation of the conventional frequency synthesizer used in the system. The systems of these patents are in widespread use commercially and permit direct selection, with automatic fine tuning adjustment, of any desired VHF channel which the viewer wishes to observe. In addition, the signal seek adaptation disclosed in the '535 patent couples all of the advantages of the frequency synthesizer tuning system of the '953 patent with the desirability of providing bidirectional signal seek operation.
While the systems disclosed in the foregoing patents operate in a highly satisfactory manner to accomplish the desired results of accurate tuning without the necessity of fine tuning adjustments, the circuitry for accomplishing the desired results is somewhat complex. It is desirable to reduce the circuit complexity and the number of signal detectors for accomplishing these results without compromising the accuracy of operation of the system.
SUMMARY OF THE INVENTION
Accordingly, it is an object of this invention to provide an improved tuning system for a television receiver.
It is an additional object of this invention to provide an improved frequency synthesizer tuning system for a television receiver.
It is another object of this invention to provide an improved frequency synthesizer tuning system for a television receiver which includes a provision for adjusting the synthesizer loop for frequency offsets in the received signal with a minimum number of signal detectors.

It is a further object of this invention to tune the local RF oscillator of a television receiver to the correct frequency for a selected channel with a frequency synthesizer tuning system, and automatically to change the reference frequency of the synthesizer system, or adjust the count of a programmable divider that produces a signal that divides the frequency of the local oscillator of the tuner, if the AFT signal produced by the AFT frequency discriminator of the receiver is outside a predetermined range corresponding to correct tuning.
It is still another object of this invention to provide an improved frequency synthesizer tuning system for a television receiver which operates to adjust the synthesizer loop for frequency offsets in the received signal over a relatively wide pull in range in response to the output of the receiver frequency discriminator by changing the division ratio of a programmable frequency divider in the reference oscillator leg or local oscillator leg of the synthesizer loop at a first relatively high rate from an initial nominal value to a pre-established maximum in one direction, and then resetting the division ratio to a second nominal value once the maximum is reached and continuing to incrementally change the division ratio in the same direction from the second nominal value until a properly tuned condition is indicated by the output of the receiver AFT frequency discriminator, followed by control at a lower rate of operation to maintain tuning during transmitting station drifts.
In accordance with a preferred embodiment of this invention, the frequency synthesizer tuning system for a television receiver includes a stable reference oscillator and a voltage controlled local oscillator in the tuner. A programmable frequency divider is connected between the output of the reference oscillator and one input to a phase comparator, the other input of which is supplied by the output of the local oscillator. The output of the phase comparator then comprises a control signal which is supplied to the local oscillator to control the frequency of its operation.
A counter circuit is connected to the programmable frequency divider for initially setting the divider to a predetermined division ratio upon selection of a desired channel by the viewer. The counter then operates to change the programmable fraction of the division ratio at a first relatively high rate in a direction controlled by the output from the receiver picture carrier discriminator in the absence of a predetermined signal output derived from the discriminator. A control means causes the counter circuit to count in this direction until it is determined that a station is tuned or a predetermined maximum count is attained if no station is correctly tuned, thereupon resetting the counter circuit to a count which is a predetermined amount less than the maximum predetermined count. Counting is continued in the same predetermined direction from the new lesser count to continuously change the programmable fraction of the frequency divider in accordance with the state of operation of the counter.

The high rate operation of the counter is terminated by the control means in response to a predetermined signal from the output of the discriminator, indicating that a station is correctly tuned, or after a fixed time-out interval; so that the system automatically adjusts for frequency offsets of the received signal which otherwise would cause the station to be mistuned if a conventional frequency synthesizer tuning system were used. After termination of the high rate operation of the counter, it is switched to a lower rate operation for maintaining tuning during transmitting station drifts.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a block diagram of a television receiver employing a preferred embodiment of the invention;
FIG. 2 is a detailed block diagram of a portion of the circuit of the preferred embodiment shown in FIG. 1;
FIG. 3 is a detailed circuit diagram of a portion of a circuit shown in FIG. 1;
FIG. 4 is a flow chart of the control sequence of operation of the circuit shown in FIG. 1 and 2; and
FIG. 5 shows a waveform and time/frequency chart, respectively, useful in explaining the operation of the circuit shown in FIGS. 1, 2 and 3.
DETAILED DESCRIPTION
Referring now to the drawings, the same reference numbers are used throughout the several figures to designate the same or similar components.
FIG. 1 is a block diagram of a television receiver, which may be a black and white or color television receiver. Most of the circuitry of this receiver is conventional, and for that reason it has not been shown in FIG. 1. Added to the conventional television receiver circuitry of FIG. 1, however, is a frequency synthesizer tuning system, in accordance with a preferred embodiment of the invention, which is capable of automatically changing the reference frequency when a frequency offset exists in the received signal for a particular channel.
Transmitted composite television signals, either received over the air or distributed by means of a master antenna TV distribution system, are received by an antenna 10 or on antenna input terminals to the receiver. As is well known, these composite signals include picture and sound carrier components and synchronizing signal components, with the composite signal applied to an RF and tuner stage 11 of the receiver. The stage 11 includes the conventional RF amplifiers and tuner sections of the receiver, including a VHF oscillator section and a UHF oscillator section. Preferably, the UHF and VHF oscillators are voltage controlled oscillators, the freuency of operation of which are varied in response to a tuning voltage applied to them to effect the desired tuning of the receiver.
The output of the RF and tuner stages 11 is applied to the remainder of the television receiver 14, which includes the IF amplifier stages for supplying conventional picture (video) and sound IF signals to the video and sound processing stages of the receiver 14. The circuitry of the receiver 14 may be of any conventional type used to separate, amplify and otherwise process the signals for application to a cathode ray tube 16 and to a loudspeaker 17 which reproduce the picture and sound components, respectively, of the received signal.
The receiver 14 also includes a conventional AFT or automatic fine tuning discriminator circuit and additionally may include a synch separator circuit for producing an output in response to the presence of vertical synchronizatin pulses, a picture carrier detection circuit, and an automatic gain control (AGC) amplifier. Outputs representative of these sensor components are shown as being coupled over a group of lead 20 to sensory circuitry 22, which in turn couples outputs representative of the operation of these various sensor circuits to a microprocessor unit 23 for controlling the operation of the microprocessor unit.
The microprocessor unit 23 is utilized in the system of FIG. 1 for controlling the operation of a frequency synthesizer tuning system capable of automatic offset correction. When the viewer desires to select a new channel, he enters the desired channel number into a channel selection keyboard 25. There are a number of different keyboards which may be employed to accomplish this function, and the particular design is not important to this invention. The channel selector keyboard 25 also may include switches or keys for initiating a signal seek function in either the "up" or "down" direction.
Information represented by the selection of channel numbers on the keyboard 25 is supplied to the microprocessor unit 23 which provides output signals over a corresponding set of leads 27 to the tuners (local oscillators) 11 to effect the appropriate band switching control for the tuners 11 in accordance with the particular channel which has been selected. In addition, the keyboard 25, operating through the microprocessor unit 23, provides output signals which operate a channel number display 29 to provide an appropriate display of the selected channel number to the viewer.
The microprocessor M3870 unit 23 also processes the signals which are used to operate the channel number display 29 through a multiplexing circuit operation to decode the selected channel number into a parallel encoded signal. This signal is applied to corresponding inputs of the count-down counter or programmable frequency divider 31 to cause the division number of the divider 31 to relate to the divided down frequency of the tuner local oscillators connected to the input of the divider 31 through a prescaler divider circuit 32 to the frequency of the reference oscillator 34. Thus, the division number or division ratio of the local oscillator frequency obtained from the output of the programmable divider 31 is appropriately related to the frequency of the reference crystal oscillator 34.

The output of the oscillator 34 also is applied through a countdown circuit or programmable frequency divider 35. Conventional frequency synthesizer techniques are employed; and the microprocessor unit 23 automatically compensates, through appropriate code converter circuitry, for the non-uniform channel spacing of the television signals. It has been found most convenient to cause the programmable frequency divider 31 to divide by numbers corresponding directly to the oscillator frequency of the selected channel, for example, 101, 107, 113 . . . up to 931.
In accordance with the time division multiplex operation of the microprocessor 23, the count of the programmable frequency divider 35 initially is adjusted to a fixed count by the application of appropriate output signals from the microprocessor unit 23 to a point selected to be at or near the mid-point of the operating range of the programmable frequency divider 35. Thus, the output of the divider 35 is a stable reference frequency (because the input is from the reference crystal oscillator 34) which is used to establish initially and to maintain tuning of the receiver to the selected channel.
The output of the programmable divider 35 is applied to one of two inputs of a phase comparator circuit 37. The other input to the phase comparator circuit 37 is supplied from the selected one of the VHF or UHF oscillators in the tuner stages 11 through the programmable frequency divider 31. The phase comparator circuit 37 operates in a conventional manner to supply a DC tuning control signal through a phase locked loop filter circuit 39 and over a lead 40 to the oscillators in the tuner system 11 to change and maintain their operating frequency.
With the exception of the use of the microprocessor unit 23, the operation of the system which has been described thus far is that of a relatively conventional frequency synthesizer system incorporated into a television receiver. This system is similar to the system of the '953 patent. As in the system of that patent, the system shown in FIG. 1, when the transmitted station or station received on a master antenna distribution system provides the station or channel signals at the proper frequency, operates as a relatively conventional frequency synthesizer system. If, however, there is a frequency offset in the received signal to cause the carrier of the received signal to be displaced from the frequency which it should have to some other frequency, it is possible that the system would give the appearance of mistuning to the received station. The microprocessor 23, operating in conjunction with the sensory circuitry 22, is employed in conjunction with the countdown or programmable frequency divider circuit 35 to eliminate this disadvantage and still retain the advantages of frequency synthesizer tuning.
Reference now should be made to FIG. 2 which shows details of the interface between the keyboard 25, the microprocessor unit 23, and the circuitry used in the frequency synthesizer portions of the system. A commercially available microprocessor which has been used for the microprocessor 23, and which forms the basis for the diagramatic representation of the microprocessor in FIG. 2, is the Matsushita Electronics Corporation MN1402 four-bit single-chip microcomputer. This microcomputer has two, four-bit parallel input ports labeled "A" and "B". In addition, three output ports, a five-bit output port "C" and two four-bit output ports "D" and "E" are provided. The internal configuration of the microcomputer 23 includes an arithmetic logic unit (ALU), a read only memory (ROM) for storing instructions and constants, and a random access memory (RAM) used for data memory, arranged into four files, each file containing 16 four-bit words. These words are selected by X and Y registers and this memory is used, for example, for timers, counters, etc., and also is used to hold intermediate results. To facilitate an understanding of the operation of the system, a portion of this memory is shown in FIG. 2 as a clock 81 and a reversible counter 82 connected between the "B" input port and the "D" output port. The microcomputer 23 is programmed to permit it to operate in conjunction with the remainder of the circuits shown in FIG. 2. The programming techniques are standard, and the microcomputer 23 itself is a standard commercially available circuit component.
There are several system parameters that must be selected in the operation of the system shown in FIG. 2. The selection of the nominal frequency of the two signals that feed the phase comparator circuit 37 is an example. Channel selection is provided by changing the frequency division ratio of the selector counter 31 which divides the local oscillator signal after this signal is passed through a prescaler circuit 32 and a divide-by-two divider circuit 41. The nominal frequency from the programmable frequency divider 31 (selector counter) is selected so that the local oscillator (tuner) 11 can be set exactly on frequency for all channels.

Since the frequency divider 31 is able to divide only by integer numbers, one distinct frequency possibility in the range of one KHz is obtained, another in the range of two KHz, etc. A choice must be made as to which of these values is optimum. Each value yields the nominal frequency of all of the 82 channels by simply multiplying by an appropriate integer for each channel. To simplify the phase locked loop filtering problem by the filter 39, it is desirable that the frequencies of the signals supplied to the phase comparator 37 are as high as possible. This permits rapid acquisition of a new channel along with a very clean DC control signal to adjust the local oscillator. A trade-off for this, however, must be made to permit fine tunning adjustment of the local oscillator automatically to correctly tune in stations which are off their assigned frequency, or to manually provide this feature, if desired. The two-speed operation of the system in accordance with the present invention allows a better trade-off to be made by allowing rapid acquisition and then a slower speed for precise tuning.
A compromise solution which is utilized in the circuit of FIG. 2 is to cause the frequency division chain from the local oscillator 11 in the tuner to the phase comparator 37 to be composed of the fixed divide-by-256 prescaler 32, and a fixed divide-by-4 division, which is accomplished by the divider 41 at the input of the counter 31 and a second divider 42 at the output of the counter 31. The variable frequency divider counter 31 then is loaded by means of three latch circuits 44, 45 and 46 at an appropriate time by the time division multiplex operation of the microcomputer 23 and a number that programs the programmable frequency divider counter 31 to divide by the numerical value of the frequency of the local oscillator in MHz for the channel selected. For example, if the receiver is to be tuned to channel 2, which has a nominal local oscillator frequency of 101 MHz, the programmable frequency divider 31 is set to divide by 101. If the receiver is to be tuned to channel 83, which has a nominal local oscillator frequency of 931 MHz, the programmable frequency divider 31 is set to divide by 931. In both cases, the variable divider 31 produces a 1 MHz signal. However, because of the fixed divide-by-256 and the two fixed divide-by-two dividers in series with the programmable divider 31, an output frequency of 976.5625 Hz is supplied from the output of the divider 42 to the upper input of the phase comparator 37.

The division ratio of the selector counter 31 is established by appropriate output signals from the latch circuits 44, 45 and 46, as mentioned above. The initial operation for changing, or maintaining, the division ratio of the divider 31 is established by an entry of the two digits of the selected channel number in the keyboard 25. The microcomputer 23 operates as a time division multiplex system for continuously monitoring the input ports and the output ports to control the operation of the remainder of the system. The selection of the two digits of the desired channel number is affected by a time division multiplex iscanning of the outputs of the D output port of microcomputer 23 and providing that information at the A input port. From here the information is translated again to the D output ports to the appropriate drivers of the channel number display circuit 29 and to the latches 44, 45 and 46, and to a pair of similar four bit latches 49 and 50 which control the divider ratio of the counter 35.
Although the D output ports of the microcomputer 23 are connected in common to all of these various portions of the circuit, the selection of which of the latches are enabled to respond to the particular output signals appearing on the D output ports at any given time is effected through the C and E output ports of the microcomputer 23 in a time division multiplex fashion. A decoder circuit 52, connected to the lowermost three outputs of the E output port of the microcomputer 23, is used to apply unique decoding signals at different times in the time division multiplex sequence of operation of the microcomputer 23 to the five latch circuits 44, 45, 46, 49 and 50, respectively. At any given time in the sequence, only one of these latch circuits is enabled for operation. A latch load signal is applied from the upper output (EO3) at each cycle of operation of the signals appearing on the E output port to set the latch circuit which is enabled by the output of the decoding circuit 52 with the data appearing on the other inputs to the latch circuit. This data simultaneously appears on the four outputs of the D output port of the microcomputer 23.
Thus, in rapid sequence, the latch circuits 44, 45 and 46 are set to store the division number corresponding to the selected channel entered onto the keyboard 25, and the latch circuits 49 and 50 are each operated to set the programmable divider reference counter 35 to a center or nominal count, which is always the same upon the selection of a new channel on the keyboard 25. Similarly, the two right-hand outputs of the C output port (CO6 and CO5) enter the two digits of the selected channel number in the drivers of the display circuit 29 at the proper time in the binary encoded sequence when these digits appear on the four-bit binary encoded representation of the D output port. This results in a visual display of the channel number selected.
In addition to the selection of a channel number directly by the keyboard 25, the keyboard also may include an additional switch 56, which is scanned in the time division multiplex sequence to determine if the receiver is placed in a "seek" mode of operation (when the signal seek capability is incorporated into such a receiver). Operating in conjunction with the signal seek switch 56 are a pair of "up" and "down" seek direction input switches shown with a graphic representation of the seek directions on the keyboard 25. A further provision is provided by two keys labeled "U" and "D", which are used for "manual" fine tuning of the receiver in the "up" or "down" directions depending upon which of the two keys U or D has been operated. The keyboard 25 includes one additional switch 58 which may be used to disable the automatic fine tuning (AFT) portion of the circuit by rendering the microcomputer insensitive to the signal output from the AFT circuit, in a manner described more fully subsequently.
As is apparent from the foregoing, the microcomputer 23 provides the intelligence, decision making, and control for the system operation. It is a complete self contained computer. The decisions or signal inputs upon which the microcomputer 23 bases its operation include, in addition to the inputs from the keyboard 25, inputs on sensory inputs into the B input port and into the SNS1 and SNS0 inputs as shown in FIG. 2. These input signals are used to provide an indication to the microcomputer 23 of the presence or absence of a received signal; and if the presence of such a signal is indicated, the inputs provide a further indication of the accuracy of the tuning of the receiver to that signal. If the system is being operated solely in a manual mode of operation (AFT switch 58 open), the microcomputer 23 disregards all of this sensory information and tunes to the frequency allocation of the channel selected in the manner described above. The system will stay tuned to this condition, operating as a conventional frequency synthesizer, whether or not a station is present in the received signal.
When the system is placed in its automatic mode of operation (similar to the mode of operation of the above mentioned '953 patent), the counter 82, integrally formed as part of the microcomputer 23, continuously adds or subtracts one number at a time from the nominal value or programmable division fraction entered into the programmable frequency divider 35 at the outset of each new channel number selection when frequency offset (mistuning) is present. The counter 82 is driven at a relatively high counting rate by clock pulses from the clock 81 during this initial or forced search mode of operation. Thus, automatic offset correction is provided for any channel which is off its assigned frequency. The offset correction automatically adjusts the frequency of the local oscillator by changing the division ratio of the signal from the reference oscillator 35 applied to the lower input of the phase comparator 37. By doing this, the output of the phase comparator 37 applied to the local oscillator 11 varies to cause the oscillator to be tuned in the proper direction to compensate for the transmitting station mistuning.
When the system is operating in its automatic mode of operation, the microcomputer 23 responds to the sensor information applied to it on its B input ports and on the S1 input port shown in FIG. 2. These inputs are obtained from the various outputs of the operational amplifiers shown connected to the corresponding input ports in the detailed circuit of FIG. 3. Depending upon whether the receiver is provided with a signal seek feature or not, one or more of the sensory inputs of the circuit of FIG. 3 are used. The system shown in the drawings has a capability of correcting for frequency offsets larger than 1.5 MHz on channels 2 and 7 and approximately 2 MHz on channels 6 and 13. The remainder of the channels have a range between these two values.
If the receiver is not tuned properly, the micromputer 23 executes the localized search of the tuning range mentioned above. Since there is a necessary settling down time for the tuning of a television receiver immediately following selection of a new channel, a time interval of 250 milliseconds has been selected to prevent any localized search or offset frequency correction until the expiration of this "settling down" time period. If, at the end of this 250 millisecond time interval, a properly tuned station is present, this is indicated by the sensory outputs from the television receiver and no localized search is effected to change the division ratio or programmable divider count in the reference counter 35 for a system that also has signal seek.
A system with no signal seek capability is described later that requires less sensory input but which uses a time period where a forced search is required directly after the settling time interval.

Upon termination of the 250 millisecond settling down period, the microcomputer 23 is rendered responsive to the sensory input signals on its sensory input signal ports. In the simplest form, only the output of the frequency discriminator 60 (FIG. 3) applied to three comparators 61, 62 and 63 is used to provide the necessary tuning information to the microcomputer 23. The outputs of these comparators are applied to the B12 and B11 inputs of the microcomputer.

The comparator 61 simply is a conventional comparator for determining whether or not the output of the frequency discriminator is positive or negative, as indicated in the upper waveform of FIG. 5. The comparators 62 and 63 are each adjusted with appropriate reference input levels to provide a narrow window centered about the center tuning frequency (fc) of the receiver. If the tuning of the receiver, as indicated by the output of the frequency discriminator 60, is outside this window on either side of the central axis shown in FIG. 5, one output condition is indicated on the input terminal B11 of the microcomputer. Only when the tuning frequency is within the tuning window, indicative of a properly tuned receiver, is the appropriate input applied to the microcomputer input terminal B11. This input overrides any other input that may be present on the input terminal B12 and is indicative of a properly tuned receiver. The input from the frequency discriminator 60, as applied to the microcomputer on its input port B12, is used to determine the direction of operation of the counter 82 of the microcomputer for the localized search count signals applied to the latch circuits 49 and 50 to change the count of the reference programmable divider counter 35 on a step-by-step basis.
The lower graph of FIG. 5 plots the relative frequency of the local oscillator 11 to the received signal frequency with respect to time. The various arrows are used to indicate the manner of operation of the counter 82 in the microcomputer 23 in conjunction with the reference counter 35 for adjusting for any mistuning conditions which may exist after the initial station selection has been effected in the manner described above.
If the receiver is properly tuned, the outputs from the comparators 62 and 63 of FIG. 3 which are combined together and applied to the input port B11 of the microcomputer 23, provide an indication that the tuning is within the properly tuned center frequency window. As a consequence, no further operation of the microcomputer to change any of the outputs applied to the latch circuits 49 and 50 for the duration of this condition is effected. On the other hand, if the receiver is mistuned on either side of the proper tuning frequency, the various operating characteristics shown in FIG. 5 are effected.
Assume initially that the receiver is capable of making tuning adjustments over a range of fc plus Δf to fc minus Δf, as indicated in the top waveform of FIG. 5. Three specific examples of mistuning will then be considered. Initially, assume that the local oscillator is mistuned relative to the received signal to a frequency f1 as shown in the lower graph of FIG. 5. In this condition, the outout of the frequency discriminator 60 is positive since this signal frequency lies to the lefthand side of the center or properly tuned region of operation of the discriminator. Under this condition of the operation, the input signal applied to the sensor port B12 of the microcomputer 23 is such that the microcomputer counter 82 is caused to advance in a positive direction to change the programmable division ratio or count of the reference counter 35 in a manner to force the output of the phase comparator 37 to adjust the frequency of the local oscillator until the proper tuning indicated at point B in the lower graph of FIG. 5 is reached. The time interval for accomplishing this result is measured from the upper end of the arrow representative of the frequency f1 to the point B.
Now assume that the receiver mistuning is to a frequency f2 which as shown in FIG. 5 as located on the righthand-side of the center axis fc. In this condition, the discriminator output is negative. This is reflected in the output of the comparator 61 applied to the input port B12 of the microcomputer 23. The polarity of this signal is identified by the microcomputer 23 to cause the counter 82 in it to operate in the reverse direction. As this count is applied on a step-by-step basis through the latch circuits 49 and 50 to the reference counter 35, the division ratio or count of the reference counter (divider) 35 is changed. As a result, the reference oscillator signal applied to the phase comparator 37 causes the phase comparator 37 output to drive the local oscillator frequency in a direction opposite to that considered in the first example. This is shown by the vector interconnecting the top of the arrow representative of f2 to point A on the time/frequency graph of FIG. 5.
As discussed in the general discussion above, whenever the tuning frequency reaches the narrow window on either side of fc, the outputs of the comparators 62 and 63 provide the necessary indication on the sensory input port terminal B11 to cause termination of the operation of the counter 82 in the microcomputer 23. Then the reference counter 35 remains set to the count attained just prior to the appearance of this input signal on the input port B11 of the microcomputer 23.
A third mistuning condition can exist, and ordinarily this condition results in an ambiguity which cannot be corrected simply by responding to the signal polarity at the output of the frequency discriminator. This is indicated by the mistuned condition where the difference between the local oscillator frequency f3 and the transmitter frequency is such that the signal f3 lies in the range to the right of the negative portion of the discriminator output shown in the upper waveform of FIG. 5. In this condition, the associated sound causes the discriminator output to be positive; so that the television receiver normally would attempt to tune toward the next adjacent channel and away from the properly tuned center frequency of the channel which is desired. The output of the discriminator 60 in this situation is the same as it was in the first example considered for frequency f1; so that the counter 82 of the microprocessor 23 operates to change the count in the reference counter 35 in a manner to cause the local oscillator frequency to go higher toward a frequency f3 +Δf, as shown in FIG. 5.
A predetermined number of counts of the counter 82 in the microcomputer 23 are necessary for the microcomputer to count through the frequency range Δf, and this range is selected to be within the pull in or operating range of the system. Once this count has been attained, the microcomputer counter 82 immediately is reset back to a count which corresponds to a frequency 2 Δf lower than the frequency attained by the maximum count. This is indicated in FIG. 5 by the frequency f3-Δf. Because the microcomputer counter 82 is limited to counting a number of counts equal to Δf, this new frequency now is on the lefthand side of the center line fc, shown in both waveforms of FIG. 5. This places the local oscillator frequency at a point such that the frequency discriminator output is the positive output shown on the lefthand-side of the upper waveform of FIG. 5. Counting continues in the same direction as previously. This time, however, it is in a proper direction to bring about correct tuning; and when the center frequency is reached, the output of the comparators 62 and 63 cause the microcomputer 23 to stop its count. The proper tuning point attained is indicated at point C on the graph of the lower part of FIG. 5.
Because the counter 82 of the microcomputer is limited to a maximum count equivalent to Δf above its initial count and thereupon is reset to a new count equivalent to 2 Δf lower than the maximum count, it is not necessary to utilize any other sensory inputs in order to properly tune the receiver over a wide pull in range (as much as plus or minus 2 MHz). Only the output of the conventional frequency discriminator 60 is used to provide the necessary sensory inputs.

The counter 82 of the microcomputer 23 is operated by the clock 81 during the foregoing sequence of operation, immediately following the selection of a new channel by the operation of the keyboard 25, at a fast or high speed operation. Typically, the counter steps are 10 milliseconds per step; so that there are no initial visual effects which can be noticed by an observer of the television screen of the receiver being tuned. The maximum forced search period is approximately 900 milliseconds in duration. At the end of this time interval, a timer in the microcomputer 23 causes a signal to be applied through the outputs of the E output port to the decoder circuit 52 indicative of the completion of this time interval. The decoder 52 then applies a pulse on an output lead connected to the B13 input of the B input port of the microcomputer 23. This pulse is sensed by the microcomputer 23 and is applied to the clock 81 to change the clock rate to a much slower rate, approximately one-third (1/3) or one-fourth (1/4) the rate used previously during the forced search mode of operation. This then permits the system to accomodate station drifts which normally occur at a very slow rate during the transmission and reception of a television signal. As a consequence, it is possible to use more filtering in the filter 39 on the tuning line (FIG. 1) and employ a smaller frequency window for the channel verification sensed by the circuitry shown in FIG. 3. The result is a more precise tuning from the receiver than is otherwise possible if only a high speed operation of the clock 81 is utilized.
When the channel once again is changed by operation of the keys in the keyboard 25 or operation of the channel selection circuitry from a remote control unit, this new channel input is sensed by the microcomputer 23 from the signals applied to the A input port and the clock 81 is reset to its fast time or the forced search mode of operation; and the process resumes.
Instead of employing an additional decoding function in the decoder 52, a separate decoder also could be connected to the outputs of the D output ports to feed back the signal to the B13 input terminal of the B input port of the microcomputer 23. The operation of the system to change the rate or frequency of the pulses applied by the clock 81 to the counter 82 otherwise is the same as described above.
Although applicant has found that it is preferable to correct for mistuning or frequency offsets by adjusting the count or division ratio of the counter 35, such offset adjustments also could be effected by adjusting the count in the counter 31 in the local oscillator signal line. The operation in such a case is the same as described above for adjusting the count in the counter 35.
If the receiver is to be used with an automatic signal seek mode of operation, however, additional sensory inputs are necessary. These inputs operate in conjunction with the output of the frequency discriminator 60. The operation of the microcomputer 23 in controlling the count of the reference programmable frequency counter divider 35 is the same as described above. The additional sensory inputs simply are used in conjunction with the outputs of the comparators 62 and 63 to signal the microcomputer 23 to assure that tuning is to a picture channel rather than an adjacent sound channel. This is accomplished by utilizing the output of the synchronizing signal separator 65 which is applied to a comparator 67 to produce an output signal to the SNS1 sensory input of the microcomputer 23 only when vertical synchronizing signal components are present.
In addition, the output of a picture carrier detector 69 is applied to the input of a comparator 70 to produce an output to the B10 sensory input of the microcomputer 23. If the picture carrier detector 69 is producing an output indicative of the presence of a carrier, but no output is being obtained from the vertical synch separator 65 at the same time, the system is mistuned to a sound carrier and the microcomputer 23 is permitted to continue its localized search until a properly tuned station is found. Only when there is coincidence of signals from the picture carrier detector 69, the synch signal separator 65, and the automatic frequency discriminator window as determined by the comparators 62 and 63, is the microcomputer operation terminated to indicate that a properly tuned channel is present.
Further insurance of tuning the receiver only to a strong signal also can be provided by the addition of an AGC amplifier 72. This is connected to a comparator 74 coupled to the B10 input port along with the output of the picture carrier detector comparator 70. When the AGC amplifier 72 is used as a sensory input, the microcomputer operation, when the system is used in a signal seek mode, is only terminated to indicate reception of a valid signal when that signal is strong enough to produce the desired output from the comparator 74. The signal level which is acceptable is set by a potentiometer 75.
It should be noted that when the system is operated in a signal seek mode, the sensory inputs must indicate the reception of a properly tuned signal within a pre-established time period. If no signal is sensed by the various sensory input circuits operating in conjunction with one another as described above, the microcomputer 23 automatically steps to the next channel number and repeats the sequence of operation described above. This is when it is placed in its signal seek mode of operation. If signal seek is not employed, the additional sensory circuits 65, 69 and 72 are not necessary, and the inputs to the microcomputer which are provided from these sensory circuits are not utilized. The sensory signal input which is used both for a receiver without a signal seek capability of operation and for a receiver which has a signal seek mode of operation in it, is the output of the frequency discriminator 60 operating in conjunction with the comparators 61, 62 and 63 as described above.
As indicated above, the wideband method of tuning precisely to an incoming signal that is at the wrong frequency described here only needs the frequency discriminator sensory information. The method that uses the additional sensors described above is needed to make this system operate compatibly with signal seek but it is not restricted to seek operation.

For a system that does not use signal seek operation, only the frequency discriminator sensory input is required for proper operation. The discriminator 60 is used for both fine tuning direction information and to produce a frequency window to indicate the presence of a correctly tuned station (channel verification). Initially, after a channel change, there is a 250 millisecond settling time, the same as the operation described above with compatible seek. After that, however, comes a period of time where a forced localized search is produced by the microcomputer 23. The forced search is needed to insure that the system will correctly tune to stations that initially may be tuned to the undesired zero voltage crossover in the right half of the upper curve of FIG. 5. Such signals may be within the frequency window of the discriminator 60; and if a search is not forced, this system will not correctly tune. The compatible seek system described previously correctly tunes the local oscillator without a forced search, because the picture carrier detector and vertical detector do not give an output for this situation and the system automatically goes into its search mode of operation. However, the non-seek system does not have a picture carrier sensor input and must be forced to search for an initial period of time sufficient to allow the system to tune up to its maximum frequency and then reset (loop) back to a frequency of 2 Δf lower. Then it is tuned to the positive left half portion of the discriminator curve (FIG. 5) and the frequency window created by the discriminator 60 is sufficient to insure proper tuning. If the discriminator output produced by the desired incoming signal created an initial situation that produces the correct tuning direction information, i.e., in the left half of the curve of FIG. 5, or in the right half portion that gives the correct direction and

frequency window information, the forced search would not be needed. However, the forced search will produce a correct tuning situation anyway. In these cases, the tuning either is correct to begin with or correct tuning is reached quickly. Then, even though the forced search is active, it simply alternates up and down through the correct tuning point because each time the receiver is tuned a little high in frequency, it produces a negative output from the discriminator 60; and the tuning direction signal causes the system to tune down in frequency.

Then, a positive discriminator output is produced, and the system tunes up in frequency. This continues until the forced search is removed by time-out of the microcomputer 23 (a fraction of a second). At such time, the receiver is correctly tuned by the frequency window of the discriminator to be very near fc. The system cannot tune to the undesired discriminator crossover shown in the right half portion of FIG. 5 because the polarity of the tuning direction signal always causes it to tune away from that point.
The fast time or forced search operation of the system can be terminated in a different way other than the preestablished time-out period described above in conjunction with the operation of the circuit shown in FIG. 2. Generally, it is desirable to build into the system (or program into the system by means of software) such a maximum time-out period to effect the operation which has been described above to terminate the search and cause the clock 81 thereafter to operate in a low speed mode of operation. Termination also can be accomplished by sensing the number of changes in the direction sensor input applied to the B12 terminal of the B input port to cause the search to be terminated when this direction changes three times (or more). By doing this, any flicker that might be observed on the screen of the television receiver is minimized, since the forced search still takes place at the high rate of application of clock pulses from the clock 81 to the counter 82 in the same manner described above.
Termination of the search, however, also may be effected by means of a search terminate counter 78 (FIG. 3), which is advanced by pulses applied to it each time the output of the comparator 61 changes its sign (indicative of a change in direction for the counter 82) as applied to it through the B12 input port, as described earlier. After three of these changes, or some other number if desired, an output pulse is obtained from the search terminate counter 78 and is applied to the SNS0 input of the microcomputer 23. This causes the operation of the clock 81 to be switched to its low speed mode of operation to terminate the fast or "forced search" mode of operation. The next time a new channel number is entered on the keyboard 25, a reset pulse is applied to the search terminate counter 78 to reset it to its original or zero count, thereby readying it for another sequence of operation. It is apparent that the search terminate counter 78 may not always be operated to terminate the count, since the time-out interval which is sensed by the decode circuit 52 and applied to the B13 input port of the microcomputer 23 may occur before there are three changes of direction of the search. In any event, the next time a new channel number is entered into the keyboard 25, the search terminate counter 78 is reset; so that it is irrelevant whether this counter reaches a full count or not to effect the termination of the forced search operation of the system.
FIG. 4 shows the control sequence of the system which is stored in the ROM (Read Only Memory) of the microcomputer 23. The microcomputer 23 operates by always running through the flow sequence, via loops L1, L2 and L3. Loop L1 corresponds to a new channel selection by two digit number entry. Loop L2 corresponds to channel number increment or decrement by an up or down key operation, respectively, or by seek operation. Loop L3 corresponds to fine tuning, either manual or automatic. To obtain exact timing for system control, the microcomputer 23 receives a standard timing pulse from the output of the reference counter 35 divided in a divide-by-five counter 80 and applied to the A13 input port of the microcomputer 23. The control functions which are programmed into the microcomputer 23, as indicated in the flow chart of FIG. 4, are outlined in the following paragraphs.
Channel Number Correction: An invalid two digit channel number entry (0, 1, 84, 99) is corrected. When the operation of the receiver is in the signal seek mode, the next channel up from 83 is channel 2, and the next lower channel from channel 2 is 83.
PLL Control I: For a given channel number, a corresponding binary code for the PLL selector counter 31 is derived as described previously. For UHF channels, the local oscillator frequency separation between two adjacent channels is 6 MHz and the code for PLL is generated by the microcomputer 23 through means of a simple calculation. This code then is transferred from the microcomputer 23 to the latches 44, 45 and 46 as described previously.
PLL Control II: This routine of the microcomputer 23 is used to transfer the fine tuning data to the latches 49 and 50 which control the count of the reference counter 35 in the PLL circuit.
Channel Number Display: The channel number is transferred from the microcomputer 23 to the driver latches of the display driver circuit 29.
Key Input Detection: The keyboard is arranged as the matrix circuit shown in FIG. 2. ROM programming for scanning and acknowledging a keyboard entry only after successive indications provides protection against false entry due to contact bounce. The four data output lines of the D output port of the microcomputer 23 are used to transfer data to the phase lock loop section of the circuit and to the display circuit 29, as well as for scanning the keyboard matrix circuit.
Time Count: The microcomputer 23 receives a basic timing pulse of approximately 200 Hz from the output of the divider 80 and performs various controls for each timing pulse. By way of example, sensing for the vertical synch input (when the system is used with a signal seek capability) on the input port SNS1 takes place every 2.5 milliseconds. Automatic seek timing is selected to be 133 milliseconds for UHF channels. All of these timing pulses are derived from the basic synchronization timing pulse applied to the microcomputer on the A13 input port from the output of the divider 80. Various other timing values used in the microcomputer to properly time multiplex sequence the operation are derived from this basic timing pulse.
Sensor Input Detection: As described previously, the output of the comparators shown in FIG. 3 reflect the status of the tuning of the television receiver. If no signal seek mode of operation is used, only the frequency discriminator or AFT discriminator 60 is necessary. When a system is being used in a signal seek mode, a proper television signal receipt is indicated by the presence of a vertical synch signal at the output of the synch signal separator 65 and corresponding outputs are applied to the input leads B10 and B11 (high level input signals) indicative of tuning to the "correct tuned" frequency discriminator window and reception of a picture carrier. As stated previously, the signal present on the B12 input lead is used to determine the direction of tuning when the receiver is operated in its automatic mode.
Mode Detection: The status of the seek and automatic/manual (A/M) switches are detected. If the A/M switch (not shown) is in its automatic position, automatic seek and offset correction are active. If only the seek switch is on, only seek is performed. If the A/M switch is in manual, manual fine tuning (MFT) is active.
Automatic Mode: If the TV receiver is not properly tuned for VHF channels in automatic, the local oscillator frequency is shifted automatically toward proper tuning. The fine tuning data is generated in the microcomputer 23 and is transferred to the latches 49 and 50 for the reference counter 35 in the PLL circuit.
Manual Fine Tuning (MFT) Control: The local oscillator frequency is shifted by pushing the fine tuning up (U) or down (D) pushbutton or switch. This MFT control can be applied to VHF channels as well as to UHF channels.
Channel Up/Down: When a channel up (upward pointing arrow) or down (downward pointing arrow) key closure in the keyboard 25 is detected, or upon a direct access to an unused channel, this routine is activated and the system will advance to the next channel in the selected direction.
The foregoing embodiment of the invention which has been described above and which is illustrated in the drawings is to be considered illustrative of the invention, which is not limited to the specific embodiment selected for this purpose. For example, hard-wired logic could be used to achieve the various circuit operations which are accomplished by the microcomputer 23 in conjunction with the other portions of the system. The relative ease of programming and debugging the microcomputer 23, however, make it much simpler to implement the system operation with the microcomputer than with hard-wired logic. With respect to the sensor circuit inputs to the system, an added degree of operating assurance can be provided by the addition of a sound carrier sensor in addition to the picture carrier sensor shown in FIG. 3. If this feature is desired, the output of the comparator for the sound carrier is combined with the outputs of the comparators 70 and 74 at the input terminal B10 of the B input port of the microcomputer 23. Because of the manner of the circut operation which has been described previously, however, the addition of a sound carrier detector to the system is not considered necessary, even for a system operating in the signal seek mode of operation. This is in contrast to conventional television receivers having a signal seek operation, in which detection of the sound carrier generally is a necessity to insure that mistuning of the receiver to an adjacent sound carrier does not take place.