MIVAR 26" C803 CHASSIS TV2018 + TV2036/1 + TV2039/2 INTERNAL VIEW.

The MIVAR CHASSIS TV2018 + TV2036/1 + TV2039/2 was introducing the 30AX CRT TUBE BY PHILIPS and was replacing all previous models with THYRISTOR HORIZONTAL DEFLECTION TECHNOLOGY BASED CHASSIS TYPES WITH 20AX CRT TUBE.

AND It's the first MIVAR using a new tuning system replacing earlyer types based around ITT Chipset ASIC's but was anyway controlled by an ultrasonic remote system.


On bottom floor part the st-by unit TV2320


The tuning control box MIVAR RT7 is based around a SGS combination chipset:
M193A MADE BY:

SGS is Società Generale Semiconduttori - Aquila Tubi E Semiconduttori (SGS-ATES, "Semiconductor General Society - Tubes and Semiconductors Aquila"), later SGS Microelettronica, a former Italian company now merged into STMicroelectronics
SGS Microelettronica and Thomson Semiconducteurs were both long-established semiconductor companies. SGS Microelettronica originated in 1972 from a previous merger of two companies:

• ATES (Aquila Tubi e Semiconduttori), a vacuum tube and semiconductor maker headquartered in the Abruzzese city of l'Aquila, who in 1961 changed its name into Azienda Tecnica ed Elettronica del Sud and relocated its manufacturing plant in the outskirts of the Sicilian city of Catania
• Società Generale Semiconduttori (founded in 1957 by Adriano Olivetti).

MIVAR RT7:
The tuning circuits operate in conjunction with the M193 (Electronic Program Memory), from which they take the voltage and band information in a digital serial mode. The 7 most significant digits of voltage information coming from the M193 are digitally converted into a 64 step variable pulse width giving either positive and negative polarity outputs for easy and versatile interfacing. The variable length strip is displayed over 11 lines of a half frame picture with nine vertical graduations of 31 lines. The vertical position of the strip can be adjusted with an external potentiometer over the whole screen. The 2 digits of band information determine the number of rectangles appearing on the screen under the tuning strip. The rectangles are displayed over 11 lines of a half frame picture. Automatic display is provided when the Electronic Program Memory is in the Search Mode;display on manual command is also possible. The M191 is the standard version. The M091 is alternatively for displaying the tuning voltage when the automatic search is made by scanning the band in a reverse way (i.e. from 30 to OV) as is required by the French standard. The M091 displays 30V (maximum length of the strip) when the M193 Electronic Program Memory transmits information corresponding to OV. It displays OV when the M193 transmits information corresponding to 30V. It displays OV when the M193 transmits information corresponding to 30V. The M091 and M191 are constructed in N-channel silicon gate technology and are available in a 16 pin dual in-line plastic package. ABSOLUTE MAXIMUM RATINGS* Vdd* Supply voltage -0.3 to 20 V v, Input voltage -0.3 to 20 V I. Input current -5 mA Vo (off) Off-state output voltage 20 V lo Output current (except pins 12-13) 5 mA (pins 12-13) 15 mA Pfot Total package power dissipation 500 mW Tstg Storage temperature -65 to 150 °C Top Operating temperature 0 to 70 °C * Stresses above those listed under "Absolute Maximum Ratings" may cause permanent damage to the device. This is a stress rating only and functional operation of the device at these or any other condition above those indicated in the operational sections of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. * All voltages are with respect to Vss (GND).


A number of digital tuning systems for controlling a voltage controlled oscillator to generate a local oscillator signal for tuning a radio or television receiver are known. These digital tuning systems may be generally categorized as being either of the frequency synthesizer, voltage synthesizer or voltage sweep type.
Frequency synthesizers are typically closed loop. One type of frequency synthesizer includes a phase or frequency comparator for generating the control voltage for a local oscillator signal by comparing the phase and/or frequency of a variable frequency signal derived by the frequency division of the local oscillator signal and a relatively stable reference frequency signal. The frequency of the loop and thereby the frequency of the local oscillator signal is determined by division factors of fixed and programmable frequency dividers in the loop. The programmable divider is controlled in response to binary signals representing the number of a selected channel to determine the particular local oscillator frequency. Another type of frequency synthesizer includes a counter for counting in cycles of a voltage controlled local oscillator signal and a count comparator for comparing the number accumulated by the counter with a number derived from binary signals representing the channel number of a selected channel to develop a local oscillator control voltage. In either system channel numbers of selected channels can be readily displayed in response to the binary signals representing the number of the selected channels. Although such frequency synthesizers are advantageous in that the frequencies of the local oscillator signal are relatively accurate because of the closed loop nature of the systems, such systems are relatively expensive due to the cost of the high speed dividers and counters they necessarily employ.
Voltage synthesizers are typically open loop systems and generally include a memory having a plurality of tuning voltage memory locations for storing binary signals representing the tuning voltages for each of the channels that a user may select. The channel numbers of selected channels can be readily displayed, for example, in response to binary signals representing the channel numbers and utilized to address corresponding tuning voltage memory locations. Although such voltage synthesizers are advantageous in that they are relatively inexpensive compared with frequency synthesizers because they do not require high speed frequency dividers and counters, they tend to be less accurate because the required precision and resolution in converting the binary signals stored in the tuning voltage memory locations to the corresponding tuning voltages is not readily attainable in open loop systems.
Many tuning systems of the voltage sweep type are known. Basically, they all generate a ramp-like tuning voltage which is utilized to sweep the frequency of the local oscillator signal. In its simplest form, the magnitude of the tuning voltage is increased or decreased under user control by means of a potentiometer or the like until the user determines that an acceptable station has been reached. Signal sweeping systems are also known in which the magnitude of a tuning voltage is changed until a carrier is automatically detected. Such sweep systems are advantageous in that they are relatively accurate compared to voltage synthesizers since the tuning voltage is continuously adjusted until an acceptable channel is located and are relatively inexpensive compared to frequen

cy synthesizers since they do not require high speed frequency dividers and counters. However, since the tuning voltage is not derived as part of an operation involving the use of binary signals representing the number of a selected channel, additional apparatus must be provided for channel identification.
Related U.S. Pat. Nos. 4,317,225 and 4.254,506 both filed on May 30, 1979 and both assigned like the present application to RCA Corporation, disclose channel identification apparatus for a voltage sweep type tuning system in which the magnitude of the tuning voltage is compared with boundary voltages having magnitudes corresponding to magnitudes of the tuning voltage at frequencies between the tuning voltage ranges associated with respective adjacent channels which are stored in respective memory locations of a memory. The present invention concerns a carrier detector useful in a voltage sweep type tuning system. A preferred embodiment of the present invention is described, by way of example, with reference to the same voltage sweep type tuning system disclosed in the above identified related U.S. patents. Apparatus are also known for monitoring the channel to which a receiver is tuned by examining the tuning voltage. In these systems the tuning voltage for a selected channel to which the receiver is already tuned is compared with voltages having magnitudes corresponding to the magnitudes of the tuning voltage to tune respective channels stored in memory locations of a memory. The memory locations are successively addressed until there is at least an approximate equality between the tuning voltage and one of the stored voltages. The number of the selected channel is derived from the address of the memory location at which the approximate equality existed. Such systems may be used by television rating services to identify a limited number of channels in a particular viewing area. However, they are not particularly well suited for television receivers to identify all of the channels in the television tuning range because of the need for greater resolution in accurately distinguishing between closely spaced channels in the latter application. Moreover, such monitoring systems are not particularly well suited for sweep type systems to display the channel numbers of channels passed before an acceptable channel is located since the tuning voltage changes until and acceptable channel is located. In sweep systems, it may be desirable to display the channel number of channels passed to reach the acceptable channel so that users have a visible indication that the system is operating and are therefore not annoyed by apparent lock of operation as an acceptable channel is sought.


CIRCUITS DESCRIPTIONS:
- Luminance + chrominance + RGB Amplifiers

(TDA2151 + TDA2140 + TDA2161) (all FAIRCHILD)


DA1180P TV HORIZONTAL PROCESSOR
DESCRIPTION
The TDA1180P is a horizontal processor circuit for
b.w. and colour monitors. It is a monolithic integrated
circuit encapsulated in 16-lead dual in-line
plastic package.

APPLICATION INFORMATION
Pin 1 - Positive supply
The operating supply voltage of the device ranges
from 10V to 13.2V
Pin 2 and 3 - Output
The outputs of TDA1180P are suitable for driving
transistor output stages, they deliver positive pulse
at Pin 3 and negative pulse at Pin 2.
The negative pulse is used for direct driving of the
output stage, while positive pulse is useful when a
driver stage is required.
The rise and fall times of the output pulses are
about 150 ns so that interference due to radiation
are avoided.
Furthermore the output stages are internally protected
against short circuit.
Pin 4 - Protection circuit input
By connecting Pin 4 of the IC to earth the output
pulses at Pin 2 and 3 are shut off ; this function has
been introduced to produced to protect the final
stages from overloads.
The same pulses are also shut off when the supply
voltage falls below 4V.
Pin 5 - Phase shifter filter
To compensate for the delay introduced by the line
final stages, the flyback pulses to Pin 6 and the
oscillator waveform are compared in the oscillatorflyback
pulse phase comparator.
The result of the comparison is a control current
which, after it has been filtered by the external
capacitor connected to Pin 5, is sent to a phase
shifter which adequately regulates the phase of the
output pulses.
The maximum phase shift allowed is: td = tp - tf
where tf is the flyback pulse duration.
Pin 5 has high input and output resistance (current
generator).
Pin 6 - Flyback input
The flyback pulse drives the high impedance input
through a resistor in order to limit the input current
to suitable maximum values.
The flyback input pulses are processed by a double
threshold circuit; this generates the blanking pulses
by sensing low level flyback voltage and the pulses
to drive the phase comparator by sensing high level
flyback voltage, therefore phase jitter caused by
ringing normally associated with the flyback pulse,
is avoided.
Pin 7 - Key and blanking pulse output
The key pulse for taking out the burst from the
chrominance signal is generated from the oscillator
ramp and has therefore a fixed phase position with
respect to the sync.
The key pulse is then added internally to the blanking
pulse obtained by correctly forming the flyback
pulse present at Pin 6.
The sum of the two signals (sandcastle pulse) is
available on low impedance at output Pin 7.

Pin 8 and 9 - Sync separators inputs
The video signal is applied by means of two distinct
biasing networks to pins 8 and 9 of the IC and
therefore to the respective vertical and horizontal
sync separators.
The latter take the sync pulses out of the video
signal and make them available to the rest of the
circuit for further processing.

Pin 10 - Vertical sync output
The vertical sync pulse, obtained by internal integration
of the synchronizing signal, is available at
this pin.
The output impedance is typically 10kW and the
lowest amplitude without load is 11V.
Pin 11 - Coincidence detector
From the oscillator waveform a gate pulse 7 ms
wide is taken whose phase position is centered on
the horizontal synchronism.
The gate pulse not only controls a logic block which
permits the sync to reach the oscillator-sync phase
comparator only for as long as its duration, but also
allows the latching and de-latching conditions of
the oscillator to be established.This function is
obtained by a coincidence detector which compares
the phase of the gate pulses with that of the
sync.
When the two signals are not accurately aligned in
time it means that the oscillator is not synchronized.
In this case the detector acts on the logic block to
eliminate its filtering effect and on the time constant
switching block to establish a high impedance on
Pin 12 (small time constant of low-pass filter).
This latter block also acts on the oscillator-sync
phase detector to increase its sensitivity and with it
the loop gain of the synchronizing system.
In this conditions the phase lock has low noise
immunity (wide equivalent noise bandwidth) and
rapid pull-in time which allows fairly short synchronization
times.
Once locking has taken place the coincidence detector
enables the logic block, causes a low impedance
on Pin 12 and reduces the sensitivity of the
phase comparator.
In these conditions the phase lock has high noise
immunity ( narrow equivalent noise bandwidth) due
to the complete elimination of interference which
occurs during the scanning period and the greater
inertia with which the oscillator can change its
frequency.
To optimize the behaviour of the IC if a video
recorder is used, the state of the detector can be
forced by connecting Pin 11 to earth or to + VS. The
characteristics of the phase lock thus correspond
to the lack of synchronization.
Pin 12 - Time constant switch, (see Pin 11)
Pin 13 - Control current output
The oscillator is synchronized by comparing the
phase of its waveform with that of the sync pulses
in the oscillator-sync phase comparator and sending
its output current I13 (proportional to the phase
difference between the two signals) to Pin 15 of the
oscillator after it has been filtered properly with an
external low-pass circuit.
The time constant of the filter can be switched
between two values according to the impedance
presented by Pin 12.
The voltage limiter at the output of the phase
comparator limits the voltage excursion on Pin 13
and therefore the frequency range in which the
oscillator remains held-in.
The output resistance of Pin 13 is:
l low when V13 > 4.3 or V13 < 1.6V
l high when 1.6V < V13 < 4.3V
To prevent the vertical sync from reaching the
oscillator-sync phase comparator along with the
horizontal sync,a signal which inhibits the phase
detector during the vertical interval is taken from
the vertical output stage; inhibition remain even if
the video signal is not present.
The free running frequenc of the oscillator is determined
by the values of the capacitor and of the
resistor connected to Pins 14 and 15 respectively.
To generate the line frequency output pulses, two
theresholds are fixed along the fall ramp of the
triangular waveform of the oscillator.
Pin14 - Oscillator (see Pin 13)
Pin 15 - Oscillator control current input (see
Pin 13)
Pin 16 - Ground

TDA2541 IF AMPLIFIER WITH DEMODULATOR AND AFC

.SUPPLYVOLTAGE : 12V TYP .SUPPLYCURRENT : 50mATYP .I.F. INPUT VOLTAGE SENSITIVITY AT
F = 38.9MHz : 85mVRMS TYP .VIDEO OUTPUT VOLTAGE (white at 10% of
top synchro) : 2.7VPP TYP .I.F. VOLTAGE GAIN CONTROL RANGE :
64dB TYP .SIGNAL TO NOISE RATIO AT VI = 10mV :
58dB TYP .A.F.C. OUTPUT VOLTAGE SWING FOR
Df = 100kHz : 10V TYP
DESCRIPTION
The TDA2540 and 2541 are IF amplifier and A.M.
demodulator circuits for colour and black and white
television receivers using PNP or NPN tuners. They
are intended for reception of negative or positive
modulation CCIR standard.
They incorporate the following functions : .Gain controlled amplifier .Synchronous demodulator .White spot inverter .Video preamplifier with noise protection .Switchable AFC .AGC with noise gating .Tuner AGC output (NPN tuner for 2540)-(PNP
tuner for 2541) .VCR switch for video output inhibition (VCR
play back).


TDA2652A Vertical deflection circuit



DESCRIPTION
The TDA2652A is a monolithic integrated circuit for vertical deflection in large screen colour television receivers.
The circuit incorporates the following functions:
· Oscillator; switch capability for 50 Hz/60 Hz operation
· Synchronization circuit
· Blanking pulse generator with guard circuit
· Sawtooth generator with buffer stage
· Preamplifier with fed-out inputs
· Output stage with thermal and short-circuit protection
· Flyback generator
· Voltage stabilizers.


APPLICATION INFORMATION
The function is described against the corresponding pin number
1, 13. Oscillator
The oscillator frequency is determined by a potentiometer at pin 1 and a capacitor at pin 13.
2. Sync input/blanking output
Combination of sync input and blanking output. The oscillator has to be synchronized by a positive-going
pulse between 1 and 12 V. The integrated frequency detector delivers a switching level at pin 12.
The blanking pulse amplitude is 20 V with a load of 1 mA.
3. Sawtooth generator output
The sawtooth signal is fed via a buffer stage to pin 3. It delivers the signal which is used for linearity control,
and drive of the preamplifier. The sawtooth is applied via a shaping network to pin 11 (linearity) and via a
resistor to pin 4 (preamplifier).
4. Preamplifier input
The DC voltage is proportional to the output voltage (DC feedback). The AC voltage is proportional to the
sum of the buffered sawtooth voltage at pin 3 and the voltage, with opposite polarity, at the feedback
resistor (AC feedback).
5. Positive supply of output stage
This supply is obtained from the flyback generator. An electrolytic capacitor between pins 7 and 5, and a
diode between pins 5 and 9 have to be connected for proper operation of the flyback generator.
6. Output of class-B power stage
The vertical deflection coil is connected to this pin, via a series connection of a coupling capacitor and a
feedback resistor, to ground.
7. Flyback generator output
An electrolytic capacitor has to be connected between pins 7 and 5 to complete the flyback generator.
8. Negative supply (ground)
Negative supply of output stage and small signal part.
9. Positive supply
The supply voltage at this pin is used to supply the flyback generator, voltage stabilizer, blanking pulse
generator and buffer stage.
10. Reference voltage of preamplifier
External adjustment and decoupling of reference voltage of the preamplifier.
11. Sawtooth capacitor
This sawtooth capacitor has been split to realize linearity control.
12. 50 Hz/60 Hz switching level
This pin delivers a LOW voltage level for 50 Hz and a HIGH voltage level for 60 Hz. The amplitudes of the
sawtooth signals can be made equal for 50 Hz and 60 Hz with these levels.







MIVAR POWER SUPPLY COMBINED WITH BU208A TRANSISTOR HORIZONTAL DEFLECTION CIRCUIT, EXPLANATION AND CONCEPT VIEW.
CHASSIS TV2018 + TV2036/1 + TV2039/2


A combination deflection circuit and switching mode power supply uses only a single switching element. Across certain diodes in this circuit is a stable voltage. A capacitor and a transformer primary are series coupled to each other and together parallel coupled across at least one of the diodes. A rectifier is coupled to the transformer secondary to provide power to other portions of a television set.








1. A line deflection circuit for generating from a direct voltage source a sawtooth current flowing through a deflection coil, said circuit comprising a parallel resonant circuit comprising said coil, a trace capacitor coupled to said coil, and a retrace capacitor coupled to said coil; a first diode coupled to said retrace capacitor, the deflection current flowing during a first part of the trace period through said first diode and during a second part of the trace period through a controllable switch, energy being applied from said direct voltage source during the trace period to a first winding arranged between said direct voltage source and the switch, and being applied through a second diode conducting during the retrace period from a second winding to the parallel resonant circuit which is connected to the switch through a third diode conducting during the second part of the trace period, at least one of the second and third diodes being shunted by the series arrangement of a capacitor and a primary winding of a current supply transformer, and means for rectifying coupled to said transformer for the direct current supply to other stages of the device. 2. A circuit as claimed in claim 1 wherein said switch comprises a transistor. 3. A circuit for generating from a direct voltage source a sawtooth current having trace and retrace periods through a deflection coil, said circuit comprising a trace capacitor, means for coupling said trace capacitor to said coil, a retrace capacitor coupled to said trace capacitor, diode coupled to said retrace capacitor, a first diode means coupled to said retrace capacitor for conveying said current during a first part of said trace period, a first winding having a first end means for coupling to said source and a second end, a controllable switch means coupled to said second end for conveying said current during a second part of said trace period, a second winding, a second diode means coupled between said first diode and said second winding for conducting during said retrace period, a third diode means coupled between said first diode and said switch for conducting during said second part of said trace period, and means for supplying direct current power comprising a transformer having primary and secondary windings, a capacitor series coupled to said primary, said primary and capacitor being parallel coupled to at least one of said second and third diodes, and a rectifier coupled to said secondary. 4. A circuit as claimed in claim 3 wherein said switch comprises a transistor.

Description:

The invention relates to a line deflection circuit for a device comprising a cathode-ray tube particularly a television receiver display tube, for generating a sawtooth current flowing through a deflection coil in which the deflection coil constitutes part of a parallel resonant circuit comprising also a trace capacitor, a retrace capacitor and a first diode, the deflection current flowing during a first part of the trace period through said first diode and during a second part of the trace period through a controllable switch, for example, a transistor, energy being applied from a direct voltage source during the trace period to a first winding arranged between said direct voltage source and the switch, and being applied through a second diode conducting during the retrace period from a second winding to the parallel resonant circuit which is connected to the switch through a third diode conducting during the second part of the trace period.

Such a circuit arrangement is known from "IEEE Transaction on Broadcast and Television Receivers", August 1972, vol. BTR-18, No. 3, pages 177 to 182. The known circuit arrangement is the combination of a transistorized line deflection stage for a television receiver and a stabilised switch mode power supply, whereby one single switching element, the above mentioned transistor is both the switching transistor and the line deflection transistor.

An object of the invention was to further develop this circuit arrangement. It was found that an alternating voltage is present at the above mentioned second and third diode, which voltage is stabilized. The object according to the invention was to utilize this available and unilaterally stabilized rectangular voltage in a particularly advantageous manner.

This object is solved in that in a line deflection circuit of the kind described in the preamble the second and/or third diode is shunted by the series arrangement of a capacitor and a primary winding of a current supply transformer serving via rectifying for the direct current supply to other stages of the device.

An embodiment of the invention is shown in the drawings and will be further described hereinafter.

FIG. 1 shows the circuit improved according to this invention.

FIG. 2 shows different voltage variations as a function of time.

For the description of FIG. 1 the description of the Figures of the previously cited known circuit may be essentially used as a reference. A transformer is denoted by T1, a primary winding is L1; it is connected through a coupling capacitor CK to a secondary winding L2. A direct voltage source is UB. Furthermore a winding L3 is provided on the transformer secondary side which may serve for the high voltage generation UH through the diode Db.

The switching transistor is TR; rectangular pulses with the line frequency and originating from a driver stage (not represented) are applied to this transistor. The entire circuit arrangement thus serves for generating a sawtooth current flowing through a deflection coil L. The deflection coil L is part of a parallel resonant circuit consisting of a retrace capacitor C2, the deflection coil L itself and a trace capacitor C3.

In the operative condition a first diode D2 which is parallel connected to the said resonant circuit conducts during a first part of the trace period and conveys the negative part of the deflection current I ₂ during the period from t1 to t3 (compare FIG. 2d). During this period the switching transistor TR is separated from the deflection circuit consisting of D2, L, C2, C3 by a third diode Dd biassed in the blocking direction.

At the instant t2 which is adjustable via the width of the rectangular pulses (compare FIG. 2f) at the base of TR, TR is rendered conducting. As a result a current can flow through L1 and TR which stores until the switch-off instant t4 the energy required for operating the circuit in L1. This energy is applied to the deflection circuit at the initiation of the retrace period t4 so as to compensate for losses. This energy storage is ended at the instant t1 of the new period.

Meanwhile the zero crossing of the deflection current occurs at instant t3. D2 is blocked. Due to the polarity change of the current I _L the third diode Dd becomes conducting and the deflection current may be taken over by the switching transistor TR. This current is superimposed uninterfered on the part of the collector current originating from the power supply function of TR.

Thus the deflection function of the circuit in addition to the power supply function is ensured. This function may be influenced by shifting the instant t2. The limits of the control range are at t1 and t3. By comparison, for example, of the voltage UA over the diode D2 in the retrace period with a reference voltage a control magnitude for t2 can be derived. A stabilisation of the deflection in case of mains voltage and beam current fluctuations is then possible.

It is often essential to provide further stages in the television display apparatus with a stabilized voltage. Conventionally such supply voltages are obtained by trace rectification on an auxiliary winding of the line transformer. In this circuit this simple possibility is not given due to the connection with the power supply function. As can be seen in FIG. 2a the secondary voltage US consists of a rectangular voltage on which the flyback pulse of the deflection circuit is superimposed. When the trace part of US is rectified no stabilized direct voltage can be obtained due to the duty cycle variations caused by the control since the value of the voltage US between the instants t ₂ and t ₄ depends on that of the voltage UB.

A flyback rectification is feasible in this case. However, due to the small conduction angle an inadmissibly high internal resistance of the obtained supply voltage is to be taken into account.

According to the invention a rectangular voltage present alternatively across the diodes D1 and D2, respectively is used. These voltages do not contain a flyback pulse FIG. 2c shows the voltage variation UN on the secondary side L5 of a transformer T2 introduced for potential separation. A primary winding L ₄ thereof is arranged in series with a capacitor C ₄ and this series arrangement shunts the diode D1. The capacitor C ₄ prevents a dc short circuit of the diode D1 by the winding L ₄ and has a capacitance which is large enough for preventing an influence upon the variation of UN. The voltage across the capacitor C ₄ is thus equal to the dc-component of the voltage across the capacitor C ₃ , which component is stabilised since the voltage UA is. The voltage across the winding L ₄ is equal to the difference between that across the diode D1 and that across the capacitor C ₄ , the first mentioned voltage being equal to U _A -U _S . The voltage UN across the winding LS, which winding has the indicated winding sense, has the variation shown in FIg. 2c and between the instants t _o and t ₂ it is equal to the stabilised dc-component of the voltage present across the capacitor C ₃ . The voltage UN is rectified with the aid of a diode DN and smoothed with the aid of a capacitor CN. The rectified voltage UL is applied to the parts of the apparatus using a low voltage which in this case are represented by a load resistor RL.

DN must have such a polarity that it conveys current during the time t _o -t ₂ . Then the rectified voltage is stabilised to the same extent as the deflection voltage. The conduction angle is large so that the internal impedance of the voltage source is low. The primary side L4 of the transformer T2 is connected to D1 as is shown in FIG. 1. D1 and DN are then conducting simultaneously so that the internal resistance of UN is further reduced. In the same manner the series arrangement of L4 and C4 in parallel with Dd is alternatively possible.

The transformer T2 may be formed with a relatively small core due to the high operating frequency. On account of the switching properties (Dd and D1 alternately conducting) the rectangular voltage cannot become larger than the direct voltage on CK (corresponds to the voltage UB). Overvoltages as a result of for example picture tube flash-overs are thus prevented.

CHASSIS TV2018 + TV2036/1 + TV2039/2 Circuit arrangement for generating a sawtooth deflection current through a line deflection coil:
1. Circuit arrangement for generating a sawtooth deflection current flowing through a line deflection coil in an image display apparatus, which circuit arrangement comprises a deflection network including trace and retrace capacitor means coupling to said coil, and a first diode coupled to said retrace capacitor through which the deflection current flows during part of the trace interval, means for conveying the deflection current during the remainder of the trace interval including a second diode and a controllable switch coupled to said diode, said switch and second diode together being coupled in parallel with the first diode, the circuit arrangement further comprising an inductive element coupled to the switch, a third diode coupled to the deflection network and to said inductive element, a transformer having a core of a magnetic material and a winding, and a capacitor coupled to said winding and to the deflection network, characterized in that the inductive element is coupled via the third diode to the series combination of the above-mentioned series capacitor and part of the transformer winding less than all of said winding.

2. Circuit arrangement as claimed in claim 1, in which the inductive element comprises a winding, characterized in that the winding of the inductive element is wound on the transformer core.

3. Circuit arrangement as claimed in claim 1, characterized in that a first capacitor is coupled in parallel with the said part of the transformer winding and a second capacitor is coupled in parallel with the remainder of the winding, the ratio between the reactances of the said capacitors being equal to the ratio between the number of turns of the said parts of the winding.

4. Circuit arrangement as claimed in claim 1 in which the inductive element has a primary winding and a secondary winding which are coupled with one another, characterized in that the ratio of the number of turns of the secondary winding to that of the primary winding is substantially equal to ##EQU19## where m is the ratio of the turns number of the part of the transformer winding between the connection to the third diode and the series capacitor to the turns number of the entire winding, α is the ratio of the amplitude of the retrace voltage to the trace voltage, and δ_max is the value of that ratio of the conduction time of the switch to the line period which is associated with the maximum value of a voltage supply source which supplies energy to the circuit arrangement.

5. A circuit arrangement as claimed in claim 1 wherein said core has two limbs, a tapped transformer winding and at least one high-voltage winding wound on one limb, a primary winding and a secondary winding wound on the other limb, the ratio of the number of turns of the secondary winding to that of the primary winding being greater than the ratio of the number of turns of the part of the transformer winding between the tapping and an end adapted to be connected to a series capacitor to the number of turns of the entire winding and being less than 1.

Description:

The invention relates to a circuit arrangement for generating a sawtooth deflection current through a line deflection coil in an image display apparatus, which circuit arrangement comprises a deflection network including the deflection coil, a trace capacitor and a retrace capacitor and a first diode through which the deflection current flows during part of the trace interval whilst during the remainder of the trace interval this current flows through a second diode and a controllable switch, which switch and which second diode are connected in parallel with the first diode, the circuit arrangement further comprising an inductive element which is connected to the switch and is coupled to the deflection network via a third diode, and a transformer which has a core of a magnetic material and a winding of which is coupled, in series with a capacitor, to the deflection network.
Such a circuit arrangement is described in "IEEE Transactions on Broadcast and Television Receivers," August 1972, volume BTR-18, Nr. 3, pages 177 to 182, and is a combination of a line deflection circuit and a switched-mode supply voltage stabilizing circuit, the controllable switch being used to perform both the said functions. This known circuit arrangement has the advantage that it can be fed with an unstabilised supply voltage and is capable of supplying a satisfactorily stabilized deflection current, a stabilized high voltage and, if desired, auxiliary voltages, the stabilization being obtained by control of the conduction time of the swtich.
When such a circuit arrangement is to be designed, amongst other problems the three following ones arise. Firstly care must be taken to ensure that the maximum voltage set up across the switch (a transistor) during the retrace interval does not exceed the permissible limit value for this element. Secondly the variation of the conduction time of the transistor must be capable of accommodating the supply voltage variations to be expected. Thirdly the (stabilized) trace capacitor voltage applied to the deflection coil during the trace interval must be selectable at will, for with a given deflection coil this voltage determines the intensity of the deflection current produced.
The said problems are not independent of one another. If, for example, the trace voltage is low, the maximum collector voltage of the transistor also is low; it may be further reduced by making the conduction time of the transistor as short as possible. It will therefore be clear that several degrees of freedom are required. One degree of freedom is available to a certain extent, namely the transformation ratio between two windings of the inductive element, one winding being connected between a terminal of the supply voltage source and the junction point of the collector and the second diode, whilst the other winding, which is coupled to the first one, is connected to the third diode, for the choice of the said ratio enables a freer choice of the trace voltage. However, the two other problems, specifically that of maximum collector voltage, are not solved thereby.
It is an object of the present invention to provide a circuit arrangement having one more degree of freedom, permitting the maximum permissible collector voltage to be freely determined, and for this purpose the circuit arrangement according to the invention is characterized in that the inductive element is connected via the third diode to the series combination of the abovementioned series capacitor and part of the transformer winding.
The introduction of a new parameter not only enables the maximum collector voltage to be reduced without the trace voltage being affected but also proves to enable a larger range of supply voltage variations to be accommodated. Hence, the step according to the invention permits of designing a circuit arrangement in which conflicting requirements can simultaneously be satisfied.
In a possible embodiment in which the inductive element has a winding the circuit arrangement is characterized in that the winding of the inductive element is wound on the transformer core.
Embodiments of the invention will now be described, by way of example, with reference to the accompanying diagrammatic drawings, in which
FIG. 1 is a circuit diagram showing schematically the basic elements of an embodiment of the circuit arrangement according to the invention,
FIG. 2 shows waveforms of voltages produced in said embodiment,
FIGS. 3a and 3b show graphs which may be used in the selection of the parameters, and
FIG. 4 is a circuit diagram of a modified part of the circuit arrangement of FIG. 1.
The circuit arrangement shown in FIG. 1 includes a driver stage Dr to which signals from a line oscillator, not shown, are applied and which delivers switching pulses to the base of a switching transistor Tr. One end of a primary winding L ₁ of a transformer T ₁ is connected to the collector of the transistor Tr, which is of the n-p-n type, the other end of the winding L ₁ being connected to the positive terminal of a direct-voltage source B to the negative terminal of which the emitter of the transistor Tr is connected. This negative terminal may be connected to the earth of the circuit arrangement.
A trace capacitor C _t is connected in series with a line deflection coil L _y of the image display apparatus, not shown further, of which the circuit arrangement of FIG. 1 forms part, the resulting series combination being shunted by a diode D ₁ having the conductive direction shown and by a retrace capacitor C _r . The capacitor C _r may alternatively be connected in parallel with the coil L _y . The said four elements represent the schematic circuit diagram including the basic elements of the deflection section only. This section may, for example, in known manner be provided with one or more transformers for mutual coupling of the elements, with devices for centering and linearity correction and the like.
A secondary winding L ₂ of the transformer T ₁ is connected to the anode of a diode D ₃ , and the anode of a diode D ₂ is connected to the junction point A of the elements D ₁ , C _r and L _y . The cathode of the diode D ₂ is connected to the collector of the transistor T _r whilst the cathode of the diode D ₃ is connected to a tapping Q on a winding L ₃ of a transformer T ₂ . One end of the winding L ₃ is connected to the point A, the other end being connected to earth via a capacitor C ₁ . The core of the transformer T ₂ carries further windings across which voltages are produced which serve as supply voltages for other components of the image display apparatus. FIG. 1 shows one of said windings, the windings L ₄ , which by means of a rectifier D ₄ produces a positive direct voltage across a smoothing capacitance C ₂ . One of said windings, for example the winding L ₄ , is the high voltage winding, so that the voltage set up across the capacitor C ₂ is the high voltage for the final accelerating anode of the display tube (not shown). The free ends of the windings L ₂ and L ₄ are connected to earth, and the winding senses of the windings shown are indicated in the Figure by polarity dots.
The operation of the circuit arrangement is similar to that described in the abovementioned paper and may be summarized as follows. During a first part of the line trace interval the diode D ₁ is conducting. The voltage across the capacitor C _t is applied to the deflection coil L _y through which a sawtooth deflection current i _y flows. At a given instant the transistor TR becomes conducting. When in about the middle of the trace interval the current i _y reverses direction the diode D ₁ is cut off, so that the current i _y then flows through the diode D ₂ and the transistor Tr. At the end of the trace interval the transistor Tr is cut off. As a result an oscillation, the retrace pulse, is produced across the capacitor C _r whilst the energy derived from the source B and stored in the winding L ₁ causes a current to flow through the diode D ₃ . When the voltage across the capacitor C _r has become zero again, the diode D ₁ becomes conducting: this is the beginning of a new trace interval. The diode D ₃ remains conducting until the transistor Tr is rendered conducting, the energy stored in the winding L ₂ being transferred to the winding L ₁ . Stabilisation is provided, for example, by feeding back the voltage across the capacitor C _t to the driver circuit Dr, in which a comparison stage and a modulator ensure that the conduction time of the transistor Tr is varied so that the said voltage and hence the amplitude of the deflection current remain constant. Compared with the known case in which the cathode of the diode D ₃ is connected to the point A instead of to the tapping Q operation is different, the difference being as follows. In the known case the current passed by the diode D ₃ flows to earth via the diode D ₁ during the first part of the trace interval. In the arrangement shown in FIG. 1, during this same part energy is stored in the series combination L ₃ , C ₁ . The voltage v _A across the capacitor C _r , the voltage v _c at the collector of the transistor T _r and the voltage v ₁ across the winding L ₁ are plotted against time in FIGS. 2 a, 2b and 2c respectively. The symbol T indicates the line period, τ ₁ indicates the retrace interval, τ ₂ that part of the period T in which the transistor Tr is non-conducting, and τ ₃ = δ T indicates the part of the period T in which this transistor is conducting. The number δ is the ratio between the time τ ₃ and the period T.
The voltage v _A consists of the retrace pulse of amplitude V during the time τ ₁ and is zero during the time τ ₂ . At the instant at which the transistor Tr is rendered conducting, i.e. the instant of transition t ₁ between τ ₂ and τ ₃ , the voltage v _C becomes substantially zero. Thus the volage V _B of the source B is set up across the winding L ₁ .
In the circuit arrangement of FIG. 1 two ratios are significant, namely the transformation ratio between the windings L ₁ and L ₂ , i.e. the ratio between the number of turns of the winding L ₁ and that of the winding L ₂ , which is equal to 1 : p, and the ratio of the turns number of the entire winding L ₃ and that of the part of this winding between the tapping Q and the end connected to the capacitor C ₁ , which ratio is 1 : m. First it will be assumed that the points Q and A coincide (m = 1).
During the time τ ₃ the voltage cross the winding L ₂ is equal to -pV _B . During the time τ ₁ the voltage v _c is equal to V/p + V _B . Let V _o be the direct voltage across the capacitor C _t , if the capacitance of this capacitor is large enough, or the direct voltage component of the voltage across this capacitor, if it has a comparatively small capacitance for the purpose of the S correction; in either case it is equal to the mean value of the voltage v _A , for no direct-voltage component can be set up across the coil L _y . The capacitor C ₁ has a large capacitance, so that a direct voltage equal to V _o is set up across it. The following equation applies: ##EQU1##
The mean value of the voltage across the winding L ₃ also is zero, so that the equation applies: ##EQU2## In this formula the integral can be substituted, Yielding V _o T = pV _B . τ ₃ , that is; V _o = pδ. V _B (1)
At given values of the ratios δ and p the diode D ₂ will conduct during the time τ ₁ . Because during this time the diode D ₃ is conducting, the windings L ₁ and L ₂ will be short-circuited by the diodes D ₂ and D ₃ , causing the retrace pulse across the capacitor C _r to be clipped and the deflection current to be distorted. U.S. Pat. Application No. 443,863 filed Feb. 19, 1974 describes steps for avoiding such an effect, for example by including in series with the diode D ₂ a transistor which is cut off during the time τ ₁ . A capacitor C ₃ is connected between the ends of the windings L ₁ and L ₂ or between tappings thereon for the purpose of preventing the occurrence of parasitic oscillations which may be produced by the leakage inductance between the said windings in a manner such that no line-frequency voltage is set up across the capacitor C ₃ . FIG. 1 shows the case where p <1.
The maximum value of the collector voltage v _c of the transistor is equal to ##EQU3## where α is the ratio V/V _o which depends upon the retrace ratio Z = τ1/T. The maximum value of V _c is obtained when V _B has its maximum value V _B max, for which δ has the value δ _min , for from the relationship (1) it follows that δ and V _B are inversely proportional to one another because the voltage V _o is maintained constant.
The voltage V _o can be chosen by choosing the ratio p, so that the deflection current _y is determined for a given deflection coil L _y . However, from the above it follows that the maximum value of the voltage V _c , which is highly critical for the transistor, is not controllable. Moreover, the relationship (1) can be written:
V _o = p δ _min . V _B max = p δ _max . V _B min, where V _B min is the minimum value of V _B for which δ = δ _max , and from which follows: ##EQU4## The ratio δ _min has its minimum value δ ₁ if the instant t ₁ coincides with the middle of the trace interval, and δ _max has its maximum value δ ₂ if the instant t ₁ coincides with the beginning t _o of the trace interval. Hence the above ratio cannot exceed 2, so that the arrangement cannot accommodate larger variations of the voltage V _B .
According to the invention the points A and Q do not coincide. The voltage across the winding L ₃ is equal to v _A - V _o so that the voltage v _Q in the point Q is equal to v _Q = V _o + m(v _A - V _o ) = mv _A + (1 - m) V _o . With the aid of the waveform of the voltage v _A of FIG. 2a the waveform of the voltage v ₁ across the winding L ₁ between the positive terminal of the source B and the collector of the transistor Tr can be plotted (FIG. 2c), allowing for the fact that the diode D ₃ is conducting during the times τ ₁ and τ ₂ .
Thus we have: ##EQU5## during time τ ₃ : v ₁ = - V _B . Writing the condition for the mean value of the voltage v ₁ being zero after some calculations yields. ##EQU6## The maximum value of the collector voltage v _c is ##EQU7## from which follows: ##EQU8## after substitution of the formula (2). It can be shown that this function steadily decreases with decrease of the ratio m. It is plotted in FIG. 3a for z = 0.2, from which follows α ≉ π/2z ≉ 7,8, and with δ _min = δ ₁ = 1/2 (1 - z) = 0.4. The Figure shows that by making m less than 1 a reduction of the maximum collector voltage is obtained and that this result is independent of the ratio p.
From the formula (2) the following relationship can be derived: ##EQU9## ##EQU10## This function also is independent of the ratio p and it increases as m decreases. It is plotted in FIG. 3b for δ _min = δ ₁ = 0.4 and δ _max = δ ₂ = 0.8 (Z = 0.2), so that the entire δ range is used, whilst the Figure shows that a larger range of supply voltage variations can be accommodated, for when m is less than 1 the ratio V _B max /V _B min exceeds 2.
Similarly to the preceding case, the voltage V _o can be determined by the choice of the ratio p. If the means described in the abovementioned U.S. Pat. Application No. 443,863 are to be dispensed with, it is found that an upper limit can be set to p. The diode D ₂ will just be conducting during the time δ ₁ if the lowest value of the voltage V _c which is found in practice, that is ##EQU11## is equal to the voltage V. In the above expression, according to the formula (2), ##EQU12## from which we can derive: ##EQU13##
The above will be explained by means of two numerical examples. If the voltage V _B can vary between 230 volts and 345 volts (with a mains voltage of 220 volts) V _B max /V _B min is less than 2, so this does not provide difficulty. If the transistor Tr is not capable of withstanding a voltage exceeding 1200 volts, it will be seen from FIG. 3a that m = 0.64. From the formula (2) it follows that ##EQU14## with δ _min = δ ₁ and ##EQU15## so that δ _max = 0.56 < δ ₂ . The formula (5) yields: ##EQU16## so that V _o = 0.87 times 161 = 140 volts.

If now the voltage V _B can vary between 115 volts and 345 volts (the mains voltage is 110 volts or 220 volts), then V _B max /V _B min = 3. FIG. 3b shows that m = 0.38, for which FIG. 3a yields V _c max = 2.9 times 345 = 1000 volts. Formula (2) yields: ##EQU17## whilst ##EQU18## so that V _o = 0.54 times 183 volts = 99 volts. Because m cannot be increased, a higher V _o if desired requires p to exceed 0.54, and hence the step according to the abovementiond Patent Application must be used.
Similarly to what is the case in U.S. Pat. Application No. 473,771, filed June 1, 1973, the cores of the transformers T ₁ and T ₂ of FIG. 1 may be one and the same core, that is to say the windings L ₁ , L ₂ and the winding L ₃ may be coupled to one another in spite of the fact that voltages of different waveforms are set up across the said windings. This is possible because the said voltage waveforms are not affected by the coupling, since the voltages V _o and V _B are "hard," that is to say they are externally impressed, and hence are not affected by the coupling. The currents flowing through the windings, however, are affected. In the lastmentioned Patent Application it is shown that the operation of the circuit arrangement is not adversely affected thereby, but on the contrary important advantages are obtained. It should be mentioned that instead of the tapping Q an additional winding may be wound on the same core as the winding L ₃ , which additional winding has a smaller number of turns than the winding L ₃ and is included between the cathode of the diode D ₃ and the junction point of L ₃ and the capacitor C ₁ .
Formula (5) shows that the ratio m should not be excessively small, because in this case the ratio p also is small, with the result that large currents flow on the secondary side of the transformer T ₁ . In addition, large currents then will flow through the leakage inductance of the said transformer, which gives rise to ringing at the instant t ₁ . Furthermore difficulties will arise in designing the abovementioned embodiment using a single transformer. If for these reasons the formula (5) is not complied with, that is to say if p is made greater than the preferred value p _max , the steps according to the abovementioned U.S. Pat. Application No. 443,863 have to be employed. This requires an additional transistor, which is expensive, or an additional diode, which does not prevent the production of a high V _c max, whilst it was the very purpose of using a low m to obtain a low V _c max.
In practice there is a leakage inductance between the two parts of the winding L ₃ . In FIG. 4, which shows only part of the circuit arrangement, this leakage inductance is shown as an inductance L ₅ between the point Q and an imaginary tapping Q' on the winding L ₃ . The inductance L ₅ prevents abrupt current transistions which in conjunction with the stray capacitances may give rise to ringing. This can be avoided by connecting a capacitor C ₄ between points A and Q and a capacitor C ₅ between the point Q and the junction point of the winding L ₃ and the capacitor C ₁ . If the ratio between the reactances of C ₄ and C ₅ is equal to that between the numbers of turns of the upper and lower parts of the winding L ₃ , no alternating voltage is set up across the inductance L ₅ so that no ringing can occur. The parallel connection of the capacitor C _r and of the network C ₄ , C ₅ together with the inductive components of the circuit arrangement results in a resonant frequency the period of which is about equal to twice the time τ ₁ .
Hereinbefore it has been assumed that the capacitance of the capacitor C ₁ is sufficiently large to enable the voltage across it to be regarded as constant (= V _o ). It should be mentioned that this is necessary only if one or more of the auxiliary voltages produced by means of windings of the transformer T ₂ are obtained by means of trace rectification.

CHASSIS TV2018 + TV2036/1 + TV2039/2 MIVAR Self-regulating deflection circuit with resistive diode biasing:
"A New Horizontal Output Deflection Circuit" by Peter L. Wessel,
A self-regulating deflection circuit includes a first inductor and switching transistor coupled across the unregulated voltage supply. A damper diode, retrace capacitor and second inductor are coupled in parallel, and the parallel combination is coupled across the transistor by a first rectifier poled to prevent current from flowing from the first inductor to the second inductor. A second rectifier is coupled between the first and second inductors for transferring energy from the first inductor to the second during the retrace interval. A control circuit coupled to the second inductor and to the base of the switching transistor controls the time during the first half of the trace interval during which the transistor conducts to allow energy to be stored in the first inductor. A storage capacitor is coupled in series with the second rectifier. Charge accumulation on the storage capacitor and resultant blocking of the second rectifier is prevented by a resistor coupled across the storage capacitor.

1. A self-regulating deflection circuit adapted to be energized from a source of unregulated direct voltage, said deflection circuit including
first inductance means;
controllable switch means including a unidirectional main current conducting path and a control electrode, said main current controlling path being serially coupled with said first inductance means across the source of unregulated direct voltage thereby forming a first series path for storing energy in said first inductance means during those intervals in which said main current conducting path is conductive;
first rectifier means;
a parallel combination of elements coupled by said first rectifier means across said main current conducting path, said parallel combination including second inductance means, damper diode means and retrace capacitance means, said first rectifier means being poled for current conduction in the same direction as said main current conducting path;
control means coupled with said second inductance means and with said control electrode for recurrently switching said main current conducting path for promoting current flow in said second inductance means during recurrent trace and retrace intervals and for maintaining the peak value of said current flow at a constant level;
second capacitance means;
second rectifier means coupled by said capacitance means with said parallel combination of elements and to a point on said first series path for transferring energy from said first inductance means to said parallel combination of elements during said retrace intervals;
wherein the improvement comprises
resistance means coupled with said second capacitance means for equalizing charge on said second capacitance means during said trace interval.
2. A circuit according to claim 1 wherein said resistance means is coupled in parallel with said second capacitance means. 3. A circuit according to claims 1 or 2 wherein said capacitance means is serially coupled with said second rectifier means. 4. A circuit according to claim 3 wherein said point on said first series path is a point along said first inductance means. 5. A circuit according to claim 4 wherein said point along said first inductance means is an end of said first inductance means. 6. A circuit according to claims 1 or 2 wherein said second rectifier means is coupled by said capacitance means with said second inductance means in said parallel combination of elements. 7. A circuit according to claims 1 or 2 wherein said second inductance means is a winding of a transformer and said second inductance means is paralleled by a deflection winding.

Description:

BACKGROUND OF THE INVENTION
This invention relates to self-regulating horizontal deflection circuits with diode steering in which one of the diodes is biased.
Horizontal deflection circuits are used in conjunction with television picture tubes in television display devices. Typically, the horizontal deflection circuit includes a magnetic winding associated with the picture tube and a switching circuit by which energy from a dc voltage source is coupled to the winding and its associated reactances. The switching circuit is synchronized with synchronizing signals associated with the information content of the video to be displayed on the picture tube. In order to avoid distorted images on the displayed raster, the size of the horizontal scanning line and the peak deflection or scanning current must be maintained constant over substantial periods of time.
Many conditions can cause the size of the horizontal scanning line to vary. If the direct energizing voltage for the horizontal deflection circuit varies, the scanning energy and hence the width of the horizontal scanning line may vary. It has in the past been customary to regulate the direct voltage applied to the horizontal deflection circuit by the use of a dissipative regulator. Requirements for low power consumption in television receivers is reducing the use of such dissipative regulators in favor of nondissipative types.
Another approach to regulating the scan width involves the use of a self-regulating deflection circuit, such as is described in the article "A New Horizontal Output Deflection Circuit" by Peter L. Wessel, which appeared in the IEEE Transactions on Broadcast and Television Receivers, August, 1972, Vol. BTR-18, No. 3, pages 117-182. The Wessel deflection circuit may be energized from an unregulated direct voltage, and uses a single switching transistor to perform the switching function for the horizontal deflection and for nondissipative switching regulation. In the Wessel circuit, the unregulated direct voltage is applied across the primary winding of a transformer by the switching transistor. The deflection winding, retrace capacitor and damper diode associated with the horizontal deflection are coupled across the collector-emitter path of the switching transistor by a first diode poled for conduction in the same direction as the collector-emitter path. A secondary winding of the transformer is coupled across the deflection winding by a second diode poled to conduct and transfer energy from the primary to the deflection winding during the retrace interval. It is desirable to eliminate the secondary winding, and thereby reduce the total number of windings.
A horizontal deflection circuit in which the secondary winding is eliminated is described in U.S. Pat. No. 3,906,307 issued Sept. 16, 1975 in the name of J. Van Hattum. However, in the Van Hattum arrangement, an additional inductor and capacitor are used. The necessity for the additional inductor negates the advantage of elimination of the secondary winding.
SUMMARY OF THE INVENTION
A self-regulating deflection circuit includes a first inductor and controllable switch serially coupled across a source of unregulated direct voltage to form a first series path for storing energy in the first inductance during the intervals in which the switch is conductive. A first rectifier couples a parallel combination of elements across the switch, the parallel combination including a second inductance, a damper diode and retrace capacitor. The first rectifier is poled for current conduction in the same direction as the switch. A control circuit coupled to the second inductance and with the switch recurrently operates the switch for promoting current flow in the second inductance during recurrent trace and retrace intervals, and maintains the peak value of the current flow at a constant level. A second rectifier is coupled by a second capacitance with the parallel combination of elements and to a point on the first series path for transferring energy from the first inductance to the parallel combination of elements during the retrace intervals. A resistance is coupled to the second capacitance for equalizing charge on the second capacitance during the trace interval.
DESCRIPTION OF THE DRAWING
FIG. 1 illustrates partially in block and partially in schematic form a portion of the deflection circuit of a television display device embodying the invention; and
FIG. 2 illustrates voltage-and current-time waveforms occurring in the arrangement of FIG. 1 during operation.
DESCRIPTION OF THE INVENTION
In FIG. 1, a power supply designated generally as 10 includes a rectifier represented by a diode 16 and a filter capacitor 18 coupled to terminals 12 and 14 adapted to be coupled to the alternating-current power mains. Unregulated direct voltage appearing across capacitor 18 energizes a horizontal deflection circuit designated generally as 20.
Deflection generator 20 includes an inductor 22 connected at one end to capacitor 18 and at the other end to the collector of an NPN switching transistor 24, the emitter of which is connected to ground. The cathode of a diode 26 is connected to the collector of transistor 24, and its anode is connected to the cathode of a damper diode 32, the anode of which is connected to ground. A retrace capacitor 28 is coupled in parallel with diode 32. A deflection winding 34 is serially coupled with an S-shaping capacitor 36, and the serial combination is coupled in parallel with capacitor 28. A primary winding 38a of a transformer 38 is coupled at a terminal 37 with the anode of diode 26. The other end of primary winding 38a is connected at a terminal 39 with one end of a storage capacitor 40, the other end of which is grounded. A high-voltage secondary winding 38b of transformer 38 has one end grounded and the other end connected to an ultor rectifier represented as a diode 44 for producing high voltage for application to the ultor of a kinescope, not shown. Another secondary winding 38c of transformer 38 has a grounded center-tap and the ends connected to rectifier diodes 46 and 48 for producing operating voltages for the low-voltage portions, not shown, of the television device.
A dc blocking capacitor 52 is serially connected with a diode 50, and the serial combination is coupled between the collector of transistor 24 and a point on winding 38a. The cathode of diode 50 is connected to winding 38a, and the anode is coupled to the collector of transistor 24. A resistor 54 has one end connected to capacitor 52 at a circuit point 56, and the other end is coupled to the end of capacitor 52 remote from point 56 so as to form a parallel connection.
A synchronized pulse-width modulator illustrated as a block 60 is coupled to capacitor 40 for sampling the voltage appearing thereacross. Modulator 60 receives horizontal synchronizing pulses illustrated as 64 at an input terminal A. Modulator 60 produces pulses in known manner, the time duration or width of which are controlled in response to the voltage across capacitor 40, and the pulses are applied by way of a conductor B to a driver circuit illustrated as a block 66. Driver 66 replicates or, if desired, shapes the pulses in a known manner and applies them to the base of switching transistor 24 to control its collector-emitter conduction in a switching manner.
The waveforms of FIG. 2 in the intervals T0-T5, T5-T10 and T10-T15 exemplify operation for low, correct, and excessive deflection energy, respectively. The interval T4-T10 is representative and will be used to describe details of the circuit operation.
In operation during the last half of the horizontal scanning or trace intervals preceding time T5, the collector-emitter path of transistor 24 is conductive, and current is increasing in inductor 22 as illustrated by waveform I22 of FIG. 2f in the interval following time T4. The current in inductor 22 flows through the collector-emitter path of transistor 24. During this same interval immediately following the time T4, which is the time of the center of the horizontal trace interval, current is flowing in deflection winding 34 as illustrated by waveform I34 of FIG. 2d, and is increasing under the impetus of the voltage on capacitor 36. The current in winding 34 flows through diode 26 and adds to the collector-emitter current flowing in transistor 24, as illustrated by waveform I24 of FIG. 2h. A current flows through winding 38a under the impetus of the voltage on capacitor 40, which current adds to the deflection current flowing through diode 26 and transistor 24. Winding 38a is in parallel with winding 34 and they may be viewed as being a single inductor through which a single current proportional to the deflection current flows. In the interval between times T4 and T5, diode 50 is reversed-biased by a voltage, poled as shown, on capacitor 52.
The deflection current and the current in inductor 22 continues to increase until a time such as T5 at which a horizontal synchronizing pulse 64 as illustrated in FIG. 2a is applied to modulator 60. Modulator 60 responds by producing a transition of voltage V60 on conductor B as illustrated in FIG. 2b. Voltage V60 causes driver 66 to render the collector-emitter path of transistor 24 nonconductive. This initiates the retrace interval, which extends from time T5 to T7. During the first portion T5-T6 of the retrace interval, winding 34 (together with winding 38a) transfers the energy stored in its magnetic field to capacitor 28 in a resonant manner, causing the voltage at circuit point 37 to rise as illustrated by V37 of FIG. 2c.
The voltage at terminal point 39 remains substantially unchanged during the retrace interval because of the filtering effect of capacitor 40. Consequently, the voltage at a point along winding 38a will rise during the retrace interval in an amount depending upon how remote the point is from circuit point 39. Thus, the voltage at the cathode of diode 50 will depend upon the exact point on winding 38a at which the cathode is connected.
When transistor 24 is rendered nonconductive at time T5, the voltage across inductor 22 rises so as to maintain the current of transistor 24 therefore rises and forces the current through capacitor 52 and forward-biased diode 50 to winding 38a and capacitor 40, resulting in an energy transfer thereto. The voltage across inductor 22 during the retrace interval determines the rate at which energy is transferred during this interval from winding 22 to winding 38a and the remainder of the deflection circuit. The voltage across winding 22 during this interval is the algebraic sum of the voltage which is then on capacitors 18, 40 and 52, the voltage produced by the inductance of winding 38a, and the forward voltage drop of diode 50. During this retrace interval, voltage is coupled from winding 38a to windings 38b and 38c for rectification and energization of the remainder of the television device.
The first half of the retrace interval ends at a time T6 as the current in windings 34 and 38a is reduced to zero and the voltage on retrace capacitor 28 peaks. Voltage V37 represents the voltage across the retrace capacitor. During the second half of the retrace interval, diode 50 continues to conduct a decreasing current as illustrated by I50 of FIG. 2i as energy is transferred to winding 38a and capacitor 40 from winding 22. Also during the second half of the retrace interval, the current in windings 34 and 38a reverses and increases to a peak at a time 27 as illustrated by I34. As the current in winding 34 increases to a peak in the negative direction, the voltage at circuit point 37 decreases towards zero as illustrated by V37 of FIG. 2c. The retrace interval ends at a time T7 as V37 reaches zero and damper diode 32 conducts.
During the first half T7-T9 of the following trace interval, the current in winding 34 decreases as its energy is transferred to capacitor 36. During a first portion T7-T8 of the trace interval, transistor 24 is maintained nonconductive. The remaining energy in winding 22 continues to cause current to flow through capacitor 52 and diode 50. The collector voltage VC24 of transistor 24 during this interval is maintained at a voltage equal to the algebraic sum of the voltage on capacitors 40 and 52, the voltage caused by winding 38a, and the forward junction potential of diode 50, as illustrated in FIG. 2e.
At a time T8, modulator 60 produces a gating pulse V60 which is coupled to transistor 24 to render it conductive. When transistor 24 becomes conductive, its collector goes to ground potential, coupling winding 22 across capacitor 18 to commence the energy storage portion of the deflection cycle. At the same time, the positive end of capacitor 52 is coupled to ground, placing a negative potential as illustrated by V56 of FIG. 2g on the anode of diode 50, which cuts it off. During the remainder of the trace interval, the increasing current in winding 22 flows through the collector-emitter path of transistor 24.
At a time T9, the deflection current in winding 34 reaches zero, and capacitor 36 has reached its maximum potential. Diode 32 becomes nonconductive. The voltage at junction point 37 rises until diode 26 becomes conductive, and current begins to flow through deflection winding 34 under the impetus of the voltage on capacitor 36. This current flows through diode 26 and the collector-emitter path of transistor 24, as illustrated by I24. The currents in windings 22 and 34 continue to increase until the end T10 of the deflection interval, at which time transistor 24 is rendered nonconductive to create a retrace voltage pulse at circuit point 37 and cause energy transfer from winding 22 to winding 38a.
In the interval between times T5 and T10, modulator 60 produces a gating pulse V60 rendering transistor 24 conductive at times during the first half of trace interval. During the interval T5-T8 in which transistor 24 is nonconductive, current in inductor 22 decreases and energy is transferred therefrom into winding 38a and capacitor 40. In the interval T8-T10 in which transistor 24 is conductive, current increases in winding 22 as it stores energy derived from the unregulated direct voltage. Time T8 is selected as that time which results in the peak value of current I22 being equal from one horizontal cycle to the net so as to maintain substantially the same transfer of energy from winding 22 to the deflection components in order to compensate for the losses during the deflection cycle. These losses include dissipative losses and energy transferred to the kinescope ultor.
In the event that the losses during successive deflection cycles exceed the energy transferred from inductor 22, less energy than desired will circulate through deflection system during each cycle, resulting in reduced raster width. The voltage across capacitor 40 will decrease as a result of this decreased energy and modulator 60 will produce a gating waveform V60 at a time T3 occurring earlier during the deflection cycle than corresponding time T8. This reduces the time T0-T3 in which current I22 decreases, and increases the interval T3-T5 in which voltage is applied to inductor 22 in a polarity to increase the current. Consequently, at a time T5 at the end of the deflection interval, the energy stored in the magnetic field of inductor 22, as measured by current I22, will exceed that at time T0. This results in an increased energy transfer which restores the circulating energy and the voltage across capacitor 40.
Similarly, when the loads on winding 38a decrease and the circulating energy increases, the voltage on capacitor 40 will increase, and modulator 60 will gate transistor 24 into conduction at a time T13 which is later relative to the deflection cycle than time T8. This allows a greater time T10-T13 in which current I22 can decrease and reduces the time T13-T15 in which the current can increase, thereby resulting in reduced current in inductor 22 at the end of the deflection cycle and reduced energy available for transfer to the deflection components, thereby restoring the voltage across capacitor 40 and maintaining the raster width. Time T13 at which transistor 24 is rendered conductive cannot be selected later than time T14 of the center of scan, because of the resulting raster distortion.
The point on winding 38a at which the cathode of diode 50 is connected may be selected at the end of winding 38a corresponding to circuit point 39. Substantial regulation results at all points along winding 38a to which the cathode of diode 50 may be connected. However, some changes in the waveforms occur. Current I222 of FIG. 2f represents the current in winding 22 when the cathode of diode 50 is coupled to circuit point 37, and current I250 of FIG. 2i represents the corresponding current in diode 50.
In the absence of resistor 54, the unidirectional current flow through capacitor 52 and diode 50 will tend to raise the voltage across capacitor 52 to a very high value in the polarity shown. If charge is allowed to accumulate on capacitor 52 in this manner, the voltage across capacitor will soon equal the maximum voltage which can occur at the collector of transistor 24, and diode 50 will cease to conduct during the retrace intervals, no energy will be transferred to the deflection components to compensate for the losses during the deflection cycle, and the circuit will cease to operate.
Resistor 54 is provided as a path for preventing accumulation of excess charge across capacitor 52. As the voltage across capacitor 52 increases, the rate at which charge is drained away through resistor 54 also increases. The end of resistor 52 remote from circuit point 56 can be coupled to any point of reference potential, such as B+ or ground, in order to achieve the desired discharge of capacitor 52. Reduced power dissipation results from coupling resistor 54 in parallel with capacitor 52, as illustrated in FIG. 1. With this arrangement, circuit point 56 takes on a negative potential during those portions of the horizontal scanning interval in which transistor 24 is conductive as illustrated by V56.
Other embodiments of the invention will be apparent to those skilled in the art. In particular, the positions of serially coupled diode 50 and capacitor 52 may be interchanged. Impedance-matching considerations may require either the collector of transistor 24 or the serial combination of diode 50 and capacitor 52 to be coupled to a tap on winding 22.