LOEWE F875 SENSOTRONIC CHASSIS F800 (543/490) INTERNAL VIEW.

A very singular chassis technology:

The LOEWE F875 SENSOTRONIC CHASSIS 543/490 is realized with the F800 LOEWE Chassis serie and in the Line / Horizontal deflection output like a COLOR TELEVISION CHASSIS with EHT output with Tripler and Horizontal output with thyristor tech.

Thyristor used:

RCA 16490
RCA 16491





















Furthermore this chassis was introducing the improved LOEWE IF UNIT based around TELEFUNKEN TDA440.























INTEGRATED circuits are slowly but surely taking over more and more of the circuitry used in television sets even B/W.
The first step, some many years ago now, was to wrap the 6MHz intercarrier sound strip into a neat package such as the TAA350 or TAA570. Then came the "jungle" i.c. which took over the sync separator and a.g.c. operations. Colour receiver decoder circuitry was the next obvious area to be parcelled up in i.c. form,  two i.c. decoder and the more sophisticated Philips four i.c. design was coming on the scene. The latter is about to be superseded by a three i.c. version in which the TBA530 and TBA990 are replaced by the new TCA800 which provides chrominance signal demodulation, matrixing, clamping and preamplification, with RGB outputs of typically 5V peak -to -peak.
To improve performance a number of sets adopted a synchronous detector i.c.-the MC1330P -for vision demodulation, which of course overcomes the problem of quadrature distortion. In one monochrome chassis this i.c. is partnered by a complete vision i.f. strip i.c., the MC1352P. In the timebase section the TBA920 sync separator/line generator i.c. has found its way into several chassis was a Texas's SN76544N 07 i.c. which wraps up the sync separator and both the field and line timebase generators has come into use. Several monochrome portables have had in use a high -power audio output i.c. as the field output stage. Audio i.c.s are of course common, and in several  chassis the Philips TCA270 has put in an appearance. This device incorporates a synchronous detector for vision demodulation, a video preamplifier with noise inversion and the a.g.c. and a.f.c. circuits. The   development to be adopted in a production chassis was that remarkable Plessey i.c., the SL437F, which combines the vision i.f. strip, vision demodulator, a.g.c. system and the intercarrier sound channel.

SGS-Aces Range
Now, from the, at the time,  Italian Development Division of SGS-Ates, comes a new range of i.c.s which SGS  will set a standard pattern for TV chassis IN 1975. How this range combines to provide a complete colour receiver is shown in Fig. 1. The only sections of the receiver left in discrete component form are the video output stages, the tuner, the a.f.c. circuit and of course the line output stage and power supplies. It will be seen that the colour decoder section is split up as in the Philips three i.c. design. The TDA1150 chrominance and burst channel carries out the same functions as the TBA560, the TDA1140 reference section the same functions as the TBA540 and the TDA1160 chrominance demodulator/matrix- ing i.c. the same functions as Philips's new TCA800. It looks therefore as if this basic decoder pattern could become widely established. The other five i.c.s in the range are common to both colour and monochrome receivers. Particularly interesting are the TDA1170 which comprises a complete monochrome receiver field timebase-for colour set use an output stage using discrete com- ponents is suggested-and the TDA440 which incorporates the vision i.f. strip, vision detector and a.g.c. circuitry. The intercarrier sound i.f. strip is neatly packed away with the audio circuitry in the TDA1190 while the TDA1180 sync separator/line oscillator i.c. is a very similar animal to the now well known TBA920. The fifth i.c., the TBA271, is a stabiliser for the varicap tuner tuning supply. The novel i.c.s in this family then were the TDA 440, TDA1170 and the TDA1190 and we shall next take a closer look at each of these.

Vision IF IC:
The TDA440 vision i.f. strip i.c. is housed in a 16 -pin plastic pack with a copper frame. There is a three -stage vision i.f. amplifier with a.g.c. applied over two stages, synchronous vision demodulator, gated a.g.c. system and a pair of video signal pre amplifiers which provide either positive- or negative - going outputs. Fig. 2 shows the i.c. in block diagram form. It is possible to design a very compact i.f. strip using this device and very exact performance is claimed. Note that apart from the tuned circuits which shape the passband at the input the only tuned circuit is the 39.5MHz carrier tank circuit in the limiter/demodulator section. The only other adjustments are the tuner a.g.c. delay potentiometer and a potentiometer (the one shown on the right-hand side) which sets the white level at the demodulator. This of course gives ease of setting up, a help to setmaker and service department alike. For a sensitivity of 200/4V the output is 3.3V peak - to -peak, giving an overall gain in the region of 82 to 85dB. The a.g.c. range is 55dB, a further 30 to 40dB being provided at the tuner. The tuner a.g.c. output is intended for use with a pnp transistor or pin diode tuner unit: an external inverter stage is required with the npn transistor tuner units generally used. discrete component video output stage; in a colour In a monochrome set the output would be fed to a design the output is fed to the chrominance section of the TDA1150 and, via the luminance delay line, to the luminance channel in the TDA1150. Also of course in both cases to the sync separator which in this series of i.c.s is contained in the TDA1180.

Field Timebase IC :
The TDA1170 field timebase i.c. is shown in block diagram form in Fig. 3. The i.c. is housed in a 12 -pin package with copper frame and heat dissipation tabs. It is capable of supplying up to 1.6A peak -to -peak to drive any type of saddle -wound scanning yoke but for a colour receiver it is suggested that the toroidal deflection coil system developed by RCA is used. In this case the i.c. acts as a driver in conjunction with a complementary pair of output transistors. The yoke current in this case is in the region of 6A. The TDA1170 is designed for operation with a nominal 22V supply. It can be operated at up to 35V however. A voltage doubler within the i.c. is brought into action during the flyback time to raise the supply to 70V. Good frequency stability is claimed and the yoke current stability with changes in ambient temperature is such that the usual thermistor in series with the field coils is not required. For monochrome receiver use the power supplied to the yoke would be 0-83W for a yoke current of lA peak -to -peak with a 1012 coil impedance and 20V supply. As the power dissipation rating of the i.c. is 2.2W no further heatsink is required. For use in a colour receiver with a toroidal coil impedance of 1.6Ohm the scanning current would be 7A peak -to -peak. The power supplied to the yoke may be as much as 6.5W while the dissipation in the i.c. would be up to 2-3W. In this case a simple heatsink can be formed from a thin copper sheet soldered to the heat fins- an area of about 3-4 sq. in. should be adequate. The sync circuit at the input gives good noise immunity while the difference between the actual and ideal interlace is less than 0-3% of the field amplitude. Because of the high output impedance a relatively low value (1/iF or less) output coupling capacitor can be used. This means that mylar types instead of electrolytics can be used, reducing the problems of linearity and amplitude stability with respect to temperature and ageing. The external controls shown in Fig. 3 are hold, height and linearity (from left to right).

Complete Sound Channel:
The TDA1190 sound channel (see Fig. 4) is housed in a 12 -pin package. Possible radiation pick-up and thermal feedback risks have been avoided by careful layout of the chip. This pack also has a copper frame, with two cooling tabs which are used as the earthing terminals. The built-in low-pass filter overcomes radiation problems and with a response 3dB down at 3MHz allows for a flat amplitude response throughout the audio range: this particular feature will appeal to hi-fi enthusiasts as well since it makes the i.c. a good proposition for f.m. radio reception. The d.c. volume control has a range of 100dB. The external CR circuit (top, Fig. 4) sets the closed - loop gain of the power amplifier. The external feedback capacitor network (right) provides a.f. bandwidth and frequency compensation while the CR circuit across the output limits any r.f. which could cause severe audio distortion. The TDA1190 does not require an extra heatsink when operating in normal ambient temperatures-up to 55°C-because of the new technique of soldering the chip directly on to the copper frame that forms part of the external tabs. By doing this, SGS-Ates have reduced the thermal resistance of the device to 12°C per watt. The device can dissipate up to 2.2W at 55°C without using an external heatsink other than the printed circuit pad (about 2 sq. in.) which is soldered to the tab. The output stages of the TDA1190 are in quasi - complementary mode (with patented features), eliminating the need for bootstrap operation without loss of power. The absolute maximum output power is 4.2W with a supply voltage of 24V and a nominal loudspeaker impedance of 1612. At 12V and 812 an output of 1.8W can be achieved. Total harmonic distortion is 0.5% for 1 mV f.m. input and 2W output into 1611 at 24V. Satisfactory operation is possible over a voltage supply range of 9 to 28V, making this versatile i.c. suitable for a wide range of applications. The whole audio circuit can be mounted on a p.c.b. 2in. x 25in. without a heatsink.

Mounting: The complete family of i.c.s has been designed so that it can be incorporated in very small and simple printed circuit modules. The use of a copper frame assists in improving the thermal stability as well as facilitating the mounting of the i.c.s on the board. Where an extra heatsink is required this can be a simple fin added to the mounting tabs or a metal clamp on the top of the pack. SGS claim that insta- bility experienced with conventional layouts in colour receivers has been eliminated provided their recommendations are observed.

Power Supplies:
A simple power supply circuit without sophisticated stabilisation can be used. The requirements are for outputs ranging between 10V and 35V with adequate decoupling and smoothing. It was  possible to provide only three supply lines to feed the whole receiver system-plus of course the high- voltage supplies required by the c.r.t. The power supply requirements are simplified since the TDA1170 incorporates a voltage regulator for its oscillator, the TDA440 incorporates a regulator for the vision i.f. strip and the TDA1190 a regulator for the low -voltage stages and the d.c. volume control.


LOEWE-OPTA Start-delay overload protection arrangement for a thyristor sweep circuit:An input switching thyristor of a deflection circuit is selectively excited at a sweep rate by a transistorized driver stage. A separate protective thyristor is disposed in a DC power source that supplies operating bias to the driver stage and to the switching thyristor via a supply capacitor. Initially, the supply capacitor is partially charged through a high-ohmic shunt resistor that budges the protective thyristor, whose control electrode is coupled to the output of the switching thyristor. A portion of the resulting voltage across the partially charged supply capacitor is applied via a voltage divider and a Zener diode to the base of a threshold-operated transistor that normally blocks the driver stage. When the supply capacitor has charged to a predetermined minimum value necessary to assume turn-on of the protective thyristor, the threshold-operated transistor is triggered to permit the driver stage to operate the switching thyristor at the sweep rate. The resulting turn-on of the protective thyristor permits the rapid completion of charge of the supply capacitor to its normal operating voltage, and thereafter provides full overload protection for the deflection circuit.


1. In a communications receiver wherein (a) a first input switching thyristor of a deflection circuit in the receiver is pulsed at a sweep rate by a driver stage when the latter is excited, (b) a supply capacitor is chargeable through a second protective thyristor by-passed by a high-impedance shunt to provide operating bias to both the driver stage and the switching thyristor, and, (c) the output of the switching thyristor is coupled to the control electrode of the protective thyristor, the improvement which comprises: 2. The improvement as defined in claim 1, in which the gating means comprises a first transistor having its collector emitter path connected in series with the driver stage, and in which the coupling means comprises, in combination, a voltage divider having its input terminals connected across the supply capacitor, and means for applying the output of the voltage divider to the base of the first transistor. 3. The improvement as defined in claim 2, in which the applying means comprises a Zener diode. 4. The improvement as defined in claim 2, in which the driver comprises a second transistor, and in which the first and second transistors are of opposite conductivity type.

Description:

BACKGROUND OF THE INVENTION

One conventional horizontal sweep circuit for a communications receiver has an input switching thyristor which is excitable to trigger the flyback phase at the deflection transformer. For this purpose, the control electrode of the switching thyristor is supplied with pulses at a sweep rate via a transistorized driver stage, whose base is coupled to the output of a horizontal oscillator. A common DC power source coupled to AC mains provides operating bias to the switching thyristor and to the driver stage via an output supply capacitor.

In order to provide overload protection of the deflection circuit it has been proposed to arrange a second protective thyristor in shunt with a high-ohmic bypass resistor in the DC power sound between the main rectifier and the supply capacitor. The voltage at the output of the switching thyristor (determined by the magnitude of the operating bias) is fed back in integrated form to the control electrode of the protective thyristor to maintain the latter in a conductive stage as required. In the presence of an overload in the deflection circuit, the output voltage of the switching thyristor drops to a value insufficient to trigger the protective thyristor on, thereby effecting the removal of operating bias from the deflection circuit.

While this arrangement is generally satisfactory for protection purposes once the deflection circuit is fully operative, its design is such that during start-up of the deflection circuit the driver stage is effectively controlled independently of the DC power source. As a result, start-up of the deflection circuit can occur before the voltage amplitude at the output of the supply capacitor (and thereby at the output of the switching thyristor) has reached a value necessary to assume reliable turnon of the protective thyristor.

SUMMARY OF THE INVENTION

The present invention provides an arrangement for assuring the reliable turn-on of the protective thyristor in the DC source during the start-up of the deflection circuit. In an illustrative embodiment the collector-emitter path of a normally disabled, threshold-operated transistor gate is serially connected with the transistorized driver stage to normally prevent excitation of the latter by the supply capacitor of the DC source. Consequently, the driver stage is initially prevented from starting up the deflection circuit. The base of the threshold transistor is excited by a voltage divider coupled across the supply capacitor.

The rectified current in the DC source initially charges the supply capacitor through the high-ohmic shunt resistor that bridges the normally off protective thyristor. When the voltage across the supply capacitor has reached a predetermined minimum value necessary to assume turn-on of the protective thyristor, the transistor gate is triggered on and the driver stage is excited to supply pulses to the switching thyristor. The rectified current of the power supply can now rapidly complete the charging of the supply capacitor to its full operating voltage through the now-conductive protective thyristor.

BRIEF DESCRIPTION OF THE DRAWING

The invention is further set forth in the following detailed description taken in conjunction with the appended drawing, in which:

FIG. 1 -- is a combined block and schematic diagram of a start-delay deflection circuit protection arrangement in accordance with the invention; and

FIG. 2 -- is a schematic diagram showing in more detail a threshold circuit useful in the arrangement of FIG. 1.

DETAILED DESCRIPTION

Referring now to the drawing, FIG. 1 illustrates a position of a communication receiver 49 having a deflection circuit 50, wherein a deflection transformer 12 applies a sweep voltage to the horizontal plates of a cathode ray tube (not shown). The deflection circuit includes an input switching thyristor 7 which may be employed, i.e. to establish the flyback phase of the transformer 12. The transconductive path of the thyristor 7 is shunted by a diode 8 of the opposite polarity. The output of the thyristor 7 is coupled through a series inductor 9 and a storage capacitor 10 to a forward sweep 11, which is schematically shown as having a forward sweep thyristor 51 shunted by an oppositely poled diode 52. The output of the circuit 11 is coupled to a primary winding 12A of the transformer 12, which is returned to ground via a capacitor 13.

In the normal operation of the deflection circuit 50, the control electrode of the switching thyristor 7 is supplied with pulses at the sweep rate from a driver stage 23 embodied by a PNP transistor. The base of the transistor 23 is coupled via an isolating capacitor 27 to the output of a horizontal oscillator 28. The required pulses for exciting the switching thyristor 7 are supplied to its control electrode via the collector of the transistor 23 through capacitor 31. The base of the transistor 23 is returned to ground through a resistor 29.

Operating bias for the transistor 23 and the deflection circuit 50 is supplied from a capacitor 6 disposed at the output of a DC power source 56, whose input terminals coupled to conventional AC mains are shown. In particular, the output of the capacitor 6 is coupled via a voltage divider 25, 26 to the emitter of the driver transistor 23; such emitter is returned to ground via capacitor 24. The output of the capacitor 6 is also coupled through a coil 14 to the deflection circuit 50 for excitation of the transconductive path of the thyristor 7.

The power supply 56 includes a diode 1 whose rectified output is effective to change the capacitor via a resistor 5 and the parallel combination of a normally disabled protective thyristor 2 and a high-impedance shunt resistor 3. In a conventional manner, the control electrode of the thyristor 2 is coupled via resistor 15 and diode 16 to the output of the switching thyristor 7, whose voltage amplitude is determined by the potential on the capacitor 6. Such output voltage is integrated by a capacitor 17 to yield an excitation potential for the thyristor 2 that is greater than its turn-on potential, so that thyristor 2 is ideally rendered conductive during alternate half cycles of the AC means. The capacitor 17 is returned to ground via a capacitor 4.

Theoretically, the protective thyristor 2 is switched to its "on" condition as soon as the driver transistor 23 applies excitation pulses to the control electrode of the transistor 7. In practice, the horizontal oscillator 28 that controls the transistor 23 is independent of the power supply 56, and there is no assurance that the voltage developed across the capacitor 6 and topped off the output of the switching thyristor 7 during turn-on of the deflection circuit will be sufficient to ignite the protective thyristor 2. In accordance with the invention, facilities are provided for preventing turn-on of the deflection circuit 50 until the voltage across the capacitor 6 has reached a predetermined minimum value necessary to assure the turn-on of the protective thyristor 2. In the embodiment shown, such facilities include a normally disabled threshold-operated NPN transistor 21 whose emitter is returned to ground and whose collector is coupled to the collector of the driver transistor 23 through a resistor 22. The base of the transistor 21 is coupled to a center tap 20 of a voltage divider 18, 19 connected in parallel with the supply capacitor 6. The parameters of the voltage divider 18, 19 and the threshold level of the transistor 21 are preferably chosen so that the transistor 21 is triggered on when the voltage at the tap 20 corresponds to the voltage across the capacitor 6 necessary to assume turn-on of the protective thyristor 2.

In operation, when the power supply 56 is first made operative, the rectified current from diode 1 partially charges the capacitor 6 through the high-impedance shunt 3. When the voltage across the capacitor 6 has risen to the point where the potential at the center tap 20 exceeds the threshold voltage of the transistor 21, the latter conducts and permits operating bias from the capacitor 6 to be applied to the driver transistor 23. The latter accordingly conducts to excite the control electrode of the switching thyristor 7 to effect turn-on of the deflection circuit 50.

Under the circumstances just described, the delayed start-up of the deflection circuit 50 is always accompanied by the ignition of the protective thyristor 2. As soon as the latter is made conductive, the rectified current from the diode 1 can rapidly complete the charging of the capacitor 6 to its normal operating voltage. The deflection circuit 50 thereafter operates in its normal, fully protected mode.

In order to increase the stability of the thresold level of the transistor 21, a Zener diode 30 (FIG. 2) may be disposed between the center tap 20 and the base of the transistor 21, as shown. Thus, triggering of the transistor 21 will occur only when the potential at the center tap 20 exceeds the breakdown level of the diode 30.

In the foregoing, the invention has been described in connection with a preferred arrangement thereof. Many variations and modifications will now occur to those skilled in the art. It is accordingly desired that the scope of the appended claims not be limited to the specific disclosure herein contained.

LOEWE-OPTA Energy stabilization in a horizontal deflection circuit for a television receiver:A thyratron-like gating device governs the current supplied to an inductance in energy-exchanging relation with the storage capacitor in a reactive horizontal sweep circuit. The gating device is triggered by a control pulse that is delayed with respect to the start of the flyback portion of the sweep cycle by an amount proportional to the magnitude of the voltage stored on the capacitor at the start of flyback. The control pulse is generated upon the coincidence of a first saw-tooth voltage triggered at the start of flyback and a second voltage derived from the magnitude of the pulse generated in a secondary winding of the deflection transformer at the start of flyback. Suitable gain control facilities limit the magnitude of the second voltage to a value that assures generation of the control pulse within the flyback interval.

1. In a deflection circuit having a repetitive sweep cycle including a first relatively short flyback interval and a second relatively long forward sweep interval wherein the circuit comprises, in combination, a first storage capacitor coupled between a supply inductor and the primary winding of a deflection transformer, first means for coupling the input of the inductor to a source of operating voltage whereby an oscillatory energy interchange occurs between the operating voltage source and the output of the deflection circuit via the inductance and the first storage capacitor, and a first thyratron-like gating means associated with the inductor, the switching time of the first gating means being varied in dependence on changes in the quantity of energy in such oscillatory energy interchange, the improvement wherein the first gating means is serially connected between the source of operating voltage and the input of the inductor, and wherein the circuit further comprises, in combination, first means for generating, during each first interval, an output control pulse of substantially fixed duration whose time of occurrence relative to the start of the associated first interval is proportional to the voltage stored on the first capacitor at the start of such first interval, and second means for coupling the output of the first generating means to the control electrode of the first gating means, whereby the time of occurrence of the output control pulse is automatically adjusted to maintain a constant dynamic current-voltage ratio at the output of the deflection transformer. 2. A circuit as defined in claim 1, in which the first generating means comprises, in combination, comparator means having first and second inputs and an output adapted to exhibit a pulse upon a coincidence of voltage amplitudes applied to the first and second inputs, second means triggered at the start of the first interval for generating a first sawtooth voltage whose magnitude reaches a first value at the conclusion of the first interval, third means for generating a second voltage proportional to the magnitude of the voltage stored on the first capacitor at the start of the first interval, third means for coupling the output of the second generating means to the first input of the comparator means, and fourth means for coupling the output of the third generating means to the second input of the comparator means. 3. A circuit as defined in claim 2, in which the third generating means further comprises, in combination, a first secondary winding of the deflection transformer, and means for rectifying the output of the first secondary winding. 4. A circuit as defined in claim 2, in which the third generating means further comprises gain control means for limiting the maximum value of the second voltage to the first value. 5. A circuit as defined in claim 3, in which the third generating means comprises, in combination, second and third gating means, second and third capacitors, means for coupling the output of the rectifying means across the second and third capacitors, respectively, means for producing a third voltage proportional to the difference between the voltage across the second capacitor and a reference value, fifth means for coupling the third voltage to the control electrode of the second gating means so that such second gating means conducts when the third voltage is within a prescribed range, sixth means for coupling the voltage across the third capacitor to the control electrode of the third gating means, and seventh means for coupling the output of the second gating means to the control electrode of the third gating means, the third gating means being made conductive upon the conduction of the second gating means. 6. A circuit as defined in claim 5, in which the deflection transformer includes a second secondary winding, and in which the circuit further comprises, in combination, a common terminal for exciting the second and third generating means, a diode for coupling the second secondary winding to one terminal of the diode, eighth means for coupling the other terminal of the diode to the common terminal, and ninth means for coupling the third capacitor to the common terminal, the diode being rendered non-conductive to decouple the second secondary winding from the common terminal in the presence of voltage across the third capacitor.

Description:

BACKGROUND OF THE INVENTION

Triggered reactive-type deflection circuits are frequently employed to effect the horizontal sweep portion of a television raster. The deflection circuit operates into the primary winding of a deflection transformer, one secondary winding of which may be coupled to the horizontal deflection plates of the picture tube.

The deflection circuit commonly includes a storage capacitor connected in oscillatory energy-exchanging relation with an input supply inductance, which in turn is coupled to a source of operating voltage. A pair of normally unoperated switching devices (typically thyristors) are connected on the output and input sides, respectively, of the capacitor. Such switching devices are separately triggerable to initiate the flyback and forward sweep portions, respectively, of the deflection circuit.

Certain deflection circuit arrangements of this type employ an additional secondary winding on the deflection transformer to trigger a regulating circuit that maintains a constant relation between dynamic deflection current changes (caused, e.g. by beam current variations in the picture tube) and the resulting changes in the voltage developed by the deflection circuit during the oscillations of energy between the operating voltage source and the deflection circuit load.

In one proposed arrangement of this type, facilities are provided for deriving, from a flyback pulse picked up by the additional secondary winding, a voltage proportional to the voltage on the storage capacitor at the start of the flyback interval. Such derived voltage is employed to vary the inductance of a regulating element coupled to the input of the deflection circuit. Such variation of the element inductance correspondingly varies the oscillatory energy exchange of the deflection circuit.

This type of control scheme has several main disadvantages. The -variable regulating element, besides being expensive to instrument, is bulky and occupies a large space in the television receiver. Moreover, this type of control scheme tends (except when operating at a particular load point such as maximum load current) to overcompensate for dynamic energy changes resulting in the application of an excess amount of correcting energy to the deflection circuit. Such excess is recycled between the capacitor in the deflection circuit and the power supply capacitor of the operating voltage source, and leads to relatively high power losses in the circuit.

SUMMARY OF THE INVENTION

The present invention provides an improved arrangement for stabilizing the energy conditions in a deflection circuit having a thyratron-like switch associated with an input storage inductance, while avoiding the above-mentioned disadvantages. In one embodiment, such thyratron-like switch (hereafter first gating device) is interposed in series between the operating voltage source and the input of the supply inductance. The first gating device is triggered at a point during the flyback interval determined by the magnitude of the flyback pulse picked off the additional secondary winding and thereby the voltage stored on the capacitor at the start of such flyback interval.

With this arrangement, the decrease in the amplitude of the flyback pulse which initially accompanies an increase in the deflection load current causes a triggering of the first gating device earlier in the flyback interval, thereby resulting in a higher current developed by the supply inductor at the end of the flyback interval. The energy increment represented by such developed current flow is transferred directly to the capacitor to increase the voltage stored thereacross during the succeeding forward sweep interval, thereby avoiding recycling losses, the necessity of "dumping" the excess energy stored in the input inductance, and the requirement of blocking diodes and similar circuitry. Since in this way the above-mentioned increase in load current effects a corresponding increase in the deflection circuit voltage, their ratio is maintained constant as desired.

A feature of the invention is the provision of facilities for converting variations in the amplitude of the flyback pulse to corresponding variations in the times of occurrence of the control pulses for the first gating device. An auxiliary saw-tooth voltage triggered at the start of the flyback interval is applied to one input of a comparator, e.g. a transistorized phase-shifting stage. To the other input of the comparator circuit is coupled a second voltage proportional to the amplitude of the flyback pulse. This second voltage is advantageously developed by a gain-controlled amplifier arranged so that the highest amplitude of the second voltage corresponds to the voltage level attained by the saw-tooth voltage at the end of the flyback interval.

Excitation of the saw-tooth generator, the comparator, and the gain-control amplifier may be accomplished over a common excitation path by either the second voltage or by a third voltage obtained from an additional auxiliary winding of the deflection transformer through a series diode. The presence of the second voltage disables such series diode to decouple the third voltage from the excitation path.

BRIEF DESCRIPTION OF THE DRAWING

The invention will be further set forth in the following detailed description taken in conjunction with the appended drawing, in which:

FIG. 1 is a combined block and schematic diagram of a regulated deflection circuit employing a thyratron-like gating device in accordance with the invention;

FIG. 2a is a composite graph showing several voltage and current wave forms in the arrangement of FIG. 1;

FIG. 2b and 2c are relative plots of the times of occurrence of trigger pulses respectively generated at the moment of triggering and at the start of the flyback interval of the gating device of FIG. 1;

FIG. 2d is a composite graph representing the production, by the arrangement of FIG. 1, of a thyratron control pulse at a given time during the flyback interval; and

FIG. 3 is a detailed schematic diagram of the circuit of FIG. 1.

DETAILED DESCRIPTION

Referring now to FIG. 1 of the drawing, there is pictorially represented a portion of a TV receiver including a picture tube 101. Several of the required operating voltages of the tube 101 and of the remainder of the receiver are supplied conventionally via a plurality of secondary windings of a transformer 8. Such windings include, e.g. a winding 10 for applying high voltage to the tube 101, a second winding 102 for exciting the horizontal deflection plates of the tube 101, and a third winding 103 for supplying filament current to the tube 101.

A primary winding 7 of the transformer 8 is driven by a triggered, substantially reactive horizontal deflection circuit 110. The circuit 110 establishes the horizontal portion of the television raster by generating a repetitive sweep voltage for application to the horizontal plates of the tube 101 via winding 102. Each cycle of such sweep voltage has a relatively short flyback portion starting at time T _O (FIG. 2a) and a relatively long forward sweep portion starting at time T ₃ .

The circuit 110 includes a main storage capacitor 6 coupled at its output to the primary winding 7, which is returned to ground via a capacitor 9. The storage capacitor 6 is coupled at its input to a main supply inductance 12 via a coil 5. The storage inductance 12 is coupled to a source of operating voltage V _B as described below.

The transconductive path of a first switching thyristor 3 is coupled between the output of capacitor 6 and ground, and is shunted by a diode 4 of opposite polarity. The transconductive path of a second switching thyristor 1 is coupled between the junction of inductors 5 and 12 and ground, and is likewise shunted by a diode 2 of opposite polarity.

The thyristor 1 is triggered at the time T _O of each sweep cycle to initiate the flyback interval. For this purpose, a suitable ignition generator 15 synchronized by a conventional horizontal sync pulse generator 14 is coupled to the control electrode of the thyristor 1. The generator 15 may be embodied by the horizontal oscillator 15 described, e.g., in U.S. Pat. No. 3,767,960 issued to P. R. Ahrens on Oct. 23, 1973.

The thyristor 3 is triggered at the instant T ₃ in each sweep cycle to terminate such flyback portion and initiate the forward sweep portion. This is accomplished by coupling an auxiliary winding 111 of the supply inductance 12 to the control electrode of the thyristor 3 over a coupling circuit 112.

In order to stabilize the load conditions of the circuit 110 in the presence of dynamic variations of load energy (e.g., via beam current variations in the tube 101), the deflection circuit is provided with facilities for maintaining the ratio of the dynamic load current variation to the variation in voltage developed by the deflection circuit at a sensibly constant value, e.g. 0.5. Such facilities include an additional secondary winding 11 of the transformer 8 designed to pick off, at the time T _O representing the start of the flyback interval, a pulse developed by the transformer 8 in response to the discharge current of the storage capacitor 6 through the now-triggered thyristor switch 1. Such discharge current is depicted as curve 121 in FIG. 2a. The amplitude of the pulse developed by the winding 11 is proportional to the voltage stored across the capacitor 6 at the time T _O ; the capacitor voltage curve is represented as curve 122 in FIG. 2a.

In accordance with the invention, such regulating facilities for the deflection circuit 110 further include a gating device 13 including a thyratron-like switch 20 (illustratively a thyristor) as its active element. The thyristor 20 is triggered, as indicated below, only during the flyback interval T _O - T ₃ . The time of occurrence T _X of the triggering instant relative to the time T _O is made proportional to the amplitude of the flyback pulse picked off the auxiliary winding 11. (The relative times of occurrence of typical pulses for triggering the thyristor 20 and the flyback-initiating thyristor 1, respectively, are depicted in FIGS. 2b and 2c.)

The triggering of the thyristor 20 serves to couple the operating voltage V _B to the inductance 12 for the remaining portion (T _X - T ₃ ) of the flyback interval. During the portion T _X - T ₃ , the energy supplied to the deflection circuit is determined in accordance with the following current developed in the inductor 12: ##EQU1## where i _LS is the inductor current (represented as curve 123 in FIG. 2a), and K is a constant related to the inductance of the inductor 12.

As indicated further in FIG. 2a, the energy increment represented by the inductor current i _LS is transferred to the storage capacitance 6 during the portion T ₃ -T _Y of the next succeeding forward sweep interval. Such transferred energy is represented by a corresponding increment of the voltage stored across the capacitor 6. Such voltage increment, in turn, leads to a corresponding increment in the amplitude of the flyback pulse picked off winding 11 at the start of the next succeeding flyback interval.

As a result of this sequence of operations, an instantaneous increase in the power extracted by the load on the deflection circuit (represented e.g. by an increase in beam current of the tube 101) will initially cause a drop in the power consumed by the deflection circuit itself and thereby a corresponding reduction in the voltage stored in the capacitor 6. The instant regulating arrangement for the deflection circuit responds to the resulting reduction in the amplitude of the next flyback pulse developed by the winding 11 to advance the time T _X of triggering of the thyristor 20 so that the latter conducts for a longer portion of the flyback interval. This causes an increase in the current through the inductance 12 at the time T ₃ . The increase in energy applied to the deflection circuit represented by such increased current is thereafter directly transferred, during the next interval T ₃ - T _Y , to the storage capacitor 6 as an increase in its stored voltage. Thus, the assumed increase in load current is effective to cause a corresponding voltage increase in the deflection circuit so that the ratio of the current change to the voltage change remains sensibly constant.

It will be appreciated that a corresponding effect will be obtained when the energy balance between the deflection circuit and the load is disturbed for any other reason, e.g. a decrease in load energy consumption (leading to a retardation of the triggering time T _X ), or a change in the level of the operating voltage V _B . In all such cases, the energy unbalance is exactly compensated without the lossy recycling of energy typical of the priorly proposed arrangements.

One suitable arrangement for converting the amplitude of the flyback pulses from the winding 11 into suitably timed control pulses for the thyristor 20 is indicated at 130 in FIG. 1. The horizontal oscillator 14 triggers, in synchronism with the ignition generator 15, a sawtooth generator 16 which provides a first linearly increasing voltage [represented by curve 131 in FIG. 2d ]. The output of the sawtooth genertor 16 is coupled to a first input of a comparator 17. A second voltage [represented by a level 132 in FIG. 2d], proportional to the amplitude of the preceding flyback pulse on winding 11, is coupled via a gain-controlled amplifier 18 to a second input of the comparator 17. As represented in FIG. 2d, the comparator exhibits an output pulse 133 at the instant T _X that the first sawtooth voltage reaches the level of the second voltage. Such output pulse is suitably shaped if desired in a pulse shaper 19 and is then applied to the control electrode of the thyristor 20.

In order to assure triggering of the thyristor 20 via the comparator 17 during the flyback interval T ₀ - T ₃ , the controlled amplifier 18 is provided with facilities for limiting the maximum value of the second voltage to a value corresponding to the voltage level of the first sawtooth wave from the generator 16 at the time T ₃ .

The converting arrangement 130, together with facilities for triggering the other thyristors 1 and 3 in the deflection circuit 110, are shown in more detail in FIG. 3. The gain-controlled amplifier 18 includes an input switching transistor 33 and an output coupling transistor 36; the latter is effective when conductive to effect the application of the second voltage to the second input of the comparator 17. In particular, the output of the winding 11 is rectified via diodes 29 and 39 and individually stored across capacitors 30 and 38, the latter via capacitor 40 and resistor 41. The output of the capacitor 38 is coupled via resistor 37 to the base of the transistor 36. The output of a voltage divider 31, whose input is coupled across capacitor 30, is coupled via Zener diode 32 to the base of switching transistor 33. The conduction condition of the transistor 33 is established, e.g. when the voltage across the capacitor 30 is in a prescribed range relative to the stabilized voltage level of the Zener diode.

The output of transistor 33 is coupled via resistor 35 to the base of transistor 36, which is triggered on when the switching transistor 33 is conductive to establish the second voltage level across a capacitor 43. By suitably selecting the voltage level at the voltage divider 31, such second voltage may be maintained within the range necessary to assure triggering of the thyristor 20 within the flyback interval as indicated before.

The comparator 17 is embodied as a phase shifter having a transistor 44. A capacitor 45 is connected from the base of transistor 44 to ground to establish the first comparator input. The above-mentioned capacitor 43, which stores the second voltage level, is connected from the emitter of transistor 44 to ground. The output of the sawtooth generator 16, whose active element is a transistor 49 and which otherwise is provided with conventional R-C and flyback circuitry, is coupled across the capacitor 45. The output of the transistor 44 is coupled to the control electrode of the thyristor 20 via pulse shaper 19 (including stages 48, 57 and 60) and a transformer 21.

Excitation of the sawtooth generator 16, the comparator circuit 17, the pulse shaper 19 and the gain-controlled amplifier 18 is provided over two separate paths: The winding 103 of the transformer 8 via a diode 63, and the portion of the rectified flyback pulse voltage developed across capacitor 38. This latter voltage serves, when present, to diasble the diode 63, so that only one of the two excitation paths is active at any given time.

In the foregoing, the invention has been described in connection with a preferred arrangement thereof. Many variations and modifications will now occur to those skilled in the art. It is accordingly desired that the scope of the appended claims not be limited to the specific disclosure herein contained.

LOEWE F875 SENSOTRONIC CHASSIS 543/490: Electron beam deflection circuit including thyristors Further Discussion and deepening of knowledge, Thyristor horizontal output circuits General concept :

1. An electron beam deflection circuit for a cathode ray tube with electromagnetic deflection by means of a sawtooth current waveform having a trace portion and a retrace portion, said circuit comprising: a deflection winding; a first source of electrical energy formed by a first capacitor; first controllable switching means comprising a parallel combination of a first thyristor and a first diode connected together to conduct in opposite directions, for connecting said winding to said first source during said trace portion, while said first switching means is turned on; a second source of electrical energy including a first inductive energy storage means coupled to a voltage supply; reactive circuit means including a combination of inductive and capacitive reactances for storing the energy supplied by said second source; second controllable switching means, substantially similar to said first one, for completing a circuit including said reactive circuit means and said first switching means, when turned on before the end of said trace portion, so as to pass through said first switching means an oscillatory current in opposite direction to that which passes through said first thyristor from said first source and to turn said first thyristor off after these two currents cancel out, the oscillatory current flowing thereafter through said first diode for an interval termed the circuit turn-off time, which has to be greater than the turn-off time of said first thyristor; wherein the improvement comprises: means for drawing, during at least a part of said trace portion, a substantial amount of additional current through said first switching means, in the direction of conduction of said first diode, whereby to perceptibly shift the waveform of the current flowing through said first switching means towards the negative values by an amount equal to that of said substantial additional current and to lengthen, in proportion thereto, said circuit turn-off time, without altering the values of the reactances in the reactive circuit which intervene in the determination of both the circuit turn-off and retrace portion time intervals.

2. A deflection circuit as claimed in claim 1, wherein said amount of additional current is greater than or equal to 5 per cent of the peak-to-peak value of the current flowing through the deflection winding.

3. A deflection circuit as claimed in claim 1, wherein said means for drawing a substantial amount of additional current through said first switching means comprises a resistor connected in parallel to said first capacitor.

4. A deflection circuit as claimed in claim 1, wherein said means for drawing an additional current is formed by connecting said first and second energy sources in series so that the current charging said reactive circuit means forms the said additional current.

5. A deflection circuit as claimed in claim 1, further including a series combination of an autotransformer winding and a second high-value capacitor, said combination being connected in parallel to said first switching means, wherein said autotransformer comprises an intermediate tap located between its terminals respectively connected to said first switching means and to said second capacitor, said tap delivering, during said trace portion, a suitable DC supply voltage lower than the voltage across said second capacitor; and wherein said means for drawing a substantial amount of additional current comprises a load to be fed by said supply voltage and having one terminal connected to ground; and further controllable switching means controlled to conduct during at least part of said trace portion and to remain cut off during said retrace portion, said further switching means being connected between said tap and the other terminal of said load.

Description:

The present invention relates to electron beam deflection circuits including thyristors, such as silicon controlled rectifiers and relates, in particular, to horizontal deflection circuits for television receivers.

The present invention constitutes an improvement in the circuit described in U.S. Pat. No. 3,449,623 filed on Sept. 6, 1966, this circuit being described in greater detail below with reference to FIGS. 1 and 2 of the accompanying drawings. A deflection circuit of this type comprises a first thyristor switch which allows the conenction of the horizontal deflection winding to a constant voltage source during the time interval used for the transmisstion of the picture signal and for applying this signal to the grid of the cathode ray tube (this interval will be termed the "trace portion" of the scan), and a second thyristor switch which provides the forced commutation of the first one by applying to it a reverse current of equal amplitude to that which passes through it from the said voltage source and thus to initiate the retrace during the horizontal blanking interval.

A undirectional reverse blocking triode type thyristor or silicon controlled rectifier (SCR), such as that used in the aformentioned circuit, requires a certain turn-off time between the instant at which the anode current ceases and the instant at which a positive bias may be applied to it without turning it on, due to the fact that there is still a high concentration of free carriers in the vicinity of the middle junction, this concentration being reduced by a process of recombination independently from the reverse polarity applied to the thyristor. This turn-off time of the thyristor is a function of a number of parameters such as the junction temperature, the DC current level, the decay time of the direct current, the peak level of the reverse current applied, the amplitude of the reverse anode to cathode voltage, the external impedance of the gate electrode, and so on, certain of these varying considerably from one thyristor to another.

In horizontal deflection circuits for television receivers, the flyback or retrace time is limited to approximately 20 percent of the horizontal scan period, the retrace time being in the case of the CCIR standard of 625 lines, approximately 12 microseconds and, in the case of the French standard of 819 lines, approximately 9 microseconds. During this relatively short interval, the thyristor has to be rendered non-conducting and the electron beam has to be returned to the origin of the scan. The first thyristor is blocked by means of a series resonant LC circuit which is subject to a certain number of restrictions (limitations as to the component values employed) due to the fact that, inter alia, it simultaneously determines the turn-off time of the circuit which blocks the thyristor and it forms part of the series resonant circuit which is to carry out the retrace. To obtain proper operation of the deflection circuit of the aforementioned Patent, especially when used for the French standard of 819 lines per image, the values of the components used have to subject to very close tolerances (approximately 2%), which results in high costs.

The improved deflection circuit, object of the present invention, allows the lengthening of the turn-off time of the circuit for turning the scan thyristor off, without altering the values of the LC circuit, which are determined by other criteria, and without impairing the operation of the circuit.

According to the invention, there is provided an electron beam deflection circuit for a cathode ray tube with electromagentic deflection by means of a sawtooth current waveform having a trace portion and a retrace portion, said circuit comprising: a deflection winding; a first source of electrical energy formed by a first capacitor; first controllable switching means comprising a parallel combination of a first thyristor and a first diode, connected together to conduct in opposite directions, for connecting said winding to said first source during said trace portion when said first switching means is turned on; a second source of electrical energy including a first inductive energy storage means coupled to a voltage supply; reactive circuit means including a combination of inductive and capacitive reactances for storing the energy supplied by the said second source; a second controllable switching means, substantially identical with the first one, for completing a circuit including said reactive circuit means and said first switching means, when turned on, so as to pass through said first thyristor an oscillatory current in the opposite direction to that which passes through it from said first source and to turn it off after these two currents cancel out, the oscillatory current then flowing through said first diode for an interval termed the circuit turn-off time which has to be greater than the turn-off time of said first thyristor; and means for drawing duing at least a part of said trace portion a substantial amount of additional current from said first switching means in the direction of conduction of said first diode, whereby said circuit turn-off time is lengthened in proportion to the amount of said additional current, without altering the values of the reactances in the reactive circuit by shifting the waveform of the current flowing through said first switching means towards the negative by an amount equal to that of said additional current.

A further object of the invention consists in using the supplementary current in the recovery diode of the first switching means to produce a DC voltage which may be used as a power supply for the vertical deflection circuit of the television receiver, for example.

The invention will be better understood and other features and advantages thereof will become apparent from the following description and the accompanying drawings, given by way of example, and in which:

FIG. 1 is a schematic circuit diagram partially in bloc diagram form of a prior art deflection circuit according to the aforementioned Patent;

FIG. 2 shows waveforms of currents and voltages generated at various points in the circuit of FIG. 1;

FIG. 3 is a schematic diagram of a deflection circuit according to the invention which allows the principle of the improvement to be explained;

FIG. 4 is a diagram of the waveforms of the current through the first switching means 4, 5 of the circuit of FIG. 3;

FIG. 5 is a circuit diagram of another embodiment of the circuit according to the invention;

FIG. 6 is a schematic representation of the preferred embodiment of the circuit according to the invention; and

FIG. 7 shows voltage waveforms at various points of the high voltage autotransformer 21 of FIG. 6.

In all these Figures the same reference numerals refer to the same components.

FIG. 1 shows the horizontal deflection circuit described and claimed in the U.S. Pat. No. 3,449,623 mentioned above, which comprises a first source of electrical energy in the shape of a first capacitor 2 having a high capacitance C ₂ for supplying a substantially constant voltage Uc ₂ across its terminals. A first terminal of the first capacitor 2 is connected to ground, whilst its second terminal which supplies a positive voltage is connected to one of the terminals of a horizontal deflection winding shown as a first inductance 1. A first switching means 3, consisting of a first reverse blocking triode thyristor 4 (SCR) and a first recovery diode 5 in parallel, the two being interconnected to conduct current in opposite directions, is connected in parallel with the series combination formed by the deflection winding 1 and the first capacitor 2. The assembly of components 1, 2, 4 and 5 forms the final stage of the horizontal deflection circuit in a television receiver using electromagnetic delfection.

The deflection circuit also includes a drive stage for this final stage which here controls the turning off of the first thyristor 4 to produce the retrace or fly-back portion of the scan during the line-blanking intervals i.e. while the picture signal is not transmitted. This driver stage comprises a second voltage source in the shape of a DC power supply 6 which delivers a constant high voltage E. The negative terminal of the power supply 6 is connected to ground and its positive terminal to one of the terminals of a second inductance 7 of relatively high value, which draws a substantially lineraly varying current from the power supply 6 to avoid its overloading. The other terminal of the second inductance 7 is connected, on the one hand, to the junction of the deflection winding 1 and the first switching means 3 by means of a second inductance 8 and a second capacitor 9 in series and, on the other hand, to one of the terminals of a second controllable bi-directionally conducting switching means 10, similar to the first one 3, including a parallel combination of a second thyristor 11 and a second recovery diode 12 also arranged to conduct in opposite directions.

The respective values of the third inductance 8 (L ₈ ) and of the second capacitor 9 (C ₉ ) are principally selected so that, on the one hand, one half-cycle of oscillation of the first series resonant circuit L ₈ - C ₉ , (i.e. π √ L ₈ ^. C ₉ ) is longer than the turn-off time of the first thyristor 4, but still is as short as possible since this time interval determines the speed of the commutation of the thyristor 4, and, on the other hand, one half-cycle of oscillation of another series resonant circuit formed by L ₁ , L ₈ and C ₉ , i.e. π √ (L ₁ + L ₈ ) ^. C ₉ , is substantially equal to the required retrace time interval (i.e. shorter than the horizontal blanking interval).

The gate (control electrode) of the second thyristor 11 is coupled to the output of the horizontal oscillator 13 of the television receiver by means of a first pulse transformer 14 and a first pulse shaping circuit 15 so that it is fed short triggering pulses which are to turn it on.

The gate of the first thyristor 4 fed with signals of a substantially rectangular waveform which are negative during the horizontal blanking intervals, is coupled to a winding 16 by means of a second pulse shaping circuit 17, the winding 16 being magnetically coupled to the second inductance 7 to make up the secondary winding of a transformer of which the inductance 7 forms the primary winding. It will be noted here that it is also possible to couple the secondary winding 16 magnetically to a primary winding connected to a suitable output (not shown) of the horizontal oscillator 13.

The operation of a circuit of this type will be explained below with reference to FIG. 2 which shows the waveforms at various points in the circuit of FIG. 1 during approximately one line period.

FIG. 2 is not to scale since one line period (t ₇ - t ₀ ) is equal to 64 microseconds in the case of 625 lines and 49 microseconds in the case of 819 lines, while the durations of the respective horizontal blanking intervals are approximately 12 and 9.5 microseconds.

Waveform A shows the form of the current i _L1 passing through deflection winding 1, this current having a sawtooth waveform substantially linear from t ₀ to t ₃ and from t ₅ to t ₇ , and crossing zero at time instants t ₀ and t ₇ , and reaching values of + I _1m and - I _1m , at time instants t ₃ and t ₅ respectively, these being its maximum positive and negative amplitudes.

During the second half of the trace portion of the horizontal deflection cycle, that is to say from t ₀ to t ₃ , the thyristor 4 of the first switching means 3 is conductive and makes the high value capacitor 2 discharge through the deflector winding 1, which has a high inductance, so that current i _L1 increases linearly.

A few microseconds (5 to 8 μ s) before the end of the trace portion, i.e. at time instant t ₁ , the trigger of the second thyristor 11 receives a short voltage pulse V _G11 which causes it to turn on as its anode is at this instant at a positive potential with respect to ground, which is due to the charging of the second capacitor 9 through inductances 7 and 8 by the voltage E from the power supply 6.

When thyristor 11 is made conductive at time t ₁ , on the one hand, inductance 7 is connected between ground and the voltage source 6 and a linearly increasing current flows through it and, on the other hand, the reactive circuit 8, 9 forms a loop through the second and first switching means 10 and 3, thus forming a resonant circuit which draws an oscillatory current i ₈ ,9 of frequency ##EQU1##

This oscillatory current i ₈ ,9 will pass through the first switching means 3, i.e. thyristor 4 and diode 5, in the opposite direction to that of current i _L1 . Since the frequency f ₁ is high, current i ₈ ,9 will increase more rapidly than i _L1 and will reach the same level at time t ₂ , that is to say i ₈ ,9 (t ₂ ) = -i _L1 (t ₂ ) and these currents will cancel out in the thyristor 4 in accordance with the well known principle of forced commutation. After time instant t ₂ , current i ₈ ,9 continues to increase more rapidly than i _L1 , but the difference between them (i ₈ ,9 - i _L1 ) passes the diode 5 (see wave form B) until it becomes zero at time instant t ₃ which is the turn off time instant of the first switching means 3, at which the retrace begins.

The interval between the time instant t ₂ and t ₃ , i.e. (t ₃ -t ₂ ), during which diode 5 is conductive and the thyristor is reverse biased will be termed in what follows the circuit turn-off time and it should be greater than the turn-off time of the thyristor 4 itself since the latter will subsequently become foward biased (i.e. from t ₃ to t ₅ ) by the retrace or flyback pulse (see waveform E) which should not trigger it.

At time instant t ₃ , the switching means 3 is opened (i ₄ and i ₅ are both zero -- see waveforms B and C) and the reactive circuit 8, 9 forms a loop through capacitor 2 and the deflection coil 1 and thus a series resonant circuit including (L ₁ + L ₈ ) and C ₉ , C ₂ being of high value and representing a short circuit for the flyback frequency ##EQU2## thus obtained.

The retrace which stated at time t ₃ takes place during one half-cycle of the resonant circuit formed by reactances L ₁ , L ₈ and C ₉ , i.e. during the interval between t ₃ and t ₅ . In the middle of this interval i.e. at time instant t ₄ , both i _L1 (waveform A) and i ₈ ,9 (waveform D) pass through zero and change their sign, whereas the voltage at the terminals of the first switching means 3 (V ₃ , waveform E) passes through a maximum. Thus, from t ₄ onwards, thyristor 11 will be reverse biased and diode 12 will conduct the current from the resonant circuit 1, 8 and 9 in order to turn the second thyristor 11 off.

At time instant t ₅ , when current i _L1 has reached - I _1m and when voltage v ₃ falls to zero, diode 5 of the first switching means 3 becomes conductive and the trace portion of scan begins.

Current i ₈ ,9 nevertheless continues to flow in the resonant circuit 8, 9 through diodes 5 and 12, which causes a break to appear in waveform D at t ₅ , and a negative peak to appear in waveform D and a positive one in waveform B in the interval between t ₅ and t ₆ , these being principally due to the distributed capacities of coil 1 or to an eventual capacitor (not shown) connected in parallel to the first switching means 3.

At time instant t ₆ , diode 12 of the second switching means 10 ceases to conduct after having allowed thyristor 11 time to become turned off completely.

The level of current i ₈ ,9 at time instant t ₅ (i.e. I _c ) as well as the negative peak I _D12 in i ₈ ,9 and the positive peak I _D5 in i ₅ depend on the values of L ₈ and C ₉ in the same way as does the turn-off time of the circuit (t ₃ - t ₂ ). If, for example, L ₈ and C ₉ , are increased I _D5 increases towards zero and this could cause diode 5 to be cut off in an undesirable fashion. I _c also increases towards zero, which is liable to cause diode 12 to be blocked and thyristor 11 to trigger prematurely.

From the foregoing it can be clearly seen that the choice of values for L ₈ and C ₉ is subject to four limitations which prevent the values from being increased to lengthen the turn-off time of the driver circuit of first switching thyristor 4 so as to forestall its spurious triggering.

Waveform F shows the voltage v _G4 obtained at the gate of thyristor 4 from the secondary winding 16 coupled to the inductor 7. This voltage is positive from t ₀ to t ₁ and from t ₆ to t ₇ and is negative between t ₂ and t ₆ i.e. while the second switching means 10 is conducting.

The present invention makes the lengthening of the turn-off time of thyristor 4 possible without altering the parameters of the circuit such as inductance 8 and capacitor 9.

In the circuit shown in FIG. 3, which illustrates the principle of the present invention, means are added to the circuit in FIG. 1 which enable the turn-off time to be lengthened by connecting a load to diode 5 so as to increase the current which flows through it during the time that it is conductive. These means are here formed by a resistor 18 connected in parallel with a capacitor 20 (which replaces capacitor 2) which is of a higher capacitance so that, in practice, it holds its charge during at least one half of the line period. FIG. 4, which shows the waveform of the current in the first switching means 3 for a circuit as shown in FIG. 3, makes it possible to explain how this lenthening of the turn-off time is achieved.

In FIG. 4, the broken lines show the waveform of the current in the first switch device 3 in the circuit of FIG. 1, this waveform being produced by adding waveforms B and C of FIG. 2. The current i ₄ above the axis flows through thyristor 4 and current i ₅ below the axis flows through diode 5. When the capacitance C ₂₀ of the capacitor in series with the deflector coil is increased to some tens of microfarads (C ₂ having been of the order of 1 μ F) and when there is connected in parallel with capacitor 20 a resistor 18 the value of which is calculated to draw a strong current I _R18 from capacitor 20, that is to say a current at least equal to 0,1 I _m (I _m being of the order of some tens of amperes), current I _R18 is added to that i ₅ which flows through diode 5 without in any way altering the linearity of the trace portion nor the oscillatory commutation of thyristor 4 which is brought about by the resonant circuit L ₈ , C ₉ .

The fact of loading capacitor C ₂₀ by means of a resistor 18 thus has the effect of permanently displacing the waveform of the current in the negative direction by I _R18 . Thus, during the trace portion of the scan, the transfer of the current from the diode 5 to the thyristor 4 begins at time t ₁₀ instead of t ₀ , that is to say with a delay proportional to I _R18 . The effect of the triggering pulse delivered by the horizontal oscillator (13 FIG. 1) to the second thyristor 11 at time instant t ₁ , will be to start the commutation process of the first thyristor 4 when the current it draws is less by I _R18 than that i ₄ (t ₁ ) which it would have been drawing had there been no resistor 18. Because of this, the turn-off time of the thyristor 4 proper, which as has been mentioned increases with the maximum current level passing throught it, is slightly reduced. Moreover, because the oscillatory current i ₈ ,9 (FIG. 2) from circuit L ₈ , C ₉ which flows through thyristor 4 in the opposite direction is unchanged, it reaches a value equal to that of the current i _L1 (FIG. 1) flowing in the coil 1 in a shorter time, that is to say at time t ₁₂ . Diode 5 will thus take the oscillatory current i ₈ ,9 (FIG. 2) over in advance with respect ro time instant t ₂ and will conduct it until it reaches zero value at a time instant t ₁₃ later than t ₃ , the amounts of advance (t ₂ - t ₁₂ ) and delay (t ₁₃ - t ₃ ) being practically equal.

It can thus be seen in FIG. 4 that the circuit turn-off time T _R of a circuit according to the invention and illustrated by FIG. 3 is distinctly longer than that T _r of the circuit in FIG. 1. This increase in the turn-off time (T _R - T _r ) depends on the current I _R18 and increases therewith.

It should be noted at this point that the current I _R18 produces a voltage drop at the terminals of the resistor the only effect of which is to heat up the resistor since the level of this voltage (40 to 60 volts) does not necessarily have a suitable value to be used as a voltage supply for other circuits in an existing transistorised television receiver.

In accordance with one embodiment of the invention, illustrated in FIG. 5, an application is proposed for the additional current which is to be drawn through diode 5. In FIG. 5, the positive terminal of capacitor 20 is connected by a conductor 19 to the negative pole of the power supply 6 and the voltage at the terminals of capacitor 20 is thus added to that E from the source 6.

In the preferred embodiment of the present invention, which is shown in FIG. 6, it is possible to cause a supplementary current of a desired value to flow through the first diode 5 while obtaining a voltage which has a suitable value for use in another circuit in the television receiver.

If the voltage at the terminals of capacitor 20 in FIG. 3 is not a usable value, it is possible to connect in parallel with the series circuit comprising the deflector coil 1 and the capacitor 2 in FIG. 1, i.e. in parallel with the terminals of the first switching means 3, a series combination of an autotransformer 21 and a high value capacitor 22 (comparable with capacitor 20 in FIGS. 3 and 5). The autotransformer 21 has a tap 23 is suitably positioned between the terminal connected to capacitor 22 at the tap 24 connected to the first switching means 3. This autotransformer 21 may be formed by the one conventionally used for supplying a very high voltage to the cathode ray tube, as described for example in U.S. Pat. No. 3,452,244; such a transformer comprises a voltage step-up winding between taps 24 and 25, which latter is connected to a high voltage rectifier (not shown).

The waveform of the voltage at the various points in the autotransformer is shown in FIG. 7, in which waveform A shows the voltage at the terminals of capacitor 22, waveform B the voltage at tap 24 and waveform C the voltage at tap 23 of the autotransformer 21.

The voltage V _c22 at the terminals of capacitor 22 varies slightly about a mean value V _cm . It is increasing while diode 5 is conducting and decreasing during the conduction of the thyristor 4.

The voltage v ₂₄ at tap 24 follows substantially the same curve as waveform E in FIG. 2, that is to say that during the retrace time interval from t ₁₃ to t ₅ to a positive pulse called the flyback pulse is produced and, during the time interval while the first switching means 3 is conducting, the voltage is zero. The mean valve of the voltage v ₂₄ at tap 24 of the auto-transformer 21 is equal to the mean value V _cm of the voltage at the terminals of capacitors 2 and 22.

Thus, there is obtained at tap 23 a waveform which is made up, during the retrace portion, of a positive pulse whose maximum amplitude is less than that of v ₂₄ at tap 24 and, during the trace portion, of a substantially constant positive voltage, the level V of which is less than the mean value V _cm of the voltage v _c22 at the terminals of capacitor 22. By moving tap 23 towards terminals 24 the amplitude of the pulse during fly-back increases while voltage V falls and conversely by moving tap 23 towards capacitor 22 voltage V increases and the amplitude of the pulse drops.

In more exact terms, the voltage V at tap 23 is such that the means value of v ₂₃ is equal to V _cm . It has thus been shown that by choosing carefully the position of tape 23, a voltage V may be obtained during the trace portion of the scan, which may be of any value between V _cm and zero.

This voltage V is thus obtained by periodically controlled rectification during the trace portion of the scan. For this purpose an electronic switch is used to periodically connect the tap 23 of trnasformer winding 21 to a load. This switch is made up of a power transistor 26 whose collector is connected to tap 23 and the emitter to a parallel combination formed by a high value filtering capacitor 27 and the load which it is desired to supply, which is represented by a resistor 28. The base of the transistor 26 receives a control voltage to block it during retrace and to unblock it during the whole or part of the trace period. A control voltage of this type may be obtained from a second winding 29 magnetically coupled to the inductance 7 of the deflection circuit and it may be transmitted to the base of transistor 26 by means of a coupling capacitor 30 and a resistor 31 connected between the base and the emitter of transistor 26.

It may easily be seen that the DC collector/emitter current in transistor 26 flows through the first diode 5 of the first switching means 3 via a resistor 28 and the part of the winding of auto-transformer 21 located between taps 23 and 24.

Experience has shown that a circuit as shown in FIG. 6 can supply 24 volts with a current of 2 amperes to the vertical deflection circuit of the same television set, the voltage at the terminals of capacitor 22 being from 50 to 60 volts.

It should be mentioned that, when the circuit which forms the load of the controlled rectifier 26, 27 does not draw enough current to sufficiently lengthen the circuit turn-off time T _R , an additional resistor (not shown) may be connected between the emitter of transistor 26 and ground or in parallel to capacitor 22, which resistor will draw the additional current required.

INTEGRAL THYRISTOR-RECTIFIER DEVICEA semiconductor switching device comprising a silicon controlled rectifier (SCR) and a diode rectifier integrally connected in parallel with the SCR in a single semiconductor body. The device is of the NPNP or PNPN type, having gate, cathode, and anode electrodes. A portion of each intermediate N and P region makes ohmic contact to the respective anode or cathode electrode of the SCR. In addition, each intermediate region includes a highly conductive edge portion. These portions are spaced from the adjacent external regions by relatively low conductive portions, and limit the conduction of the diode rectifier to the periphery of the device. A profile of gold recombination centers further electrically isolates the central SCR portion from the peripheral diode portion.

That class of thyristors known as controlled rectifiers are semiconductor switches having four semiconducting regions of alternate conductivity and which employ anode, cathode, and gate electrodes. These devices are usually fabricated from silicon. In its normal state, the silicon controlled rectifier (SCR) is non-conductive until an appropriate voltage or current pulse is applied to the gate electrode, at which point current flows from the anode to the cathode and delivers power to a load circuit. If the SCR is reverse biased, it is non-conductive, and cannot be turned on by a gating signal. Once conduction starts, the gate loses control and current flows from the anode to the cathode until it drops below a certain value (called the holding current), at which point the SCR turns off and the gate electrode regains control. The SCR is thus a solid state device capable of performing the circuit function of a thyratron tube in many electronic applications. In some of these applications, such as in automobile ignition systems and horizontal deflection circuits in television receivers, it is necessary to connect a separate rectifier diode in parallel with the SCR. See, for example, W. Dietz, U. S. Pat. Nos. 3,452,244 and 3,449,623. In these applications, the anode of the rectifier diode is connected to the cathode of the SCR, and the cathode of the rectifier is connected to the SCR anode. Thus, the rectifier diode will be forward biased and current will flow through it when the SCR is reverse biased; i.e., when the SCR cathode is positive with respect to its anode. For reasons of economy and ease of handling, it would be preferable if the circuit function of the SCR and the associated diode rectifier could be combined in a single device, so that instead of requiring two devices and five electrical connections, one device and three electrical connections are all that would be necessary. In fact, because of the semiconductor profile employed, many SCR's of the shorted emitter variety inherently function as a diode rectifier when reverse biased. However, the diode rectifier function of such devices is not isolated from the controlled rectifier portion, thus preventing a rapid transition from one function to the other. Therefore, it would be desirable to physically and electrically isolate the diode rectifier portion from that portion of the device which functions as an SCR.





Gating circuit for television SCR deflection system AND REGULATION / stabilization of horizontal deflection NETWORK CIRCUIT with Transductor reactor / Reverse thyristor energy recovery circuit.GRUNDIG SUPER COLOR 6236/30 SERIE 30F22 CHASSIS 29301-114.68(07)
In a television deflection system employing a first SCR for coupling a deflection winding across a source of energy during a trace interval of each deflection cycle and a second SCR for replenishing energy to the source of energy during a commutation interval of each deflection cycle, a gating circuit for triggering the first SCR. The gating circuit employs a voltage divider coupled in parallel with the second SCR which develops gating signals proportional to the voltage across the second SCR.


1. In a television deflection system in which a first switching means couples a deflection winding across a source of energy during a trace interval of each deflection cycle and a second switching means replenishes energy to said source of energy during a commutation interval of each deflection cycle, a gating circuit for said first switching means, comprising:
capacitive voltage divider means coupled in parallel with said second switching means for developing gating signals proportional to the voltage across said second switching means; and
means for coupling said voltage divider means to said first switching means to provide for conduction of said first switching means in response to said gating signals.
2. A gating circuit according to claim 1 wherein said voltage divider includes first and second capacitors coupled in series and providing said gating signals at the common terminal of said capacitors. 3. A gating circuit according to claim 2 wherein said first and second capacitors are proportional in value to provide for the desired magnitude of gating signals. 4. A gating circuit according to claim 3 wherein said means for coupling said voltage divider means to said first switching means includes an inductor. 5. A gating circuit according to claim 4 wherein said inductor and said first and second capacitors comprise a resonant circuit having a resonant frequency chosen to shape said gating signal to improve switching of said first switching means.

Description:

BACKGROUND OF THE INVENTION
This invention relates to a gating circuit for controlling a switching device employed in a deflection circuit of a television receiver.

Various deflection system designs have been utilized in television receivers. One design employing two bidirectional conducting switches and utilizing SCR's (thyristors) as part of the switches is disclosed in U.S. Pat. No. 3,452,244. In this type deflection system, a first SCR is

employed for coupling a deflection winding across a source of energy during a trace interval of each deflection cycle, and a second SCR is employed for replenishing energy during a commutation interval of each deflection cycle. The first SCR is commonly provided with gating voltage by means of a separate winding or tap of an input reactor coupling a source of B+ to the second SCR.

Various regulator system designs have been utilized in conjunction with the afore described deflection system to provide for uniform high voltage production as well as uniform picture width with varying line voltage and kinescope beam current conditions.
One type regulator system design alters the amount of energy stored in a commutating capacitor coupled between the first and second SCR's during the commutating interval. A regulator design of this type may employ a regulating SCR and diode for coupling the input reactor to the source of B+. With this type regulator a notch, the width of which depends upon the regulation requirements, is created in the current supplied through the reactor and which notch shows up in the voltage waveform developed on the separate winding or tap of the input reactor which provides the gating voltage for the first SCR. The presence of the notch, even though de-emphasized by a waveshaping circuit coupling the gating voltage to the first SCR, causes erratic control of the first SCR.
SUMMARY OF THE INVENTION
In accordance with one embodiment of the invention, a gating circuit of a television deflection system employing a first switching means for coupling a deflection winding across a source of energy during a trace interval of each deflection cycle and a second switching means for replenishing energy to said source of energy during a commutation interval of each deflection cycle includes a voltage divider means coupled in parallel with the second switching means for developing gating signals proportional to the voltage across the second switching means. The voltage divider means are coupled to the first switching means to provide for conduction of the first switching means in response to the gating signals.
A more detailed description of a preferred embodiment of the invention is given in the following description and accompanying drawing of which:
FIG. 1 is a schematic diagram, partially in block form, of a prior art SCR deflection system;
FIG. 2 is a schematic diagram, partially in block form, of an SCR deflection system of the type shown in FIG. 1 including a gating circuit embodying the invention;
FIG. 3 is a schematic diagram, partially in block form, of one type of a regulator system which employs an SCR as a control device and which is suitable for use with the SCR deflection system of FIG.2;
FIG. 4 is a schematic diagram, partially in block form, of another type of a regulator system suitable for use with the deflection circuit of FIG. 2; and
FIG. 5 is a schematic diagram, partially in block form, of still another type of a regulator system suitable for use with the SCR deflection system of FIG. 2.
DESCRIPTION OF THE PREFERRED EMBODIMENT
FIG. 1 is a schematic diagram, partially in block form, of a prior art deflection system of the retrace driven type similar to that disclosed in U.S. Pat. No. 3,452,244. This system includes a commutating switch 12, comprising a silicon controlled rectifier (SCR) 14 and an oppositely poled damper diode 16. The commutating switch 12 is coupled between a winding 18a of an input choke 18 and ground. The other terminal of winding 18a is coupled to a source of direct current voltage (B+) by means of a regulator network 20 which controls the energy stored in the deflection circuit 10 when the commutating switch is off, during an interval T3 to T0' as shown in curve 21 which is a plot of the voltage level at the anode of SCR 14 during the deflection cycle. A damping network comprising a series combination of a resistor 22 and a capacitor 23 is coupled in parallel with commutating switch 12 and serves to reduce any ringing effects produced by the switching of commutating switch 12. Commutating switch 12 is coupled through a commutating coil 24, a commutating capacitor 25 and a trace switch 26 to ground. Trace switch 26 comprises an SCR 28 and an oppositely poled damper diode 30. An auxiliary capacitor 32 is coupled between the junction of coil 24 and capacitor 25 and ground. A series combination of a horizontal deflection winding 34 and an S-shaping capacitor 36 are coupled in parallel with trace switch 26. Also, a series combination of a primary winding 38a of a horizontal output transformer 38 and a DC blocking capacitor 40 are coupled in parallel with trace switch 26.
A secondary of high voltage winding 38b of transformer 38 produces relatively large amplitude flyback pulses during the retrace interval of each deflection cycle. This interval exists between T1 and T2 of curve 41 which is a plot of the current through windings 34 and 38a during the deflection cycle. These flyback pulses are applied to a high voltage multiplier (not shown) or other suitable means for producing direct current high voltage for use as the ultor voltage of a kinescope (not shown).
An auxiliary winding 38c of transformer 38 is coupled to a high voltage sensing and control circuit 42 which transforms the level of flyback pulses into a pulse width modulated signal. The control circuit 42 is coupled to the regulator network 20.
A horizontal oscillator 44 is coupled to the gate electrode of commutating SCR 14 and produces a pulse during each deflection cycle slightly before the end of the trace interval at T0 of curve 21 to turn on SCR 14 to initiate the commutating interval. The commutating interval occurs between T0 and T3 of curve 21. A resonant waveshaping network 46 comprising a series combination of a capacitor 48 and an inductor 50 coupled between a winding 18b of input choke 18 and the gate electrode of trace SCR 28 and a damping resistor 52 coupled between the junction of capacitor 48 and inductor 50 and ground shapes the signal developed at winding 18b (i.e. voltage waveform 53) to form a gating signal voltage waveform 55 to enable SCR 28 for conduction during the second half of the trace interval occurring between T2 and T1' of curve 41.
The regulator network 20, when of a type to be described in conjunction with FIG. 3, operates in such a manner that current through winding 18a of input choke 18 during an interval between T4 and T5 (region A) of curves 21, 53 and 55 is interrupted for a period of time the duration of which is determined by the signal produced by the high voltage sensing and control circuit 42. During the interruption of current through winding 18a a zero voltage level is developed by winding 18b as shown in interval T4 to T5 of curve 53. The resonant waveshaping circuit 46 produces the shaped waveform 55 which undesirably retains a slump in region A corresponding to the notch A of waveform 53. The slump in waveform 55 applied to SCR 28 occurs in a region where the anode of SCR 28 becomes positive and where SCR 28 must be switched on to maintain a uniform production of the current waveshape in the horizontal deflection winding 34 as shown in curve 41. The less positive amplitude current occurring at region A of waveform 55 may result in insufficient gating current for SCR 28 and may cause erratic performance resulting in an unsatisfactory raster.
FIG. 2 is a schematic diagram, partially in block form, of a deflection system 60 embodying the invention. Those elements which perform the same function in FIG. 2 as in FIG. 1 are labeled with the same reference numerals. FIG. 2 differs from FIG. 1 essentially in that the signal to enable SCR 28 derived from sampling a portion of the voltage across commutating switch 12 rather than a voltage developed by winding 18b which is a function of the voltage across winding 18a of input choke 18 as in FIG. 1. This change eliminates the slump in the enabling signal during the interval T4 to T5 as shown in curve 64 since the voltage across the commutating switch 12 is not adversely effected by the regulator network 20 operation.
A series combination of resistor 22, capacitor 23 and a capacitor 62 is coupled in parallel with commutating switch 12, one terminal of capacitor 62 being coupled to ground. The junction of capacitors 23 and 62 is coupled to the gate electrode of SCR 28 by means of the inductor 50. The resistor 52 is coupled in parallel with capacitor 62.
Capacitors 23 and 62 form a capacitance voltage divider which provides a suitable portion of the voltage across commutating switch 12 for gating SCR 28 via inductor 50. The magnitude of the voltage at the junction of capacitors 23 and 62 is typically 25 to 35 volts. It can, therefore, be seen that the ratio of values of capacitors 23 and 62 will vary depending on the B+ voltage utilized to energize the deflection system. Capacitors 23 and 62 and inductor 50 form a resonant circuit tuned in a manner which provides for peaking of the curve 64 between T4 and T5. This peaking effect further enhances gating of SCR 28 between T4 and T5.
Since the waveshape of the voltage across commutating switch 12 (curve 21) is relatively independent of the type of regulator system employed in conjunction with the deflection system, the curve 64 also is independent of the type of regulator system.
When commutating switch 12 switches off during the interval T3 to T0' curve 21, the voltage across capacitor 62 increases and the voltage at the gate electrode of SCR 28 increases as shown in curve 64. As will be noted, no slump of curve 64 occurs between T3 and T5 because there is no interruption of the voltage across commutating switch 12.



















FIG. 3 is a schematic diagram, partially in block form, of one type of a regulator system which may be used in conjunction with the invention. B+ is supplied through a regulator network 20 which comprises an SCR 66 and an oppositely poled diode 68. The diode is poled to provide for conduction of current from B+ to the horizontal deflection circuit 60 via winding 18a of input choke 18. Current flows through the diode during the period T3 to T4 of curve 21 FIG. 1 after which current tries to flow through the SCR 66 from the horizontal deflection circuit to B+ since the commutating capacitor 25 is charged to a voltage higher than B+.
The horizontal deflection circuit 60 produces a flyback pulse in winding 38a of the flyback transformer 38 which is coupled to winding 38c. The magnitude of the pulse on winding 38c determines how long the signal required to switch SCR 66 on is delayed after T4 curve 21 FIG. 1. If the flyback pulse is greater than desirable, the SCR 66 turns on sooner than if the flyback pulse is less than desirable and provides a discharge path for current in commutating capacitor 25 back to the B+ supply. In this manner a relatively constant amplitude flyback pulse is maintained.
FIG. 4 is a schematic diagram, partially in block form, of another well-known type of a regulator system which may be used in conjunction with the invention shown in FIG. 2. B+ is coupled through winding 18a of input choke 18 and through a series combination of windings 70a and 70b of a saturable reactor 70 and a parallel combination of a diode 72 and a resistor 74 to the horizontal deflection circuit 60. Diode 72 is poled to conduct current from the horizontal deflection circuit 60 to B+.
Flyback pulse variations are obtained from winding 38c of the horizontal output transformer 38 and applied to a voltage divider comprising resistors 76, 78 and 80 of the high voltage sensing and control circuit 42. A portion of the pulse produced by winding 38c is selected by the position of the wiper terminal on potentiometer 78 and coupled to the base electrode of a transistor 82 by means of a zener diode 84. The emitter electrode of transistor 82 is grounded and a DC stabilization resistor 85 is coupled in parallel with the base-emitter junction of transistor 82. When the pulse magnitude on winding 38c exceeds a level which results in forward biasing the base-emitter junction of transistor 82, current flows from B+ through a resistor 86, a winding 70c of saturable reactor 70 and transistor 82 to ground. Due to the exponential increase of current in winding 70c during the period of conduction of transistor 82, the duration of conduction of transistor 82 determines the magnitude of current flowing in winding 70c and thus the total inductance of windings 70a and 70b. The current in winding 70c is sustained during the remaining deflection period by means of a diode 88 coupled in parallel with winding 70c and poled not to conduct current from B+ to the collector electrode of transistor 82. A capacitor 90 coupled to the cathode of diode 88 provides a bypass for B+. Windings 70a and 70b are in parallel with input reactor 18a and thereby affect the total input inductance of the deflection circuit and thereby controls the transfer of energy to the deflection circuit. The dotted waveforms shown in conjunction with a curve 21' indicate variations from a nominal waveform provided at the input of horizontal deflection circuit 60 by the windings 70a and 70b.













FIG. 5 is a schematic diagram of yet another type of a regulator system which may be used in conjunction with the invention. B+ is coupled through a winding 92a and a winding 92b of a saturable reactor to the horizontal deflection circuit 60. Windings 92a and 92b are used to replace the input choke 18 shown in FIGS. 1 and 2 while also providing for a regulating function corresponding to that provided by regulating network 20.
Flyback pulse variations are obtained from winding 38c and applied to the high voltage sensing and control circuit 42 as in FIG. 4. Current flows from B+ through resistor 86, a winding 92c and transistor 82 to ground. As in FIG. 4 the duration of the conduction of transistor 82 determines the energy stored in winding 92c and thus the total inductance of windings 92a and 92b which control the amount of energy transferred to the deflection circuit during each horizontal deflection cycle. The variations in waveforms of curve 21', shown in conjunction with FIG. 4, are also provided at the input of horizontal deflection circuit 60 by windings 92a and 92b.

For various reasons including cost or performance, a manufacturer may wish to utilize a particular one of the regulators illustrated in FIGS. 3, 4 and 5. Regardless of the choice, the gating circuit according to the invention may be utilized therewith advantageously by providing improved performance and the possibility of cost savings by eliminating taps or extra windings on the wound components which heretofore normally provided a source of SCR gating waveforms.

LOEWE F875 SENSOTRONIC  CHASSIS F800  (543/490)  CONTACTLESS TOUCH SENSOR PROGRAM CHANGE KEYBOARD CIRCUIT ARRANGEMENT FOR ESTABLISHING A CONSTANT POTENTIAL OF THE CHASSIS OF AN ELECTRICAL DEVICE WITH RELATION TO GROUND :

Circuit arrangement for establishing a reference potential of a chassis of an electrical device such as a radio and/or TV receiver, such device being provided with at least one contactless touching switch operating under the AC voltage principle. The device is switched by touching a unipole touching field in a contactless manner so as to establish connection to a grounded network pole. The circuit arrangement includes in combination an electronic blocking switch and a unidirectional rectifier which separates such switch from the network during the blocking phase.


1. A circuit arrangement for establishing, at the chassis of an electrical device powered by a grounded AC supply network, a reference potential with relation to ground, said device having at least one contactless touching switch operating on the AC voltage principle, the switch being operated by touching a unipole touching field in a contactless manner, said arrangement comprising an electronic switch for selectively blocking the circuit of the device from the supply network, a half-wave rectifier including a pair of diodes individually connected in series-aiding relation between the terminals of the supply network and the terminals of the device for separating the electronic blocking switch from the supply network during a blocking phase defined by a prescribed half period of the AC cycle, and a pair of condensers individually connected in parallel with the respective diodes. 2. A circuit arrangement according to claim 1, wherein the capacitances of the two condensers are of equal magnitude.

Description:

This invention relates to a circuit arrangement for establishing a constant reference potential on the chassis of an electrical instrument such as a radio and/or a TV receiver. Such instrument includes at least one contactless touching switch operating under the AC voltage principle, whereby by touching a single pole touching field the contactless switch is operated.

In electronic devices, for example TV and radio receivers, there are used in ever increasing numbers electronic touching switches for switching and adjusting the functions of the device. In one known embodiment of this type of touching switch, which operates on a DC voltage principle, the function of the electronic device, is contactlessly switched by touching a unipole touching field, the switching being carried out by means of an alternating current voltage. When using such a unipole touching electrode, one takes advantage of the fact that the AC current circuit is generally unipolarly grounded. In order to close the circuit by touching the touching surface via the body of the operator to ground, it is necessary to provide an AC voltage on the touching field. In one special known embodiment there is employed a known bridge current rectifier for the current supply. This type of arrangement has the drawback that the chassis of the device changes its polarity relative to the grounded network pole with the network frequency. With such construction considerable difficulties appear when connecting measuring instruments to the device, such difficulties possibly eventually leading to the destruction of individual parts of the electronic device.

In order to avoid these drawbacks, the present invention provides a normal combination of a unidirectional rectifier with an electronic blocking switch that separates the chassis of the electronic device from the network during the blocking phase. In accordance with the present invention, the polarity of the chassis of the electronic device does not periodically change, because the electronic device is practically separated from the network during the blocking phase of the unidirectional rectifier by means of the electronic blocking switch.

In a further embodiment of the invention a further rectifier is connected in series with the unidirectional rectifier in the connection between the circuit and the negative pole of the chassis. Such further rectifier is preferably a diode which is switched in the transfer direction of the unidirectional rectifier. According to another feature of the invention there are provided condensers, a respective condenser being connected parallel with each of the rectifiers. Preferably the two condensers have equal capacitances. Because of the use of such condensers, which are required because of high frequency reasons, during the blocking phase there is conducted to the chassis of the electronic device an AC voltage proportional to the order of capacitances of the condensers. Thus there is placed upon the touching field in a desired manner an AC voltage, and there is thereby assured a secure functioning of the adjustment of the device when such touching occurs.

In the embodiment of the invention employing two rectifiers there is the further advantage that over a bridging over of the minus conduit of the rectifier that is connected between the network and the negative pole of the chassis connection, no injuries can be caused by a measuring instrument in the electronic device itself and in the circuit arrangement connected thereto.

In the accompanying drawing:

The sole FIGURE of the drawing is a circuit diagram of a preferred embodiment of the invention.

In the illustrated embodiment the current supply part of the device, shown at the left, is connected via connecting terminals A and B to an AC voltage source, the terminal B being grounded at 8. The current supply part consists of a unidirectional rectifier in the form of a diode 1 with its anode connected to the terminal I, the cathode of diode 1 being connected to one input terminal 9 of an electronic device 2. In the device 2 there is also arranged a sensor circuit 3, shown here mainly as a block, circuit 3 being shown as including a pnp input transistor the emitter of which is connected to an output terminal 11 of the device 2. The collector of such transistor is connected to the other output terminal 12 of the device 2. The base of the transistor is connected by a wire 13 to a unipolar touching field 4 which may be in the form of a simple metal plate instead of the pnp transistor shown, the sensor circuit itself may consist of a standard integrating circuit which controls, among other things, the periodic sequential switching during the touching time of the touching field 4. All of the circuits of the electronic device 2 are isolated in a known manner from the chassis potential. Between the network terminal B and the negative pole 10 of the chassis there is arranged in the direction opposite that of diode 1 a further diode 5, the anode of diode 5 being connected to the terminal 10, and the cathode of diode 5 being connected to the terminal B of the current supply. To provide for HF type bridging of the diodes 1 and 5 there are arranged condensers 6 and 7 respectively, which are connected in parallel with such diodes.

The invention functions by reason of the fact that in an AC network separate devices radiate electromagnetic waves which produce freely traveling fields in the body of the person who is operating and/or adjusting the device, thereby producing an alternating current through his body to ground, as indicated by the - line at the right of the circuit diagram. If now the person operating the device touches the switching field 4, then the pnp type input transistor of the sensor circuit 3, which is placed on a definite reference potential (for example 12 Volts) and is connected with the negative halfwave of the AC voltage potential, is made conductive. There is thereby released a control command in the sequential switching, for example, for switching the electronic device to the next receiving channel. It is understood that the most suitable connection is formed between ground and the touching field 4 by means of a wire. By the use of such wires it would be assured that in all cases the base of the transistor in circuit 3 is connected to ground. This would, however, not permit anyone to operate the switch without the use of an auxiliary means such as a wire. It will be assumed that the touching almost always results directly via the almost isolated human body. For this reason the AC current fields are necessary, because otherwise there cannot always be provided a ground contact. Thus this connection is established via the body resistance of the person carrying out the touching of the switch.

The positive half wave of the alternating current travels to the terminal 9 of the electronic device 2 after such current has been rectified and smoothed by the devices 1, 6. Such positive halfwave is also conducted to the sensor circuit 3. The thus formed current circuit is closed by way of the chassis of the electronic device 3, the diode 5, and the terminal B. When there is a negative halfwave of the alternating current delivered by the current supply, both diodes 1 and 5 remain closed so that the chassis of the device 2 remains separated from the network during the blocking phase. Nevertheless, by means of condensers 6 and 7 the chassis is placed in a definite network potential, which depends on the relationship of the order of magnitude of the two condensers 6 and 7. When the capacitances of such condensers are equal, there is placed upon the chassis of the device 2 the constant reference potential, and simultaneously there is present via the sensor circuit 3 the required AC voltage at the touching field 4 for adjusting the function or functions of the device 2 upon the touching of the touching field 4.

The reference character 15 indicates a terminal or point at which the potential of the chassis of the device 2 may be measured. As above explained, the diode 5 causes the potential of the chassis at 15 to be separated from the network ground when a negative AC halfwave arrives. It will be noted that the return conduit of the circuit is held at a fixed chassis potential. The input transistor of the sensor circuit 3 remains, however, locked because it is subjected to a DC current of about 12 volts. If now, by means of touching the touching field 4, the chassis potential is connected to ground, then the transistor switches through and releases a switching function.

If the connecting terminals AB of the current source are exchanged, as by changing the plug, then there is still secured the condition that the chassis of the device is separated from the network ground via the diode, in this case the diode 1. The reference potential of the chassis consequently remains constant and the changing AC fields which are superimposed on the condensers can produce in the touching human body an AC current voltage due to the fields which are radiated by the device.

A suitable sensor which may be employed for the circuit 3 herein may be a sensor known as the "SAS 560 Tastatur IS," manufactured and sold by Siemens AG.

It is to be understood that the present invention is not limited to the illustrated environment. They can also be used in electronic blocking switch including a Thyristor circuit, which in the same manner separates the electronic device during the blocking phase from the network rectifier. With such Thyristor circuit the drawbacks described in the introductory portion of the specification of known circuit arrangements are also avoided.

Although the invention is illustrated and described with reference to a plurality of preferred embodiments thereof, it is to be expressly understood that it is in no way limited to the disclosure of such a plurality of preferred embodiments, but is capable of numerous modifications within the scope of the appended claims.

LOEWE F875 SENSOTRONIC  CHASSIS F800  (543/490)  Voltage multiplier:

A voltage multiplier formed by a pair of end contacted layer capacitors. The layer capacitors may be arranged side by side or may be stacked one upon the other. In each case a series of slits are cut in the capacitor to divide the unit into a plurality of individual capacitors, each being integrally connected by an unslit web. The diodes which are part of the voltage multiplier are then arranged in substantially parallel relation to extend across the capacitor body to make electrical contact between the end faces thereof.

1. A voltage multiplier having two rows of serially connected capacitors and terminals between capacitors on each row, a diode connecting each capacitor terminal on one row with a capacitor terminal on another row, all of said diodes being connected in series to pass pulses of one polarity only, said diodes being arranged to charge each capacitor, said two rows of serially connected capacitors consisting of end contacted layer capacitors which are integrally joined to one another, said end contacted layer capacitors being split into individual capacitors by slots which extend from one end face of the capacitor through the capacitive zone thereof to a point short of the opposite end face, the individual capacitors having unslotted portions of the metal coatings which serially connect each other, all of the terminals of a row of serially connected capacitors being formed in an end face, whereby said diodes may be readily connected from the terminals on the end face of one row of capacitors to the terminals on the end face of the other row of capacitors. 2. A voltage multiplier in accordance with claim 1 wherein said two rows of capacitors are arranged in side by side relation and spaced from one another, the majority of said diodes being arranged substantially parallel to one another. 3. A voltage multiplier in accordance with claim 1 wherein the terminals of each of the individual capacitors of each capacitor row lie substantially in the same plane. 4. A voltage multiplier in accordance with claim 1 wherein the two rows of capacitors are congruent in physical design, arranged on top of each other as a single integral unit separated by an intermediate insulating layer, the diodes arranged on a single side of the integral capacitor unit and extend perpendicularly to the end contact layers and are welded to the narrow edges of said end contact layer. 5. A voltage multiplier in accordance with claim 4 wherein said end contact layers are formed by the Schoop process. 6. A voltage multiplier in accordance with claim 1 wherein the two capacitor rows are arranged in spaced side by side relation, the end contact surfaces being arranged at the top of each row and lying generally in a single plane, the contact surfaces being arranged in a direction generally perpendicular to the longitudinal direction of the foils forming the capacitors, the majority of said diodes being arranged substantially parallel to one another and having their connection wires welded to the contact surfaces of said capacitor rows. 7. A voltage multiplier in accordance with claim 1 wherein one of said capacitor rows is formed on top of the other, said rows being separated by an insulating intermediate layer which does not project beyond the end contact layers, both rows of capacitors being subject in common to the Schoop process, the diodes spanning the end contact layers and being welded to the narrow edges thereof, the diodes extending in a direction generally parallel to the direction of the foils forming the capacitors. 8. A voltage multiplier in accordance with claim 1 wherein said diodes are silicon diodes. 9. A voltage multiplier comprising first and second end contacted layer capacitors, said capacitors being stacked upon each other, a plurality of substantially parallel slits extending alternately from opposite end faces of the stacked capacitor assembly to points intermediate of but not through the assembly, thereby electrically dividing the capacitor assembly into a number of individual capacitors which are serially connected by the remaining unslit portion of the metal coatings and which form the conductive layers thereof, an intermediate insulating layer separating the first and second end contacted capacitors, and a plurality of diodes arranged substantially parallel to each other and generally in a single plane which defines one side of the capacitor assembly, the diodes extending between the end contacts of the capacitor assembly and making electrical contact with the narrow edges of those end contacts. 10. A voltage multiplier in accordance with claim 9 wherein said first and second capacitors, being stacked to form an assembly, have opposite end faces contacted in common by the Schoop process, and the diodes are caused to span the capacitor assembly to make electrical contact between said opposite end faces. 11. A voltage multiplier in accordance with claim 1 wherein the two rows of serially connected capacitors are stacked one above the other and are separated by an intermediate insulating layer, said capacitors being end contacted by the schoop process, the intermediate insulating layer extending outwardly of the schoop layer, only one row of capacitors on each side thereof having free edge zones, diodes being arranged substantially parallel to said slots and making electrical contact with opposite end contact surfaces of said rows of capacitors. 12. A voltage multiplier in accordance with claim 11 wherein the portion of the intermediate insulating layer which extends outwardly of the schoop layer is free of schoop metal.

Description:

BACKGROUND OF THE INVENTION

1. Description of the Prior Art

Voltage multipliers of the type involved in the present invention are normally used in color television receivers to produce the high voltage for the picture tube anode. The known electrical voltage multipliers of this type are normally manually assembled from individual components and are soldered together. Such an arrangement is time consuming and it is not always possible to automate the soldering process in such a method of construction.

2. Field of the Invention

The field of art to which this invention pertains is solid state voltage multipliers and in particular to such voltage multipliers which are formed of an integral arrangement of layer type capacitors end contacted by the Schoop process.

SUMMARY OF THE INVENTION

It is an important feature of the present invention to provide an improved structure for a voltage multiplier.

It is another feature of the present invention to provide a voltage multiplier using a layer type capacitor arrangement.

It is a principle object of the present invention to provide an improved solid state voltage multiplier which is subject to easy production techniques.

It is another object of the present invention to provide a voltage multiplier formed of a layer type capacitor which is manufactured in such a way as to permit a series of diodes used in the multiplier to be readily easily soldered to the capacitor terminals in a single plane.

It is an additional object of the present invention to provide a voltage multiplier as described above wherein the capacitors of the multiplier consist of two layer type capacitors, each being slit in such a way as to form two sets of series connected individual capacitors.

It is also an object of this invention to provide a voltage multiplier as described above wherein the two sets of individual capacitors are arranged in side by side spaced relation.

It is a further object of the present invention to provide a voltage multiplier as described above wherein the two sets of serially connected individual capacitors are stacked one upon the other, and the diodes used in the multiplier are caused to span the body of the capacitor assembly and make electrical contact between the end faces thereof.

These and other features, advantages and objects of the present invention will be understood in greater detail from the following description and the associated drawings wherein reference numerals are utilized as to designate the preferred embodiments.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 shows a circuit diagram of a voltage multiplier in accordance with the present invention.

FIG. 2 shows a voltage multiplier in accordance with the present invention wherein two separate capacitor networks are used and arranged in side by side spaced relation.

FIG. 3 shows an arrangement having electrical characteristics similar to FIGS. 1 and 2 but wherein the capacitor networks or sets are stacked upon one another in a spaced saving arrangement.

FIG. 4 is a diagrammatic cross sectional view of the arrangement shown in FIG. 3.

FIG. 5 shows two capacitor networks which are separated by an insulating layer which extends beyond the Schoop layers.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

The present invention relates to an electric voltage multiplier of the type used in television receivers to produce the high anode voltage for the picture tube. Such multipliers usually consist of two rows of serially connected capacitors with a plurality of diodes connecting the capacitor terminals on one row with the like terminals on the other row. The diodes are connected in series to pass a unidirectional pulse and are arranged in such a way as to charge each of the capacitors.

By virtue of the present invention, there is provided a smaller physical design for a voltage multiplier and a physical construction which is easily suited to automation processes.

This is accomplished in the present invention by providing the two rows of serially connected capacitors to be formed from end contacted layer capacitors which are integrally joined together. The metal coatings of each layer of the capacitors have free edges at opposite sides, and the capacitor layers are split by a series of slits which form individual capacitors connected electrically in series. The slits pass from one end face to a point just beyond the capacitor zone of the capacitor. In each case the individual capacitors are connected in series with the other capacitors by an inner coating which is not severed in making the slits.

Capacitor networks of the type described are known from the German text laid open to public inspection No. 1,764,861. It is not possible however to achieve a simple production technique for voltage multipliers by the use of the teachings of this patent. Only by means of the combination of features of the present invention such as using an inner series connection has it been possible to develop an arrangement where all contact surfaces lie on the same end face of the capacitor network. In this way it has been possible to achieve a physical construction which enables the taking advantage of the most automated production techniques. Furthermore the use of capacitor networks which are based upon the principle of layer capacitors provides the added advantage that it is possible to use silicon diodes which are considerably smaller than the selenium rectifiers heretofore used.

In one arrangement of the invention, the two capacitor networks are arranged next to one another and the diodes are arranged substantially parallel to one another except for one of the diodes at the end of the circuit. This enhances the possibility of using automating processes. For the same reason, it is advisable for the contact surfaces to lie in a single plane which is possible according to the present invention.

A particularly space saving embodiment of the invention is provided in which the two capacitor networks are arranged one above another in a stacked manner and are separated from one another by an insulating intermediate layer which may extend beyond the Schoop layer as shown in FIG. 5.

Also, the projecting of the insulating layer and the removal of Schoop metal from the layers can be avoided if the two capacitor networks are stacked in such a way and separated by an insulating layer which does not project beyond the Schoop layers, if in the region of one end face, all the coatings of one capacitor network has free edge zones exposed. In such an arrangement each of the coatings is displaced in relation to the adjacent coatings. In such a manner the Schoop layer only covers the coatings of one of the two capacitor networks and the coatings of the other network is safely insulated from the Schoop layer by the free edge zones. At the same time, a mechanically stable structure is formed, since the Schoop layer also secures the parts of the end faces which they do not electrically contact.

A construction which is particularly advantageous is one in which the two capacitor networks are congruent and the diodes are arranged on one side of the capacitor network and welded to the Schoop layers. In this case the diodes are arranged perpendicularly to the end contact layers.

A simple process for the production of voltage multipliers according to the present invention is such that the capacitor networks are arranged in side by side spaced relation and the contact surfaces are generally in a single plane. In this arrangement the diodes are arranged in a direction perpendicular to the longitudinal direction of the foils in the capacitor, and the diodes span the end faces of the structure. A diode arranged at the beginning or end of the unit can be positioned at an angle, while the other diodes are arranged in a generally parallel orientation.

The preferred embodiment of the invention which is particularly small designed voltage multipliers consist of stacking the two layer capacitors upon each other and separating them by an insulating layer which does not project over the ends. The slits which are then formed on the capacitor layers, divide the capacitors into individual elements, and the diodes are placed in position on the top of the arrangement in such a way that the terminals of the diodes connect the end faces of the capacitor layers.

Referring to the drawings for greater detail FIG. 1 shows a schematic of a voltage multiplier according to the present invention in which capacitors 1 form one serially connected set and capacitors 2 form a second serially connected set. The capacitor network 4 of FIG. 2 is formed from the capacitors 1 of FIG. 1. The capacitor network 5 of FIG. 2 is formed in accordance with the invention from the capacitors 2 of FIG. 1. In FIG. 1 the diodes 3 are connected from the terminals of the capacitors 1 to the terminals of capacitors 2 as shown to produce a series diode arrangement which charges each of the capacitors as is well understood in the art of voltage multipliers.

The contact surface 6 (FIG. 2) is grounded, while the contact surface 7 is connected to the pulse input. Contact surface 8 is the tap for the high voltage output and the contact surface 9 serves to contact the diodes to two of the capacitors of one capacitor network and the contact surface 22 serves to contact the last capacitor of the capacitor network 5 to two of the diodes.

As in FIG. 2, the diodes 3 are placed on the contact surfaces 7, 8, 22, and 9 and are electrically connected thereto by spot welding. Slots 10 are provided and filled with synthetic material in the course of encasing the arrangement so that they possess the requisite dielectric strength. During operation these slots are connected at least temporarily with the full voltage of a capacitor in the case of television, for example, the voltage across one of these slots may be 8.5 K.V.

In FIG. 3, an arrangement is shown where the capacitor networks 4 and 5 are congruent and are stacked one above the other. In this case, the outer flanks 11 of the capacitor networks 5 are not used. The two end faces of the capacitor networks 4 and 5 are entirely covered with Schoop layers. The diodes 3 are arranged generally in a parallel layout with respect to the slots 10. The diodes connect the opposite contact surfaces 13 and 14 of the Schoop layers.

As illustrated in FIG. 4, the contact surfaces 13 and 14 contact only the corresponding metal coatings 15 and 17 of one of the capacitor networks 4 and 5. The coating 16 of the capacitor network 4 contacts neither of the two Schoop layers 13 and 14. However, these coatings extend beyond the slot depth 19, so that after the completion of the capacitor network, electrically conductive arms remain outside the slots and are integrally joined to the blind coatings. Accordingly, the blind coatings 18 of the capacitor 5 project beyond the slot depth 20. Thus, it is only possible to contact such a capacitor network on one end side.

The diodes 3 can thus only be connected by welding their two terminals to in each case one contact surface 13 and 14 in accordance with FIG. 1. This construction also must be sealed in order to achieve the required dielectric strength.

The intermediate layer 21 does not project beyond the end faces of the capacitor network 4 and 5 and is covered by the Schoop layer. It prevents sparkovers in the region of the cut edges and breakdowns through the dielectric, since, particularly in the use of a multiple inner series connection, the dielectric does not possess a dielectric strength sufficient to support the voltage of the overall capacitor.

In FIG. 5 the two capacitor networks 4 and 5 are separated from each other by an insulating intermediate layer 21 which extends beyond the Schoop layers 13 and 14. The Schoop layers 13 and 14 are interrupted by this intermediate layer and the upper and lower part of the Schoop layers 13 and 14 can always be contacted and wired separately.

LOEWE F875 SENSOTRONIC  CHASSIS F800  (543/490)  ELECTRICAL COMPONENT PROTECTED AGAINST HIGH TENSION, PARTICULARLY FOR COLOR TELEVISION RECEIVERS AND METHOD OF ITS PRODUCTION:Voltage multipliers of the type involved normally used in color television receivers to produce the high voltage for the picture tube anode:

A grid-shaped electrical component is formed by molding a plurality of electrically interconnected capacitors and diodes, physically forming a grid structure, within a synthetic casting resin. The electrical components form the cores of the struts of the grid structure and each strut is provided with a plurality of indentations in the synthetic resin in the area of the walls of the components.

1. A grid-shaped electrical component protected against high tension comprising a plurality of molded struts extending in substantially two grid directions and in at least one grid direction oblique thereto, each of said struts formed of synthetic resin and having a large wall strength, a plurality of electrically interconnected electrical components individually cast within and forming the respective cores or said struts, the components extending in one of said grid directions disposed to lie in one plane, the components extending in the other grid direction disposed to lie in another plane parallel to said one plane, the components extending in the oblique direction disposed to lie in one of said planes, said components comprising a plurality of capacitors disposed in the struts which extend in one grid direction and a plurality of rectifiers which are disposed in said struts which extend in the other grid direction and in the oblique grid direction, said capacitors forming serially connected rows of capacitors and said rectifiers connected to the ends of said capacitors, a first group of conductors disposed at one end of said grid structure and having the individual conductors thereof connected to the junctions of said electrical components and each including a portion extending substantially parallel to the respective capacitors and a portion extending through said grid structure generally perpendicular to the first-mentioned portions, and a second group of electrical conductors at the other end of said grid structure connected to the junctions of said electrical components thereat, said grid structure further including a base having mounting shoulders formed thereon, said grid structure together with said base and said conductors forming a self-supporting structure for permitting the passage of cooling air between said struts.

Description:

BACKGROUND OF THE INVENTION

1. Field of the Invention

This invention relates to a grid-shaped electrical component having grid struts extending substantially in two main grid directions, and more particularly to grid-shaped components which are protected against high tension, particularly for color television receivers.

2. Description of the Prior Art

Heretofore, the prior art recognized a conventional expedient to provide a color television receiver with a line transformer to generate the high voltage required for the picture tube. This line transformer was advantageously utilized to provide high voltage pulses which were then rectified prior to being employed at the picture tube. Construction of the line transformer for protection against high voltages, as well as the associated rectifier arrangement, has proven difficult in situations wherein the transformer was required to deliver the full value of the required high voltage. This is particularly true in the case of color television receivers since a direct current voltage of approximately 25kv. is ordinarily required for proper operation of the picture tube.

In view of the foregoing it is therefore advisable to design the line transformer for a lower voltage and to generate a direct current voltage of the required magnitude by utilization of a multiplier cascaded with the line transformer. For example, it is significant that the line transformer delivers recoil pulses with an amplitude of 8.5kv., from which a DC voltage of 25kv. can be obtained in a multiplier cascade having 5 silenium rectifiers and four or five capacitors.

In electrical equipment technology, however, there is always the problem, as here in a multiplier cascade, to design components free from brush discharge and protected against high tension, so that neither adjacent components nor the operating personnel can be harmed. To this end, it is generally known in the art to combine such units which are exposed to high tension into a single component which can be built in or exchanged as a one unit component both during original manufacture of the apparatus and in case of maintenance or repair.

Arrangements of the type initially mentioned avoid a majority of the inconvenience experienced heretofore by components manufactured to meet the above conditions. Thus, for example, the corresponding components are frequently cast with plastic into one compact block. However, in addition to the problem of an effective heat dissipation there is the further difficulty that the wall strength of the grouting mass between the components is not sufficiently constant, or that the metallic connecting elements of the components or the wiring may appear on the surface and cause glow defects or flash-overs.

To avoid the latter drawbacks it is also an old expedient in the art to fixedly arrange the components of the unit on a base plate and to cast the entire unit in a beaker consisting of a material which is combined with the grouting mass. However, these solutions have the drawback that casting in a beaker is relatively expensive and that as a result of these steps the problem of cooling the component parts is rendered more difficult.

Although the component parts of the type initially mentioned have great advantages from the electrical point of view as well as with regard to the manufacturing cost, there is also an additional difficulty as a result of the size of the components thus produced. In many cases, particularly in the interior of electrical apparatus such as, for example, color television sets, the measurements of which one seeks to reduce through the use of integrated switching circuits, it is frequently undesirable or even quite impossible to accommodate the component parts.

Hence, it is an object of the invention to produce an electrical component protected against high tension which has the advantages of a component of a type mentioned initially, but which has smaller dimensions than heretofore known in employing the same individual components.

SUMMARY OF THE INVENTION

According to the invention, the component parts are disposed in a molded grid structure wherein certain component parts lie in one grid direction, other component parts lie in another grid direction, and still other components are disposed obliquely to the two main grid directions. Each of the individual components are molded in a strut of the grid structure and the components parts which are disposed in one grid direction and those disposed obliquely to the grid directions are located in one plane, while the component parts disposed in the other grid direction lie in a plane parallel to the first-mentioned plane. Through this arrangement with the component parts lying in different planes, one can substantially reduce the space required for the apparatus in question without adversely affecting the advantages of prior known constructions.

As shown in relatively long experiments, it is very important for an effective operation of the apparatus that the wall strength of the synthetic resin enveloping the component parts is substantially equally large throughout the entire apparatus. It has namely been shown that the important point is that the thickness of the layer of synthetic resin varies only relatively little throughout the apparatus, and in all respects the variation is not excessive, since otherwise there may be the danger that the sealing layer will crack upon cooling. The danger of a crack formation may also be responsible for the fact that with the component parts known heretofore and case in the form of a block with or without a beaker, only adverse results were obtained since the plastics suitable for the electrical components with respect to high tension do not lend themselves to casting into blocks.

A new type of apparatus which is protected against high tension in accordance with the principles of the present invention is particularly suitable for a high tension multiplier cascade in color television sets of the type mentioned initially which contains a plurality of rectifier elements and capacitors. A further development of the invention provides that the capacitors lie parallel one behind the other in an upper plane, that in the second plane lying therebeneath a portion of the rectifier elements is spaced at a distance from the other plane and lie perpendicular to the capacitors, and that in a diagonal line of each rectangle defined by two capacitors and two rectifiers there is arranged an additional rectifier in the second lower plane. For the electrical connection of the component part with the elements of a corresponding circuit, it is provided that electrical conductors are guided outwardly from adjacent ends of the rows of capacitors, the conductors first extending a distance within the synthetic casting resin substantially parallel to the capacitors at one end of the apparatus, and at the opposite end of the apparatus the electrical conductors extend from the rows of capacitors whose ends are connected with a rectifier element and guided outwardly therefrom through the envelope of synthetic resin.

BRIEF DESCRIPTION OF THE DRAWINGS

Other objects, features and advantages of the invention will become apparent from the following detailed description taken in conjunction with the accompanying drawings, in which:

FIG. 1 is a plan view, partially in section, illustrating apparatus constructed according to the invention; and

FIG. 2 is a sectional view taken along the line II--II of FIG. 1.

DESCRIPTION OF THE PREFERRED EMBODIMENT

FIG. 1 illustrates a high tension cascade constructed in accordance with the principles of the present invention to produce the accelerating voltage required for color television tubes amounting to approximately 25 kv. consisting of five silenium rectifiers 21, 31 and four capacitors 22, fifth capacitor not cast within the component part must be separately connected thereto. Therefore, this capacitor is not shown in the drawing. The reference numerals 25, 26 and 27 designate the electrical conductors which are guided outwardly through the synthetic resin 23. The electrical conductors 25 and 26 on the base side of the component part are first pulled a distance upwardly in the synthetic resin casing so that when they exit the component part (see FIG. 2) the conductors are already sufficiently separated from the base side which is generally connected with the base frame. The indentations 24 in the synthetic resin compound are shown somewhat exaggerated schematically, since the indentations generally end, at least partly, on the wall portions of the component parts which are protected against high tension. The indentations 24 are produced during the manufacture of the cascade in accordance with the invention, wherein the interconnected electrical components which form a loose grid are placed in a mold consisting of a material, such as polyethylene or polypropylene, which does not combine with the casting resin, for example, epoxy resin. Within this mold there are disposed spacing blocks which are preferably formed directly on the mold and which are arranged and dimensioned such that they assure a sufficient space between live metal parts of the electrical components and the inner wall of the mold when the loose grid is installed. Owing to the tolerances required for a simple insertion of the grid of component parts, not all component parts will rest against all spacing blocks provided for the centering thereof so that not all of the indentations 24 in the synthetic resin compound 23 reach as far as the high tension resistant wall of the component part accommodated within the corresponding strut. The fact is illustrated schematically in the drawing where some of the indentations do not reach quite as far as the component parts.

As apparent from FIG. 2, a plastic strip 28 is provided on the narrow side (base side) of the component part remote from the high tension conductor 27, which interconnects the parallel rows of capacitors and wherein recesses 29 are provided by means of which the component part can be screwed onto the apparatus or attached in a different manner.

The invention is not limited to the embodiment shown. For example, is it also possible to form separate feet on the component for attachment purposes. It is likewise possible to slide the capacitors 22 still further inwardly by way of the rectifiers whereby a lateral contacting of the connection conductors of the capacitors is advantageous so as to make the lateral expansion of the component part still smaller. It is however, essential that the component forms a punctured grid structure since in this way there is, first, comparatively uniform wall strengths assured and, second, the cooling of the component elements can be separately effected without forcing the heat to first penetrate a larger layer of synthetic resin.

When employed in television sets, particularly color television sets, component parts in accordance with the invention may be built into advantage in horizontal position in so-called knapsacks, where as a result of their grid-shaped construction, they not only do not impede the air circulation caused by heating of the other component parts of the set, but are at the same time effectively cooled as a result of the air currents flowing therethrough.

Changes and modifications may be made of the invention within the scope and spirit of the appended claims which define what is believed to be new and desired to have protected by Letters Patent.

HOW THYRISTOR LINE DEFLECTION OUTPUT SCAN STAGES WORK:

INTRODUCTION:

The massive demand for colour television receivers in Europe/Germany in the 70's  brought about an influx of sets from the continent. Many of these use the thin -neck (29mm) type of 110° shadowmask tube and the Philips 20AX CRT Tube, plus the already Delta Gun CRT . 

Scanning of these tubes is accomplished by means of a toroidally wound deflection yoke (conventional 90° and thick -neck 110° tubes operate with saddle -wound deflection coils). The inductance of a toroidal yoke is very much less than that of a saddle -wound yoke, thus higher scan currents are required. The deflection current necessary for the line scan is about 12A peak -to -peak. This could be provided by a transistor line output stage but a current step-up transformer, which is bulky and both difficult and costly to manufacture, would be required. 

An entirely different approach, pioneered by RCA in America and developed by them and by ITT (SEL) in Germany, is the thyristor line output stage. In this system the scanning current is provided via two thyristors and two switching diodes which due to their characteristics can supply the deflection yoke without a step-up transformer (a small transformer is still required to obtain the input voltage pulse for the e.h.t. tripler). The purpose of this article is to explain the basic operation of such circuits. The thyristor line output circuit offers high reliability since all switching occurs at zero current level. C.R.T. flashovers, which can produce high current surges (up to 60A), have no detrimental effects on the switching diodes or thyristors since the forward voltage drop across these devices is small and the duration of the current pulses short. If a surge limiting resistor is pro- vided in the tube's final anode circuit the peak voltages produced by flashovers seldom exceed the normal repetitive circuit voltages by more than 50-100V. This is well within the device ratings.

Brief Basics: LINE Scan output stages operate on the same basic principle whether the active device used is a valve, transistor or thyristor. As a starting point, let's remind ourselves of this principle, which was first developed by Blumlein in 1932. The idea in its simplest form is shown in Fig. 1. The scan coils, together with a parallel tuning capacitor, are connected in series with a switch across the h.t. supply. When the switch is closed - (a) - current flows through the coils, building up linearly as required to deflect the beam from the centre to the right-hand side of the screen. At this point the switch is opened. The coils and the capacitor then form a resonant circuit. The magnetic fields generated around the coils during the preceeding forward scan as current flowed through them when the switch was closed now collapse, charging the capacitor - (b). As a result of the resonant action the capacitor next discharges, driving current through the coils in the opposite direction - (c). Once more magnetic fields are generated around the coils. This resonant action lasts for one half -cycle of oscillation, during which the beam is rapidly deflected from the right- hand side to the centre and then to the left-hand side of the screen. The flyback is thus complete. If the switch is now closed again further oscillation is prevented and, as the magnetic fields around the coils collapse, a decaying current flows through them in the direction shown at (d). This decaying current flow deflects the beam from the left-hand side of the screen back towards the centre: the period during which this occurs is often referred to as the energy recovery part of the scanning cycle. When the current has decayed to zero we are back at the situation shown at (a): the current through the coils reverses, driving the beam to the right-hand side of the screen. This is a very efficient System, since most of the energy drawn from the supply is subsequently returned to it. There is negligible resistance in the circuit, so there is very little power loss.

 Basic Transistor Circuit:

 In Blumlein's day valves had to be used to perform the switching action. Two were required since a valve is a unidirectional device, and as we have seen current must flow through the switch in both directions. Nowadays we generally use a transistor to perform the switching action, arranging the circuit along the lines shown in Fig. 2. The line output transformer T is used as a load for the transistor and as a simple means of generating the e.h.t. and other supplies required by the receiver. The scan -correction capacitor Cs also serves as a d.c. block. Capacitor Ct tunes the coils during the flyback when the transistor is cut off. During the forward scan Cs first charges, then discharges, via the scan coils, thus providing deflection from the left- hand side to the right-hand side of the screen. One advantage of a transistor is that it can conduct in either direction. Thus unless we are operating the stage from an 1.t. line of around 11V - as in the case of many small -screen portables - we don't need a second switching device. With a supply of 11-12V a shunt efficiency diode - connected in parallel with the transistor, cathode to collector and anode to emitter, is required because the linearity is otherwise unacceptable. Another advantage of a transistor compared to a valve is that it is a much more efficient switch. When a transistor is saturated both its junctions are forward biased and its collector voltage is then at little more than chassis potential. The anode voltage of a saturated pentode however is measured in tens of volts, and this means that there is considerable wasteful dissipation. Thyristor Switch If what we need is an efficient switch, why not use a thyristor??? 

Thyristors are even more efficient switches than transistors. They are more rugged, can pass heavy currents, and are insensitive to the voltage overloads that can kill off transistors. In addition, in the sort of circuit we are about to look at the power supply requirements can be simplified (a line output transistor must be operated in conjunction with a stabilised power supply: this is not necessary in the thyristor circuit since regulation can be built in). In the nature of things however there must be disadvantages as well - and there are! First, a thyristor will not act as a bidirectional switch. 

There is no great problem here however: all we need do is to shunt it with a parallel efficiency diode. More awkward is the fact that once a  thyristor has been triggered on at its gate it cannot be switched off again by any further action taken in its gate circuit. In fact it's this problem of operating the thyristor switch that is responsible for the complexity of thyristor line output circuits. 

A thyristor can be switched off only by reducing the current through it below the "hold on" value, either by momentarily removing the voltage across the device or by passing an opposing current through it in the opposite direction - this latter technique is used in practical thyristor line output circuits. Once the reverse current through the thyristor is about equal to the forward current flowing through it the net current falls below the "hold on" value and the thyristor switches off.

 Basic Thyristor Circuit:

 There is more than one way of arranging a thyristor line output stage. Only one basic circuit has been used so far however, though as you'd expect there are differences in detail in the circuits used by different setmakers. The basic circuit was first devised and put into production by RCA in the USA in the late 1960s. It was subsequently popularised in Europe by ITT, and many continental setmakers have used it, mainly in colour receiver chassis fitted with 110° delta gun c.r.t.s. They include Finlux, Grundig, Saba, Siemens and ASA. Korting use it in their 55636 chassis which is fitted with a 90° PIL tube, while Grundig continue to use it in their latest sets which use the Mullard/Philips 20AX tube. 

Amongst Japanese setmakers, Sharp use it in their Model C1831H which is fitted with a Toshiba RIS tube. 

Reduced to its barest essentials, the circuit takes the form shown in Fig. 3. To start with this looks strange indeed! The right-hand side however is simply the equivalent of the scanning section of the transistor circuit shown in Fig. 2, with TH2 and D2 replacing the transistor as the bidirectional switch.  

The tuning capacitor however is returned to chassis via the left-hand side of the circuit - in consequence there is no d.c. path between the right-hand and left-hand sides of the circuit. L1 provides a load. The efficiency diode D2 conducts during the first part of the forward scan, after which TH2 is switched on to drive the beam towards the right-hand side of the screen. The purpose of the left-hand side of the circuit, the bidirectional switch TH1/D1 and L2, together with the tuning capacitor Ct, is to switch TH2 off and to provide the flyback action.

 The output from the line oscillator consists  of a brief pulse to initiate the flyback. It occurs just before the flyback time (roughly 3µS before) and is applied to the gate of TH1, switching it on. When this happens L2 is connected to chassis and current flows into it, discharging Ct (previously charged from the h.t. line). L2 is called the commutating coil, and forms a resonant circuit with Ct. Thus when TH1 is switched on a sudden pulse builds up and this is used to switch off TH2. In addition to tuning L2, Ct tunes the scan coils to provide the usual flyback action. 

Roughly speaking therefore D2 and TH2 conduct alternately during the forward scan and are cut off during the flyback, while TH1 is triggered on just before the flyback, TH1 and D I subsequently conducting alternately during the flyback and then cutting off when the efficiency diode takes over. 

 Thyristor Line Scan Practical Circuit:

 A more practical arrangement is shown in Fig. 4. A secondary winding L3 is added to Ll to provide the trigger pulse for TH2: L4, C4 and R I provide the pulse shaping required. The tuning capacitor Ct is rearranged as a T network: this is done to reduce the voltage across the individual capacitors and enable smaller values to be used, all in the interests of economy. And finally a transformer is coupled to the circuit by C5 to make use of the flyback pulse for e.h.t. generation and to provide other supplies. In many recent chassis THUD 1 and TH2/D2 are encapsu- lated together, in pairs. In practical circuits L1 and L2 generally consist of a single transformer - often a transductor is used, for convenience rather than for the transductor characteristics. This makes practical circuits look at first glance rather different to the basic form shown in Figs. 3 and 4. A further winding is often added to the transformer to provide a supply for other parts of the receiver, making the circuit look even more confusing. In addition e.h.t. regulation, pincushion distortion correction and beam limiting circuitry is required, and protection circuits may be incorporated.

Scanning Sequence:  It's time to look at the basic scanning sequence in more detail, basing the description on Figs. 3 and 4. We'll start at the beginning of the flyback. TH2 and D2 have just been switched off - we'll come to how this is done later - while  TH1 which was triggered on by a pulse from the line oscillator is still conducting. Energy is stored in the scan coils in the form of magnetic fields. As these collapse, a decaying current flows via the coils, Cs, Ct, L2 and TH 1. When this current falls to zero the charge on Ct will have reversed and TH 1 will switch off. This completes the first half of the flyback. The left-hand plate of Ct is charged negatively, while its right-hand plate carries a positive charge. D1 is now biased on and Ct discharges back into the scan coils to give the second half of the flyback. Current is flowing via D1, L2, Ct, Cs and the scan coils. At the end of this period the circuit energy will have been transferred once again to the scan coils - in the form of magnetic fields. One complete half cycle of oscillation will have occurred, returning the beam to the left-hand side of the screen. With Ct discharged, D 1 switches off. The oscillation tries to continue in the negative direction, but we then get the normal efficiency diode action, i.e. D2 conducts shorting out the tuned circuit. As the fields around the coils collapse a linearly decaying current flows via the coils, Cs and D2. This gives us the first part of the forward scan. Towards the centre of the screen TH2 is switched on by the pulse obtained from L3 and the current in the scan coils reverses to complete the scan.  

 Switching the Scan Thyristor Off:  The tricky part is when it comes to switching TH2 off. As we have seen, TH1 is triggered on about 3fitS before the end of the forward scan. Prior to this Ct will have been charged to the h.t. potential via L 1 and L2. When TH1 conducts, current flows via TH1, L2, Ct and TH2 (which is on remember). Because of the tuned circuit formed by L2 and Ct, the current builds up rapidly in the form of a pulse - the commutating pulse shown in Fig. 5. When this current, which flows through TH2 in the opposite direction to the scan current, exceeds the scan current TH2 switches off. Once TH2 cuts off D2 is able to conduct - it is no longer reverse biased - which it does for a short period to provide an earth return path for the remaining duration of the commutating pulse and also to enable the scan to be completed (Cs discharging via the scan coils). When the reverse, commutating current falls below the scan current D2 switches off and we then get the flyback action as the magnetic fields around the coils collapse.

 Power Transferring ; during the forward scan Ct is charged via L1 and L2, its right-hand plate being held at little above

 through the conduction of D2 and then TH2. During the flyback, when TH1 and D1 conduct alternately, connecting the junction L1, L2 to chassis, Ct supplies energy to the scan part of the circuit. The Practical Circuit so much then for the basic circuit and its action. Turning now to a practical circuit, Fig. 6 shows the thyristor line output stage used in the Grundig SuperColor  Models 5011 and 6011. Ty511/Di511 form the flyback switch, T1 is the input/commutating transformer, C516/7/8 comprise the tuning capacitance, Di518 is the efficiency diode and Ty518 the forward scan thyristor. The scan -correction capacitor Cs is C537. As can be seen, the line output transformer circuit is quite conventional. The main complication arises because of the need to provide width/e.h.t. stabilisation. In a valve line output stage it is a simple matter to achieve stabilisation by using a v.d.r. in a feedback circuit to alter the bias on the output pentode. We can't do this with a transistor line output stage, so we have to operate this in conjunction with a stabilised supply. There is a subtle but quite simple method of applying stabilisation to a thyristor line output stage however. As we have seen, the energy supplied to the output side of the circuit is provided by the tuning capacitors when they discharge during the flyback period. During the forward scan they charge via the input coil - or transformer as it is in practice. Now if we shunt the transformer's input winding with a transductor we can control the inductance in series with the tuning capacitors and in consequence the charging time of the capacitors and hence the power supplied to the output side of the circuit.

EHT/Width Stabilisation:

 The stabilising transductor in Fig. 6 is Td 1, whose load windings are connected in series with R504/Di504 across the input winding of T1. The transductor's control winding is driven by transistor Tr506, which senses the h.t. voltage (via R506) and the amplitude of the signal at tag d on the line output transformer. R508 in the transistor's base circuit enables the e.h.t. to be set to the correct voltage (25kV). 

Other Circuit Details: A fourth winding on Ti feeds the 1.t. rectifier and stabiliser which provide the supply for the low -power circuits in the receiver. The trigger pulse winding also feeds a stabilised 1.t. supply circuit (21V). 

EW pincushion distortion correction is applied by connecting the load windings of a second transductor (Td2) across a section of the line output transformer's primary winding. By feeding a field frequency waveform to the control winding on this transductor the line scanning is modulated at field frequency. There is a simple but effective safety circuit in this Grundig line output stage. If the voltage at tag c on the line output transformer rises above 68V zener diode Di514 conducts, triggering thyristor Ty511 into conduction with the result that the cut-out operates. C517 is returned to chassis via a damped coil (L517) so that the voltage transient when the efficiency diode cuts off is attenuated. Likewise L512/C512/R512 are added to suppress the voltage transient when the flyback thyristor Ty511 cuts off. The balancing coil L516 is included to remove unwanted voltage spikes produced by the thyristors. 

At the end........This Grundig circuit is representative of the way in which thyristor line output circuits are used in practice. There are differences in detail in the thyristor line output stages found in other setmakers' chassis, but the basic arrangement will be found to be substantially 

Servicing / Throubleshooting / Repairing Thyristor Line Scan Timebases Crt Deflections circuits:

LARGELY due to advances in colour c.r.t. scan coil design, the thyristor line output stage has become obsolete laready in the 1981's.

 It  was a very good system to use where the line scan coils require large peak currents with only a moderate flyback voltage - an intrinsic characteristic of toroidally wound deflection coils.

it was originally devised by RCA. Many sets fitted with 110°, narrow -neck delta -gun tubes used a thyristor line output stage - for example those in the Grundig and Saba ranges and the Finlux Peacock , Indesit, Siemens, Salora, Metz, Nordmende, Blaupunkt, ITT, Seleco, REX, Mivar, Emerson, Brionvega, Loewe, Galaxi, Stern, Zanussi, Wega, Philco. The circuit continued to find favour in earlier chassis designed for use with in -line gun tubes, examples being found in the Grundig and Korting ranges - also,  Indesit, Siemens, Salora, Metz, Nordmende, Blaupunkt, ITT, Seleco, REX, Mivar, Emerson, Brionvega, Loewe, Galaxi, Stern, Zanussi, Wega, Philco the Rediffusion Mk. III chassis. Deflection currents of up to 13A peak -to -peak are commonly encountered with 110° tubes, with a flyback voltage of only some 600V peak  to peak. The total energy requirement is of the order of 6mJ, which is 50 per cent higher than modern 110° tubes of the 30AX and S4 variety with their saddle -wound line scan coils.   The only problem with this type of circuit is the large amount of energy that shuttles back and forth at line frequency. This places a heavy stress on certain components. Circuit losses produce quite high temperatures, which are concentrated at certain points, in particular the commutating combi coil. This leads to deterioration of the soldered joints around the coil, a common cause of failure. This can have a cumulative effect, a high resistance joint increasing the local heating until the joint becomes well and truly dry -a classic symptom with some Grundig / Emerson sets. The wound components themselves can be a source of trouble, due to losses - particularly the combi coil and the regulating transductor. Later chassis are less prone to this sort of thing, partly because of the use of later generation, higher efficiency yokes but mainly due to more generous and better design of the wound components. The ideal dielectric for use in the tuning capacitors is polypropylene (either metalised or film). It's a truly won- derful dielectric - very stable, with very small losses, and capable of operation at high frequencies and elevated temperatures. It's also nowadays reasonably inexpensive. Unfortunately many earlier chassis of this type used polyester capacitors, and it's no surprise that they were inclined to give up. When replacing the tuning capacitors in a thyristor line output stage it's essential to use polypropylene types -a good range of axial components with values ranging from 0.001µF to 047µF is available from RS Components, enabling even non-standard values to be made up from an appropriate combination. Using polypropylene capacitors in place of polyester ones will not only ensure capacitor reliability but will also lower the stress on other components by reducing the circuit losses (and hence power consumption).

       Numerous circuit designs for completely transistorized television receivers either have been incorporated in commercially available receivers or have been described in detail in various technical publications. One of the most troublesome areas in such transistor receivers, from the point of View of reliability and economy, lies in the horizontal deflection circuits.

       As an attempt to avoid the voltage and current limitations of transistor deflection circuits, a number of circuits have been proposed utilizing the silicon controlled rectifier (SCR), a semiconductor device capable of handling substantially higher currents and voltages than transistors.

       The circuit utilizes two bi-directionally conductive switching means which serve respectively as trace and commutating switches. Particularly, each of the switching means comprises the parallel combination of a silicon controlled rectifier (SCR) and a diode. The commutating switch is triggered on shortly before the desired beginning of retrace and, in conjunction with a resonant commutating circuit having an inductor and two capacitors, serves to turn off the trace switch to initiate retrace. The commutating circuit is also arranged to turn oft the commutating SCR before the end of retrace.  

Circuit Operation:

The basic thyristor line output stage arrangement used in all these chassis is shown in Fig. 1 - it was originally devised by RCA. The part to the right of the tuning capacitance acts in exactly the same manner as a transis- tor line output stage, with the scan thyristor Th2 replacing the transistor. The thyristor is switched on about half way through the forward scan, the efficiency diode D2 provid- ing the initial part of the line scan (left-hand side of the screen). The scan coils and line output transformer (used to generate the e.h.t. plus various other supply lines and pulse waveforms as required) are a.c. coupled, via the scan -correction capacitor C5 and C6 respectively. The problem with a thyristor is that it can be turned on at its gate but not off. To switch a thyristor off, the current flowing through it must be reduced below a value known as the hold -on current. This is the main function of the components on the left-hand side - the line generator, the flyback thyristor with its parallel diode and the commutat- ing coil. During the forward scan, the tuning capacitors are charged from the h.t. line via the input and commutat- ing coils. The line generator produces a pulse to trigger the flyback thyristor Th1- this occurs just before the actual flyback. When Thl1 switches on, the junction of the  input coil and the commutating coil is momentarily con- nected to chassis. The tuning capacitance and the com- mutating coil then resonate, producing a pulse which draws current via the scan thyristor. Since this current flow is in the opposite direction to the scan current flow, the two cancel and the current flowing via the scan thyris- tor falls below the hold -on current. Th2 is thus switched off, and the scan coils resonate with the tuning capaci- tance to provide the flyback action. So much for the basic action. A secondary winding coupled to the input coil produces a pulse to switch the scan thyristor on, in conjunction with the shaping/delay network Ll, C4, R1. The tuning capacitors are usually arranged in the T formation shown to reduce the values required and the voltages developed across them. In practical circuits the input and commutating coils are usually combined in a single unit which for obvious reasons is generally known as the combi coil. The main point not so far mentioned is stabilisation. There are two approaches to this. In earlier circuits a transductor was included in parallel with the input coil to vary the impe- dance in series with the tuning capacitance. This was driven by a transistor which was in turn controlled by feedback from the line output transformer. A more efficient technique is used in later circuits, with a current dumping thyristor in series with the input coil. Practical Circuit As a typical example of the earlier type of circuit, Fig. 2 shows the thyristor line output stage used in the Grundig 5010/5011/6010/6011 series. Td1 is the regulating transductor which is driven by Tr506. Ty511 is the flyback thyristor (commutating thyristor might be a better name), Ty518 the scan thyristor, Di518 the efficiency diode and C516/7/8 the tuning capacitance. The scan coils are cou- pled via C537, while C532 provides coupling between the primary winding of the line output transformer and chas- sis. A transductor (Td2) is used for EW raster correction. The combi coil also feeds 1.t. rectifiers from its secondary windings. 

Component Problems: The only problem with this type of circuit is the large amount of energy that shuttles back and forth at line frequency. This places a heavy stress on certain components. Circuit losses produce quite high temperatures, which are concentrated at certain points, in particular the combi coil. This leads to deterioration of the soldered joints around the coil, a common cause of failure. This can have a cumulative effect, a high -resistance joint increasing the local heating until the joint becomes well and truly dry -a classic symptom with some Grundig sets. The wound components themselves can be a source of trouble, due to losses - particularly the combi coil and the regulating transductor. Later chassis are less prone to this sort of thing, partly because of the use of later generation, higher efficiency yokes but mainly due to more generous and better design of the wound components. The ideal dielectric for use in the tuning capacitors is polypropylene (either metalised or film). It's a truly won- derful dielectric - very stable, with very small losses, and capable of operation at high frequencies and elevated temperatures. It's also nowadays reasonably inexpensive. Unfortunately many earlier chassis of this type used polyester capacitors, and it's no surprise that they were inclined to give up. When replacing the tuning capacitors in a thyristor line output stage it's essential to use poly- propylene types -a good range of axial components with values ranging from 0.001µF to 047µF is available from RS Components, enabling even non-standard values to be made up from an appropriate combination. Using polypropylene capacitors in place of polyester ones will not only ensure capacitor reliability but will also lower the stress on other components by reducing the circuit losses (and hence power consumption). The thyristors are also liable to fail, as are their parallel diodes. Earlier devices were less reliable than their successors. Since all thyristor line output stages operate in the same way and under similar conditions, the use of later types of thyristors and diodes in earlier circuits is a matter of mechanical rather than electrical con- siderations. One important point should be noted: the scan thyristor is a faster device and often has a higher voltage rating than the flyback thyristor. The simplest course is to keep in stock some of the later scan thyristors that incorporate an efficiency diode - suitable types are the RCA S3900SF and the Telefunken TD3-800H. The Telefunken device is in a TO66 package (and can be obtained quite cheaply) while the RCA type is in a TO220 package. Either type can be used in the scan or flyback positions and can also be used as a replacement for the regulating thyristor used in later designs instead of a transductor. Whenever replacing a thyristor in the line output stage it's good practice to replace the parallel diode at the same time. Using one of the above recom- mended devices will do this automatically, since the thyristor and its parallel diode share the same encapsulation - always remember to remove the old diode when this is a separate device however, as some can exhibit high -voltage leakage/breakdown which is not evident from a quite check with the Avo. Apart from the wound components (including the line output transformer), the thyristors and their parallel diodes and the tuning capacitors several other com- ponents are prone to failure. These include the tripler, scan/flyback rectifier diodes used to provide various supply lines, surge limiting resistors, the scan coil coup- ling/scan correction capacitor (replace with a metalised polypropylene type) and regulator components such as the thyristor in later types and the transductor driver transistor in earlier circuits. 

Basic Fault Conditions: At one time every engineer must have scratched his head and cursed the new-fangled idea of the thyristor line output stage. That they are awkward to service is a fallacy however. The usual symptom of a fault in the line output stage is the cutout tripping. All chassis that use a thyristor line timebase incorporate a trip of some sort. The type varies from chassis to chassis. Early Grundig sets have a mechanical cutout; the Saba H chassis uses a thyristor and solenoid to open the mains on/off switch; a common arrangement consists of a thyristor in series with the h.t, line and a control transistor which shorts the thyristor's gate and cathode in the event of excessive current demand (this gives audible tripping at about 2Hz). Some sets incorporate both excess current and over -voltage trips, but most have just the former. 

There are two basic fault conditions: when the excess current trip is activated and the set goes dead, or no e.h.t. with the trip not activated. The first condition is usually due to a line timebase fault, the most common being a short-circuit flyback thyristor or its parallel diode. A straightforward resistance check will sort this out. If this is not the case, short-circuit the scan thyristor by soldering a wire link between its anode and cathode. This will prevent any drive to the scan coils and the line output transformer. If the tripping stops, the fault could be due to the tripler, the line output transformer, a rectifier diode fed from a winding on the latter or a short in a circuit supplied by a scan rectifier diode. If the trip continues to operate and the flyback thyristor/diode is not the culprit, the most likely causes are incorrect drive to this thyristor - if possible check with a scope against the waveform given in the manual - or a rectifier diode fed from the combi coil. As an example of the latter, Fig. 3 shows the arrangement used in the Finlux Peacock: the electronic trip will operate if either D503 or D504 goes short-circuit, a fairly common fault on these sets. The diodes can also go open-circuit/high resistance to give the no sound with field collapse symp- tom, but that's another story ( referring to the diodes as D603/4 ). When the set is dead, h.t. is present and the trip is not activated, suspect the following: the scan thyristor, the efficiency diode, the line output transformer, the scan - correction capacitor, or lack of drive to the scan thyristor. Dry -joints can be the cause of any of these basic fault conditions, depending on the actual circuit and where the dry -joint has occurred. 

Other Symptoms: Hairline cracks in the ferrite core of a wound com- ponent can give rise to strange symptoms since this upsets the delicate balance of the tuning arrangements. There will usually be excessive current which will probably cause the trip to operate. Alternatively the fault may be incorrect line frequency which cannot be set by the line hold control. This fault can also give rise to excessive e.h.t., which can in turn produce a chain reaction of des- truction, e.g. the tripler is a common victim as are the two line output stage thyristors. Excessive e.h.t. leading to instant destruction of these components may also be due to open -circuit line scan coils or the connections to them. A quick resistance check done on the board itself will eliminate both the coils and the leads/connectors. Excessive e.h.t. with foldover in the centre of the screen and cooking in the tube's first anode supply net- work occurs in the Grundig 5010 series when L515 in the scan thyristor's trigger circuit (see Fig. 2) goes short- circuit. The reason for this situation is that the thyristor is triggered on early. Another common fault in these sets is failure of Di504/R504 - failure of one seems to affect the other, so both should be replaced. The usual symptom is fuzzy verticals and a sawtooth effect on diagonals. The trip may operate, possibly after period of operation. These components set up the transductor's operating bias. Linearity problems are usually caused by the regulator circuit, which can also be responsible for line "hunting". In the event of lack of width in the earlier type of circuit, check for dry -joints in the regulator circuit and suspect the control transistor. Foldover on the left-hand side of the screen can be caused by an open -circuit flyback diode. Foldover at the centre of the screen with greatly reduced width is the symptom when the efficiency diode goes open -circuit - the trip may or may not operate. Unusual interference patterns on the screen, best viewed with the contrast control turned to minimum and the brightness control advanced until a distinctly visible but not over bright white raster is obtained, can be due to the tripler if there's curved patterning on the extreme left- hand side of the screen, the regulator clamp diode (Di505 in Fig. 2) if there's curved interference just to the left of centre, or the flyback thyristor drive circuit if there's a single vertical line of patterning about four fifths of the way to the right of the screen.

The aim of this article has been to provide a general guide to servicing rather than to list faults common to particular models. Much useful information on individual 
chassis with thyristor line output stages has appeared in previous issues of  Obsolete Technology Tellye !- refer to the following as required: Search with the tag Thyristors at the bottom of the post to select all posts with this argument on various fabricants.

LOEWE F875 SENSOTRONIC  CHASSIS F800  (543/490)  Television Voltage multiplier arrangement with capacitor rolls surrounded by diodes:

A voltage multiplier includes a plurality of capacitors and diodes in an integral unit. The capacitors are combined in capacitor rolls surrounded by diodes on the outside of the diodes can be positioned between two capacitor rolls. AC and dc-voltage-operated capacitors can be combined in separate or the same rolls. The ac capacitors may be located inside within the roll and the dc capacitors on the outside of the same roll. The capacitor plates are made of aluminum foil with intermediate polystyrene and polyester layers. A common capacitor electrode plate may be used for adjacent capacitors.


1. A voltage multiplier comprising a plurality of series connected diodes and a plurality of capacitors, each capacitor being connected between opposite ends of a pair of said diodes and including an intermediate thermoplastic dielectric layer and a pair of metal foil electrode layers on opposite sides of said dielectric layer, said electrode and dielectric layers being rolled into a plurality of overlapping layers including said plurality of capacitors within a common roll, said diodes being connected to said electrode layers and being disposed about opposite outer sides of said roll. 2. The device of claim 1 including means applying a.c. and d.c. voltages to different respective groups of said capacitors. 3. The device of claim 1 wherein said plurality of diodes surround the sides and one end of said roll, the other end of said roll having external connections thereto, and a common thermoplastic cover encapsulating said diodes and capacitors, said external connections extending from one end of said cover. 4. The device of claim 1 including means applying a.c. and d.c. voltages to different respective groups of said capacitors within said common roll, said a.c. voltage capacitors being within the inner layers and said d.c. voltage capacitors being within the outer layers. 5. The device of claim 1 wherein said roll has external connections to the two ends thereof. 6. The device of claim 1 wherein said thermoplastic dielectric layer includes two outer layers of polystyrene and a layer of polyester therebetween. 7. The device of claim 1 wherein one electrode layer is common to two capacitors within said roll. 8. The device of claim 6 wherein said electrode layers are of aluminum.

Description:

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a voltage multiplier arrangement with diodes and capacitors wherein at least two capacitors are combined into a unit.

2. Description of the Prior Art

Voltage multiplier arrangements serve to produce high voltages and are particularly useful for the operation of television picture tubes. The diodes and capacitors are connected in separate series paths with a capacitor in parallel with two series diodes and are generally embedded in plastic for voltage protection.

In one known arrangement, the diodes and capacitors are disposed in a lattice configuration to keep the volume to a minimum. At each long side, two capacitors are arranged one behind another in the longitudinal direction; a diode is located at each short side and in the middle therebetween, and an additional diode is disposed in each diagonal direction. Thus, the length of this arrangement is essentially determined by the length of the capacitors, while the width is determined by the length of the diodes.

In another known arrangement, in order to meet the requirement for optimum utilization of the space available and for technical simplification, the ac-voltage-operated capacitors are connected in series and potted to form a unit, and the same is done with the dc-voltage-operated capacitors.

SUMMARY OF THE INVENTION

The primary object of the present invention is to provide a simplified voltage multiplier that occupies less space and insures the necessary voltage protection.

According to the invention a plurality of capacitors are combined in a unit which forms a capacitor roll. The space occupied by the potted elements is reduced by at least one-half that of known arrangements.

According to one feature of the invention, the capacitors are combined into two separate capacitor rolls and the diodes are disposed between said rolls.

In a variation of the invention, all capacitors are combined into one capacitor roll and surrounded by diodes on three sides.

The invention will now be explained in further detail with reference to the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a circuit diagram of a voltage tripler using the present novel arrangement;

FIG. 2 schematically shows the structure of the arrangement of FIG. 1;

FIG. 2a shows the actual physical arrangement of the elements of FIG. 2;

FIG. 3 is a circuit diagram of a voltage doubler as in the present invention;

FIG. 4 schematically shows the structure of the arrangement of FIG. 3;

FIG. 4a shows the physical arrangement of the elements of FIG. 4; and

FIG. 5 is a section through a portion of a laminated structure of a capacitor roll.

DESCRIPTION OF THE PREFERRED EMBODIMENT

In FIGS. 1 and 2, the diodes of the voltage tripler are designated by the reference numerals 1, 2, 3, 4, and 5, and the reference numerals 6 and 7 denote ac-voltage-operated capacitors which come into operation only during charging, while 8, 9, and 10 are dc-voltage-operated capacitors. The tripler is constructed in the form of a five-stage cascade circuit, the capacitor of the first stage, formed by the diode 1 and the capacitor 8, being grounded, and is used, for example, to generate the high voltage for operating color picture tubes. The capacitors 6 and 7 of the ac-voltage portion of the circuit are combined into a single capacitor roll 11, and the capacitors 8, 9, and 10 of the dc-voltage portion are combined in another capacitor roll 12. As shown in FIGS. 2 and 2a, the diodes 1, 2, 3, 4, and 5 are disposed centrally between capacitor rolls 11 and 12. The arrangement is potted in a case 13 filled with plastic. The ac-voltage input terminal 14, the dc-voltage output terminal 15, the ground terminal 16, and an additional terminal 17 are brought out on one side of the case 13. In addition, a terminal 18 is provided at a diode 19 connected to the ac-voltage input terminal 14.

The high voltages necessary for operating the black-and-white picture tube in a television receiver are generated with a voltage doubler. In FIGS. 3, 4, and 4a, the diodes are designated by the reference numerals 20, 21, and 22, while 23 denotes an ac-voltage-operated capacitor, and 24 and 25 the dc-voltage-operated capacitors of a three-stage cascade circuit. The alternating voltage to be doubled is applied to the terminal 28, and the dc voltage to be doubled appears at the terminal 29. The terminal 30 represents the ground terminal, to which is connected the capacitor of the first stage of the cascade, formed by the diode 20 and the capacitor 24. Both the ac-voltage-operated capacitor 23 and the dc-voltage-operated capacitors 24 and 25, are all combined into one capacitor roll 26. As shown in FIGS. 4 and 4a, the capacitor roll 26 is disposed centrally and surrounded by diodes 20, 21, and 22 on three sides. The arrangement is potted in a case 27 filled with plastic. The terminals 28, 29, 30 are brought out on the fourth, free side. The capacitor 25, shown with a broken line in FIG. 3, is dispensed with in the arrangement of FIG. 4. This is primarily replaced by the self-capacitance of the picture tube when the multiplier arrangement is connected into the television receiver.

The individual capacitors used in the present multiplier arrangements are designed as high-voltage capacitors having a laminated dielectric of polystyrene and polyester, and metal foil electrode plates. As shown in FIG. 5, the plates of a capacitor roll of this type are preferably aluminum foils 31, 32. Disposed between these foils are two polystyrene foils 33 with an intermediate polyester foil 34. The various terminals may be brought out at one or the other end of the roll or at both ends. At least two capacitor rolls of this laminated structure are wound over one another so that at least one multiple capacitor roll is obtained.

Since ac-voltage- and dc-voltage-operated capacitors may be combined into one capacitor roll, as shown in the embodiments of FIGS. 3 and 4, and as can also be done in the embodiments of FIGS. 1 and 2, the ac-voltage-operated capacitors are advantageously located inside within the roll. With such an arrangement, the corona discharge at the surface of the plastic-filled case 13 or 27 is greatly reduced.

In addition, the various possibilities of combining the capacitors, in conjunction with the diodes, make it possible to achieve particular input capacitances for voltage multipliers. For example, the input capacitance of the voltage multiplier arrangement is increased if at least one dc-voltage-operated and at least one ac-voltage-operated capacitor are combined into one roll.