GRUNDIG SUPER COLOR W6330 CHASSIS 29301-114.46(03) INTERNAL VIEW.

GRUNDIG SUPER COLOR W6330  CHASSIS 29301-114.46(03)  COLOR AMPLIFIER WITH Constant bandwidth RGB output amplifiers having simultaneous gain and DC output voltage control :
A color television receiver includes conventional circuitry for processing and detecting a received color television signal. Three chrominance-luminance matrices combine detected color difference and luminance signals forming color red, blue and green video signals. Emitter follower coupling stages apply the color video signals individually to each

of three output amplifiers which in turn drive the cathode electrodes of a unitized gun CRT. Potentiometers couple the emitter electrodes of the output amplifiers to a source of operating potential providing a simultaneous signal gain and DC output voltage adjustment for each amplifier during CRT color temperature setup. A voltage divider controls the voltage applied to the common screen grid electrode of the CRT providing a master setup adjustment.

1. In a color televison receiver, for processing and displaying a received television signal bearing modulation components of picture information, having a cathode ray tube including a trio of electron source means producing individual electron beams impinging an image screen to form three substantially overlying images and in which the respective operating points and relative conduction levels of said electron source means determine the color temperature of the reproduced image, the combination comprising:
master conduction means, coupled to said trio of electron source means simultaneously varying said conduction levels;
a plurality of substantially equal bandwidth amplifiers, each coupled to a different one of said electron source means, separately influencing said conduction levels;
low output impedance signal translation means recovering said picture information and supplying it to each of said plurality of amplifiers; and
separate adjusting means individually coupled to at least two of said amplifiers for simultaneously producing predetermined same sense variations in gain and DC output voltage of its associated amplifier while preserving said bandwidths.
2. The combination set forth in claim 1, wherein the transconductance and cutoff voltage of each of said electron source means bear a predetermined relationship and wherein said simultaneous predetermined variations in gain and DC output voltage are determined by said transconductance-cutoff voltage relationship. 3. The combination set forth in claim 2, wherein said plurality of amplifiers each include a gain and DC output voltage determining impedance and wherein each of said separate adjusting means include:
a variable impedance, coupling said gain and DC output voltage determining impedance of said associated amplifier to a source of bias current and forming a shunt path for signals within said amplifier.
4. The combination set forth in claim 3, wherein each of said electron source means include a cathode electrode and wherein each of said amplifiers include:
a transistor having input, common, and output electrodes, said output electrode being coupled to said electron source means cathode.

5. The combination set forth in claim 4, wherein said gain and DC output voltage determining impedance is coupled to said common electrode. 6. The combination set forth in claim 5, wherein said input, common, and output electrodes of said transistors are defined by base, emitter, and collector electrodes, respectively. 7. The combination set forth in claim 6, wherein said gain and DC output voltage determining impedance includes a resistor coupling said emitter electrode to ground and wherein said variable impedance includes:
a resistive control, having a variable resistance, coupling said emitter electrode to a source of operating potential.
8. The combination set forth in claim 7, wherein said three electron source means include control grid and screen grid electrodes common to said three electron guns and wherein variations of cathode electrode voltages permit changes of said relative conduction levels and said respective operating points. 9. The combination set forth in claim 8, wherein said master conduction means includes a variable bias potential source coupled to said common screen grid electrode.

Description:

BACKGROUND OF THE INVENTION
This invention relates to color television receivers and in particular to cathode ray tubes (CRT) drive systems therefor. Each of the several types of color television cathode ray tubes in current use includes a trio of individual electron sources producing distinct electron beams which are directed toward an image screen formed by areas of colored-light-emitting phosphors deposited on the inner surface of the CRT. The phosphors emit light of a given additive primary color (red, blue or green) when struck by high energy electrons. A "delta" electron gun arrangement, in which the electron sources comprise three electron guns disposed at the vertice

s of an equilateral triangle, having its base oriented in a horizontal plane and its apex above or below the base plane, may be used. Alternatively, the three electron sources may be "in line", that is, positioned in a horizontal line. In either case, the three beams produced are subjected to deflection fields and scan the image screen in both the horizontal and vertical directions thereby forming three substantially overlying rasters.
The phosphor deposits forming the image screen may alternatively comprise round dots, elongated areas, or uninterrupted vertical lines. A parallax barrier or shadow mask, defining apertures generally corresponding to the shape of the phosphor areas, is interposed between the electron guns and the image screen to "shadow" or block each phosphor area from electrons emitted from all but its corresponding electron gun.
A color television signal includes both luminance (monochrome) and chrominance (color) picture components. In the commonly used RGB drive systems the separately processed luminance and chrominance information is matrixed (or combined) before application to the CRT cathodes. Three output amplifiers apply the respective red, blue and green video signals thus produced for controlling the respective electron source currents.
The luminance components have substantially the same effect on all three electron sources whereas the color components are differential in nature, causing relative changes in electron source currents. In the absence of video signals, the combined raster should be a shade of grey. At high gun currents, the grey is very near white and at low settings, it is near black. The "color", commonly called color temperature, of the monochrome raster depends upon the relative contributions of red, blue and green light. At high color temperatures, the raster may appear blue and at low color temperatures it may appear sepia. While the most pleasing color temperature is largely a matter of design preference, ideally the receiver should not change color temperature under high and low brightness nor for high and low frequency picture information.
Generally, the electron sources comprise individual electron guns each including separately adjustable cathode, control grid and screen grid electrodes and a desired color temperature is achieved by adjustment of each electrode voltage during black and white setup. While the exact setup procedure employed varies with the manufacturer and specific CRT configuration, all manufacturers attempt to achieve consistent color temperature throughout the usable range of CRT beam current variations.
A typical color temperature adjustment involves setting the low light color temperature condition of each electron gun by adjusting its screen grid electrode voltage to produce the required DC conditions between electron guns at minimum beam currents. A high light or dive adjustment at increased CRT beam current is then made to insure consistent color temperature. In receivers utilizing CRT's with separately adjustable screen grid electrode voltages, the drive adjustment may take the form of a minor change in signal gain of the output amplifiers. The process is, in essence, one of configuring the operating points of the three electron guns to conform to three substantially identical output amplifiers.
The recently developed economical "unitized gun" type CRT has a combined electron source structure in which three common control grids and three common screen grids are used with the cathodes being the only electrically separate electrodes. The greatly simplified and more economical unitized gun structure, however, imposes some restrictions on the circuitry used to drive the electron sources. Perhaps most significant is the absence of

the flexibility previously provided by individually adjustable screen grid electrode voltages. Due in part to the inverse relationship between electron source transconductance, which may be thought of as "gain" of the electron source, and cutoff voltage, the typical individual low level color temperature or equal cutoff adjustment described above also performs the additional function of establishing nearly equal transconductances for the three electron sources. As a result only minor relative changes in electron source currents occur at higher CRT beam currents.
Color temperature adjustment in a receiver with a unitized gun CRT involves a somewhat different process, namely, configuring the drive and bias applied to each of the gun cathodes to accommodate differences in relative electron source characteristics which, without the equalizing effect of separate screen electrode adjustments, may be considerable.
Initially television receivers using unitized gun CRT's utilized a variable DC voltage divider operative upon each output amplifier to provide adjustment of the DC cutoff voltage. Drive, or signal gain, adjustment to accommodate differences in electron source transconductances was generally accomplished by separate individual gain controls operative on each of the output amplifiers.
However, the more recently developed unitized gun systems combine the DC voltage (cutoff) and signal gain (drive) adjustments for each electron source by simultaneously varying the signal gain and DC voltage in the same direction in a predetermined relationship. One such system used three CRT coupling networks each of which includes a variable impedance simultaneously operative on both the amplitude of coupled signal and DC voltage. Another system uses a variable collector load impedance for each of the output amplifiers, making use of the changes in amplifier signal gain and DC output voltage resulting from collector load variations.
While such systems provide an adequate range of adjustment to achieve color temperature setup using a reduced number of controls, they often degrade image quality. Ideally, the luminance portion of the signal is applied uniformly to each of the three electron sources. Although the relative signal amplitudes may be varied to accommodate transconductance differences between electron sources, it is desirable that each applied signal be an otherwise identical replica of the others. The variable impedance elements in the voltage divider networks and variable collector loads of the prior art interact with the capacities inherent in the output amplifiers and electron gun structures to produce unequal bandwidths for the different color video signals, which cause color changes in their high frequency components (which correspond to detailed picture information). The resulting effect upon the displayed image is similar in appearance to the well-known "color fringing" or misconvergence effect.
OBJECTS OF THE INVENTION
It is an object of the present invention to provide an improved color television receiver.
It is a further object of this invention to provide a novel CRT color temperature setup system.
SUMMARY OF THE INVENTION
In a color television receiver, for processing and displaying a received television signal bearing modulation components of picture information, a

cathode ray tube includes three electron source means producing individual electron beams which impinge an image screen to form three substantially overlying images. The respective operating points and relative conduction levels of the electron source means determine the color temperature of the reproduced image. Master conduction means, coupled to the three electron source means, simultaneously vary the conduction levels and a plurality of substantially equal bandwidth amplifiers, each coupled to a different one of the electron source means, separately influence the conduction levels. Low output impedance signal translation means recover the picture information and supply it to each of the amplifiers. Separate adjusting means are individually coupled to at least two of the amplifiers for simultaneously producing predetermined variations in the gain and DC output voltage of the amplifiers while preserving the bandwidths.
BRIEF DESCRIPTION OF THE DRAWING
The drawing shows a partial-schematic, partial-block diagram representation of a color television receiver constructed in accordance with the present invention.
DESCRIPTION OF THE PREFERRED EMBODIMENT
Referring to the drawing, a signal processor 10 includes conventional circuitry (not shown) for amplifying a received television signal and detecting the modulated components of luminance and chrominance information therein. The output of signal processor 10 is coupled to a luminance amplifier 11 and a chrominance processor 30. Luminance amplifier 11 is conventional and includes circuitry controlling brightness and contrast of the luminance signal. The output of luminance amplifier 11 is coupled to three luminance-chrominance matrices 12, 13 and 14. Chrominance processor 30 includes conventional chrominance information detection circuitry for providing three color difference or color-minus-luminance output signals (R-Y, G-Y and B-Y) which are individually coupled to luminance-chrominance matrices 12, 13 and 14, respectively. The signal from luminance amplifier 11 is combined with the color-minus-luminance signals from chrominance processor 30 to form the respective red, green and blue video signals which are coupled to the R, G and B output amplifiers 15, 16 and 17, respectively. The outputs of amplifiers 15, 16 and 17 are coupled to the cathode electrodes 23, 24 and 25, respectively, of a CRT 20 having an image screen 21. A voltage divider, formed by a series combination of resistors 83 and 84, is coupled between a source of operating potential +V₂ and ground. The junction of resistors 83 and 84 is connected to a common control grid electrode 28 and to ground by a filter capacitor 85 which provides a signal bypass. A potentiometer 80 and a resistor 81 are series coupled between a source of operating potential +V₁ and ground, forming another voltage divider. The junction of potentiometer 80 and resistor 81 is connected to common screen grid electrode 29 and to ground by a bypass capacitor 82. Cathode electrodes 23-25, control grid electrode 28 and screen grid electrode 29 are part of a unitized gun structure in CRT 20 with the control grid and screen grid being common to each of the three electron sources defined by the separate cathode electrodes.
While luminance-chrominance matrices 12 and 13 are shown in block form, it should be understood that they are identical to the detailed structure of matrix 14. Similarly, red output amplifier 15 and green output amplifier 16 are identical to the detailed structure of blue output amplifier 17. Further, the receiver shown is understood to include conventional circuitry for horizontal and vertical electron beam deflection together with means deriving a CRT high voltage accelerating potential, all of which have, for clarity, been omitted from the drawing.
Luminance-chrominance matrix 14 includes a matrix transistor 40 having an emitter electrode 41 coupled to ground by a resistor 55 and by a series combination of resistors 46 and 47, a base electrode 42 coupled to the output of luminance amplifier 11, and a collector electrode 43 coupled to a source of operating potential +V₃ by a resistor 45. The B-Y output of chroma processor 30 is connected to the junction of resistors 46 and 47. An emitte

r-follower transistor 50 has an emitter electrode 51 coupled to ground by a resistor 56, a base electrode 52 connected to the collector of matrix transistor 40, and a collector electrode 53 connected to +V₃.
Blue amplifier 17 includes an output transistor 60 having an emitter electrode 61 coupled to ground by a series combination of resistors 67 and 68, a base electrode 62 connected to the emitter of transistor 50, and a collector electrode 63 coupled to +V₂ by a resistor 66. A series combination of a potentiometer 70 and a resistor 69 couples the junction of resistors 67 and 68 to +V₃. Collector 63, which is the output of amplifer 17, is connected to cathode 25 of CRT 20.
During signal reception, the separately processed luminance and B-Y color difference signals are applied to matrix transistor 40. The combined signal developed at its collector 43 forms the blue video signal which controls the blue electron beam in CRT 20 and represents the relative contribution of blue light in the image produced.
The blue video signal at collector 43 is coupled via transistor 50 to base 62 of output transistor 60. The low source impedance of emitter follower transistor 50 obviates any detrimental effects upon the blue video signal due to loading at the input to amplifier 17 caused by gain or frequency dependent input impedance variations of amplifier 17. The blue video signal applied to base 62 is amplified by transistor 60 to a level sufficient to control the conduction of its respective electron source.
During color temperature setup, a predetermined setup voltage (corresponding to black) is applied to matrices 12, 13 and 14. The voltage on common screen grid electrode 29 is adjusted, by varying potentiometer 80 which together with resistor 81 and capacitor 82 form master conduction means, to cause a low brightness raster to appear on image screen 21. As will be seen, adjustment of potentiometer 70 and the corresponding potentiometers in amplifiers 15 and 16 establish the correct combination of DC electron source cathode voltages and output amplifier gains to produce the selected color temperature at both low and high CRT beam currents.
Amplifier 17 includes a common emitter transistor stage in which the impedance coupled to emitter electrode 6 is a gain and DC output voltage determining impedance. Signal gain is approximately equal to the ratio of the collector impedance (resistor 66), to this gain and DC voltage determining impedance (ignoring the effects of capacities associated with the tr

ansistor and the electron gun which will be considered later). Because the source of operating potential +V₃ coupled to potentiometer 70 forms a good AC or signal ground, the series combination of resistor 69 and potentiometer 70 are effectively in parallel with resistor 68 and the total impedance coupling emitter 61 to signal ground comprises resistor 67 in series with this combination of resistors 68 and 69 and potentiometer 70. Variations in this impedance caused by adjustment of potentiometer 70 changes the ratio of collector to emitter impedances and thereby the gain of amplifier 17. If potentiometer 70 is varied to present increased resistance, gain is reduced and if varied to present decreased resistance, gain is increased.
The DC voltage at collector 63 of transistor 60 is determined by the product of the collector resistance and quiescent collector current (current in the absence of applied signal) and V₂. The voltage at base 62 is established by the emitter voltage of transistor 50. Variations in the resistance of potentiometer 70 cause variations in current flow in the series path including potentiometer 70 and resistors 69 and 68. The voltage developed across resistor 68 is supplied to emitter 61 through resistor 67.
In the absence of signal, the DC voltage at base 62 is constant and the relative voltage between base 62 and emitter 61, which controls the conduction level of transistor 60, is a function of the voltage at emitter 61. Increases in the resistance of potentiometer 70 reduce the emitter voltage, increase the relative base-emitter voltage of transistor 60, and increase collector current. The increased collector current develops a greater voltage drop across collector resistor 66 and reduces the DC voltage at collector 63 (and cathode 25). Conversely, a decrease in the resistance of potentiometer 70 increases the voltage at emitter 61, reducing the relative base-emitter voltage and decreasing collector current. The smaller voltage drop across resistor 66 increases the DC voltage at collector 63 and cathode 25.
Thus, increasing the resistance of potentiometer 70 produces proportionate simultaneous reduction of the DC voltage applied to cathode 25 and the voltage gain of amplifier 17, whereas decreasing the resistance of potentiometer 70 produces proportionate simultaneous increase of the DC voltage and signal gain. As mentioned above, amplifiers 15 and 16 are identical to amplifier 17. In practice only two of the three output amplifiers require adjustment to achieve color temperature setup. However, greater flexibility and optimum use of amplifier signal handling capability is realized if all three output amplifiers are adjustable.
As previously mentioned capacities associated with transistor 60, cathode 25 and corresponding interconnections (such as those used to couple collector 63 to cathode 25) are effectively in parallel with collector load resistor 66 forming a partially reactive "coupling network" which exhibits a frequency characteristic (bandwidth) affecting signals coupled therethrough. In practice, the other coupling networks have identical bandwidths and affect their signals in an equal manner. The setup control adjustments of the present invention do not change the characteristics of these coupling networks and the uniformity of signal coupling for the different color signals is preserved. In contrast, conventional adjustment circuitry (whether variable collector load or voltage divider) place variable impedances within these couplings. The varied adjustments of these impedances to effect color temperature control adjustment disturb the bandwidth characteristics of the coupling networks causing differential variations in the individual color video signals.
What has been shown is an RGB CRT drive system which includes output amplifiers each having a single control which simultaneously achieves changes of the DC output voltage and signal gain of the amplifier in a predetermined relationship. The bandwidths of all three output amplifiers and their associated coupling networks remain substantially undisturbed by these control adjustments during CRT color temperature setup.
While particular embodiments of the invention have been shown and described, it will be obvious to those skilled in the art that changes and modifications may be made without departing from the invention in its broader aspects, and, therefore, the aim in the appended claims is to cover all such changes and modifications as fall within the true spirit and scope of the invention.


GRUNDIG SUPER COLOR W6330  CHASSIS 29301-114.46(03)  LINE DEFLECTION WITH THYRISTOR SWITCH TECHNOLOGY OVERVIEW.(ZEILEN ABLENKUNG MIT THYRISTORS SCHALTUNG)


 

GRUNDIG SUPER COLOR W6330  CHASSIS 29301-114.46(03)  INTEGRAL THYRISTOR-RECTIFIER DEVICE:

BSTCC0146 7J R  BSTCC0643 7F H (SIEMENS)

A semiconductor switching device comprising a silicon controlled rectifier (SCR) and a diode rectifier integrally connected in parallel with the SCR in a single semiconductor body. The device is of the NPNP or PNPN type, having gate, cathode, and anode electrodes. A portion of each intermediate N and P region makes ohmic contact to the respective anode or cathode electrode of the SCR. In addition, each intermediate region includes a highly conductive edge portion. These portions are spaced from the adjacent external regions by relatively low conductive portions, and limit the conduction of the diode rectifier to the periphery of the device. A profile of gold recombination centers further electrically isolates the central SCR portion from the peripheral diode portion.
That class of thyristors known as controlled rectifiers are semiconductor switches having four semiconducting regions of alternate conductivity and which employ anode, cathode, and gate electrodes.
 
These devices are usually fabricated from silicon. In its normal state, the silicon controlled rectifier (SCR) is non-conductive until an appropriate voltage or current pulse is applied to the gate electrode, at which point current flows from the anode to the cathode and delivers power to a load circuit. If the SCR is reverse biased, it is non-conductive, and cannot be turned on by a gating signal. Once conduction starts, the gate loses control and current flows from the anode to the cathode until it drops below a certain value (called the holding current), at which point the SCR turns off and the gate electrode regains control. The SCR is thus a solid state device capable of performing the circuit function of a thyratron tube in many electronic applications. In some of these applications, such as in automobile ignition systems and horizontal deflection circuits in television receivers, it is necessary to connect a separate rectifier diode in parallel with the SCR. See, for example, W. Dietz, U. S. Pat. Nos. 3,452,244 and 3,449,623. In these applications, the anode of the rectifier diode is connected to the cathode of the SCR, and the cathode of the rectifier is connected to the SCR anode. Thus, the rectifier diode will be forward biased and current will flow through it when the SCR is reverse biased; i.e., when the SCR cathode is positive with respect to its anode. For reasons of economy and ease of handling, it would be preferable if the circuit function of the SCR and the associated diode rectifier could be combined in a single device, so that instead of requiring two devices and five electrical connections, one device and three electrical connections are all that would be necessary. In fact, because of the semiconductor profile employed, many SCR's of the shorted emitter variety inherently function as a diode rectifier when reverse biased. However, the diode rectifier function of such devices is not isolated from the controlled rectifier portion, thus preventing a rapid transition from one function to the other. Therefore, it would be desirable to physically and electrically isolate the diode rectifier portion from that portion of the device which functions as an SCR.

(Thyristor Horizontal Ablenkung steuerung)

Horizontal deflection circuit



























Description:



1. A horizontal deflection circuit for generating the deflection current in the deflection coil of a television picture tube wherein a first switch controls the horizontal sweep, and wherein a second switch in a so-called commutation circuit with a commutating inductor and a commutating capacitor opens the first switch and, in addition, controls the energy transfer from a dc voltage source to an input inductor, characterized in that the input inductor (Le) and the commutating inductor (Lk) are combined in a unit designed as a transformer (U) which is proportioned so that the open-circuit inductance of the transformer is essentially equal to the value of the input

inductor (Le), while the short-circuit inductance of the transformer (U) is essentially equal to the value of the commutating inductor (Lk), and that the second switch (S2) is connected in series with the dc voltage source (UB) and a first winding (U1) of the transformer (U). 2. A horizontal deflection circuit according to claim 1, characterized in that the transformer (U) operates as an isolation transformer between the supply (UB) and the subcircuits connected to a second winding. 3. A horizontal deflection circuit according to claim 1, characterized in that the second switch (S2) is connected between ground and that terminal of the first winding (U1) of the transformer (U) not connected to the supply potential (+UB). 4. A horizontal deflection circuit according to claim 1, characterized in that a capacitor (CE) is connected across the series combination of the first winding (U1) of the transformer and the second switch (S2). 5. A horizontal deflection circuit according to claim 1, characterized in that the second winding (U2) of the transformer (U) is connected in series with a first switch (S1), the commutating capacitor (Ck), and a third, bipolar switch (S3) controllable as a function of the value of a controlled variable developed in the deflection circuit. 6. A horizontal deflection circuit according to claim 5, characterized in that the third switch (S3) is connected between ground and the second winding (U2) of the transformer. 7. A horizontal deflection circuit according to claim 2, characterized in that the isolation transformer carries a third winding via which power is supplied to the audio output stage of the television set. 8. A horizontal deflection circuit according to claims 2, characterized in that the voltage serving to control the first switch (S1) is derived from a third winding of the transformer.

Description:

The present invention relates to a horizontal deflection circuit for generating the deflection current in the deflection coil of a television picture tube wherein a first switch controls the horizontal sweep, and wherein a second switch in a so-called commutation circuit with a commutating inductor and a commutating capacitor opens the first switch and, in addition, controls the energy transfer from a dc voltage source to an input inductor.
German Auslegeschrift (DT-AS) No. 1,537,308 discloses a horizontal deflection circuit in which, for generating a periodic sawtooth current within the respective deflection coil of the picture tube, in a first branch circuit, the deflection coil is connected to a sufficiently large capacitor serving as a current source via a first controlled, bilaterally conductive switch which is formed by a controlled rectifier and a diode connected in inverse parallel. The control electrode of the rectifier is connected to a drive pulse source which renders the switch conductive during part of the sawtooth trace period. In that arrangement, the sawtooth retrace, i.e. the current reversal, also referred to as "commutation", is initiated by a second controlled switch.
The first controlled switch also forms part of a second branch circuit where it is connected in series with a second current source and a reactance capable of oscillating. When the first switch is closed, the reactance, consisting essentially of a coil and a capacitor, receives energy from the second current source during a fixed time interval. This energy which is taken from the second current source corresponds to the circuit losses caused during the previous deflection cycle.
As can be seen, such a circuit needs two different, separate inductive elements, it being known that inductive elements are expensive to manufacture and always have a certain volume determined by the electrical properties required.
The object of the invention is to reduce the amount of inductive elements required.

The invention is characterized in that the input inductor and the commutating inductor are combined in a unit designed as a transformer which is proportioned so that the open-circuit inductance of the transformer is essentially equal to the value of the input inductor, while the short-circuit inductance of the transformer is essentially equal to the value of the commutating inductor, and that the second switch is connected in series with the dc voltage source and a first winding of the transformer.
This solution has an added advantage in that, in mass production, both the open-circuit and the short-circuit inductance are reproducible with reliability.
According to another feature of the invention, the electrical isolation between the windings of the transformer is such that the transformer operates as an isolation transformer between the supply and the subcircuits connected to a second winding or to additional windings of the transformer. In this manner, the transformer additionally provides reliable mains isolation.
According to a further feature of the invention, the second switch is connected between ground and that terminal of the first winding of the transformer not connected to the supply potential. This simplifies the control of the switch.

According to a further feature of the invention, to regulate the energy supply, the second winding of the transformer is connected in series with the first switch, the commutating capacitor, and a third, bipolar switch controllable as a function of the value of a controlled variable developed in the deflection circuit.

The advantage gained by this measure lies in the fact that the control takes place on the side separated from the mains, so no separate isolation device is required for the gating of the third switch. Further details and advantages will be apparent from the following description of the accompanying drawings and from the claims. In the drawings,
FIG. 1 is a basic circuit diagram of the arrangement disclosed in German Auslegeschrift (DT-AS) No. 1,537,308;
FIG. 2 shows a first embodiment of the horizontal deflection circuit according to the invention, and
FIG. 3 shows a development of the horizontal deflection circuit according to the invention.
FIG. 1 shows the essential circuit elements of the horizontal deflection circuit known from the German Auslegeschrift (DT-AS) No. 1,537,308 referred to by way of introduction.
Connected in series with a dc voltage source UB is an input inductor Le and a bipolar, controlled switch S2. In the following, this switch will be referred to as the "second switch"; it is usually called the "commutating switch" to indicate its function.
In known circuits, the second switch S2 consists of a controlled rectifier and a diode connected in inverse parallel.
The second switch S2 also forms part of a second circuit which contains, in addition, a commutating inductor Lk, a commutating capacitor Ck, and a first switch S1. The first switch S1, controlling the horizontal sweep, is constructed in the same manner as the above-described second switch S2, consisting of a controlled rectifier and a diode in inverse parallel. Connected in parallel with this first switch is a deflection-coil arrangement AS with a capacitor CA as well as a high voltage generating arrangement (not shown). In FIGS. 1, 2, and 3, this arrangement is only indicated by an arrow and by the reference characters Hsp. The operation of this known horizontal deflection circuit need not be explained here in detail since it is described not only in the German Auslegeschrift referred to by way of introduction, but also in many other publications.
FIGS. 2 and 3 show the horizontal deflection circuit modified in accordance with the present invention. Like circuit elements are designated by the same reference characters as in FIG. 1.
FIG. 2 shows the basic principle of the invention. The two inductors Le and Lk of FIG. 1 have been replaced by a transformer U. To be able to serve as a substitute for the two inductors Le and Lk, the transformer must be proportioned in a special manner. Regardless of the turns ratio, the open-circuit inductance of the transformer is chosen to be essentially equal to the value of the input inductor Le, and the short-circuit inductance of the transformer is essentially equal to the value of the commutating inductor Lk.
To permit the second switch S2 to be utilized for the connection of the dc voltage source UB, it is included in the circuit of that winding U1 of the transformer connected to the dc voltage UB.
In principle, it is of no consequence for the operation of the switch S2 whether it is inserted on that side of the winding U1 connected to the positive operating potential +UB or on the side connected to ground. In practice, however, the solution shown in FIGS. 2 and 3 will be chosen since the gating of the controlled rectifier is less problematic in this case.

In compliance with pertinent safety regulations, the transformer U may be designed as an isolation transformer and can thus provide mains separation, which is necessary for various reasons. It is known from German Offenlegungschrift (DT-OS) No. 2,233,249 to provide dc isolation by designing the commutating inductor as a transformer, but this measure is not suited to attaining the object of the present invention.
If the energy to be taken from the dc voltage source is to be controlled as a function of the energy needed in the horizontal deflection circuit and in following subcircuits, the embodiment of the horizontal deflection circuit of FIG. 3 may be used.
The circuit including the winding U2 of the transformer U contains a third controlled switch S3, which, too, is inserted on the grounded side of the winding U2 for the reasons mentioned above. This third switch S3, just as the second switch S2, is operated at the frequency of a horizontal oscillator HO, but a control circuit RS whose input l is fed with a controlled variable is inserted between the oscillator and the switch S3. Depending on this controlled variable, the controlled rectifier of the third switch S3 can be caused to turn on earlier. A suitable controlled variable containing information on the energy consumption is, for example, the flyback pulse capable of being taken from the high voltage generating circuit (not shown). Details of the operation of this kind of energy control are described in applicant's German Offenlegungsschrift (DT-OS) No. b 2,253,386 and do not form part of the present invention.
With mains isolation, the additional, third switch S3 shown here has the advantage of being on the side isolated from the mains and eliminates the need for an isolation device in the control lead of the controlled rectifier.
As an isolation transformer, the transformer U may also carry additional windings U3 and U4 if power is to be supplied to the audio output stage, for example; in addition, the first switch S1 may be gated via such an additional winding.
The points marked at the windings U1 and U2 indicate the phase relationship between the respective voltages. Connected in parallel with the winding U1 and the second switch S2 is a capacitor CE which completes the circuit for the horizontal-frequency alternating current; this serves in particular to bypass the dc voltage source or the electrolytic capacitors contained therein.
If required, a well-known tuning coil may be inserted, e.g. in series with the second winding U2, without changing the basic operation of the horizontal deflection circuit according to the invention.



GRUNDIG SUPER COLOR W6330  CHASSIS 29301-114.46(03)    Electron beam deflection circuit including thyristors Further Discussion and deepening of knowledge, Thyristor horizontal output circuits:

1. An electron beam deflection circuit for a cathode ray tube with electromagnetic deflection by means of a sawtooth current waveform having a trace portion and a retrace portion, said circuit comprising: a deflection winding; a first source of electrical energy formed by a first capacitor; first controllable switching means comprising a parallel combination of a first thyristor and a first diode connected together to conduct in opposite directions, for connecting said winding to said first source during said trace portion, while said first switching means is turned on; a second source of electrical energy including a first inductive energy storage means coupled to a voltage supply; reactive circuit means including a combination of inductive and capacitive reactances for storing the energy supplied by said second source; second controllable switching means, substantially similar to said first one, for completing a circuit including said reactive circuit means and said first switching means, when turned on before the end of said trace portion, so as to pass through said first switching means an oscillatory current in opposite direction to that which passes through said first thyristor from said first source and to turn said first thyristor off after these two currents cancel out, the oscillatory current flowing thereafter through said first diode for an interval termed the circuit turn-off time, which has to be greater than the turn-off time of said first thyristor; wherein the improvement comprises: means for drawing, during at least a part of said trace portion, a substantial amount of additional current through said first switching means, in the direction of conduction of said first diode, whereby to perceptibly shift the waveform of the current flowing through said first switching means towards the negative values by an amount equal to that of said substantial additional current and to lengthen, in proportion thereto, said circuit turn-off time, without altering the values of the reactances in the reactive circuit which intervene in the determination of both the circuit turn-off and retrace portion time intervals.

2. A deflection circuit as claimed in claim 1, wherein said amount of additional current is greater than or equal to 5 per cent of the peak-to-peak value of the current flowing through the deflection winding.

3. A deflection circuit as claimed in claim 1, wherein said means for drawing a substantial amount of additional current through said first switching means comprises a resistor connected in parallel to said first capacitor.

4. A deflection circuit as claimed in claim 1, wherein said means for drawing an additional current is formed by connecting said first and second energy sources in series so that the current charging said reactive circuit means forms the said additional current.

5. A deflection circuit as claimed in claim 1, further including a series combination of an autotransformer winding and a second high-value capacitor, said combination being connected in parallel to said first switching means, wherein said autotransformer comprises an intermediate tap located between its terminals respectively connected to said first switching means and to said second capacitor, said tap delivering, during said trace portion, a suitable DC supply voltage lower than the voltage across said second capacitor; and wherein said means for drawing a substantial amount of additional current comprises a load to be fed by said supply voltage and having one terminal connected to ground; and further controllable switching means controlled to conduct during at least part of said trace portion and to remain cut off during said retrace portion, said further switching means being connected between said tap and the other terminal of said load.

Description:

The present invention relates to electron beam deflection circuits including thyristors, such as silicon controlled rectifiers and relates, in particular, to horizontal deflection circuits for television receivers.

The present invention constitutes an improvement in the circuit described in U.S. Pat. No. 3,449,623 filed on Sept. 6, 1966, this circuit being described in greater detail below with reference to FIGS. 1 and 2 of the accompanying drawings. A deflection circuit of this type comprises a first thyristor switch which allows the conenction of the horizontal deflection winding to a constant voltage source during the time interval used for the transmisstion of the picture signal and for applying this signal to the grid of the cathode ray tube (this interval will be termed the "trace portion" of the scan), and a second thyristor switch which provides the forced commutation of the first one by applying to it a reverse current of equal amplitude to that which passes through it from the said voltage source and thus to initiate the retrace during the horizontal blanking interval.

A undirectional reverse blocking triode type thyristor or silicon controlled rectifier (SCR), such as that used in the aformentioned circuit, requires a certain turn-off time between the instant at which the anode current ceases and the instant at which a positive bias may be applied to it without turning it on, due to the fact that there is still a high concentration of free carriers in the vicinity of the middle junction, this concentration being reduced by a process of recombination independently from the reverse polarity applied to the thyristor. This turn-off time of the thyristor is a function of a number of parameters such as the junction temperature, the DC current level, the decay time of the direct current, the peak level of the reverse current applied, the amplitude of the reverse anode to cathode voltage, the external impedance of the gate electrode, and so on, certain of these varying considerably from one thyristor to another.

In horizontal deflection circuits for television receivers, the flyback or retrace time is limited to approximately 20 percent of the horizontal scan period, the retrace time being in the case of the CCIR standard of 625 lines, approximately 12 microseconds and, in the case of the French standard of 819 lines, approximately 9 microseconds. During this relatively short interval, the thyristor has to be rendered non-conducting and the electron beam has to be returned to the origin of the scan. The first thyristor is blocked by means of a series resonant LC circuit which is subject to a certain number of restrictions (limitations as to the component values employed) due to the fact that, inter alia, it simultaneously determines the turn-off time of the circuit which blocks the thyristor and it forms part of the series resonant circuit which is to carry out the retrace. To obtain proper operation of the deflection circuit of the aforementioned Patent, especially when used for the French standard of 819 lines per image, the values of the components used have to subject to very close tolerances (approximately 2%), which results in high costs.

The improved deflection circuit, object of the present invention, allows the lengthening of the turn-off time of the circuit for turning the scan thyristor off, without altering the values of the LC circuit, which are determined by other criteria, and without impairing the operation of the circuit.

According to the invention, there is provided an electron beam deflection circuit for a cathode ray tube with electromagentic deflection by means of a sawtooth current waveform having a trace portion and a retrace portion, said circuit comprising: a deflection winding; a first source of electrical energy formed by a first capacitor; first controllable switching means comprising a parallel combination of a first thyristor and a first diode, connected together to conduct in opposite directions, for connecting said winding to said first source during said trace portion when said first switching means is turned on; a second source of electrical energy including a first inductive energy storage means coupled to a voltage supply; reactive circuit means including a combination of inductive and capacitive reactances for storing the energy supplied by the said second source; a second controllable switching means, substantially identical with the first one, for completing a circuit including said reactive circuit means and said first switching means, when turned on, so as to pass through said first thyristor an oscillatory current in the opposite direction to that which passes through it from said first source and to turn it off after these two currents cancel out, the oscillatory current then flowing through said first diode for an interval termed the circuit turn-off time which has to be greater than the turn-off time of said first thyristor; and means for drawing duing at least a part of said trace portion a substantial amount of additional current from said first switching means in the direction of conduction of said first diode, whereby said circuit turn-off time is lengthened in proportion to the amount of said additional current, without altering the values of the reactances in the reactive circuit by shifting th

e waveform of the current flowing through said first switching means towards the negative by an amount equal to that of said additional current.

A further object of the invention consists in using the supplementary current in the recovery diode of the first switching means to produce a DC voltage which may be used as a power supply for the vertical deflection circuit of the television receiver, for example.

The invention will be better understood and other features and advantages thereof will become apparent from the following description and the accompanying drawings, given by way of example, and in which:

FIG. 1 is a schematic circuit diagram partially in bloc diagram form of a prior art deflection circuit according to the aforementioned Patent;

FIG. 2 shows waveforms of currents and voltages generated at various points in the circuit of FIG. 1;

FIG. 3 is a schematic diagram of a deflection circuit according to the invention which allows the principle of the improvement to be explained;

FIG. 4 is a diagram of the waveforms of the current through the first switching means 4, 5 of the circuit of FIG. 3;

FIG. 5 is a circuit diagram of another embodiment of the circuit according to the invention;

FIG. 6 is a schematic representation of the preferred embodiment of the circuit according to the invention; and

FIG. 7 shows voltage waveforms at various points of the high voltage autotransformer 21 of FIG. 6.

In all these Figures the same reference numerals refer to the same components.

FIG. 1 shows the horizontal deflection circuit described and claimed in the U.S. Pat. No. 3,449,623 mentioned above, which comprises a first source of electrical energy in the shape of a first capacitor 2 having a high capacitance C ₂ for supplying a substantially constant voltage Uc ₂ across its terminals. A first terminal of the first capacitor 2 is connected to ground, whilst its second terminal which supplies a positive voltage is connected to one of the terminals of a horizontal deflection winding shown as a first inductance 1. A first switching means 3, consisting of a first reverse blocking triode thyristor 4 (SCR) and a first recovery diode 5 in parallel, the two being interconnected to conduct current in opposite directions, is connected in parallel with the series combination formed by the deflection winding 1 and the first capacitor 2. The assembly of components 1, 2, 4 and 5 forms the final stage of the horizontal deflection circuit in a television receiver using electromagnetic delfection.

The deflection circuit also includes a drive stage for this final stage which here controls the turning off of the first thyristor 4 to produce the retrace or fly-back portion of the scan during the line-blanking intervals i.e. while the picture signal is not transmitted. This driver stage comprises a second voltage source in the shape of a DC power supply 6 which delivers a constant high voltage E. The negative terminal of the power supply 6 is connected to ground and its positive terminal to one of the terminals of a second inductance 7 of relatively high value, which draws a substantially lineraly varying current from the power supply 6 to avoid its overloading. The other terminal of the second inductance 7 is connected, on the one hand, to the junction of the deflection winding 1 and the first switching means 3 by means of a second inductance 8 and a second capacitor 9 in series and, on the other hand, to one of the terminals of a second controllable bi-directionally conducting switching means 10, similar to the first one 3, including a parallel combination of a second thyristor 11 and a second recovery diode 12 also arranged to conduct in opposite directions.

The respective values of the third inductance 8 (L ₈ ) and of the second capacitor 9 (C ₉ ) are principally selected so that, on the one hand, one half-cycle of oscillation of the first series resonant circuit L ₈ - C ₉ , (i.e. π √ L ₈ ^. C ₉ ) is longer than the turn-off time of the first thyristor 4, but still is as short as possible since this time interval determines the speed of the commutation of the thyristor 4, and, on the other hand, one half-cycle of oscillation of another series resonant circuit formed by L ₁ , L ₈ and C ₉ , i.e. π √ (L ₁ + L ₈ ) ^. C ₉ , is substantially equal to the required retrace time interval (i.e. shorter than the horizontal blanking interval).

The gate (control electrode) of the second thyristor 11 is coupled to the output of the horizontal oscillator 13 of the television receiver by means of a first pulse transformer 14 and a first pulse shaping circuit 15 so that it is fed short triggering pulses which are to turn it on.

The gate of the first thyristor 4 fed with signals of a substantially rectangular waveform which are negative during the horizontal blanking intervals, is coupled to a winding 16 by means of a second pulse shaping circuit 17, the winding 16 being magnetically coupled to the second inductance 7 to make up the secondary winding of a transformer of which the inductance 7 forms the primary winding. It will be noted here that it is also possible to couple the secondary winding 16 magnetically to a primary winding connected to a suitable output (not shown) of the horizontal oscillator 13.

The operation of a circuit of this type will be explained below with reference to FIG. 2 which shows the waveforms at various points in the circuit of FIG. 1 during approximately one line period.

FIG. 2 is not to scale since one line period (t ₇ - t ₀ ) is equal to 64 microseconds in the case of 625 lines and 49 microseconds in the case of 819 lines, while the durations of the respective horizontal blanking intervals are approximately 12 and 9.5 microseconds.

Waveform A shows the form of the current i _L1 passing through deflection winding 1, this current having a sawtooth waveform substantially linear from t ₀ to t ₃ and from t ₅ to t ₇ , and crossing zero at time instants t ₀ and t ₇ , and reaching values of + I _1m and - I _1m , at time instants t ₃ and t ₅ respectively, these being its maximum positive and negative amplitudes.

During the second half of the trace portion of the horizontal deflection cycle, that is to say from t ₀ to t ₃ , the thyristor 4 of the first switching means 3 is conductive and makes the high value capacitor 2 discharge through the deflector winding 1, which has a high inductance, so that current i _L1 increases linearly.

A few microseconds (5 to 8 μ s) before the end of the trace portion, i.e. at time instant t ₁ , the trigger of the second thyristor 11 receives a short voltage pulse V _G11 which causes it to turn on as its anode is at this instant at a positive potential with respect to ground, which is due to the charging of the second capacitor 9 through inductances 7 and 8 by the voltage E from the power supply 6.

When thyristor 11 is made conductive at time t ₁ , on the one hand, inductance 7 is connected between ground and the voltage source 6 and a linearly increasing current flows through it and, on the other hand, the reactive circuit 8, 9 forms a loop through the second and first switching means 10 and 3, thus forming a resonant circuit which draws an oscillatory current i ₈ ,9 of frequency ##EQU1##

This oscillatory current i ₈ ,9 will pass through the first switching means 3, i.e. thyristor 4 and diode 5, in the opposite direction to that of current i _L1 . Since the frequency f ₁ is high, current i ₈ ,9 will increase more rapidly than i _L1 and will reach the same level at time t ₂ , that is to say i ₈ ,9 (t ₂ ) = -i _L1 (t ₂ ) and these currents will cancel out in the thyristor 4 in accordance with the well known principle of forced commutation. After time instant t ₂ , current i ₈ ,9 continues to increase more rapidly than i _L1 , but the difference between them (i ₈ ,9 - i _L1 ) passes the diode 5 (see wave form B) until it becomes zero at time instant t ₃ which is the turn off time instant of the first switching means 3, at which the retrace begins.

The interval between the time instant t ₂ and t ₃ , i.e. (t ₃ -t ₂ ), during which diode 5 is conductive and the thyristor is reverse biased will be termed in what follows the circuit turn-off time and it should be greater than the turn-off time of the thyristor 4 itself since the latter will subsequently become foward biased (i.e. from t ₃ to t ₅ ) by the retrace or flyback pulse (see waveform E) which should not trigger it.

At time instant t ₃ , the switching means 3 is opened (i ₄ and i ₅ are both zero -- see waveforms B and C) and the reactive circuit 8, 9 forms a loop through capacitor 2 and the deflection coil 1 and thus a series resonant circuit including (L ₁ + L ₈ ) and C ₉ , C ₂ being of high value and representing a short circuit for the flyback frequency ##EQU2## thus obtained.

The retrace which stated at time t ₃ takes place during one half-cycle of the resonant circuit formed by reactances L ₁ , L ₈ and C ₉ , i.e. during the interval between t ₃ and t ₅ . In the middle of this interval i.e. at time instant t ₄ , both i _L1 (waveform A) and i ₈ ,9 (waveform D) pass through zero and change their sign, whereas the voltage at the terminals of the first switching means 3 (V ₃ , waveform E) passes through a maximum. Thus, from t ₄ onwards, thyristor 11 will be reverse biased and diode 12 will conduct the current from the resonant circuit 1, 8 and 9 in order to turn the second thyristor 11 off.

At time instant t ₅ , when current i _L1 has reached - I _1m and when voltage v ₃ falls to zero, diode 5 of the first switching means 3 becomes conductive and the trace portion of scan begins.

Current i ₈ ,9 nevertheless continues to flow in the resonant circuit 8, 9 through diodes 5 and 12, which causes a break to appear in waveform D at t ₅ , and a negative peak to appear in waveform D and a positive one in waveform B in the interval between t ₅ and t ₆ , these being principally due to the distributed capacities of coil 1 or to an eventual capacitor (not shown) connected in parallel to the first switching means 3.

At time instant t ₆ , diode 12 of the second switching means 10 ceases to conduct after having allowed thyristor 11 time to become turned off completely.

The level of current i ₈ ,9 at time instant t ₅ (i.e. I _c ) as well as the negative peak I _D12 in i ₈ ,9 and the positive peak I _D5 in i ₅ depend on the values of L ₈ and C ₉ in the same way as does the turn-off time of the circuit (t ₃ - t ₂ ). If, for example, L ₈ and C ₉ , are increased I _D5 increases towards zero and this could cause diode 5 to be cut off in an undesirable fashion. I _c also increases towards zero, which is liable to cause diode 12 to be blocked and thyristor 11 to trigger prematurely.

From the foregoing it can be clearly seen that the choice of values for L ₈ and C ₉ is subject to four limitations which prevent the values from being increased to lengthen the turn-off time of the driver circuit of first switching thyristor 4 so as to forestall its spurious triggering.

Waveform F shows the voltage v _G4 obtained at the gate of thyristor 4 from the secondary winding 16 coupled to the inductor 7. This voltage is positive from t ₀ to t ₁ and from t ₆ to t ₇ and is negative between t ₂ and t ₆ i.e. while the second switching means 10 is conducting.

The present invention makes the lengthening of the turn-off time of thyristor 4 possible without altering the parameters of the circuit such as inductance 8 and capacitor 9.

In the circuit shown in FIG. 3, which illustrates the principle of the present invention, means are added to the circuit in FIG. 1 which enable the turn-off time to be lengthened by connecting a load to diode 5 so as to increase the current which flows through it during the time that it is conductive. These means are here formed by a resistor 18 connected in parallel with a capacitor 20 (which replaces capacitor 2) which is of a higher capacitance so that, in practice, it holds its charge during at least one half of the line period. FIG. 4, which shows the waveform of the current in the first switching means 3 for a circuit as shown in FIG. 3, makes it possible to explain how this lenthening of the turn-off time is achieved.

In FIG. 4, the broken lines show the waveform of the current in the first switch device 3 in the circuit of FIG. 1, this waveform being produced by adding waveforms B and C of FIG. 2. The current i ₄ above the axis flows through thyristor 4 and current i ₅ below the axis flows through diode 5. When the capacitance C ₂₀ of the capacitor in series with the deflector coil is increased to some tens of microfarads (C ₂ having been of the order of 1 μ F) and when there is connected in parallel with capacitor 20 a resistor 18 the value of which is calculated to draw a strong current I _R18 from capacitor 20, that is to say a current at least equal to 0,1 I _m (I _m being of the order of some tens of amperes), current I _R18 is added to that i ₅ which flows through diode 5 without in any way altering the linearity of the trace portion nor the oscillatory commutation of thyristor 4 which is brought about by the resonant circuit L ₈ , C ₉ .

The fact of loading capacitor C ₂₀ by means of a resistor 18 thus has the effect of permanently displacing the waveform of the current in the negative direction by I _R18 . Thus, during the trace portion of the scan, the transfer of the current from the diode 5 to the thyristor 4 begins at time t ₁₀ instead of t ₀ , that is to say with a delay proportional to I _R18 . The effect of the triggering pulse delivered by the horizontal oscillator (13 FIG. 1) to the second thyristor 11 at time instant t ₁ , will be to start the commutation process of the first thyristor 4 when the current it draws is less by I _R18 than that i ₄ (t ₁ ) which it would have been drawing had there been no resistor 18. Because of this, the turn-off time of the thyristor 4 proper, which as has been mentioned increases with the maximum current level passing throught it, is slightly reduced. Moreover, because the oscillatory current i ₈ ,9 (FIG. 2) from circuit L ₈ , C ₉ which flows through thyristor 4 in the opposite direction is unchanged, it reaches a value equal to that of the current i _L1 (FIG. 1) flowing in the coil 1 in a shorter time, that is to say at time t ₁₂ . Diode 5 will thus take the oscillatory current i ₈ ,9 (FIG. 2) over in advance with respect ro time instant t ₂ and will conduct it until it reaches zero value at a time instant t ₁₃ later than t ₃ , the amounts of advance (t ₂ - t ₁₂ ) and delay (t ₁₃ - t ₃ ) being practically equal.

It can thus be seen in FIG. 4 that the circuit turn-off time T _R of a circuit according to the invention and illustrated by FIG. 3 is distinctly longer than that T _r of the circuit in FIG. 1. This increase in the turn-off time (T _R - T _r ) depends on the current I _R18 and increases therewith.

It should be noted at this point that the current I _R18 produces a voltage drop at the terminals of the resistor the only effect of which is to heat up the resistor since the level of this voltage (40 to 60 volts) does not necessarily have a suitable value to be used as a voltage supply for other circuits in an existing transistorised television receiver.

In accordance with one embodiment of the invention, illustrated in FIG. 5, an application is proposed for the additional current which is to be drawn through diode 5. In FIG. 5, the positive terminal of capacitor 20 is connected by a conductor 19 to the negative pole of the power supply 6 and the voltage at the terminals of capacitor 20 is thus added to that E from the source 6.

In the preferred embodiment of the present invention, which is shown in FIG. 6, it is possible to cause a supplementary current of a desired value to flow through the first diode 5 while obtaining a voltage which has a suitable value for use in another circuit in the television receiver.

If the voltage at the terminals of capacitor 20 in FIG. 3 is not a usable value, it is possible to connect in parallel with the series circuit comprising the deflector coil 1 and the capacitor 2 in FIG. 1, i.e. in parallel with the terminals of the first switching means 3, a series combination of an autotransformer 21 and a high value capacitor 22 (comparable with capacitor 20 in FIGS. 3 and 5). The autotransformer 21 has a tap 23 is suitably positioned between the terminal connected to capacitor 22 at the tap 24 connected to the first switching means 3. This autotransformer 21 may be formed by the one conventionally used for supplying a very high voltage to the cathode ray tube, as described for example in U.S. Pat. No. 3,452,244; such a transformer comprises a voltage step-up winding between taps 24 and 25, which latter is connected to a high voltage rectifier (not shown).

The waveform of the voltage at the various points in the autotransformer is shown in FIG. 7, in which waveform A shows the voltage at the terminals of capacitor 22, waveform B the voltage at tap 24 and waveform C the voltage at tap 23 of the autotransformer 21.

The voltage V _c22 at the terminals of capacitor 22 varies slightly about a mean value V _cm . It is increasing while diode 5 is conducting and decreasing during the conduction of the thyristor 4.

The voltage v ₂₄ at tap 24 follows substantially the same curve as waveform E in FIG. 2, that is to say that during the retrace time interval from t ₁₃ to t ₅ to a positive pulse called the flyback pulse is produced and, during the time interval while the first switching means 3 is conducting, the voltage is zero. The mean valve of the voltage v ₂₄ at tap 24 of the auto-transformer 21 is equal to the mean value V _cm of the voltage at the terminals of capacitors 2 and 22.

Thus, there is obtained at tap 23 a waveform which is made up, during the retrace portion, of a positive pulse whose maximum amplitude is less than that of v ₂₄ at tap 24 and, during the trace portion, of a substantially constant positive voltage, the level V of which is less than the mean value V _cm of the voltage v _c22 at the terminals of capacitor 22. By moving tap 23 towards terminals 24 the amplitude of the pulse during fly-back increases while voltage V falls and conversely by moving tap 23 towards capacitor 22 voltage V increases and the amplitude of the pulse drops.

In more exact terms, the voltage V at tap 23 is such that the means value of v ₂₃ is equal to V _cm . It has thus been shown that by choosing carefully the position of tape 23, a voltage V may be obtained during the trace portion of the scan, which may be of any value between V _cm and zero.

This voltage V is thus obtained by periodically controlled rectification during the trace portion of the scan. For this purpose an electronic switch is used to periodically connect the tap 23 of trnasformer winding 21 to a load. This switch is made up of a power transistor 26 whose collector is connected to tap 23 and the emitter to a parallel combination formed by a high value filtering capacitor 27 and the load which it is desired to supply, which is represented by a resistor 28. The base of the transistor 26 receives a control voltage to block it during retrace and to unblock it

during the whole or part of the trace period. A control voltage of this type may be obtained from a second winding 29 magnetically coupled to the inductance 7 of the deflection circuit and it may be transmitted to the base of transistor 26 by means of a coupling capacitor 30 and a resistor 31 connected between the base and the emitter of transistor 26.

It may easily be seen that the DC collector/emitter current in transistor 26 flows through the first diode 5 of the first switching means 3 via a resistor 28 and the part of the winding of auto-transformer 21 located between taps 23 and 24.

Experience has shown that a circuit as shown in FIG. 6 can supply 24 volts with a current of 2 amperes to the vertical deflection circuit of the same television set, the voltage at the terminals of capacitor 22 being from 50 to 60 volts.

It should be mentioned that, when the circuit which forms the load of the controlled rectifier 26, 27 does not draw enough current to sufficiently lengthen the circuit turn-off time T _R , an additional resistor (not shown) may be connected between the emitter of transistor 26 and ground or in parallel to capacitor 22, which resistor will draw the additional current required.

GRUNDIG SUPER COLOR W6330  CHASSIS 29301-114.46(03)   Gating circuit for television SCR deflection system AND REGULATION / stabilization of horizontal deflection NETWORK CIRCUIT with Transductor reactor / Reverse thyristor energy recovery circuit.In a television deflection system employing a first SCR for coupling a deflection winding across a source of energy during a trace interval of each deflection cycle and a second SCR for replenishing energy to the source of energy during a commutation interval of each deflection cycle, a gating circuit for triggering the first SCR. The gating circuit employs a voltage divider coupled in parallel with the second SCR which develops gating signals proportional to the voltage across the second SCR.


1. In a television deflection system in which a first switching means couples a deflection winding across a source of energy during a trace interval of each deflection cycle and a second switching means replenishes energy to said source of energy during a commutation interval of each deflection cycle, a gating circuit for said first switching means, comprising:
capacitive voltage divider means coupled in parallel with said second switching means for developing gating signals proportional to the voltage across said second switching means; and
means for coupling said voltage divider means to said first switching means to provide for conduction of said first switching means in response to said gating signals.
2. A gating circuit according to claim 1 wherein said voltage divider includes first and second capacitors coupled in series and providing said gating signals at the common terminal of said capacitors. 3. A gating circuit according to claim 2 wherein said first and second capacitors are proportional in value to provide for the desired magnitude of gating signals. 4. A gating circuit according to claim 3 wherein said means for coupling said voltage divider means to said first switching means includes an inductor. 5. A gating circuit according to claim 4 wherein said inductor and said first and second capacitors comprise a resonant circuit having a resonant frequency chosen to shape said gating signal to improve switching of said first switching means.

Description:

BACKGROUND OF THE INVENTION
This invention relates to a gating circuit for controlling a switching device employed in a deflection circuit of a television receiver.

Various deflection system designs have been utilized in television receivers. One design employing two bidirectional conducting switches and utilizing SCR's (thyristors) as part of the switches is disclosed in U.S. Pat. No. 3,452,244. In this type deflection system, a first SCR is

employed for coupling a deflection winding across a source of energy during a trace interval of each deflection cycle, and a second SCR is employed for replenishing energy during a commutation interval of each deflection cycle. The first SCR is commonly provided with gating voltage by means of a separate winding or tap of an input reactor coupling a source of B+ to the second SCR.

Various regulator system designs have been utilized in conjunction with the afore described deflection system to provide for uniform high voltage production as well as uniform picture width with varying line voltage and kinescope beam current conditions.
One type regulator system design alters the amount of energy stored in a commutating capacitor coupled between the first and second SCR's during the commutating interval. A regulator design of this type may employ a regulating SCR and diode for coupling the input reactor to the source of B+. With this type regulator a notch, the width of which depends upon the regulation requirements, is created in the current supplied through the reactor and which notch shows up in the voltage waveform developed on the separate winding or tap of the input reactor which provides the gating voltage for the first SCR. The presence of the notch, even though de-emphasized by a waveshaping circuit coupling the gating voltage to the first SCR, causes erratic control of the first SCR.
SUMMARY OF THE INVENTION
In accordance with one embodiment of the invention, a gating circuit of a television deflection system employing a first switching means for coupling a deflection winding across a source of energy during a trace interval of each deflection cycle and a second switching means for replenishing energy to said source of energy during a commutation interval of each deflection cycle includes a voltage divider means coupled in parallel with the second switching means for developing gating signals proportional to the voltage across the second switching means. The voltage divider means are coupled to the first switching means to provide for conduction of the first switching means in response to the gating signals.
A more detailed description of a preferred embodiment of the invention is given in the following description and accompanying drawing of which:
FIG. 1 is a schematic diagram, partially in block form, of a prior art SCR deflection system;
FIG. 2 is a schematic diagram, partially in block form, of an SCR deflection system of the type shown in FIG. 1 including a gating circuit embodying the invention;
FIG. 3 is a schematic diagram, partially in block form, of one type of a regulator system which employs an SCR as a control device and which is suitable for use with the SCR deflection system of FIG.2;
FIG. 4 is a schematic diagram, partially in block form, of another type of a regulator system suitable for use with the deflection circuit of FIG. 2; and
FIG. 5 is a schematic diagram, partially in block form, of still another type of a regulator system suitable for use with the SCR deflection system of FIG. 2.
DESCRIPTION OF THE PREFERRED EMBODIMENT
FIG. 1 is a schematic diagram, partially in block form, of a prior art deflection system of the retrace driven type similar to that disclosed in U.S. Pat. No. 3,452,244. This system includes a commutating switch 12, comprising a silicon controlled rectifier (SCR) 14 and an oppositely poled damper diode 16. The commutating switch 12 is coupled between a winding 18a of an input choke 18 and ground. The other terminal of winding 18a is coupled to a source of direct current voltage (B+) by means of a regulator network 20 which controls the energy stored in the deflection circuit 10 when the commutating switch is off, during an interval T3 to T0' as shown in curve 21 which is a plot of the voltage level at the anode of SCR 14 during the deflection cycle. A damping network comprising a series combination of a resistor 22 and a capacitor 23 is coupled in parallel with commutating switch 12 and serves to reduce any ringing effects produced by the switching of commutating switch 12. Commutating switch 12 is coupled through a commutating coil 24, a commutating capacitor 25 and a trace switch 26 to ground. Trace switch 26 comprises an SCR 28 and an oppositely poled damper diode 30. An auxiliary capacitor 32 is coupled between the junction of coil 24 and capacitor 25 and ground. A series combination of a horizontal deflection winding 34 and an S-shaping capacitor 36 are coupled in parallel with trace switch 26. Also, a series combination of a primary winding 38a of a horizontal output transformer 38 and a DC blocking capacitor 40 are coupled in parallel with trace switch 26.
A secondary of high voltage winding 38b of transformer 38 produces relatively large amplitude flyback pulses during the retrace interval of each deflection cycle. This interval exists between T1 and T2 of curve 41 which is a plot of the current through windings 34 and 38a during the deflection cycle. These flyback pulses are applied to a high voltage multiplier (not shown) or other suitable means for producing direct current high voltage for use as the ultor voltage of a kinescope (not shown).
An auxiliary winding 38c of transformer 38 is coupled to a high voltage sensing and control circuit 42 which transforms the level of flyback pulses into a pulse width modulated signal. The control circuit 42 is coupled to the regulator network 20.
A horizontal oscillator 44 is coupled to the gate electrode of commutating SCR 14 and produces a pulse during each deflection cycle slightly before the end of the trace interval at T0 of curve 21 to turn on SCR 14 to initiate the commutating interval. The commutating interval occurs between T0 and T3 of curve 21. A resonant waveshaping network 46 comprising a series combination of a capacitor 48 and an inductor 50 coupled between a winding 18b of input choke 18 and the gate electrode of trace SCR 28 and a damping resistor 52 coupled between the junction of capacitor 48 and inductor 50 and ground shapes the signal developed at winding 18b (i.e. voltage waveform 53) to form a gating signal voltage waveform 55 to enable SCR 28 for conduction during the second half of the trace interval occurring between T2 and T1' of curve 41.
The regulator network 20, when of a type to be described in conjunction with FIG. 3, operates in such a manner that current through winding 18a of input choke 18 during an interval between T4 and T5 (region A) of curves 21, 53 and 55 is interrupted for a period of time the duration of which is determined by the signal produced by the high voltage sensing and control circuit 42. During the interruption of current through winding 18a a zero voltage level is developed by winding 18b as shown in interval T4 to T5 of curve 53. The resonant waveshaping circuit 46 produces the shaped waveform 55 which undesirably retains a slump in region A corresponding to the notch A of waveform 53. The slump in waveform 55 applied to SCR 28 occurs in a region where the anode of SCR 28 becomes positive and where SCR 28 must be switched on to maintain a uniform production of the current waveshape in the horizontal deflection winding 34 as shown in curve 41. The less positive amplitude current occurring at region A of waveform 55 may result in insufficient gating current for SCR 28 and may cause erratic performance resulting in an unsatisfactory raster.
FIG. 2 is a schematic diagram, partially in block form, of a deflection system 60 embodying the invention. Those elements which perform the same function in FIG. 2 as in FIG. 1 are labeled with the same reference numerals. FIG. 2 differs from FIG. 1 essentially in that the signal to enable SCR 28 derived from sampling a portion of the voltage across commutating switch 12 rather than a voltage developed by winding 18b which is a function of the voltage across winding 18a of input choke 18 as in FIG. 1. This change eliminates the slump in the enabling signal during the interval T4 to T5 as shown in curve 64 since the voltage across the commutating switch 12 is not adversely effected by the regulator network 20 operation.
A series combination of resistor 22, capacitor 23 and a capacitor 62 is coupled in parallel with commutating switch 12, one terminal of capacitor 62 being coupled to ground. The junction of capacitors 23 and 62 is coupled to the gate electrode of SCR 28 by means of the inductor 50. The resistor 52 is coupled in parallel with capacitor 62.
Capacitors 23 and 62 form a capacitance voltage divider which provides a suitable portion of the voltage across commutating switch 12 for gating SCR 28 via inductor 50. The magnitude of the voltage at the junction of capacitors 23 and 62 is typically 25 to 35 volts. It can, therefore, be seen that the ratio of values of capacitors 23 and 62 will vary depending on the B+ voltage utilized to energize the deflection system. Capacitors 23 and 62 and inductor 50 form a resonant circuit tuned in a manner which provides for peaking of the curve 64 between T4 and T5. This peaking effect further enhances gating of SCR 28 between T4 and T5.
Since the waveshape of the voltage across commutating switch 12 (curve 21) is relatively independent of the type of regulator system employed in conjunction with the deflection system, the curve 64 also is independent of the type of regulator system.
When commutating switch 12 switches off during the interval T3 to T0' curve 21, the voltage across capacitor 62 increases and the voltage at the gate electrode of SCR 28 increases as shown in curve 64. As will be noted, no slump of curve 64 occurs between T3 and T5 because there is no interruption of the voltage across commutating switch 12.



















FIG. 3 is a schematic diagram, partially in block form, of one type of a regulator system which may be used in conjunction with the invention. B+ is supplied through a regulator network 20 which comprises an SCR 66 and an oppositely poled diode 68. The diode is poled to provide for conduction of current from B+ to the horizontal deflection circuit 60 via winding 18a of input choke 18. Current flows through the diode during the period T3 to T4 of curve 21 FIG. 1 after which current tries to flow through the SCR 66 from the horizontal deflection circuit to B+ since the commutating capacitor 25 is charged to a voltage higher than B+.
The horizontal deflection circuit 60 produces a flyback pulse in winding 38a of the flyback transformer 38 which is coupled to winding 38c. The magnitude of the pulse on winding 38c determines how long the signal required to switch SCR 66 on is delayed after T4 curve 21 FIG. 1. If the flyback pulse is greater than desirable, the SCR 66 turns on sooner than if the flyback pulse is less than desirable and provides a discharge path for current in commutating capacitor 25 back to the B+ supply. In this manner a relatively constant amplitude flyback pulse is maintained.
FIG. 4 is a schematic diagram, partially in block form, of another well-known type of a regulator system which may be used in conjunction with the invention shown in FIG. 2. B+ is coupled through winding 18a of input choke 18 and through a series combination of windings 70a and 70b of a saturable reactor 70 and a parallel combination of a diode 72 and a resistor 74 to the horizontal deflection circuit 60. Diode 72 is poled to conduct current from the horizontal deflection circuit 60 to B+.
Flyback pulse variations are obtained from winding 38c of the horizontal output transformer 38 and applied to a voltage divider comprising resistors 76, 78 and 80 of the high voltage sensing and control circuit 42. A portion of the pulse produced by winding 38c is selected by the position of the wiper terminal on potentiometer 78 and coupled to the base electrode of a transistor 82 by means of a zener diode 84. The emitter electrode of transistor 82 is grounded and a DC stabilization resistor 85 is coupled in parallel with the base-emitter junction of transistor 82. When the pulse magnitude on winding 38c exceeds a level which results in forward biasing the base-emitter junction of transistor 82, current flows from B+ through a resistor 86, a winding 70c of saturable reactor 70 and transistor 82 to ground. Due to the exponential increase of current in winding 70c during the period of conduction of transistor 82, the duration of conduction of transistor 82 determines the magnitude of current flowing in winding 70c and thus the total inductance of windings 70a and 70b. The current in winding 70c is sustained during the remaining deflection period by means of a diode 88 coupled in parallel with winding 70c and poled not to conduct current from B+ to the collector electrode of transistor 82. A capacitor 90 coupled to the cathode of diode 88 provides a bypass for B+. Windings 70a and 70b are in parallel with input reactor 18a and thereby affect the total input inductance of the deflection circuit and thereby controls the transfer of energy to the deflection circuit. The dotted waveforms shown in conjunction with a curve 21' indicate variations from a nominal waveform provided at the input of horizontal deflection circuit 60 by the windings 70a and 70b.













FIG. 5 is a schematic diagram of yet another type of a regulator system which may be used in conjunction with the invention. B+ is coupled through a winding 92a and a winding 92b of a saturable reactor to the horizontal deflection circuit 60. Windings 92a and 92b are used to replace the input choke 18 shown in FIGS. 1 and 2 while also providing for a regulating function corresponding to that provided by regulating network 20.
Flyback pulse variations are obtained from winding 38c and applied to the high voltage sensing and control circuit 42 as in FIG. 4. Current flows from B+ through resistor 86, a winding 92c and transistor 82 to ground. As in FIG. 4 the duration of the conduction of transistor 82 determines the energy stored in winding 92c and thus the total inductance of windings 92a and 92b which control the amount of energy transferred to the deflection circuit during each horizontal deflection cycle. The variations in waveforms of curve 21', shown in conjunction with FIG. 4, are also provided at the input of horizontal deflection circuit 60 by windings 92a and 92b.
For various reasons including cost or performance, a manufacturer may wish to utilize a particular one of the regulators illustrated in FIGS. 3, 4 and 5. Regardless of the choice, the gating circuit according to the invention may be utilized therewith advantageously by providing improved performance and the possibility of cost savings by eliminating taps or extra windings on the wound components which heretofore normally provided a source of SCR gating waveforms.

GRUNDIG SUPER COLOR W6330  CHASSIS 29301-114.46(03)  UNITS:
 

- NF Baustein / Audio Amplifier: 29301-004.01 with TBA800

- Horizontal-Baustein / Line Osc + Synch : 29301-008.02 with TBA920

- Sicherung Bst / Safety Unit 29301-410.01

- OW Dioden Baustein / E/W Correction diode Mod. Unit: 29301-041.01

- Regelbaustein / regulator unit: 20301-035.01



VERTIKAL BAUSTEIN/ FRAME DEFLECTION OSC DRIVER : 29301-009.03
















TBA920 line oscillator combination
DESCRIPTION
The line oscillator combination TBA920 is a monolithic
integrated circuit intended for the horizontal deflection of the black and white
and colour TV sets
picture tube.

FEATURES:
SYNC-PULSE SEPARATION
OPTIONAL NOISE INVERSION
GENERATION OF A LINE FREQUENCY VOL-
TAGE BY MEANS OF AN OSCILLATOR
PHASE COMPARISON BETWEEN SYNC-
PULSE AND THE OSCILLATOR WAVEFORM
PHASE COMPARISON BETWEEN THE OS-
CILLATOR WAVEFORM AND THE MIDDLE OF
THE LINE FLY-BACK PULSE
AUTOMATIC SWITCHING OF THE VARIABLE
TRANSCONDUCTANCE AND THE VARIABLE
TIME CONSTANT TO ACHIEVE NOISE SUP-
PRESSION AND, BY SWITCHING OFF, POS-
SIBILITY OF TAPE-VIDEO-REGISTERED RE-
PRODUCTION
SHAPING AND AMPLIFICATION OF THE OS-
CILLATOR WAVEFORM TO OBTAIN PULSES
FOR THE CONTROL OF DRIVING STAGES IN
HORIZONTAL, DEFLECTION CIRCUITS
USING EITHER TRANSISTORS OR THYRISTORS.
 THE TBA920 SYNC/TIMEBASE IC It has been quite common for some time for sync separation to be carried out in an i.c. but until 1971 this was as far as i.c.s had gone in television receiver timebase circuitry. With the recent introduction of the delta featured 110°  colour series however i.c.s have gone a step farther since this chassis uses a TBA920 as sync separator and line generator. A block diagram of this PHILIPS /Mullard  i.c. is shown in Fig. 1.
The video signal at about 2-7V peak -peak is fed to the sync separator section at pin 8, the composite sync waveform appearing at pin 7.
The noise gate switches off the sync separator when a positive -going input pulse is fed in at pin 9, an external noise limiter circuit being required .
The line sync pulses are shaped by R1 /C1 /C2/R2 and fed in to the oscillator phase detector section at pin 6.
The line oscillator waveform is fed internally to the oscillator phase detector circuit which produces at pin 12 a d.c. potential which is used to lock the line oscillator to the sync pulse frequency, the control potential being fed in at pin 15. The oscillator itself is a CR type whose waveform is produced by the charge and discharge of the external capacitor (C7) connected to pin 14. The oscillator frequency is set basically by C7 and R6 and can be varied by the control potential appearing at pin 15 from pin 12 and the external line hold control. Internally the line oscillator feeds a triangular waveform to the oscillator and flyback phase detector sections and the pulse width control section. The coincidence detector section is used to set the time constant of the oscillator phase detector circuit. It is fed internally with sync pulses from the sync separator section, and with line flyback pulses via pin 5. When the flyback pulses are out of phase with the sync pulses the impedance looking into pin 11 is high (21(Q). When the pulses are coincident the impedance falls to about 150Q and the oscillator phase detector circuit is then slow acting. The effect of this is to give fast pull -in when the pulses are out of sync and good noise immunity when they are in sync. The coincidence detector is controlled by the voltage on pin 10. When the sync and flyback pulses are in sync C3 is charged: when they are out of sync C3 discharges via R3. VTR use has been taken into consideration here. With a video recorder it is necessary to be able to follow the sync pulse phase variations that occur as a result of wow and flutter in the tape transport system, while noise is much less of a problem. For use with a VTR therefore the network on pin 10 can simply be left out so that the oscillator phase detector circuit is always fast acting. A second control loop is used to adjust the timing of the pulse output obtained from pin 2 to take into account the delay in the line output stage. The fly back phase detector compares the frequency of the flyback pulses fed in at pin 5 with the oscillator signal which has already been synchronised to the sync pulse frequency.
Any phase difference results in an output from pin 4 which is integrated and fed into the pulse width control section at pin 3. The potential at pin 3 sets the width of the output pulse obtained at pin 2: with a high positive voltage (via R11 and R12) at pin 3 a 1:1 mark -space ratio out- put pulse (32/us on, 32/us off) will be produced while a low potential at pin 3 (negative output at pin 4) will give a 16us output pulse at  the same frequency. The action of this control loop continues until the fly- back pulses are in phase with a fixed point on the oscillator waveform: the flyback pulses are then in phase with the sync pulses and delays in the line output stage are compensated. The output obtained at pin 2 is of low impedance and is suitable for driving valves, transistors or thyristors: R9 is necessary to provide current limiting.


 
 THE TBA800, TBA810 AUDIO integrated circuits:
AUDIO integrated circuits are being increasingly used in television chassis and certainly represent the simplest approach to improving the audio side of a TV set. A number of such i.c.s have appeared during the 70's.

Here describes the use of two fairly recent ones, the SGS-ATES TBA800 and TBA8I0S. Both devices can provide reasonably high outputs into a suitable loudspeaker-the TBA800 will give up to 5W and the TBA810S up to 7W.
The main difference between them being that the TBA800 is a somewhat higher voltage, lower current device. The TBA800 is used in the current Grundig and ASA 110° colour chassis while the Finlux 110' colour chassis uses a TBA810. In each of these chassis the audio i.c. is driven from a TBA120 intercarrier sound i.c. The TBA800 and TBA810S can also be used as the field output stage in 110' monochrome chassis with c.r.t.s of up to l7in. and as the field driver stage in larger screen monochrome sets.
The TBA800 is designed to provide up to 5W into a 16 Ohm load when operated from a 24V supply. It is encapsulated in the type cf quad -in -line case shown in Fig. I: the tabs at the centre are to assist in cooling the device and must be earthed. The TBA800 can be operated from power supply voltages up to the absolute maximum permissible value of 30V. It is best to regard 24V as being the upper limit however in order to provide an adequate safety margin and prevent possible damage during voltage surges. The minimum power supply voltage recommended by the manufacturers is 5V, but the power output is then less than 0-5W. The quiescent current taken by the TBA800 is typically 9mA from a 24V supply-no device of this type should draw more than 20mA. When an input signal is applied the current increases considerably- up to about 1.5A at full power. Two circuits for use with the TBA800 are shown in Figs. 2 and 3 and give comparable performance. The circuit shown in Fig. 2 is somewhat simpler but that
shown in Fig. 3 enables one side of the loudspeaker to be connected to chassis. The input resistance of the TBA800 is quite high (typically 5 MOhm) but a resistor must be connected between the input pin 8 and chassis otherwise the out- put stage will not operate with the correct bias. In the circuits shown the volume control VR1 provides this function: the bias current that flows through it is typically 1 microA (maximum 5 microA). The average voltage at the output pin 12 is half the supply potential. The loudspeaker must be capacitively coupled therefore and the low frequency response will be worse as this capacitor is decreased in value. The output coupling capacitor C4 in Fig. 2 also provides the bootstrap connection to pin 4. In Fig. 3 an additional capacitor (C9) is required for this purpose.
In both circuits the value of R1 controls the amount of feedback and thus the gain. The output signal is fed back to pin 6 via an internal 7 kOhm resistor. If R1 is reduced in value the gain will increase but the frequency response will be affected and the distortion will rise. With the component values shown the voltage gain of both circuits is typically 140 (43dB) which is quite adequate for most audio applications. R3 in Fig. 3 is necessary only if the power supply voltage is fairly low (less than about 14V).
C2 smooths the power supply input and C1 is connected between pin 1 and chassis to provide r.f. decoupling and help prevent instability. If mains hum is present on the supply line with the circuit shown in Fig. 3 capacitor C8 should be included between pin 7 and chassis. The circuits shown have a level frequency response (within ±3dB) between about 40Hz and 20kHz. If you wish to reduce the upper 3dB level to about 8kHz C5 can be increased to about 560pF. The total harmonic distortion provided by these circuits remains fairly constant at about 0.5% until the power output reaches 3W: it then rises rapidly with power level as shown in Fig. 4.
The TBA800 can be operated from a 13V supply to feed up to 2.5W into an 80 load or from a 17V supply to feed the same power into a 160 load without an additional heatsink. If more output power is required the cooling tabs must be connected to a heatsink. Two methods of mounting the TBA800 are shown in Figs. 5 and 6. In Fig. 5 the device is inserted into a circuit board and a heatsink is soldered to the same points as the tabs: this has the disadvantage that the heatsink extends above the board though on the other hand the whole board can be used for the construction of the circuit. In Fig. 6 the tabs are soldered directly to a suitable area of copper on the board: this method has the disadvantage that about two square inches of the board are not available for component mounting. It is generally best to make soldered connections to the pins of the device since this ensures good heat dissipation with minimum unwanted feedback. Observe the usual heat precautions when soldering. The pins can however be carefully bent so that they will fit into a 16 -pin dual -in -line socket.
The TBA810S has the same type of encapsulation as the TBA800 and the connections are also as shown in Fig. 1 except that there is no internal connection to pin 3. An alternative version, the TBA810AS, has two horizontal tabs with a hole in each (see Fig. 7) so that a heatsink can be bolted on. Some readers may find it easier to bolt a heatsink to a TBA810AS than to solder the TBA810S tabs. TBA810 devices can provide 7W of audio power to a 40 loudspeaker when operated from a I6V supply. Fig. 8 shows the change in maximum output power with different supply voltages. As a 4.5W output can be obtained with a 12V supply the TBA810 is much more suitable than the TBA800 for use with battery operated equipment. The TBA810 can provide output currents up to 2.5A.
Two circuits for use with TBA810 devices are shown in Figs. 9 and 10: they are very similar to the circuits shown in Figs. 2 and 3 though some of the capacitor values are larger because of the lower output impedance. The two circuits have comparable performance but that shown in Fig. 10 gives somewhat better results at low supply voltages (down to 4V). In either circuit R2 may be replaced with a 100k0 volume control. The bias current flowing in the pin 8 circuit is typically
0-4 microA and the input resistance 5M 0 (the value of R2 must be much less however to ensure correct bias.
 The gain decreases as the value of R1 is increased for the same reason as with the TBA800. The values of R1, C3 and C7 affect the high -frequency response. With the values shown the response is level within ±3dB from about 40Hz to nearly 20kHz. Fig. 11 shows values of C3 plotted against R1 where the frequency is 3dB down at 10kHz and 20kHz and C7 is five times C3. The output distortion with these circuits is about 0.3% for outputs up to 3W rising to about 1% at 4W, 3% at 5W and 9% at 6W with a 14.4V supply voltage. The voltage gain is typically 70 times (37dB). Although this value is half that obtained with the TBA800 the input voltage required to produce a given output power is about the same for both types. This is because a smaller output voltage is required to drive a 40 load at a certain power level than is required to drive a 160 load.
The TBA810S may be mounted in the same way as the TBA800. One way of mounting the TBA810AS is shown in Fig. 12. It is simpler however to bolt flat heatsinks to the tabs.
Devices of this type will be destroyed within a fraction of a second if the power supply is accidentally con- nected with reversed polarity. When experimenting therefore it is wise to include a diode in the positive power supply line to prevent any appreciable reverse current flowing in the event of incorrect power supply connection. The diode can be removed once the circuit has been finalised. The TBA800 is likely to be destroyed if the output is accidentally shorted to chassis. The TBA810S and TBA810AS however are protected from damage in the event of such a short-circuit even if this remains for a long time (but note that the earlier TBA8I0 and  TBA810A versions did not contain internal circuitry to provide this protection). The TBA800 is not protected against overheating but the TBA810S and TBA810AS incorporate a thermal shutdown circuit.
For this reason the heat- sinks used with the TBA810S and TBA810AS can have a smaller safety factor than those used with the TBA800. If the silicon chip in a TBA810S or TBA810AS becomes too hot the output power is temporarily reduced by the internal thermal shutdown circuit. As with all high -gain amplifiers great care should be taken to keep the input and output circuits well separated otherwise oscillation could occur. The de- coupling capacitors should be soldered close to the i.c. -especially the 0 1pF decoupling capacitor in the supply line (this should be close to pin I).
Field Output Circuit:
 Fig. 13 shows a suggested field output stage for monochrome receivers with 12-17in. 110° c.r.t.s using the TBA81OS. For safe working up to 50°C ambient temperature each tab of the device must be soldered to a square inch of copper on the board. The peak -to - peak scanning current is 1.5A, the power delivered to the scan coils 0.47W, power disspipation in the TBA810S 1 8W, scan signal amplitude 4.1V, flyback amplitude 5V and the maximum peak -to -peak current available in the coils 1.75A.

TDA2521 synchronous demodulator for PAL

GENERAL DESCRIPTION
The TDA2521 is a monolithic integrated circuit designed as a synchronous demodulator for PAL color television receivers. It includes an 8.8 MHz oscillator and divider, to generate two 4.4 MHz reference signals, and provides color difference output.
The TDA2521 is intended to interface directly with the TDA251O with a minimum of external components and is constructed on a single silicon chip using the Fairchild Planar
epitaxial process.

ABSOLUTE MAXIMUM RATINGS
Supply Voltage 14 V
Internal Power Dissipation 600 mW ORDER INFQRMATIQN
Operating Temperature Range —2O°C to +6O°C TYPE PART NO.
Storage Temperature Range —55°C to +125°C 2521 TDA2521
Pin Temperature iSo|dering 10 si 260°C

Planar is a patented Fairchild process













TDA2510 CHROMINANCE COMBINATION

GENERAL DESCRIPTION —
The TDA2510 is a monolithic integrated circuit designed for the function of a color television receiver. It Is designed to Interface directly with the TDA2521, using a minimum number of external components.
TDA251O is constructed on a single silicon chip using the Fairchild Planar‘ epitaxial process.

ABSOLUTE MAXIMUM RATINGS

supply Voltage 15 V
Collector voltage of chroma output transistor (pin 7) 20 V
(PD I 100 mW max)
Collector current of chroma output transistor (pin 7) 20 mA
Collector current of color killer output transistor (pin 11) 10 mA
Power dissipation 500 mW
Operating temperature range —25°C 10 +6O°
Storage temperature range *55°C to +12!-3°C

- RGB Baustein / RGB Matrix : 29301-046.01 with TCA660 TBA530 (PHILIPS)

TEXAS INSTRUMENTS Solid state television channel selection system:
GRUNDIG Tuning Memory / Suchlauf Abstimmung overview.
GRUNDIG SUPER COLOR W6330  CHASSIS 29301-114.46(03)  HERE below the description of the technology of the Texas Instruments Incorporated Tuning system used in GRUNDIG Super Color television Tuning circuits (STATIONS SUCHLAUF / STATION MEMORY) , and discloses various embodiments of solid state television channel selection systems.

It's shown and explained the theory on which is based all the technology of the SUCHLAUF BAUSTEIN / STATION MEMORY 29301-045.13 (SEARCH UNIT) BASED AROUND TEXAS INSTRUMENTS CHIPSET and fitted in GRUNDIG SUPER COLOR CHASSIS DESIGN IN THE 70's.








An automatic TV tuner circuit responsive to a tuning voltage for tuning over a desired frequency reception range. The tuning voltage is generated by a staircase voltage generator. A discriminator produces an output signal having a nominal value when the receiver is tuned to a transmitted broadcast frequency and at a discriminator output having a predetermined difference from the nominal value, a sequence control circuit halts the operation of the staircase generator and regulates the tuning voltage in a continuous manner under control of the discriminator output signal to complete the tuning process.
1. A circuit for generating a tuning voltage for tuning a signal receiver having an input stage tunable over a desired reception frequency range by means of the tuning voltage and which includes discriminator means for producing an output signal having a selected value when the receiver is tuned to a nominal frequency; said circuit including staircase voltage generator means for generating a stepwise increasing tuning voltage for application to said input stage; frequency control circuit means operably connectable to said staircase voltage generator means for adjusting the output voltage of said staircase voltage generator in dependence on said discriminator means output signal in a sense to tune said receiver to said nominal frequency; sequence control circuit means for stopping stepping operation of said staircase voltage generator means when the value of said discriminator output signal has a predetermined difference from said selected value and for effecting said operable connection of said frequency control circuit means when stepping operation

of said staircase voltage generator means has been stopped by said sequence control circuit means; and wherein said frequency control circuit means has a control range at least sufficiently wide to include tuning frequencies attained in successive steps of said staircase voltage.

2. A circuit according to claim 1, wherein said staircase voltage generator means includes scanning counter means for continuously sweeping its counting cycle, information counter means fed with stepping pulses by said sequence control circuit means, and comparator means for comparing the counts of said scanning and information counter means to produce an output voltage as long as the count of said scanning counter means is lower than the count of said information counter means, said output pulses having a duration which increases with the count of the information counter means and which remains constant when stepping operation of said staircase voltage generator means has been stopped; and means connecting the output of said comparator means to the input of integrator means for integrating the output voltage of said comparator means to produce said stepwise increasing tuning voltage, said tuning voltage having a level which depends on the existing count of the information counter when stepping operation of said staircase voltage generator means has been stopped.

3. A circuit according to claim 2, further including storage means for storing the count of said information counter means when said staircase voltage generator has been stopped by said sequence control circuit means and for subsequent re-entry of said count into said information counter.

4. In a television broadcast receiver including means responsive to received video signals to produce sync pulses and line flyback pulses, an input stage tunable over a desired reception frequency range by a tuning voltage applied to a tuning input of said input stage; staircase voltage generator means for generating a stepwise increasing voltage for application to said tuning input; and discriminator means for producing an output signal, said output signal having a predetermined value when said receiver is tuned to a nominal broadcast frequency, the improvement characterized by:
frequency control means connected to receive said discriminator output signal and responsive to enablement for continuously adjusting the output voltage from said staircase voltage generator, said frequency control circuit means having a control range sufficiently wide to include at least the tuning frequencies attained in successive steps of said staircase voltage;
sequence control circuit means including control logic means for stopping stepping operation of said staircase voltage generator means in response to a selected difference of said discriminator output signal from said predetermined value and, when enabled, for enabling said frequency control circuit means until said discriminator output signal has said predetermined value;
coincidence circuit means having an output connected to said control logic means and responsive to coincidence between said sync pulses and said line flyback pulses to effect said enablement of said control logic means, and responsive to non-coincidence between said sync pulses and said line flyback pulses to cause said control logic means to restart operation of said staircase voltage generator means.


5. A circuit according to claim 4, wherein said coincidence circuit includes a capacitor chargeable to a predetermined value to indicate presence of said coincidence, and a transistor circuit responsive to coincidence of said sync and line flyback pulses to pass charging current to said capacitor.

6. A circuit according to claim 4, wherein said coincidence circuit includes a capacitor chargeable to a predetermined value to indicate presence of said coincidence, a first transistor for passing charging current to said capacitor, a second transistor shunting a control input of said first transistor and switchable to an off condition in the presence of a sync pulse, and a third transistor shunting said control input of said first transistor and switchable to an off condition in the presence of a line flyback pulse.

7. A circuit according to claim 6, including discharge transistor means connected in parallel with said charging capacitor for switching to a conductive state in response to presence of a sync pulse in the absence of a line flyback pulse.

8. In a television broadcast receiver including means responsive to received video signals to produce sync pulses and line flyback pulses, an input stage tunable over at least one desired reception frequency range by a tuning voltage applied to a tuning input of said input stage;
staircase voltage generator means for generating a stepwise increasing voltage for application to said tuning input for coarse tuning;
discriminator means responsive to received signals for producing an output signal, said output signal having a selected value when said receiver is tuned to a nominal broadcast frequency;
frequency control means responsive to enablement for adjusting the output voltage from said staircase voltage generator in a continuous manner for fine tuning until said discriminator output voltage has said predetermined value, said frequency control means having a control range sufficently wide to include at least the tuning frequencies attained in successive steps of said staircase voltage;
gating means having an output connected for enabling said frequency control means, said gating means including a switch input and an enabling input;
sequence control circuit means including control logic means having a first output connected to apply stepping pulses to said staircase voltage generator and a second output connected to the switch input of said gating means;
first trigger means responsive to a discriminator output voltage having a predetermined difference from said selected value to input said control logic means for disabling said first output;
second trigger means for inputting said control logic means to actuate said switching input of said gating means, and to confirm disablement of said first output;
coincidence circuit means responsive to coincidence of said sync pulses and said line flyback pulses to trigger said second trigger means and to actuate said enabling input of said gating means.


9. A circuit according to claim 8, wherein said staircase voltage generator means includes scanning counter means for continuously sweeping its counting cycle, information counter means, the first output of said control logic means connected for stepping said information counter, compartor means for comparing the counts of said scanning and information counter means to produce an output voltage as long as the count of said scanning counter means is lower than the count of said information counter means, said output pulses having a duration which increases with the count of the information counter means and which remains constant when stepping operation of said staircase voltage generator means has been stopped, and means connecting the output of said comparator means to the input of integrator means for integrating the output voltage of said comparator means to produce said stepwise increasing tuning voltage, said tuning voltage having a level which depends on the existing count of the information counter when stepping operation of said staircase voltage generator means has been stopped; and storage means for storing the count of said information counter means when said staircase voltage generator has been stopped by said sequence control circuit means and for subsequent re-entry of said count into said information counter.

10. A circuit according to claim 8, wherein said coincidence circuit includes a capacitor chargeable to a predetermined value to indicate presence of said coincidence, a first transistor for passing charging current to said capacitor, a second transistor shunting a control input of said first transistor and switchable to an off condition for the duration of a sync pulse, a third transistor shunting said control input of said first transistor and switchable to an off condition for the duration of a line flyback pulse, and discharge transistor means connected in parallel with said charging capacitor for switching to a conductive state in response to presence of a sync pulse in the absence of a line flyback pulse.

11. A circuit according to claim 8, wherein said frequency control means comprises an AFC circuit of said receiver.

Description:

The invention relates to a circuit arrangement including a staircase voltage generator, for generating a tuning voltage for tuning a broadcast receiver whose input stage can be tuned through by means of the tuning voltage over a desired reception frequency range and which contains a discriminator whose output signal has a given nominal value when the receiver is tuned to a nominal frequency.
In such a circuit arrangement, it has been suggested to reduce the rate of variation of the output voltage of the staircase voltage generator if the output voltage of the discriminator indicates that the setting of the input stage is approaching tuning to a transmitter radiating a nominal frequency. The rate of variation is finally reduced to zero when the input stage is tuned exactly to the transmitter frequency. As can be readily seen, the steps of the staircase voltage must be very low to ensure exact tuning to any given nominal frequency. Voltage steps that are too high could have the result that the receiver is tuned to a frequency which is still below the nominal frequency, while it becomes tuned to too high a frequency in the next following step. Generating the finely tuned staircase voltage requires relatively high circuit complexity, which is further increased by the fact that the speed at which the individual steps are traversed must be reduced when approaching the tuning to a transmitter radiating the nominal frequency.
The invention is concerned with a circuit arrangement of the indicated type such that the required circuit complexity, and thus the costs may be reduced.
In a preferred embodiment of the invention, a sequence control circuit stops the staircase voltage generator when the value of the output signal of the discriminator has a predetermined difference from the nominal value. A frequency control circuit triggered by the sequence control circuit when the staircase voltage generator is stopped, regulates the voltage produced by the stopped staircase generator in dependence on the discriminator output signal to a value tuning the receiver to the nominal frequency, the control range of the frequency control circuit being at least sufficiently wide that it includes the range between two tuning frequencies attained in successive steps of the staircase voltage.
In such a circuit arrangement embodying the invention, the operation of the staircase voltage generator is stopped when the discriminator output signal indicates the predetermined approach of the tuning to a transmitter radiating the nominal frequency. The tuning voltage generated in the stopped state of the staircase voltage generator can then be so varied by means of a frequency control circuit so that the receiver is finely tuned to the nominal frequency. The control range of the frequency control circuit is sufficiently wide that it includes the range between two tuning frequencies attained in successive steps of the staircase voltage.
The coarse tuning achieved by means of the circuit using the steps of the tuning voltage produced by the staircase voltage generator, and the subsequent fine tuning by means of the similarly acting frequency control circuit have the result that high circuit complexity is no longer required for tuning the receiver. Production costs of a circuit embodying the invention can therefore be reduced.
In an advantageous embodiment of the invention, the staircase voltage generator includes a scanning counter continuously passing through its counting cycle, and an information counter fed from the sequence control circuit with stepping pulses. A comparator compares the counts of the two counters and produces an output voltage as long as the count of the scanning counter is less than the reading of the scanning information counter. The output of the comparator is connected to the input of an integrator which integrates the output voltage of the comparator and and produces the tuning voltage at its output.
Furthermore the count of the information counter, attained when the staircase voltage generator is stopped by the sequence control circuit, can be stored and fed again into the information counter when needed. This has the result that the receiver

, by feeding an information counter count stored during its search run into the information counter, can be tuned immediately to this transmitter, without requiring a further search run. This is of particular advantage if a receiver is tuned through in its first operation in a search run over a predetermined frequency range, so that the transmitters located in this range can be determined. The counts of the information counter attained in the reception of these transmitters can be stored, so that the receiver can be tuned subsequently to a desired transmitter by feeding the stored information into the information counter.
For tuning a television receiver, the sequence control circuit embodying the invention is preferably so designed that it enables the frequency control circuit, after the operation of the staircase voltage generator has been stopped, only when there is coincidence between the sync pulses of the received video signal and line flyback pulses of the and that it starts the operation staircase voltage generator again in the absence of coincidence. This ensures that the television receiver is fine-tuned only to true picture transmitters where there is coincidence between the sync pulses and the line flyback pulses.
In circuit arrangement embodying the invention, the coincidence circuit used to determine the coincidence between the sync pulses and the line flyback pulses advantageously may contain a charging transistor which is arranged in the charging circuit of a charging capacitor, and which is conductive only when a sync pulse and a line flyback pulse appear in the coincidence circuit, a predetermined voltage value on the charging capacitor indicating the presence of coincidence.
An embodiment of the invention will now be described by way of example and in greater detail to show how the invention may be put into effect, reference being made to the accompanying drawings, in which:
FIG. 1 shows a block circuit diagram of a circuit arrangement embodying the invention,
FIG. 2 shows a circuit diagram of the coincidence circuit shown in FIG. 1.
The circuit arrangement illustrated in FIG. 1 serves to tune a television receiver to a nominal transmission frequency. The input stage of this television receiver receives over antenna 2 the RF-signal. This input stage 1 includes a conventional discriminator 3 which produces a signal having a predetermined nominal value when the receiver is tuned to the nominal frequency.
The tuning voltage for input stage 1 is supplied by a staircase voltage generator 4. This staircase voltage generator 4 includes an oscillator 5 which generates continuous output-pulses and feeds them to the stepping input 6 of a scanning counter 7. The scanning counter 7 thus sweeps its counting cycle continuously.
Furthermore, the staircase voltage generator 4 contains an information counter 8 which is operated only when required, as will be described below, and which is supplied at its stepping input 9 with pulses whose recurrence frequency is lower than the output pulses of the oscillator 5.
The counts of the scanning counter 7 and of the information counter 8 can be compared with each other in a comparator 10. This comparator 10 is so designed that it produces an output voltage at its output 11 whenever the count of the scanning counter 7 is lower than that of the information counter 8. The output voltage of the comparator 10 is integrated by an integrator 12 and fed to the input 14 of input stage 1 for tuning purposes. The integrator 12 is so designed that the value of the voltage at its output 13 depends not only on the output voltage of the comparator 10, but also on a signal fed to an input 15.
The signal at input 15 influencing the output voltage of the integrator is supplied by a frequency control circuit 16, generally also called an AFC-circuit, which in known manner produces at its output 17 a control voltage which depends on the difference between the discriminator output voltage and a fixed reference voltage. The frequency control circuit 16 can be gated or made inoperative by a switching signal fed to a gate input 18.
The operating sequence of the circuit arrangement illustrated by FIG. 1 is controlled by a sequence control circuit 19. This sequence control circuit 19 receives at the input 20 line flyback pulses of the television receiver, and its input 21 receives sync pulses from the video amplifier. At the input 22 of the sequence control circuit 19 is applied the reference voltage which is also fed to the frequency control circuit 16. At the input 23 is applied the output voltage from the discriminator 3. The sequence control circuit 19 includes a coincidence circuit 24 which triggers a Schmitt-trigger 25 to produce a coincidence signal indicating coincidence between the sync pulses and the line flyback pulses. Another Schmitt trigger 26 produces an output signal when the discriminator output voltage has a predetermined difference from the reference voltage. The output signals of the two Schmitt triggers 25 and 26 are applied to the inputs 28 and 27 of a control logic 29. This control logic 29 is so designed that it feeds the pulses derived from the scanning counter 7, which are fed to its input 30 and whose recurrence frequency is lower than that of the pulses at the stepping input 6 of the scanning counter 7, over an output 31 to the stepping input 9 of the information counter 8 and interrupts the supply of pulses when a signal from Schmitt trigger 26 appears at input 27, and the coincidence signal of Schmitt trigger 25 indicates the presence of coincidence between the line flyback pulses and the sync pulses. The supply of pulses to the stepping input 9 of the counter 8 is resumed, however, if the presence of coincidence is not indicated within a period required for checking the coincidence, by producing the coincidence signal on Schmitt trigger 25 after the pulse supply has been interrupted. But if coincidence is indicated, the control logic 29 generates at output 32 a signal which has the effect that the switching signal is applied together with the coincidence signal over an AND-circuit 33 to the enable input 18 of the frequency control circuit 16.
The counter reading attained when the supply of stepping pulses is interrupted to the input 9 of the information counter 8 can be stored in a store 34. The storage position can be selected by means of addressing inputs A, B, C, and D. This store also permits feeding of a previously stored counter reading into the information counter 8. The transmission path for the exchange of information between the information counter 8 and the store 34 is indicated schematically by the line 35. The control signals for carrying out the exchange of information are supplied over the line 36 by the control logic 29.
The manner of operation of the circuit arrangement represented in FIG. 1 is as follows. The circuit arrangement is to be used to tune the television broadcast receiver to a frequency radiated from a television transmitter. Input stages of television receivers are initially set to a certain frequency range before actual tuning to the nominal frequency within the set range. To this end the circuit arrangement represented in FIG. 1 includes three trip switches 37, 38, and 39, which, when tripped, set the input stage to the VHF-I range, the VHF-III range and the UHF range respectively. Assuming that the desired nominal frequency to which the receiver is to be tuned is in the VHF-I range, trip switch 37 is actuated. This has the result that information characterizing this range is fed into the band store 40 which effects the switching of input stage 1 to the desired range. Together with the switching, a trigger signal is applied over line 42 from band store 40 to input 43 of control logic 29, so that the latter is put into operation.
As mentioned above, the control logic 29 receives at input 30 pulses which are derived from the continuously recirculating scanning counter 7 and whose repetition frequency is lower than the repetition frequency of the stepping pulses of this counter, which are supplied by oscillator 5. The pulses at input 30 of control logic 29 are fed to the stepping input 9 of information counter 8 from band store 40, after they have been triggered by means of the signal at input 43. The count of this information counter 8 is therefore increased in synchronism with the pulses at the stepping input 9.
The counts of the scanning counter 7 and of the information counter 8 are compared continuously with each other by the comparator 10. As long as the count of the scanning counter 7 is less than the count of the information counter 8, comparator 10 produces at its output 11 an output signal with a high voltage value. If the count of the scanning counter 7 is greater than that of the information counter 8, the output signal of comparator 10 assumes the voltage value zero.
In the comparison of the readings of the continuously recirculating scanning counter 7 with that of the information counter 8 running at a lower stepping frequency, pulses appear at output 11 of comparator 10 whose duration increases with an increase of the count of the information counter 8. If we assume, for example, that the count of the information counter 8 has a low value, then the time interval in which the count of the scanning counter 7 is less than that of the information counter 8 is short. For this short time, a signal with a high voltage value is given off at output 11 of comparator 10. If the information counter 8 has a higher count, it takes longer until identity between counts of the two counters is achieved and, thus, until the signal at the comparator output 11 assumes the voltage value 0. The longest time for the high voltage produced at output 11 of comparator 10 is when the information counter 8 has attained its highest reading. The pulses produced by comparator 10 thus increase in duration with increasing count of the information counter 8.
The output pulses of comparator 10 are integrated by integrator 12, so that the latter produces at output 13 a voltage rising in steps. This rising voltage arrives as tuning voltage at the input 14 of the input stage 1, which is tuned through in the range selected by means of trip switch 37.
When a transmitter is received in the course of the tuning, discriminator 3 produces in known manner an output voltage with an S-shaped characteristic whose deviation from a reference value serving as a nominal value indicates how accurately the input stage is tuned to the transmitter frequency being received. Exact tuning has been achieved when there is not deviation anymore.
The output voltage of discriminator 3 is fed to Schmitt trigger 26 in the sequence control circuit 19 to which a reference voltage is applied from input 22 and which produces a control pulse when the discriminator output voltage has a predetermined difference from the applied reference voltage, and thus from the nominal value. The control pulse from Schmitt trigger 26 is fed to input 27 of the control logic 29. The control logic 29 then interrupts the supply of stepping pulses to the stepping input 9 of the information counter 8.
In the coincidence circuit 24 of the sequence control circui

t 19, the sync pulses and line flyback pulses are checked for coincidence. The coincidence of these pulses is a criterion that the transmitter received is a picture transmitter, as desired. If coincidence is found, Schmitt trigger 25 triggered by coincidence circuit 24 generates a coincidence signal which is fed to the input 28 of control logic 29 and to one input of the AND-circuit 33.
The control logic 29 has only temporarily interrupted the supply of stepping pulses to the information counter 8 during the reception of a control pulse at input 27, and interruption of this supply is confirmed only when the coincidence signal appears at the input 28 after a delay necessary to check the coincidence. The appearance of the control pulse at input 27 and the delayed appearance of the coincidence signal at input 28 have the effect that the information counter 8 is definitely stopped, and a switching signal is produced at output 32 of the control logic 29. The switching signal at outputs 32 and 36 then arrives at the second input of the AND-circuit 33, which is thus enabled, triggering the frequency control circuit 16 over the line 44 leading to the trigger input 18.
Stopping of the information counter 8 has the result that output pulses of equal pulse duration are produced at the output 11 of comparator 10 with each passage of the counting period of the information counter 8, so that the integrator 12 produces at the output 13 a tuning voltage level which depends on the existing count of the information counter 8. Following stoppage of the information counter 8, triggering of the frequency control circuit 16 permits the turning voltage at the output 13 of the integrator 12 to be varied.
The frequency control circuit 16 can be a conventional frequency control circuit which generates a frequency control voltage at output 17 based on the deviation of the output voltage of discriminator 3 from a reference voltage. This control voltage is fed to the integrator 12 at the input 15; it effects a variation of the tuning voltage produced, so that input stage 1 is finely tuned to the transmitter received.
The control range of the frequency control circuit 16 must be so dimensioned that it comprises the range between two tuning frequencies attained in successive steps of the staircase voltage at the output 13 of the integrator circuit 12. Any value of the tuning voltage between the individual steps of the staircase voltage can thus be attained by means of the frequency control circuit 16, so that exact tuning to any desired nominal frequency is possible.
It can be seen that the tuning of the input stage to the nominal frequency takes place substantially in two steps, namely in a coarse-tuning obtained substantially with digital means, and in subsequent silimar fine-tuning.
The count of the information counter 8, which is obtained by the control logic 29 in the interruption of its step-by-step action, can be stored in a store 34. The desired store position can be selected over the addressing inputs A to D. The storage process is controlled by the control logic 29.
By pressing again on trip switch 37, the above described transmitter search can be continued. When the next transmitter is found, the same processes take place as described above. The count of the information counter 8 attained at the location of the next transmitter can likewise be stored again in store 34.
Since, with the storage of the respective readings of the information counter 8, the information on the respective frequency band contained in band store 40 has also been stored away, not only the stored counter reading arrives in the information counter during the addressing of store 34 for the selection of the respective frequency, but also the band information stored in band store 40. This has the effect that the input stage is set to this frequency range. The reading input of the information counter 8 has the result that the tuning voltage associated with this reading is given off at the output 13 of integrator 12, so that the input stage is coarse-tuned to the desired transmitter. This coarse-tuning thus effects the triggering of the frequency control circuit 16, when it approaches a transmitter, just as in the search run, which then effects the fine tuning to the desired transmitter.
As known, the frequency ranges which can be covered with a certain tuning range, are not equal in the VHF-ranges and in the UHF-ranges. The same variation of the tuning voltage leads in the UHF-range, to a greater variation of the tuning frequency than in the VHF-range. In order to take this fact into account, a switching is effected in Schmitt trigger 26 when trip switch 30 selecting the UHF-range is actuated. This switching has the effect on the control pulse produced by Schmitt trigger 26 that the frequency interval between the nominal frequency and the frequency at which the information counter 8 is stopped, is greater in the UHF-range than in the VHF-range.
FIG. 2 shows an embodiment of the circuit 24 for determining the coincidence between the sync pulses and the line flyback pulses. The video signal is fed to the input 45 and the line flyback signals are applied to the input 46. The sync pulses proper appear at the collectors of the Darlington pair transistors T1 and T2 which together with the input circuit consisting of the resistances R1 and R2 and the capacitors C1 and C2 serve to clip the sync pulses from the video signal. A resistance R3 leads from the connecting points of the capacitors C1 and C2 and of the resistance R2 to the positive terminal of the supply voltage.
The collectors of the transistors T1 and T2 are connected over a resistance R4 with the positive terminal of the supply voltage, and over two base current setting resistances R5, R7 with the base terminals of the transistors T3 and T4. The resistances R6 and R8 serve to shunt charge carriers from the base terminals of the transistors T3 and T4 to ground. Resistances R9 and R10 respectively lead from the collectors of the transistors T3 and T4 to the positive terminal of the supply voltage. Connected to the collector of transistor T3 is the base of a transistor T5 whose collector is connected directly to the positive terminal of the supply voltage and whose emitter is connected by a resistance R11 to the output of the coincidence circuit.
The line flyback pulses are fed from input 46 jointly over a resistance R12 and a capacitor C3 to the base terminals of two transistors T6 and T7 over two base current setting resistances R13 and R14. For shunting charge carriers, resistances R15 and R16 are connected to the base terminals of these transistors. The collector of transistor T7 is connected by a resistance R17 to the positive terminal of the supply voltage. Furthermore this collector is connected by a resistance R18 to the base of a transistor T8 whose emitter is connected to ground and whose collector is connected to the base of transistor T5. The collector of transistor T6 is connected to the collector of transistor T4 and to the base of the transistor T9 whose emitter is connected to ground, and whose collector is connected by a resistance R19 to the output 47 of the coincidence circuit. Between the output 47 and ground is connected a capacitor C4.
When the television receiver is in operation, line flyback pulses are constantly fed to the input 46. These line flyback pulses make the transistors T6 and T7 conductive. Transistor T8 is cut off. In the time periods between the line flyback pulses, transistor T8 is always conductive and transistor T6 is cut off.
As a criterion that the receiver is tuned to a television transmitter, the fact is used that in such a tuning to a television transmitter, the sync pulses transmitted by the latter coincide in time with the line flyback pulses in the television receiver or appear within the line flyback pulses. The presence of coincidence between the line flyback pulses and the sync pulses thus means that the receiver is tuned to a television transmitter.
In the tuning to a television transmitter, the sync pulses clipped by the transistors T1 and T2 from the video signal at the input 45 have the effect that the transistors T3 and T4 are blocked. Since transistor T8 had already been blocked by the line flyback pulses at the input 46, transistor T5 can switch into the conductive state, so that a charging current can flow over transistor T4 to capacitor C4. When the charging voltage at capacitor C4 exceeds the threshold value of Schmitt trigger 25, the latter gives off a control voltage which indicates the coincidence between the sync pulses and the line flyback pulses.
Transistor T9, which in the conductive state provides a discharge circuit for capacitor C4, is always blocked when at least one of the transistors T4 and T6 is conductive. This is naturally the case when there is coincidence, that is, when transistor T5 is conductive and permits the charging of capacitor C4.
In order to prevent that a charging voltage is formed on capacitor C4 by some breakdown in the absence of coincidence, the coincidence circuit of FIG. 2 is so designed that capacitor C4 is discharged in the time periods between line flyback pulses. This is achieved in the following manner: When a noise signal appears in the time interval between two line flyback pulses, which is so great that it simulates a sync pulse, the transistors T3 and T4 are blocked. Transistor T5 becomes thus non-conductive, since it is still held in the cut-off state by the conductive transistor T8. Since transistor T6 is always cut off in the time intervals between line flyback pulses, the cut off state of transistor T4 caused by the noise signal effects switching of transistor T9 into the conductive state. A charging voltage on capacitor C4 can therefore be by-passed by resistor R19 and the collector-emitter junction of transistor T9. Tests have shown that such noise signals appear relatively frequently, so that transistor T9 is switched several times into the conductive state in the periods between line flyback pulses. This prevents a charging voltage from building up on capacitor C4, which could trigger a response of Schmitt trigger 25 connected to output 47.
The same effect, namely the discharge of capacitor C4, is also realized by true sync pulses not simulated by noise signals, which fall into the periods between line flyback pulses. Such sync pulses appear in the non-synchronized state when the video signal received has not yet synchronized the oscillator generating the line flyback pulses.
Summarizing it can thus be said that the sync pulses act as charging pulses in the synchronized state, that is, in the case of coincidence which leads to the charging of capacitor C4, while they act in the non-synchronized state as discharge pulses which enable a discharge circuit for capacitor C4.

The specification discloses various embodiments of solid state television channel selection systems. The systems provide sequential and/or parallel access of television channels by operation of simple pushbutton or sense touch switches on the control panel of the television set, as well as sequential or parallel access of the channels through operation of remote control units. The system enables selected television channels to be skipped during the sequential access mode. The system enables the operator to selectively choose which VHF and UHF channels may be selected by the system.
(The photo is referred to the GRUNDIG:Station-Memory / Tuning Search - Program Memory)




1. A broadcast receiver tuning system comprising:
a memory for storing digital tuning words,
means responsive to said digital tuning words for tuning said receiver to a desired position,
programming switches operable to change the stored digital tuning words in said memory, and
speed control means operable in response to said switches for providing fine tuning of said receiver by slow progressive change of said words in said memory for a preset time interval and thereafter providing course tuning of said receiver by faster progressive change of said words in said memory.


2. The tuning system of claim 1 wherein said broadcast receiver comprises a television set.

3. The tuning system of claim 2 wherein said speed control means provides different tuning speeds for VHF channel tuning than for UHF channel tuning.

4. A television tuning system comprising:
means on the exterior of a television set for selecting desired channels,
address generator means responsive to said means for selecting for generating digital address signals,
memory means storing digital tuning words and responsive to said address signals for outputting selected ones of said tuning words,
means for converting said digital tuning words into analog tuning control signals,
programming switches on the exterior of the television set for changing the stored digital tuning words in said memory means, and
speed control means operable in response to initial actu

ation of said programming switches to provide fine tuning by slowly changing said stored digital tuning words in said memory means during a preset time interval and for thereafter providing coarse tuning by rapidly changing said stored digital tuning words in said memory means until said programming switches are deactivated.


5. The tuning system of claim 4 wherein said programming switches comprise:
up and down switches, and
skip circuitry responsive to actuation of both said switches to cause a predetermined television channel to be skipped.


6. The tuning system of claim 5 wherein said skip circuitry is operable to store logic zeroes at a memory location in said memory means to cause skipping of a channel.

7. The tuning system of claim 4 wherein said speed control means provides slower fine and coarse tuning speeds for UHF channel selection than for VHF channel selection.

8. The tuning system of claim 4 wherein said speed control means again provides fine tuning when said programming switches are deactivated.

Description:

This invention relates to television channel selection, and more particularly relates to a solid state system for selecting television channels and for displaying the selected channel.
It is well known that problems commonly occur in conventional rotary mechanical switch tuning systems which are presently utilized to select channels in television sets. For example, such mechanical rotary switches are subject to mechanical failure and inferior performance due to the inherent unreliability of the switch contacts. In addition, such rotary mechanical switches have not been able to provide parallel channel access, or the direct selection of a desired channel without the requirement of sequentially moving through unwanted channels. Moreover, such rotary mechanical switches have been bulky and expensive.
It has been heretofore proposed to eliminate the problems associated with rotary mechanical switches by the utilization of electronic circuitry. However, such previously developed electronic channel selection systems have not been sufficiently flexible to enable widespread use for a variety of different types of television sets and applications. For example, certain previously developed systems have required extremely uniform varactor tuning diodes to enable channel tuning, therby allowing insufficient tolerances for conventional variances between varactor diodes. Other previously developed systems have not been sufficiently modular to enable a selection of various types of channel access or displays. Moreover, previously developed electronic channel selectors have not been sufficiently economical to fabricate and have required uneconomical printed circuit boards or other uneconomical fabrication techniques or construction. For example, certain prior systems have required expensive potentiometers for each channel desired to be tuned.
In addition, previously developed electronic television tuning systems have not satisfactorily satisfied recent regulatory requirements which call for a television tuner to provide a comparable capability and quality of tuning for both VHF and UHF stations. Specifically, such prior tuning systems have not enabled selection and display of a selected group of precise UHF channels, nor have the prior systems provided means for easily changing selected UHF channels.
Iin accordance with the present invention, a solid state system is provided for tuning a television set which includes an array of switches each corresponding to a predetermined television channel. An address generator is responsive to the operation of the switches for generating multibit digital address words each corresponding to one of the switches. A tuning memory includes a random access memory for storing digital tuning words and for outputting the tuning words in response to the address words. Circuitry converts the tuning words into analog signals and controls a varactor diode ttuner in order to select the desired television channel.
In accordance with another aspect of the invention, a television tuning system includes a matrix array of switches for selecting a desired television channel. An address generator generates a unique binary address corresponding to the selected channel. A tuning memory stores a plurality of digital tuning words and outputs one digital tuning word in response to each unique binary address. A digital-to-analog converter converts the binary tuning words into an analog signal for providing channel tuning. Circuitry is responsive to the unique binary address for generating one of three band selection signals. A diode tuner is responsive to the analog signal and the band selection signal for tuning the desired television channel automatically.
In accordance with yet another aspect of the invention, a sequential access television channel tuning system includes switches for generating up and down scan indications. An up/down couonter is responsive to the switches for counting clock signals and for generating binary address signals. A memory generates unique digital tuning words in response to the address signals. Circuitry is responsive to the tuning words for tuning to a desired television channel.
In accordance with another aspect of the invention, a television channel tuning system includes a matrix array of switches operable to select a desired television channel. An address generator generates a unique binary address corresponding to the selected channel. A memory stores a plurality of digital tuning words each representative of a different television channel. A converter converts the tuning words into analog tuning levels. Circuitry is operable to selectively change the digital tuning words stored in the memory. Circuitry is also operable to cause selected ones of the digital addresses to be skipped during sequential access tuning of the system.
In accordance with yet another aspect of the invention, a combined sequential and parallel access television channel tuning system includes a matrix array of channel selection switches each operable to provide parallel selection of a television channel. Up and down channel selection switches are operably to provide sequential selection of television channels, the up and down switches being connected to terminals of the matrix array. Circuitry is responsive to operation of the matrix array switches for generating unique binary address signals corresponding to the selected television channel. Circuitry connected to the matrix array is responsive to operation of the up and down switches for generating sequential binary address signals corresponding to sequential television channels. Tuning circuitry is responsive to the binary address signals for tuning a selected television channel.
For a more complete understanding of the present invention and for further objects and advantages thereof, reference is now made to the following description taken in conjunction with the accompanying drawings, in which:
FIG. 1 is a somewhat diagrammatic view of a first embodiment of a 16 channel parallel access television selection system with illuminated sense buttons and including a sequential access remote tuning device;
FIG. 2 illustrates a 20 channel sequential access television channel selection system with illuminated channel lamps;
FIG. 3 illustrates a 20 channel parallel access channel selection system with pushbutton switches and adjacent light sources;
FIG. 4 illustrates another embodiment of a 20 channel parallel access selection unit with illuminated tab displays;
FIG. 5 illustrates a sequential access channel selection system having a two digit seven segment numeric channel display;
FIG. 6 illustrates a 20 channel parallel access system with a two digit seven segment channel display;
FIG. 7 is a block diagram of s 16 channel parallel or sequential access system with a neon light illuminated tab display;
FIG. 8 is a block diagram of a 20 channel combined sequential and parallel access system with a two digit seven segment display;
FIG. 9 is a schematic diagram of an address generator for a 16 channel sequential channel selection circuit;
FIG. 10 is a state diagram of the operation of the AFC sequence counter and associated circuitry of the circuit shown in FIG. 9;
FIGS. 11a-h comprise waveforms of portions of the circuit as shown in FIG. 9 during operation;
FIG. 12 is a schematic diagram of a 16 channel parallel access address generator for use in conjunction with the sequential access circuit of FIG. 9;
FIG. 13 is a block diagram of a tuning memory for use with both the sequential access address generator of FIG. 9 and the parallel access address generator of FIG. 12;
FIG. 14 is a schematic diagram of a 20 channel combined sequential access and parallel access address generator as shown in FIG. 8;
FIG. 15 is a block diagram of a tuning memory for use with the 20 channel selection address generator of FIG. 14;
FIGS. 16a and 16b are schematic diagrams of the tuning memory of FIG. 15; and
FIGS. 17a-h are waveforms of a portion of the tuning memory as shown in FIG. 16;
Referring to the drawings, FIGS. 1-6 illustrate variations of tuning control features for a television set which are provided by the present invention. Referring to FIG. 1, a television set 10 includes a conventional television screen 12 and a control panel 14. A picture and sound control set 16 includes variable controls for controlling the volume, picture brightness, contrast, tint and color of the television picture. A microphone 18 receives ultrasonic commands from the remote control unit.
A set of 12 pushbutton or touch sense switches 20 is provided to enable selection of VHF television stations. A set of four pushbuttons or touch sense switches 22 is provided to enable selection of UHF television channels. The switches 20 may be selectively depressed to select any one of VHF channels 2-13. The numerals 2-13 may be permanently affixed to the pushbutton switches 20, or alternatively, tabs bearing indicia may be selectively affixed to or removed from the switches 20. Depression of the switches 20 causes the selected switch and the corresponding indicia to become illuminated, such that the operator will know the desired channel has been selected for display.
The switches 22 may also be provided with removable tabs bearing numerals, such that any selected group of VHF and UHF television channels may be selected in the manner to be subsequently described. In the illustrated embodiment, four UHF television stations may be selected by depression of the switches 22. Depression of one of the switches 22 causes the selected switch and the corresponding tab indicia to be illuminated. Although in the illustrated embodiment four UHF channels are provided for selection, in other embodiments of the invention to be subsequently described, the capacity for more or less UHF channel selections may be provided. As will be subsequently described, when the television set 10 is initially tuned, the desired VHF and UHF channels are set into the system and tabs bearing the desired channel indicia are inserted into the pushbutton switches 20 and 22. If desired, the selected VHF and UHF channels may be selectively changed at any time.
A remote control tuning device 24 is provided for use with the present tuning system and includes an off/on button 26. An up button 28 is provided to be depressed to enable sequential selection of the channel by moving the tuning system of the television set 10 from one channel to a higher number channel. A down button 30 is provided to be depressed to cause automatic sequential selection of television channels in the set 10 by causing the tuning of the set to move from one channel to a lower number channel.
As noted, any of the pushbutton switches 20 and 22 may be selectively depressed and the television set 10 will be automatically tuned to the desired channel. This type of selection is termed parallel access or selection. The type of series channel selection provided by the remote control unit 24 is termed sequential channel selection. It will be understood that an array of switch buttons could be provided on the remote control unit 24 to enable parallel selection of channels by the remote control unit 24. Remote control unit 24 operates according to any one of a variety of well known techniques, such as by generating acoustic signals which are detected by tuned circuits in the television set 10 to enable channel selection.
An up voltage programming pushbutton switch 32 and a down voltage programming pushbutton switch 34 are provided behind a removable panel, etc. on the television set 10 to enable initial tuning of the VHF and UHF channels which may be selected by actuation of the switches 20 and 22. In order to initially program the system to select a desired channel, the operator first disengages the channel skip circuit and AFC circuit (to be later described) and then pushes the first VHF switch 20 which corresponds to Channel 2. Both voltage programming switches 32 and 34 are then depressed for a brief period to clear the system. The switch 32 is then depressed until Channel 2 appears on screen 12. If either switch 32 or 34 is depressed longer than a preset period, as for example 8 seconds, the system switches to a fast tuning mode. When the picture appears on screen 12, the switch 32 is released and, on again pressing switch 32 or 34, the system switches back to the slow tuning mode. Switches 32 and 34 may then be "bumped" to fine tune the program on the screen 12.
When Channel 2 is tuned, the next switch 20 is depressed and the same sequence is performed to select VHF Channel 3. When all VHF channels have been programmed, the four selected UHF channels are programmed in the same way and tabs having indicia corresponding to the selected UHF channels are added to the switches 22. If desired, less VHF and more UHF channels may be programmed into the system. Once all switches 20 and 22 have been programmed, the skip circuit is again energized, and it is necessary only to depress or otherwise actuate one of the switches 20 or 22 and the set 10 will be automatically tuned to the desired channel and the actuated switch will be illuminated.
FIG. 2 illustrates a variation of the television tuning system shown in FIG. 1, and like numerals will be utilized for like and corresponding parts previously described. In this embodiment, parallel access of television channels is not provided on the front panel of the set 10, but only serial or sequential channel access is provided for 20 channels. Twelve lamps 36 are each provided with an indicia from 2-13 in order to indicate a desired VHF television channel when illuminated. A second set of lamps 38 is provided in order to indicate eight selected UHF channels. The indicia formed on lamps 36 are fixed, while the indicia for lamps 38 may be removed and changed as desired. The particular UHF channels which may be selected and illuminated by the lamps 38 are chosen by the operator by operation of an up voltage programming button 40 and down voltage programming button 42 in the manner previously described.
An up channel advance button 44 and a down channel advance button 46 may be selectively depressed by the operator in order to cause lamps 36 and 38 to be sequentially illuminated. When the lamp corresponding to the desired channel is illuminated, the button 44 or 46 is released and the set 10 will be tuned to the desired illuminated channel. It will be understood that sequential channel access may also be provided with the system shown in FIG. 2 by a remote control unit similar to that shown in FIG. 1. In addition, if desired, parallel access may be provided by remote control units having the required number of pushbutton selection switches.
As noted, initial programming of the channels associated with lamps 36 and 38 is accomplished in the manner previously described by operation of voltage programming buttons 40 and 42. In some cases, it may be desirable to skip certain channels during sequential access. To skip a channel, the skip disable switch is opened, and the channel is selected. Both buttons 40 and 42 are then simultaneously depressed for a brief period, and the skip disable switch is then closed. Subsequent operation of the tuning system will result in the skipping of that channel, and thus the lamps 36 and 38 corresponding to skipped channels will not be illuminated. In this manner, inactive or undesired channels need not be tuned through when searching for valid channels.
FIG. 3 illustrates a third embodiment of the television channel selection system according to the invention which enables selection of VHF channels by 12 pushbutton switches 50, each of which is provided with a suitable channel indicia. Twelve lamps or light emitting diodes (LEDs) 52 are disposed adjacent switches 50 and are illuminated upon depression of the associated switches 50. To enable UHF channel selection, eight pushbutton switches 54 are provided in association with eight LEDs 56 which are illuminated when the corresponding switch is depressed. Indicia tabs are operable to be inserted into the pushbutton switches 54 to designate the particular UHF channels desired to be selected. An up voltage programming switch 58 and down voltage programming switch 60 are operable during initial programming of the set 10 to tune the channels selected by pushbutton switches 50 and 54.
FIG. 4 illustrates another embodiment of a channel select and display system according to the invention. Twelve pushbutton switches 62 may be depressed to select VHF channels. Lamps 64 are associated with the pushbutton switches 62 and have indicia thereon which are illuminated when the lamp is energized by the depression of the corresponding pushbutton switch 62. Eight, or any other selected number less then eight, pushbutton switches 66 may be depressed for selection of UHF channels. Eight lamps 68 include selected indicia thereon corresponding to the desired UHF channels. The indicia may be selectively changed by removing tabs bearing the indicia and by selection of new tabs with different indicia thereon. An up voltage programming button 70 and a down voltage programming button 72 enable tuning of the desired channels to be selected by the pushbutton switches 62 and 66. It will be noted, if desired, a remote control unit similar to that shown in FIG. 1 may be utilized to control sequential selection of the channels of the system shown in FIGS. 3 and 4. Further, up and down channel selection switches may be also incorporated on the select system shown in FIGS. 3 and 4.
FIG. 5 illustrates a station select and display system wherein sequential channel selection is enabled by an up channel advance button 74 and by a down channel advance button 76. The number of the channel presently being displayed on the screen 12 is displayed in a seven segment two digit display 78. In this embodiment, the VHF channel numbers are contained in a read only memory (ROM), and are not programmable. The UHF channel numbers which may be displayed in the display 78 are chosen by operation of a display programming button 83, which is depressed until the correct number is displayed, then released. In operation, the operator pushes either the up button 74 or the down button 76 and when the desired television channel number is displayed on the display 78, the operator releases his finger and the set is tuned to the desired channel.
FIG. 6 illustrates a variation of the system shown in FIG. 5 wherein parallel channel access may be provided by the use of a switch array 84. The switch array 84 may comprise mechanical pushbutton switches or sense touch buttons with removable tabs to enable the channel number to be changed. In operation of the system shown in FIG. 6, the desired channel is selected by merely depressing or actuating a switch in the array 84 which corresponds to the desired channel. The desired channel number will appear in the display 78 and the set will be automatically tuned for display of the selected channel.
Sixteen Channel Tuner With Neon Display
FIG. 7 illustrates a 16 channel system with a neon light display. This system may be utilized to provide the functions of the systems shown in FIGS. 1-4. The terminals of a 4×4 pushbutton switch array matrix 90 are connected to a parallel address generator 92 which applies binary address signals on a multiline bus comprising four leads 94, 96, 98 and 100 to a tuning memory 102 and to a neon display 104. A second sequential address generator 93 also connects to leads 94, 96, 98 and 100. The multiline bus enables the present system to be modular, such that different displays and address generators may be easily substituted to provide a wide range of desired functions. An up pushbutton switch 106 and a down pushbutton switch 108 are connected to the address generator 93 to enable serial channel access when desired. The address generators 92 and 93 generate an AFC (Automatic Frequency Control) defeat signal via leads 110 which is utilized to eliminate AFC during tuning operation. A skip signal is applied from the tuning memory 102 to the generator 93 via lead 112 to enable predetermined tuning positions to be skipped during sequential channel selection. Switch 113 may be thrown to disengage the skip function during the set-up procedure. Switch 113 may be mechanically interlocked with the AFC switch to eliminate AFC during skip programming.
An up pushbutton switch 114 and a down pushbutton switch 116 are utilized to program the tuning memory to enable selection of desired channels. The selected binary words stored in the tuning memory 102 are applied to a digital to analog converter 103 which generates duty cycle modulated binary signals that are fed to the analog integrator, which in turn feeds the varactor diode tuner 120. Tuner 120 operates in the known manner to vary the local oscillator of the television set in order to select the desired television channels, and the RF tanks (not shown) to reject unwanted signals. Band selection signals are applied from the tuning memory 102 to the bases of transistors 122. The collectors of transistors 122 are coupled to the bases of transistors 124 to generate band selection signals UHF, HVHF and LVHF which are applied to the varactor diode tuner 120 in order to select the varactor diode set necessary to perform the desired tuning operation. The generation of the band selection signals is accomplished automatically and no action by the operator of the set is needed, except during initial programming.
The address signals generated by the address generators 92 and 93 are also applied from the multiline bus to a neon display driver 104 which enables transistors 130 and leads 132 in order to selectively energize neon light bulbs contained within a 4×4 display matrix 134. The neon display 104 is responsive to the address generated by the generators 92 and 93 such that one of the neon lamps within matrix 134 is energized to correspond with the pushbutton depressed in the matrix 90 to indicate which television channel is presently being tuned. The lamps may be incorporated into the switches 90 (FIG. 1) or placed adjacent the switches 90 (FIG. 3).
Briefly, in operation of the system shown in FIG. 7 in the parallel access mode, one of the switches in the matrix 90 is actuated. The address generator 92 detects which of the pushbutton switches was actuated and generates a four bit binary address via leads 94-100 to the tuning memory 102. In response to the digital address, the tuning memory 102 applies a binary tuning word stored in the memory to the digital-to-analog converter 103. Converter 103 converts the binary word into a duty cycle modulated binary signal, which drives integrator 118, generating an analog voltage and applies the voltage to the varactor diode set within tuner 120 which is selected by the band selection signals. Such varactor diode tuners are known in the art and generally operate as voltage variable capacitors which vary a local oscillator and RF tank frequency to provide channel tuning. The address signals generated by the generator 92 operate through the neon display driver 104 in order to energize one of the neon lamps in the array 134 in order to indicate which channel is presently being displayed.
In the operation of the system shown in FIG. 7 in the serial or sequential channel access mode, one of the up channel select buttons 106 or down channel select buttons 108 is depressed by the operator. The address generator 93 then generates a series of addresses to the tuning memory 102 such that a series of the stored binary words in the memory 102 is applied to the digital-to-analog convertor 103 and integrator 118. A series of analog voltages are then applied to the varactor diode tuner 102 such that one channel after another is selected for display. The address from the address generator 93 is also applied to the neon display driver 104 such that lamps in the array 134 are sequentially energized in order to indicate which of the channels is presently being displayed. When the operator sees that the desired channel is displayed by viewing the energized lamp in the array 134, he releases his finger from one of the buttons 106 or 108 and the set is properly tuned. As previously noted, desired channels may be programmed by operation of the up voltage programming button 114 and down voltage programming button 116 such that the channels are skipped during sequential tuning. In such a case, the tuning memory 102 operates to skip the unused position and the neon display 134 does not display the skipped channel indication.
Twenty Channel Tuner With Numeric Display
FIG. 8 illustrates a system somewhat similar to the system shown in FIG. 7, and thus like numerals are utilized for like and corresponding parts previously described. In this embodiment, a 5×4 pushbutton switch selection matrix 140 is provided such that twelve VHF and eight UHF channels may be selected. It will, of course, be understood that the display shown in FIG. 8 may be used with the system of FIG. 7, and vice versa, the illustrated systems being merely exemplary.
In the embodiment of FIG. 8, a multiline bus comprising five address lines 94, 96, 98, 100 and 101 extends between an address generator 141 and a tuning memory 143. An advantage of the system shown in FIG. 8 is that both the up channel select switch 106 and down channel select switch 108 are connected to terminals of the matrix 140, and thus do not require additional chip pin connections. Moreover, the skip signal applied via lead 112 is applied through matrix 140 to the address generator 141. The system shown in FIG. 8 may be used in a system such as shown in FIGS. 5 or 6, or alternatively may be utilized in a channel selection system utilizing both the up and down channel advance buttons 74 and 76 of FIG. 5 and the pushbutton array 84 shown in FIG. 6 on the same television set front panel.
The operation of the system shown in FIG. 8 is similar to that shown in FIG. 7, except that the five address lines from the address generator 141 are applied to a display memory 145 which generates control signals to a seven segment decoder 142. The two circuits 142 and 145 may be formed on separate chips, or may be combined in one circuit. Decoder 142 drives the two digit seven segment display 147 to provide a visual indication of the channel presently being tuned. If desired, the neon lamp display driver 104 of FIG. 7 may be used in place of the digital readout and display.
Address Generator For 16 Channel Sequential Access System
FIG. 9 illustrates in schematic detail a sequential access program generator 93 for use in the system of FIG. 7. As will be subsequently described, the circuit shown in FIG. 9 may be combined with circuitry shown in FIG. 12 to allow the selection by a manufacturer of serial and/or parallel channel access. An advantage of the circuitry shown in FIG. 9 is that the entire circuit may be formed on a single 16 pin semiconductor chip utilizing conventional integrated fabrication techniques.
Referring to FIG. 9, an up channel select pushbutton switch 150 is connected to a pin 152 of the semiconductor chip and is applied through inverting buffers 154 and 156 to a terminal of a NAND gate 158. Similarly, a down channel select pushbutton switch 160 is applied to a pin 162 of the semiconductor chip and is applied through inverting buffers 164 and 166 to a terminal of a NAND gate 168. Gates 158 and 168 are interconnected as a latch. The outputs of gates 158 and 168 are applied to the up and down inputs of a four bit up/down presettable multi-modulo counter 170. Counter 170 enables the circuit to select either six, eight, 12 or 16 television channels, as desired. In order to set the counter 170 to enable selection of a predetermined number of television channels, pins 172 and 174 are selectively grounded or opened according to a conventional code.
The outputs of the counter 170 are applied through NAND gates 176, 178, 180 and 182 to pins 184, 186, 188 and 190 to provide the four bit binary address which is applied to leads 94, 96, 98 and 100 previously shown in FIG. 7. The channel skip signal is generated from circuitry to be later described in FIG. 13 or other figures and is applied through inverting buffers 192 and 194 to an input of an OR gate 196. The output of gate 196 is applied through an invertor 198 to control the clocking of the counter 170 such that a memory address will be skipped for each input signal on the channel skip line.

An oscillator 200 generates clock signals to a three bit AFC sequence counter 202. The C output of counter 202 comprises a clock out signal which is applied to the second input of gate 196. The output of invertor 192 is applied as an extend signal to oscillator 200, and the output of invertor 194 is applied as a clear signal to counter 202. The three bit outputs from counter 202 are applied as inputs to a NAND gate 204, the output of which is applied as an enable signal to oscillator 200. The A and B outputs from counter 202 are applied to a NAND gate 206, the output of which is applied through a NAND gate 208 and invertor 210 to an input of an OR gate 212 to provide a stretched AFC defeat signal.
The outputs of invertors 156 and 166 are tied to the inputs of an OR gate 214, the output of which is applied to a NAND gate 216. The output of gate 216 is tied to inputs of NAND gates 176-182. The output of gate 216 is also applied to a pin 218 to a NAND gate 220 which is located on the chip to be described in FIG. 12. The output of gate 220 is applied through pin 222 to the input of gate 216. Gate 220 comprises a latch which enables the selection by a television manufacturer of either the serial access circuit shown in FIG. 9 or the parallel access circuit shown in FIG. 12, or both circuits. The output of gate 216 is also applied to the load input of counter 170.
FIGS. 10 and 11 assist in the explanation of the operation of the system shown in FIG. 9. FIG. 10 comprises a state diagram illustrating the various stages of operation of the AFC sequence counter system of FIG. 9 during sequential channel selection. FIG. 11 illustrates various waveforms of portions of the circuit of FIG. 9. If it is desired to operate the television set only in the sequential access mode, pin 222 is grounded such that the circuit of FIG. 9 is activated and the circuit to be described in FIG. 12 is not installed. This grounding connection will be made by the television manufacturer upon determination of the type of channel selection control desired. If the manufacturer so decides, both the circuits of FIGS. 9 and 12 will be used to provide both sequential and parallel access. Upon grounding of pin 222, gate 216 generates a logic one output which allows data to ripple through the gates 176, 178, 180 and 182.
Referring to FIG. 10, at the beginning of operation of circuitry shown in FIG. 9, the counter 202 is set at 111. When the operator depresses either the up or down channel selection buttons 150 or 160, the counter 202 cycles to state 000 generating a station change signal, and then to states 001 and 010. During the counter states 000, 001 and 010, the gates 206 and 208 decode the counter outputs and generate the AFC defeat signal through gate 212. During the remaining states 011, 100, 010 and 110, the operator decides whether or not the channel being accessed is the desired channel. If so, the operator releases the button 150 or 160 and terminates operation of the system. The oscillator 200 is enabled through the NAND gate 204 during this operator recognition time and the oscillator 200 cannot be stopped during the states of operation until the operator takes his finger off the up or down selection buttons 150 or 160. The counter 202 will only stop in state 111 in any case.
If the particular station being accessed is not the desired station, the operator maintains his finger on either buttons 150 or 160, and the counter 202 sequences to state 111 and to state 000, such that a station change is generated to counter 170 and the cycle is repeated. The counter 170 then generates a new binary address to pins 184-190. If, however, a skip signal is generated during any state, the skip signal is applied from gate 196 and invertor 198 to counter 170 such that the counter is advances one state for each skip signal. The memory address, or addresses, are then skipped and the cycle begins again for another cycle. The resulting outputs from the counter 170 for a valid station comprise a unique four bit binary word which is decoded by subsequent circuitry.
Specifically referring to sequence counter 202 and associated circuitry in FIG. 9, a logic low appearing at the output of gate 214 and which is applied to an input of NAND gate 204 forces the output of a NAND gate 204 high, thereby enabling the oscillator 200. The counter 202 will continue to count in complete cycles as long as a logic low forces a one on the enable output of gate 204. When the low applied to gate 204 is removed, the output of gate 204 will set up conditions for the counter 202 to stop the next time that logic state 111 appears at the output of counter 202, thereby terminating the cycle.
Referring to FIG. 11, waveform 11a illustrates the depression of the up channel select button 150 by the operator during the time interval t ₁ -t ₀ . The waveform shown in FIG. 11b illustrates the oscillator enable signal that is applied to oscillator 200 from gate 204. The waveform shown in FIG. 11c illustrates the capacitor charging voltage generated within the oscillator 200. The interval designated generally by arrow 230 indicates the generation of skip channel pulses while that shown as 231 illustrates one skip pulse. The waveform shown in FIG. 11d indicates the output of the oscillator 200 which is applied to clock the sequence counter 202. The waveform shown in FIG. 11e comprises the eight states of operation of the counter 202 previously described in FIG. 10. The first three states zero, one and two of each cycle provide the AFC defeat function.
The waveform shown in FIG. 11f comprises the clock output of the counter 202 which is applied to gate 196 and to the invertor 208. The clock output pulse 232 is elongated due to the generation of skip channel signals. The waveform shown in FIG. 11g comprises the channel skip inputs which are applied through inverters 192 and 194 to gate 196. As noted, the channel skip inputs are generaated by circuitry to be subsequently described such that invalid channels which do not have programs in a particular area will not be cycled through by the present circuit. During the interval designated generally by the arrow 234, five skip signals are generated such that five invalid channels are skipped. During the interval designated generally by the numeral 236, a single invalid channel is skipped. The waveform shown in FIG. 11h comprises the clock inputs which are generated by gate 196 and are inverted by inverter 198 to clock the counter 170.
During operation of the circuit of FIG. 9, the depression of up channel select button 150 causes the counter 170 to generate an upward changing sequence of four bit address words to pins 184, 186, 188 and 190. These address words are applied to the memory circuit shown in FIG. 13 in order to select memory words, which in turn cause the control of varactor diode tuners. Depression of the down channel select button 160 causes operation of the counter 170 in the down mode to cause a downward sequence of address words to be generated from the counter 170. As noted, the number of address words generated by the counter 170 is controlled by selective grounding of pins 172 and 174.
Address Generator For 16 Channel Parallel Access System
FIG. 12 illustrates a parallel access address generator 92 which may be fabricated on a single 18 pin semiconductor chip with integrated logic techniques and utilized in the system of FIG. 7. This circuit may be used in place of or in conjunction with the circuitry shown in FIG. 9. As previously noted, the circuits shown in FIG. 9 and FIG. 12 are interconnected by a common gate 220, the terminals of which may be selectively grounded to enable the television manufacturer to use either the circuitry shown in FIG. 9 or FIG. 12. If the gate terminals are cross connected, both circuits are enabled for use.
Referring to FIG. 12, a 4×4 array of sixteen touch sense switches, or any other suitable type of switches, is formed as a switch matrix 240. Four terminals of the matrix 240 are connected to a four line sense amplifying stage 242, while the remaining four terminals of the matrix 240 are connected to four line sense amplifiers 244. The outputs of amplifiers 242 are applied to a four to two line encode circuit 246. The outputs of the amplifiers 244 are applied to a four to two line encode circuit 248. The outputs of the amplifiers 242 are also applied to a NOR gate 250, the output of which is applied to a NAND gate 252. The outputs from the amplifiers 244 are applied to the inputs of a NOR gate 254, the output of which is also applied to an input of gate 252.
The two outputs from the encode circuits 246 are applied through time delays 256 and 258 to a four bit latch 260. The outputs from the encode circuit 248 are applied through time delays 262 and 264 to the latch 260. The output of gate 252 is applied to load the latch 260 and is applied to the AFC defeat circuit 266. The AFC defeat circuit 266 generates an AFC defeat signal which is applied through the gate 212 previously shown in FIG. 9. Pin 268 connects the gate 212 to the chip shown in FIG. 12, while pin 270 connects the gate to the circuit of FIG. 9. The outputs of the latch 260 are applied through NAND gates 274, 276, 278 and 280 to provide a four bit binary address at pins A, B, C and D.
In operation of the circuit shown in FIG. 12, the manufacturer may choose either or both of the address generator circuits shown in FIG. 9 and FIG. 12 by selective grounding or interconnecting of terminals of the gates 220. In some instances, the circuit shown in FIG. 9 will be utilized as the remote control channel selection circuitry, while the circuitry shown in FIG. 12 will be used as the channel selection circuitry for the set control panel. When one of the sense switches in the matrix 240 is depressed, a logic zero appears at the output of one of the amplifiers 242 and one of the amplifiers 244. Gates 250 and 254 then generate logic one outputs which operate through the gate 252 in order to initiate an AFC defeat signal from circuit 266. This eliminates the AFC of the system during the tuning operation, and also provides an extended AFC defeat signal after the customer releases the button.
The encode circuits 246 and 248 detect the output from the amplifiers 242 and 244 and transmit encoded signals through the time delays 256-264 which provide sufficient time delays to enable the latch 260 to be loaded in response to the signals generated by the gate 252. A four bit binary code is generated from the encode circuits 246 and 248 for storage in the latch 260. The latch 260 then generates a four bit digital code output through the NAND gates 274-280. The outputs from the gates 274-280 may not be transmitted to pins A-D unless the gate 220 generates a logic one which is applied to the inputs of the NAND gates 274-280. Gate 220 also operates to override the output from the circuit shown in FIG. 9.
When the circuits shown in FIGS. 9 and 12 are used concurrently, provision must be made to store the state of the circuit shown in FIG. 12 in the parallel circuit of FIG. 9. Consequently, assume that the circuit shown in FIG. 9 has been placed in the up mode and then the circuit shown in FIG. 12 is energized and a channel selected thereby. When the operator next tries to go back to the circuit shown in FIG. 9, it is desirable to start at the last channel selected by the circuitry shown in FIG. 12. Thus, the data word generated from the four bit latch 260 is also applied to the output of the circuit shown in FIG. 9. Thus, when the circuit of FIG. 9 is not active, a load signal is generated from the gate 220 and is applied through the invertor 286 to load the counter 170 with the output of the circuit shown in FIG. 12.
Tuning Memory For Sixteen Channel System
FIG. 13 is a schematic diagram of the tuning memory 102 previously shown in FIG. 7. This circuit may be utilized with either or both of the circuits shown in FIGS. 9 and 12, as well as other circuits using similar techniques. The binary address outputs from the circuits previously described in FIGS. 9 and 12 are applied to the A, B, C and D inputs of the circuit of FIG. 13 and are applied to a four to 16 line decode circuit 290. The resulting word enable signals are applied to a two bit 16 word band switch and skip random access memory (RAM) 292 and a 12 bit 16 word tuning voltage RAM 294. RAMs 292 and 294 have stored therein binary coded words which flag the television channel to be skipped, LVHF, HVHF and UHF band switch information, and which define an analog voltage level utilized to control a desired varactor diode in the tuning system. Data multiplex gates 296 control the input and output of the data stored in RAM 292. Data multiplexing gates 298 control the input and output of data stored in RAM 294. A D.C. voltage, typically from a battery, is applied to pin 300 in order to protect the storage of the RAMs 292 and 294 during the interval from set-up by the manufacturer and ultimate use by the consumer. Voltage is applied from a storage battery to pins 300 in order to provide as much as six months protection to the stored contents of the RAMs.
The output from RAM 294 is applied to a 12 bit presettable ripple up/down counter 302 which generates output data to a 12 bit data comparator 304. A 12 bit synchronous binary counter 306 also applies binary patterns to the comparator 304 which in turn generates an output through buffer 308 representative of the digital data from the RAM 294 in a predetermined output sequence. The predetermined output sequence is provided with a high ripple frequency such that the integrating capacitor in the integrating filter of the D-A convertor may be as small as possible. The output applied through invertor 308 to pin 310 is thus applied to a D-flipflop and to an integrating filter in order to provide the desired analog voltage for control of the varactor diode for tuning of the television set.
A 1 MHz clock input is applied through pin 312 and through an invertor 314 to clock the synchronous binary counter 306. Clock signals are also applied from the counter 306 to a count down and frequency select circuit 318 which generates clock signals for a tuning program generator 320. Generator 320 is loaded by operation of the voltage programming up button 322 and down button 324. The operation of the programming buttons has been previously described with respect to FIGS. 1-6. The buttons 322 and 324 are utilized to program the tuning voltage for the VHF and UHF channels which are selected by the system. Up/down clock and load signals are applied from the generator 320 to the counter 302 and are utilized to program the binary words stored in the RAM 294. Read/write signals are also generated from the generator 320 to the data multiplexing gates 298 for the RAM 294.
Program band and skip signals are applied to pin 325 to a band/skip program generator 326 which generates read/write signals to the data multiplexing gates 296 and which also generates clock signals to a two bit ripple counter 330. Band select signals and channel skip information is generated by a generator 334 and applied to pins 336, 338, 340 and 342 for application to the varactor diode tuner in the manner shown in FIG. 7. Data is transmitted to the data multiplexing gates 296 from the ripple counter 330 and is applied from the gates 296 to the generator 334.
In operation of the tuning memory circuit shown in FIG. 13, it will be assumed that the system is to be initially programmed by the manufacturer. A storage battery is connected to the pin 300 in order to protect the memory of the RAMs 292 and 294. In order to tune the first valid VHF Channel 2, both of the up and down buttons 322 and 324 are simultaneously pressed for a brief period. A signal is generated by the generator 320 such that all logic zeroes are applied to the up/down counter 302, and subsequently loaded into the RAM. Since all logic zeroes are now present in the RAM, giving a known initial condition, the up button 322 is depressed. After a set time interval such as 8 seconds, the system changes from a slow mode to a fast mode until the button is released. Inasmuch as the system started out at the logic zero level, the first channel will be Channel 2 and the operator then removes his finger. The system goes into an 8 second slow mode of operation on release of the button and the operator may alternatively operate the tuning buttons 322 and 324 in order to fine tune Channel 2. This procedure is repeated by the manufacturer until all VHF signals have been selected. The selected UHF channels may also be selected in the same manner, usually by the consumer. It should be noted for channels whose binary data word in closer to all ones than all zeroes (above the middle of a band), it would be preferable to press the down button after setting memory to zero, thereby counting down, and approaching the channel from above its frequency rather than from below as described previously.
After the consumer buys a set, it may be desired to skip certain program channels which are not available in the consumer's viewing area. In order to skip a channel, the set is addressed to the memory address of the channel to be skipped. Both the up button 322 and down button 324 are simultaneously depressed and two zeroes are entered into the RAM 292 for that channel. When the channel skip generator 334 detects two zeroes transmitted from the RAM 292, the channel skip signal is generated from generator 334 such that the channel will be skipped when operating in the sequential access mode.
In operation of the circuit, the band switch RAM 292 receives the two bit data input word from the counter 330 through the multiplexing gates 296. The word is stored in RAM 292, and later transmitted through the multiplexing gates 296 to the generator 334 in order to generate the band switch signals to the varactor tuner through pins 336-340.
The 1 MHz clock applied to pin 312 clocks the counter 306, which provides twelve outputs to the comparator 304 which changes in a synchronous fashion. The comparator 304 converts the binary word generated from the RAM 294 through the up/down counter 302 into an output signal which is applied through buffer 308. The output signal has a duty cycle equivalent to the desired D.C. level. Thus, the output of comparator 304 comprises data words having duty cycles such that when the data words are integrated, a desired analog voltage is provided. The least significant (fastest changing) bit from the synchronous binary counter 306 is matched with the most significant bit from the RAM 294 in order to provide the maximum cross coupling to provide maximum ripple frequency of the output signal applied to pin 310.
It will be understood that other types of digital scanning systems for providing inputs to the integration circuit may be utilized with the invention in place of the synchronous counter 306 and comparator 304. For example, a digital scanning system such as described in U.S. Pat. Application Ser. No. 457,664, filed Apr. 3, 1974 by Tegze Haraszti and Preston Snuggs may be utilized in the system shown in FIG. 13. Alternatively, the SN7497 Binary Rate Multiplier manufactured and sold by Texas Instruments of Dallas, Tex. may be used for the counter 306 and comparator 304.
In the U.S.A. market, the low VHF band (enabled by pin 340) comprises five VHF channels 2 through 6, while the high VHF band (enabled by pin 338) comprises seven VHF channels 7 through 13. However, the UHF band (enabled by pin 336) will be comprised of a possible seventy channels. Hence, when fine tuning during channel selection, more bits will be required to fine tune across each VHF channel than across each UHF channel. The count down and frequency select circuit 318 provides a fast or a slow tuning voltage in dependence upon which band is being enabled. When the UHF signal band signal is generated on pin 336, a signal is applied via lead 350 to the select circuit 318 such that a slower clock signal is applied to the tuning program generator 320.
Combined Serial and Parallel Access Channel Selection Circuit for Twenty
Channel Tuning System
FIG. 14 is a schematic diagram of the address generator 141 for use in the twenty channel tuning system shown in FIG. 8. An advantage of the circuitry shown in FIG. 14 is that both serial access and parallel access channel selection functions are accomplished by circuitry on a single semiconductor 18 pin integrated logic semiconductor chip.
Referring to the circuit shown in FIG. 14, a 4×5 twenty pushbutton switch array 360 is provided which may correspond to the parallel access systems shown in FIGS. 1, 4 and 6. An up sequential access button 362 and a down sequential access button 264 are connected through the matrix 360 to enable sequential channel selection. Such up and down channel selection may alternatively be accomplished with use of a remote control unit. The channel skip signal from the memory chip also enters chip 141 through the switch matrix 360.
The terminals of the array 360 are connected to a set of output buffers 366 and input buffers 368. The outputs from the input buffers 368 are applied to a five to three line encoder 370. The inputs of the output buffers 366 are connected to a two to four line decoder 372. The outputs of encoder 370 are applied to the Data C, D and E terminals of a five bit presettable up/down counter 374. The decoder 372 inputs are connected to D _A and D _B input terminals of the counter 374. The output of the counter 374 is applied to pins A-E and comprises the five bit output applied from the address generator 141 to the tuning memory 143 as shown in FIG. 8.
The outputs from the input buffers 368 are also applied to the inputs of an OR gate 376, the output of which is applied to load counter 374 and is applied through an invertor 378 to an input of a NAND gate 380. Gate 380 is connected in a latch configuration with a NAND gate 382. The output of NAND gate 382 is applied as an input to an OR gate 384, the output of which is applied through an invertor 386 to clock the counter 374. The output of gate 382 is also applied as an input to a NAND gate 388, the output of which is applied to an OR gate 390 which applies a clear signal to a three bit AFC sequence counter 392.
The A, B and C outputs of counter 392 are applied to a NAND gate 394, the output of which is applied as a clock through NAND gate 396 to counter 392. The C output of counter 392 is applied as a serial clock to the OR gate 384. The A and B outputs of counter 392 are applied through a NAND gate 398 and a NAND gate 400 and through an inverting buffer 402 to cause the generation of and AFC defeat signal.
An oscillator 404 generates clock signals which are applied to the clock input of a two bit scan counter 406. The NAND gate 388 generates a clear signal which is applied through gate 390 to the counter 392, and to the scan counter 406. The A and B outputs of counter 406 are applied to the decoder 372. The B output of counter 406 is applied as an input to gate 396, serving as the clock for counter 392.
Inputs and outputs of the output buffers 366 are applied through inverters as inputs to AND gates 408 and 410, the outputs of which are applied through inverters to an OR gate 412. The output of OR gate 412 is applied through an inverter to an input of gate 394. The output of oscillator 404 is applied through an inverter 414 to the inputs of NAND gates 416 and 418. The output of gate 416 is applied to an input of an OR gate 420 and to the input of a NAND gate 422. The output of gate 418 is applied to the second input of gate 420 and is applied to an input of a NAND gate 424. Gates 422 and 424 are interconnected in a latch configuration in order to provide up and down control signals to the counter 374.
In operation of the circuitry shown in FIG. 14, both sequential and parallel access channel selection may be accomplished. The local oscillator 404 drives the two bit scan counter 406 which is conventionally running. The two binary outputs from the counter 406 are applied to D _A and D _B terminals of the up/down counter 374 which are the parallel load data inputs of the counter 374. The outputs of the counter 406 are also applied to the two to four line decoder 372 which converts from the binary code to a single one out of four code. The outputs from the decoder 372 are applied to the output buffers 366.
The outputs of buffers 366 are high 25 percent of the time and are low 75 percent of the time. Thus, the four vertical lines of the pushbutton matrix array 360 are sequentially high when the system is in the parallel access mode. If the operator pushes one of the twenty buttons in the array 360, and one of the vertical lines is high, the corresponding horizontal line also goes high, which condition is applied through the input buffer 368 and through the five to three line encoder 370. The encoder 370 converts the signal into a binary code which corresponds to the three most significant bits of the address code and which is applied to the parallel load data inputs of the up/down counter 374. The gate 376 performs a NOR function on the zeroes on the inputs of the encoder 370 such that any input to the encoder 370 which goes low results in the corresponding three bit binary code being loaded into the counter 374. Additionally, at the same time, the two bit code applied at D _A and D _B is loaded into counter 374.
As long as the operator's finger remains on one of the pushbuttons in the array 360, a signal is generated through gates 376, 378 and 390 to clear the counter 392. Counter 392 and its associated circuitry thus operates in a similar manner as counter 202 in the circuit shown in FIG. 9. When the system is in the parallel mode, a logic one appears on line 430, and a zero thus appears at the output of gate 382. No clock signals are applied through gate 384 and inverter 386 to the up/down counter 374. The operation of the circuitry shown in FIG. 14 is thus similar to the operation of the circuitry shown in FIG. 9.
If it is desired to operate the system of FIG. 14 in the serial access mode, one of the buttons 362 or 364 is depressed in order to ground the respective vertical line through the array 360. Gate 408 or 410 detects the up/down mode by detecting the situation wherein the output buffer 366 commands the line B to go high, but when the down button 364 is depressed it cannot comply, thereby creating the down command. Similarly, gate 408 detects the up mode which occurs when the output buffer 366 commands the A line to go high, but the up button 362 is depressed, thereby grounding the line. The outputs of gates 408 and 410 are applied to gates 416 and 418 and Nanding with the narrow output pulse from the invertor 414 and the oscillator 404. The outputs of the gates 416 and 418 are applied to gate 420 which detects a logic zero on either of the gates in order to set the serial side of the latch comprising gates 380 and 382. Setting of the serial side of the latch enables clock signals to be applied through the gate 384 to the up/down counter 374 for operation of the circuit.
Other outputs of gates 408 are inverted and applied to gate 412 which detects a logic zero on either input. Detection of the logic zero by gate 412 causes the generation of the logic one which is inverted and utilized to momentarily depress, through gate 394, the application of clock signals to the counter 392. Unless counter 392 is cleared, the counter will attempt to start the AFC sequence. This operation just described is provided by the first pulse generated through the array 360 by operation of the up or down buttons 362 and 364.
In the serial mode, a latch comprising gates 380 and 382 applies a logic zero to line 430 in order to indicate to the decoder 372 that it is desired that all logic ones appear on the outputs A-D from the output buffers 366.
The latch comprising gates 380 and 382 enables application of clock pulses to counter 374 in the manner previously described. In addition, the latch opens gate 388, thereby enabling counter 392 and counter 406 to be cleared when receiving skip data. When no skip clock is available, gate 384 is opened in order to advance the counter 374 one count. Counter 374 thus operates to generate the five bit binary code on pins A-E.
If the channel selected by the operator is not a valid channel and it is desired to be skipped, a skip signal is applied through switch 434. The skip signal is large enough to overpower the circuitry shown in FIG. 14 in the same manner as the switches 362 and 364, such that the skip data flows back through line D of matrix 360 and line 436 to the OR gate 384. Gate 388 detects the skip signal and generates a logic zero, thereby reconstituting the skip signal in the serial mode. The logic zero from gate 388 clears the counter 392 and maintains the counter 406 in zero-zero state, preventing normal counting action. The latch comprising gates 380 and 382 has previously opened the gate 384 for operation in the serial channel selection or to receive the skip signals via lines 436.
When either the up button 362 or down button 364 is depressed, the counter 374 must be directed as to which direction to count. The output from either gate 416 or 418 sets one side of the latch comprising gates 422 and 424 in order to set the counter 374 in the desired count mode.
If the mode of operation of the circuit shown in FIG. 14 is changed from the serial access to the parallel access on a valid channel, no data is provided to reset counter 392. Counter 406 is scanning at this time, but is blocked by the decoder 372, and the action of line 430 from gate 380. When the operator presses one of the buttons in the array 360, the line is already high and is applied through the input buffers 368 to gate 376 in order to set the latch comprising gates 380 and 382 back to the parallel mode.
As noted, the selected vertical line in the array 360 is high 25 percent of the time and is low 75 percent of the time. The decoder 372 is noe open to decode the input data from the scan counter and to cause loading of the counter 374 with the correct data.
When the system shown in FIG. 14 is in the parallel access mode and the operator attempts to pick up an unprogrammed channel, the skip signal is applied to gate 384, which is not open and which prevents serial advance away from that channel. However, the system still performs the loading of data and gate 376 is energized such that the counter is loaded by a generated spike from gate 376. Thus, if the operator so desires, a channel that is not programmed may nevertheless be addressed.
Tuning Memory For Twenty Channel System
FIG. 15 is a schematic diagram of the tuning memory 143 (FIG. 8) for use with the circuitry shown in FIG. 14. The five binary address inputs generated from the circuitry shown in FIG. 14 or other means are applied to a five to 20 line decode 450 which applies the resulting word enable signals to a 12 bit 20 word tuning voltage RAM 452. The output of the RAM 452 is applied through multiplexing gates 453 to a 12 bit data shunt 454. The output of shunt 454 is applied to a 12 bit presettable ripple up/down counter 456. The output of counter 456 is applied to a 12 bit data comparator 458 which also receives the output from a 12 bit synchronous binary counter 460. The resulting output from the comparator 458 is applied through an invertor 461 to a D-flipflop 462 and an integrating filter 463 which generates the analog signal for control of the varactor diodes of the tuner 464.
A 1 MHz clock is applied to pin 465 to drive the synchronous binary counter 460. A signal of approximately 256 Hz is applied from the counter 460 to a count down frequency select circuit 466. The outputs from the circuit 466 apply a clock frequency to a tuning program generator 468 which applied clock and load signals to the counter 456 and which applies enable signals to the shunt 454. Operation of an up voltage programming button 470 and a down voltage programming button 472 operates a tuning mode timer 474 which generates fast or slow signals to the select circuit 466.
Output signals from the counter 456 are applied to the data multiplexing gates 453 for input to the RAM 452. Read or write signals are applied from the tuning program generator 468 to control the operation of the reading or writing of the RAM 452. The output from the counter 456 is also applied to a channel skip decode 478 which generates the channel skip output previously noted.
The decoder output from the address decode 450 is also applied to a band switch circuit 480 which generates three band switch control signals LVHF, HVHF and UHF which are applied to the varactor diodes in the manner previously described. In addition, when the output of the band switch 480 appears on a UHF output, a signal is applied via lead 482 to the frequency select circuit 466 in order to operate the slow clock mode of the select circuit 466.
Operation of the tuning memory of FIG. 15 is similar to the circuitry shown in FIG. 13, except that additional RAM storage is not required for band switching bits or skip bits. Thus, when it is desired to address signals to be stored in the RAM 452, a battery is connected to pin 486 to protect the memory of the RAM 452 and both buttons 470 annd 472 are depressed. Depression of both the buttons 470 and 472 for a brief period causes a shunt signal to be generated from the generator 468 to the 12 bit data shunt 454. This causes all logic zeroes to be applied to the output of the shunt 454. This action sets the RAM to the known all zero state. Subsequently, only the up button is depressed, and after a time interval such as 8 seconds, tuning mode timer 474 kicks into the fast mode and operates the count down frequency select 466 in the fast mode to apply a high clock frequency to the tuning program generator 468. When the first channel is detected by the operator, the operator removes the finger and the circuit moves into the slow mode such that the channel may be fine tuned by selective action of the tuning buttons. The tuning program generator then causes the desired address to be read and stored in the RAM 452.
When the operator desires to skip unused channels, both buttons 470 and 472 are depressed for a brief period at those channel locations and a zero appears on the output from the tuning program generator 468 and is detected by the 12 bit data shunt 454, causing twelve zeroes to appear as data. Therefore, all zeroes are entered into the RAM 452 at that location. The all zero data word generates a skip signal from the decoder 478 when that channel address is selected in subsequent operation.
As previously noted, the output words from the RAM 452 are applied through the shunt 454 to operate the counter 456. The outputs from counter 456 are applied to the comparator 458 along with the output from the counter 460. The comparator 458 converts the binary word stored in the RAM 452 into a duty cycle of a specific D.C. level. This duty cycle is detected by the flipflop 462 and is applied to the integrating filter 463 in order to generate the desired analog signal for control of the varactor diode tuner 464.
In order to more specifically illustrate the operation of the tuning memory shown in FIG. 15, FIGS. 16a and 16b illustrate a schematic logic diagram of the circuit shown in FIG. 15. An important aspect of the present invention is that the circuitry shown in FIG. 16 may be formed on a single semiconductor chip by the use of integrated injection logic techniques. A description of integrated injection logic may be found in the article "Integrated Injection Logic -- A New Approach to LSI", I.E.E.E. Journal of Solid-State Circuits, K. Hart and A. Slob, Vol. SC-7, No. 5, October, 1972. Thus, the five binary address inputs are applied to pins 500 and are applied through invertors to a five to twenty line decoder comprising NAND gates 502. The decoded outputs from the gates 502 are applied to a twenty word twelve bit RAM 504. For simplicity of illustration, the construction of RAM 504 is not completely shown, but one bit 506 of the RAM is illustrated in detail. The bit 506 is constructed from a plurality of interconnected transistors which generate WORD ENABLE, DATA IN, DATA OUT and DATA IN signals in the manner illustrated.
The output of the decode gates 502 are detected by NAND gates 508 and 510 in order to generate through buffers and through a NAND gate 512 the three band selection signals for the varactor diode tuner. These band selection signals control the selection of the low VHF, the high VHF and UHF bands.
The input and output from the RAM 504 is controlled by the data multiplexing gate 514 which comprises 12 stages of interconnected NAND gates 516 and 518. The read and write control signals are applied to the data multiplexing gates via lead 520, the read word being logic zero and the write word being logic one at the output of the invertor 522. The output from the data multiplexing gates is applied to the NAND gates 528 which comprise interconnections to counter 456. Counter 456 includes twelve stages each comprising a counter 526, NAND gates 528 and 530, AND gates 532 and 534 and OR gates 536. Up and down control signals are applied via lines 540 and 542. The up and down signals are generated by operation of the up and down voltage programming buttons 470 and 472 which are applied through NAND gates 544 and 546 connected in a latch configuration.
The twelve outputs from the counter 456 are applied to the twelve bit data comparator 458 (see FIG. 16a) which comprises twelve NAND gates 550. Inputs of NAND gates 550 also receive clock signals from the twelve bit synchronous counter 460 which comprises ten NAND gates 552 interconnected with twelve flipflops 554. Flip-flops 554 are driven by a 1 MHz clock signal applied to line 556.
The output from the gates 550 in the comparator 458 are applied to a NAND gate 558 which generates a data output on pin 560 which is applied to the D-flipflop and the integrator 463 which generates the analog signals of the invention.
The tuning program generator 468 comprises NAND gates 564 and 566 which receive up/down signals generated by voltage programming switches 470 and 472. The output of gate 564 is applied via lead 568 to the data shunt 454 (which is an input on NAND gates 528). The output of gate 566 is applied to D-flipflop 570, the outputs of which are connected to four interconnected flipflops 571. The Q and Q outputs of flipflops 571 are applied to NAND gates 5772, 573 and 574. The outputs of gates 572-574 generate clock, write/read and load signals. Outputs from the up and down switches 470 and 472 are also applied through a NAND gate 577 to a latch comprising NAND gates 578 and 580.
The count down frequency select circuit 466 comprises seven counter stages 590 which receive a signal of approximately 256 Hz from the counter 460 via lead 592. An output of approximately 2 Hz is applied via lead 594 to the input of a NAND gate 596. Other outputs from counter stages 590 are applied to the input of NAND gates 600, 602, 604 and 606. The outputs of gates 600-606 are applied to a NAND gate 608, the output of which applies a clock frequency via lead 609 to the D-flipflop 570.
A logic signal indicating that the UHF band has been selected by the output of gate 512 is applied via lead 610 to the inputs of gates 600 and 606. Indication of the presence of the UFH band causes the count down frequency select 466 to generate a slow clock frequency to the tuning program generator 468. The tuning program generator 468 is responsive to an up or down input from the output of gate 566 which allows a logic one on the clear line to start the count. After sixteen counts, or an 8 second delay, a logic one output is applied to lead 616 by the tuning mode timer, which comprises five stages 612, to shift the count down frequency select 466 to the fast mode for the coarse tuning feature.
Clock signals are applied from the count down frequency select 466 via leads 618 to the channel skip decode ciircuit 478. The clocks are applied to a NAND gate 620, the output of which is applied to a NAND gate 622. Outputs from the counter 456 are applied to the inputs of a NAND gate 624, the output of which is applied to a D-flipflop 626. The Q output of the D-flipflop 626 is applied to the NAND gate 622. The output from the gate 622 is buffered and is applied to pin 630 to generate the channel skip signal.
In operation of the tuning program generator 468 in FIG. 16b, if it is desired to skip a tuning position, the up and down buttons 470 and 472 are simultaneously depressed. The inverted outputs from the switches are logic ones and are applied to gate 564. Gate 577 also detects the logic ones and feeds, through an inverter, the latch comprising gates 578 and 580. The input of gate 578 is connected to the five-bit time delay counter comprising stages 612.
Due to the depression of the switches 470 and 472, a logic zero would be applied to the shunt line if not for the output of the latch gate 578. The tuning mode control counter 468 does not enter all zeroes until switches 470 and 472 have been depressed for a brief period, due to the clearing applied to the Q lines connected to the input of gate 578. After a predetermined time interval, the input of gate 578 goes to logic zero, thereby causing the output of gate 578 to become the logic one. Now the input conditions of gate 574 are satisfied and the shunt becomes enabled.
The latch comprising gates 578 and 580 debounces the signal from the switches 470 and 472. The outputs from switches 470 and 472 are sensed by gate 577. When either or both of the switches returns to ground, a logic one appears at the output of gate 577. The output is inverted to reset gate 580 of the latch, thereby removing a logic one from gate 564. This cycle is then required to start again during the next cycle of operation.
The counter stages 612 extend the fine tuning mode before going into the coarse mode. Gate 596 requires the counter to maintain counting in the fast mode after the 8 second time interval because of the feedback applied thereby. Gates 600-604 creat the generation of different fast and slow tuning speeds for UHF and VHF modes.
The operation of the circuits shown in FIGS. 15 and 16a-b may be further understood by reference to the waveforms shown in FIG. 17. FIG. 17a illustrates the depression of one of the switches 470 or 472. The waveform shown in FIG. 17b is the free clock signal generated from the NAND gate 566 and applied to the D-flipflop 570. The waveform shown in FIG. 17c is the Q output of the D-flipflop 570. The state diagram shown in FIG. 17d illustrates the states of the program counter comprising the four counter stages 571 in the tuning program generator 468. The waveform shown in FIG. 17e comprises the Q output of D-flipflop 571 which is applied to clear D-flipflop 570. The waveform of FIG. 17f is the load up/down signal generated by the output of NAND gate 572 and utilized to control the loading of the up/down counter 456. The waveform shown in FIG. 17g comprises the output of the NAND gate 573 which is applied via the clock line to the counter 456. The waveform shown in FIG. 17h comprises the RAM write signal which is applied from gate 574 to the data multiplexer gates 516 and 518 in order to command the RAM to write.
In normal operation, the shunt 454 is open and the counter 456 is continuously loaded and the RAM 504 is continuously read. When the write signal is generated as shown in FIG. 17h, the shunt is controlled and the load is disabled. The count up or down is generated as required and the write RAM, enable read, and enable load signals are generated by the tuning program generator 468.
When the waveform of FIG. 17a goes high due to depression of one of the buttons 470 or 472, and the free clock signals shown in FIG. 17b are high, the leading edge of the waveform of 17b transfers the data of FIG. 17a to the Q output of the flipflop 570. After a transition time of τ ₁ as shown in FIG. 17c, the time occurs for shifting of data.
Referring to the state diagram of FIG. 17d, during preset, the first three flipflops 571 are held high as shown in state 7. After the time interval τ ₂ , to enable operation of the flipflops, the next 500 KHz clock that is applied causes state transition to state zero as shown in FIG. 17d. Referring to FIG. 17f, the state seven is decoded by gate 572 which goes high to isolate the RAM from the up/down counter. The preset counter is then no longer parallel loading and is ready to accept the clock signals. State one is now decoded by gate 573 and the up/down counter is clocked with the data. States two, three and four shown in FIG. 17d are skipped to provide ripple time for the counter and state five is decoded by gate 574 (FIG. 17h) to execute the write instruction for the RAM. State six is not decoded to provide debounce protection. As soon as the circuitry enters state seven, gate 572 goes low and the inverted output of gate 572 goes high.
Referring to FIG. 17e, the output of the last flipflop 571 was high. After a τ ₃ delay, the output of the last flipflop 571 now goes low because of the inverted output from gate 572. This causes flipflop 570, after a time delay of τ ₄ , to provide a low output which is utilized to preset the remaining flipflops. Now the last flipflop 571, whose output just went low, after a time interval of τ ₅ goes high. This removes the clear from flip-flop 570 and the state of the circuitry recycles to the beginning.
If the operator still has his finger on one of the buttons 470 or 472, the next time that the clock shown in FIG. 17b goes high, the entire cycle is repeated. If the operator removes his finger from the buttons 470 or 472, the set is now fine tuned and the counter comprising the four stages 571 remains at state 7.
During a channel skip operation, whenever a logic one is applied on the Q output of the flipflop 626 in response to a zero from gate 624, the D-flipflop is clocked from the line 618 by the 32 Hz clock at the leading edge thereof. The Q output of the D-flipflop 626 follows the leading edge of the clock signals and follows the data from gate 624. The output of D-flipglop 626 is Nanded at gate 622 with the output of gate 620 and inverted to provide the desired output waveform. The resulting output comprises a negative going narrow pulse which is delayed to allow the RAM to settle, thereby enabling time for the circuitry to change the program counter. The remainder of the operation of the schematic circuitry shown in FIG. 16 is apparent from the previous description of FIGS. 15 and 13.
Neon Display for 16 Channel Tuning System
for driving the various channel-number displays in response to the signal generated by the address generators is disclosed in the U.S. patent application Ser. No. 508,968 filed Sept. 25, 1974 and assigned to the same assignee as the present application.
It will thus be seen that the present invention provided an improved solid state television tuning system. The present system is modular, thus enabling easy interchange of displays or channel selection switches without modification of the remainder of the circuitry. The present system does not require expensive potentiometers or bulky printed circuit board fabrication. Although the present invention has been described with respect to the tuning of the television set, it will be understood that the present circuitry is also useful for tuning of other broadcast receivers such as radio, cable television and the like. The circuitry shown in FIG. 13 required RAM storage of two bit words to provide band switch information. As noted, this band switch information is programmed by actuation of switches by the operator during initial programming of the circuitry. Another aspect of the present invention is a system wherein such programming by the operator is not required.
In such a system, a band switch logic circuit such as a band switch circuit 480 shown in FIG. 15, is connected to the circuitry of FIG. 19 to detect the contents of the ROM 704 and the RAM 706. The band switch logic circuitry would thus decode portions of the discrete address generated from the ROM 704 and RAM 706 to generate the band switching signals which are applied to the varactor diode tuner in the manner previously described. The logic circuitry could consist of simple comparison logic of two BCD decades of the display memory word, and could comprise as few as two OR gates and a connected NOR gate.

Whereas the present invention has been described with respect to specific embodiments thereof, it will be understood that various changes and modifications will be suggested to one skilled in the art, and it is intended to encompass such changes and modifications as fall within the scope of the appended claims. 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

CRT   TV EHT VOLTAGE MULTIPLIER - KASKADE COCKCROFT-WALTON CASCADE CIRCUIT FOR VOLTAGE MULTIPLICATION:

A Cockcroft-Walton cascade circuit comprises an input voltage source and a pumping and storage circuit with a series array of capacitors with pumping and storage portions of the circuit being interconnected by silicon rectifiers, constructed and arranged so that at least the capacitor nearest the voltage source, and preferably one or more of the next adjacent capacitors in the series array, have lower tendency to internally discharge than the capacitors in the array more remote from the voltage source.


1. An improved voltage multiplying circuit comprising,

2. An improved voltage multiplying circuit in accordance with claim 1 wherein said first pumping capacitor is a self-healing impregnated capacitor which is impregnated with a high voltage impregnant.

3. An improved voltage multiplying circuit in accordance with claim 1 wherein said first pumping capacitor comprises a foil capacitor.

Description:
BACKGROUND OF THE INVENTION

The invention relates in general to Cockcroft-Walton cascade circuits for voltage multiplication and more particularly to such circuits with a pumping circuit and a storage circuit composed of capacitors connected in series, said pumping circuits and storage circuit being linked with one another by a rectifier circuit whose rectifiers are preferably silicon rectifiers, especially for a switching arrangement sensitive to internal discharges of capacitors, and more especially a switching arrangement containing transistors, and especially an image tube switching arrangement.

Voltage multiplication cascades composed of capacitors and rectifiers are used to produce high D.C. voltages from sinusoidal or pulsed alternating voltages. All known voltage multiplication cascades and voltage multipliers are designed to be capacitance-symmetrical, i.e., all capacitors used have the same capacitance. If U for example is the maximum value of an applied alternating voltage, the input capacitor connected directly to the alternating voltage source is charged to a D.C. voltage with a value U, while all other capacitors are charged to the value of 2U. Therefore, a total voltage can be obtained from the series-connected capacitors of a capacitor array.

In voltage multipliers, internal resistance is highly significant. In order to obtain high load currents on the D.C. side, the emphasis in the prior art has been on constructing voltage multipliers with internal resistances that are as low as possible.

Internal resistance of voltage multipliers can be reduced by increasing the capacitances of the individual capacitors by equal amounts. However, the critical significance of size of the assembly in the practical application of a voltage multiplier, limits the extent to which capacitance of the individual capacitors can be increased as a practical matter.

In television sets, especially color television sets, voltage multiplication cascades are required whose internal resistance is generally 400 to 500 kOhms. Thus far, it has been possible to achieve this low internal resistance with small dimensions only by using silicon diodes as rectifiers and metallized film capacitors as the capacitors.

When silicon rectifiers are used to achieve low internal resistance, their low forward resistance produces high peak currents and therefore leads to problems involving the pulse resistance of the capacitors. Metallized film capacitors are used because of space requirements, i.e., in order to ensure that the assembly will have the smallest possible dimensions, and also for cost reasons. These film capacitors have a self-healing effect, in which the damage caused to the capacitor by partial evaporation of the metal coating around the point of puncture (pinhole), which develops as a result of internal spark-overs, is cured again. This selfhealing effect is highly desirable as far as the capacitors themselves are concerned, but is not without its disadvantages as far as the other cirucit components are concerned, especially the silicon rectifiers, the image tubes, and the components which conduct the image tube voltage.

It is therefore an important object of the invention to improve voltage multiplication cascades of the type described above.

It is a further object of the invention to keep the size of the entire assembly small and the internal resistance low.

It is a further object of the invention to increase pulse resistance of the entire circuit.

It is a further object of the invention to avoid the above-described disadvantageous effects on adjacent elements.

It is a further object of the invention to achieve multiples of the foregoing objects and preferably all of them consistent with each other.

SUMMARY OF THE INVENTION

In accordance with the invention, the foregoing objects are met by making at least one of the capacitors in the pumping circuit, preferably including the one which is adjacent to the input voltage source, one which is less prone to internal discharges than any of the individual capacitors in the storage circuit.

The Cockcroft-Walton cascade circuit is not provided with identical capacitors. Instead, the individual capacitors are arranged according to their loads and designed in such a way that a higher pulse resistance is attained only in certain capacitors. It can be shown that the load produced by the voltage in all the capacitors in the multiplication circuit is approximately the same. But the pulse currents of the capacitors as well as their forward flow angles are different. In particular, the capacitors of the pumping circuit are subjected to very high loads in a pulsed mode. In the voltage multiplication cascade according to the invention, these capacitors are arranged so that they exhibit fewer internal discharges than the capacitors in the storage circuit.

The external dimensions of the entire assembly would be unacceptably large if one constructed the entire switching arrangement using such capacitors.

The voltage multiplication cascade according to the invention also makes it possible to construct a reliably operating

arrangement which has no tendency toward spark-overs, consistent with satisfactory internal resistance of the voltage multiplication cascade and small dimensions of the entire assembly. This avoids the above cited disadvantages with respect to the particularly sensitive components in the rest of the circuit and makes it possible to design voltage multiplication cascades with silicon rectifiers, which are characterized by long lifetimes. Hence, a voltage multiplication cascade has been developed particularly for image tube circuits in television sets, especially color television sets, and this cascade satisfies the highest requirements in addition to having an average lifetime which in every case is greater than that of the television set.

A further aspect of the invention is that at least one of the capacitors that are less prone to internal discharges is a capacitor which is impregnated with a high-voltage impregnating substance, especially a high-voltage oil such as polybutene or silicone oil, or mixtures thereof. In contrast to capacitors made of metallized film which have not been impregnated, this allows the discharge frequency due to internal discharges or spark-overs to be reduced by a factor of 10 to 100.

According to a further important aspect of the invention, at least one of the capacitors that are less prone to internal discharges is either a foil capacitor or a self-healing capacitor. In addition, the capacitor in the pumping circuit which is adjacent to the voltage source input can be a foil capacitor which has been impregnated in the manner described above, while the next capacitor in the pumping circuit is a self-healing capacitor impregnated in the same fashion.

Other objects, features and advantages of the invention will be apparent from the following detailed description of preferred embodiments, taken in connection with the accompanying drawing, the single FIGURE of which:

BRIEF DESCRIPTION OF THE DRAWING

is a schematic diagram of a circuit made according to a preferred embodiment of the invention.

DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS

The voltage multiplier comprises capacitors C1 to C5 and rectifiers D1 to D5 connected in a cascade. An alternating voltage source UE is connected to terminals 1 and 2, said voltage source supplying for example a pulsed alternating voltage. Capacitors C1 and C2 form the pumping circuit while capacitors C3, C4 and C5 form the storage circuit.

In the steady state, capacitor C1 is charged to the maximum value of the alternating voltage UE as are the other capacitors C2 to C5. The desired high D.C. voltage UA is picked off at terminals 3 and 4, said D.C. voltage being composed of the D.C. voltages from capacitors C3 to C5. Terminal 3 and terminal 2 are connected to one pole of the alternating voltage source UE feeding the circuit, which can be at ground potential. In the circuit described here, a D.C. voltage UA can be picked off whose voltage value is approximately 3 times the maximum value of the pulsed alternating voltage UE. By using more than five capacitors, a correspondingly higher D.C. voltage can be obtained.

The individual capacitors are discharged by disconnecting D.C. voltage UA. However, they are constantly being recharged by the electrical energy supplied by the alternating voltage source UE, so that the voltage multiplier can be continuously charged on the output side.

According to the invention, in this preferred embodiment, capacitor C1 and/or C2 in the pumping circuit are designed so that they have a lower tendency toward internal discharges than any of the individual capacitors C3, C4 and C5 in the storage circuit.

It is evident that those skilled in the art, once given the benefit of the foregoing disclosure, may now make numerous other uses and modifications of, and departures from the specific embodiments described herein without departing from the inventive concepts. Consequently, the invention is to be construed as embracing each and every novel feature and novel combination of features present in, or possessed by, the apparatus and techniques herein disclosed and limited solely by the scope and spirit of the appended claims.

Inventors:Petrick, Paul (Landshut, DT)
Schwedler, Hans-peter (Landshut, DT)
Holzer, Alfred (Schonbrunn, DT)
ERNST ROEDERSTEIN SPEZIALFABRIK

US Patent References:
3714528    ELECTRICAL CAPACITOR WITH FILM-PAPER DIELECTRIC    1973-01-30    Vail    
3699410    SELF-HEALING ELECTRICAL CONDENSER    1972-10-17    Maylandt    
3463992    ELECTRICAL CAPACITOR SYSTEMS HAVING LONG-TERM STORAGE CHARACTERISTICS    1969-08-26    Solberg    
3457478    WOUND FILM CAPACITORS    1969-07-22    Lehrer    
3363156    Capacitor with a polyolefin dielectric    1968-01-09    Cox    
2213199    Voltage multiplier    1940-09-03    Bouwers et al.