NORDMENDE SPECTRA COLOR SC1032 CHASSIS F6TT Controlled power supply for a television receiver equipped with remote control:BLAUPUNKT SWITCH MODE POWER SUPPLY.Blaupunkt-Werke GmbH (Hildesheim, DT)
NORDMENDE SPECTRA COLOR Chassis F6TT Power Supply view.SMPS supply with S2530A (TOSHIBA) Chassis F6TT Power Supply view.
1. A power supply circuit for a television receiver equipped for remote control comprising, in combination:
an
on-off switch for connecting and disconnecting the television
receiver and its power supply circuit respectively to and from the
electricity supply mains;
pulse generating means arranged for energization through said on-off switch;
an isolation transformer having its primary winding supplied with the output of said pulse generating means;
a
power conversion circuit connected to the secondary winding of
said isolation transformer for energization thereby, for supplying
an operating voltage for the scanning circuits of the television
receiver and for supplying a plurality of other voltages to
said receiver, at least one of which other voltages is also
supplied to said scanning circuits;
a
remote control signal receiver for remote control of said
television receiver and controlled switching means responsive to said
remote control receiver for switching said television receiver
between a stand-by condition and an operating condition, both
said remote control receiver and said controlled switching means
being connected to a secondary winding of said isolation
transformer for energization thereby, said controlled switching
means having a switching path for connecting and disconnecting
said scanning circuits of said television receiver respectively
to and from a source of said operating voltage in said power
conversion circuit and
means for reducing energy transfer
through said pulse generating means to said isolation
transformer when said television receiver is in the stand-by
condition.
2. A power supply circuit as defined in claim 1,
in which said pulse generating means includes rectifying means
energized through said on-off switch for supplying direct
current for energization of said pulse generating means.
3. A power supply circuit as
defined in claim 2, in which said energy transfer reducing means
includes means for varying the width (duration) of pulses
generated by said pulse generating means in response to the
extent of loading of the secondary circuit of said isolating
transformer as measured in the primary circuit of said transformer.
4. A power supply circuit
as defined in claim 2, in which said pulse generating means
includes a blocking oscillator and said energy transfer reducing
means includes means for reducing the width (duration) of the
pulses generated by said blocking oscillator.
5. A power supply circuit as defined in claim
4, in which said blocking oscillator includes a switching
transistor (5) and a load measuring resistor (7) interposed in a
connection between the emitter of said switching transistor and
the receiver chassis, and in which said pulse width reducing
means is responsive to the voltage drop across said load
measuring resistor. 6. A
power supply circuit as defined in claim 5, in which said pulse
width reducing means includes a controllable resistance (10) in
the circuit of said blocking oscillator controlled in response to
the voltage drop across said load measuring resistor.
7. A power supply circuit as defined
in claim 1, in which said operating voltage connected and
disconnected to said scanning circuits by said controlled
switching means is the low voltage supply voltage (U 3') of the
line scan and picture scan oscillators of the television receiver
and in which said controlled switching means is controlled so as
to switch off said low voltage supply voltage to put the
television receiver in the stand-by condition.
8. A power supply circuit as defined in claim
7, in which said controlled switching means includes a first
switching transistor (15) at the collector of which there is
applied a direct current supply voltage (U 3) energized through
said isolating transformer and a second switching transistor
(24) for controllably short-circuiting the base bias of said
first switching transistor, whereby a stabilized low voltage (U
3') exists at the emitter of said first switching transistor
(15) when a positive signal is supplied from an operating control
of the television receiver or from said remote control receiver
to the base of said second switching transistor (24).
9. A power supply circuit as defined
in claim 7, in which said controlled switching means is
responsive independently to an overcurrent condition in the
picture tube for switching off said low voltage supply voltage
(U 3') in response to said overcurrent condition.
Description:
In recent times television receivers have frequently been provided with ultrasonic remote control devices for the purpose of offering easier control. As more and more television receivers are utilized in combination with additional equipment, it becomes increasingly necessary to connect the receivers only indirectly to the electric power mains (house wiring). In a known advantageous solution of this problem, a power supply unit includes an isolating transformer which is wired up with a blocking oscillator in the primary circuit. The blocking oscillator is supplied with a d-c voltage which is obtained by rectification of the supply voltage. Compared to the isolating transformers which are directly mains-operated, these so-called switch-mode power supply units have the advantage that they can be made in considerably smaller size, as they are operated at a significantly higher frequency, and the further advantage that they require less expensive means for rectification.
It is necessary to supply television receivers equipped with ultrasonic remote control with the possibility for a stand-by operation in which only the ultransonic receiver is supplied with power and, in some cases, also the heating current for the picture tube. Usually a separate power supply unit is provided for the ultrasonic receiver and the heating of the picture tube, a unit that includes an isolating transformer of its own, the primary winding of which is directly mains-fed. Upon transition from normal operation to stand-by operation, the power supply unit of the blocking osciallator is switched off, so that the television receiver receives only the relatively small quantity of energy required for the ultrasonic receiver and, in some cases, also for the heating of the picture tube.
Because of the required second isolating transformer, this known circuit has the disadvantages that it requires both greater space and greater expenditure.
It is the object of the present invention to develop a simplified power supply unit which does not have the above-mentioned disadvantages.
SUMMARY OF THE INVENTION
Briefly, the television receiver and the ultrasonic receiver are connected to the same isolating transformer; means for the switching from normal operation to stand-by operation and vice versa are placed in the secondary circuit of the isolating transformer, and means are arranged in the primary circuits of the isolating transformer for reducing the amount of energy made available for stand-by operation purposes.
The main advantages of the present invention are that no separate isolating transformer is required for supplying the current during the stand-by operation, and that, during the stand-by operation, it is nevertheless only the power required for this operation which is consumed.
An advantageous embodiment of the present invention obtains reduction of the energy quantum transmitted through the power supply during stand-by by reduction of the pulse width of the pulses generated by the blocking oscillator.
Another advantageous embodiment of the present invention utilizes measurement in the primary circuit of the isolating transformer of variation in load occurring in the secondary circuit as a control variable for determining the pulse width.
A further advantageous embodiment of the present invention obtains the control variable for the pulse width across a measuring resistor interposed in the connection of the emitter of the switching transistor of the blocking oscillator to the chassis.
Still another advantageous embodiment of the present invention provides that the voltage drop across the measuring resistor controls a controllable resistor.
The advantageous embodiments described above offer highly simple and advantageous possibilities for measuring the variation in load upon switching between normal and stand-by operation, as well as for the consequent control of the energy transmitted via the isolating transformer.
The possibility of a simple and inexpensive switching between normal and stand-by operation is achieved by effecting the switching between normal and stand-by operation by means of switching on or switching off, respectively, the low voltage supply of the line scan oscillator, and, especially, by a first switching transistor which short-circuits the base bias of a second switching transistor at the collector of which a direct current supply voltage is present and at the emitter of which a stabilized low voltage exists, when a positive signal is supplied from the operating control of the television receiver or from the remote control receiver to the base of the first switching transistor.
The circuit arrangements just mentioned offer the advantage that they may simultaneously be utilized as a protective circuit. This is achieved by a switching-off device for the low voltage which can also be triggered at any time by a signal built up by overcurrent in the picture tube.
BRIEF DESCRIPTION OF THE DRAWINGS
The invention is further described by way of illustrative example by reference to the annexed drawings in which:
FIG. 1 is a circuit diagram, partly in block form, of an embodiment of the invention;
FIG. 2 is a circuit diagram of one form of means for interrupting the power to the picture circuits in the stand-by condition in connection with the circuit of FIG. 1, and
FIG. 3 is a circuit diagram of one way of controlling the pulse width of the blocking oscillator 4 in response to the switching circuit 8 in the circuit of FIG. 1.
DESCRIPTION OF THE PREFERRED EMBODIMENT
An
on-off power switch 2 of the television receiver is connected
to the supply terminals 1, providing a primary operating control
for the receiver. Consquently, the supply voltage is also
present at the output of the operating control 2 when the
television receiver is turned on thereby, and arrives at a
rectifying stage 3 comprising means for rectifying and smoothing
the supply current as well as for suppressing interference. A
d-c voltage, feeding a blocking oscillator stage 4, is present
at the output of the recifying stage 3. The main part of the
blocking oscillator 4, symbolically represented in FIG. 1 by a
fragmentary circuit diagram, is a switching transistor 5, in the
load circuit of which the primary winding of an isolating
tranformer 6 is placed. A measuring resistor 7 is connected
between the emitter of the switching transistor 5 and the
chassis, across which measuring resistor a voltage is taken and
applied to a load-dependent control circuit 8. The voltage taken
at the measuring resistor 7 is fed via a resistor 9 to the base
of a transistor 10 which serves as a controllable load for the
blocking oscillator 4. A resistor 11 and a capacitor 12, each of
which is connected to chassis with its other terminal, are also
connected to the base of the transistor 10. The emitter of
transistor 10 is connected to chassis, while the collector of
the transistor 10 is connected back to the blocking oscillator
stage 4.The base of the transistor 15a is connected to a switching stage 23 responsive to a remote control ultrasonic receiver by a conductor leading to the collector of a switching transistor 24 which is connected to chassis via its emitter. The base of the switching transistor 24 is connected to an input terminal 28 leading into the television receiver via two resistors 25, 26 and a capacitor 27 connected in series, that input terminal 28 passing on switching signals from the receiver to the switching transistor 24, as will be explained in more detail below.
The cathode of a diode 29, which is connected to chassis via its anode, is connected to the junction point of the resistor 26 and the capacitor 27. The junction point of the two resistors 25, 26 is connected to chassis via a capacitor 30. The base of the switching transistor 24 is connected to chassis via a resistor 31. Furthermore, that base electrode is also connected to a terminal 32 to which an electrical switching signal is applied which is either built up in response to an ultrasonic signal received by the remote control receiver 32' or is supplied from an operating control of the television receiver. At the terminal 32, the switching transistor 24 receives the signal containing the information whether the television receiver is to work in the normal operating condition, i.e. to receive and process the sound and video signals, or in the stand-by condition in which it is substantially only the ultrasonic receiver that is supplied with current.
When a positive signal arrives at the base of the switching transistor 24, the latter becomes conductive, and causes chassis potential to be present at the base of transistor 15a. The transistor 15 is thereby blocked, and there is no longer any voltage at the terminal U 3'. Since the voltage U 3' serves as an operating voltage for the line and picture scan oscillator, the deflecting stages of the receiver cannot work and no high voltage and other related supply voltages are generated at the line circuit transformer. In consequence, by means illustrated diagrammatically in
FIG.
2, the electric circuits connected to the terminals U 1 to U 3
are interrupted. The voltages U 4 and U 5 serve for supplying
the ultrasonic receiver, i.e. they are required for the stand-by
operation.In case no counteracting means should be provided for, the variation in load would cause a voltage rise in the secondary circuit of the isolating transformer 6, which effect is, of course, not desired. Therefore, a measuring resistor is connected in the primary circuit in the emitter line of the switching transistor 5 of the blocking oscillator 6, the variation in load in the secondary circuit appearing at the measuring resistor 7 as a current variation. The current change thus produced, causes a variation in the base bias of the transistor 10, the capacitor 12 having an integrating effect to avoid undesired effects due to interference pulses and abrupt load fluctuations.
The change of the working point of the transistor 10 causes a change in the pulse width in the blocking oscillator stage 4, as more fully shown in FIG. 3, so that the energy quantum transmitted via the isolating transformer 6 is such that the required voltages are present in the secondary circuit. It should also be mentioned that the load-dependent switch 8 and the circuit of FIG. 3 are represented only by way of illustration and that many circuit arrangements may be devised by straight-forward application of known principles for controlling the pulse width.
Using the circuit diagram according to FIG. 3 it is possible in a simple manner to control the pulse width of the blocking oscillator 4 in response to the switching circuit 8.
According to the circuit diagram of FIG. 2 the terminal U1 is connected to a line scan oscillator circuit 40, the terminal U2 to a picture scan oscillator circuit 41 and the terminal U3 to a circuit 42 for a sound output stage. The circuits 40, 41, 42 get their operating voltage from the terminal U3'. If the operating voltage U3' is zero, the circuits 40, 41, 42 are interrupted. In this case the voltages at the terminals U1, U2, U3 remain.
The described circuit of this invention for controlling the voltage in the secondary circuit of the isolating transformer 6 offers the advantage that it is exclusively arranged in the primary circuit, and, therefore, permits an uncomplicated design which is easy to realize. To control the pulse width by measuring the load fluctuations at the low-end resistor of the switching transistor 5, represents a very useful means for control since, thereby the transmitted energy can effectively and easily be controlled.
The blocking oscillator stage 4 shown in detail in FIG. 3 incorporates an externally triggered blocking oscillator arranged to be triggered through an oscillator operating preferably at the line scanning frequency, which is to say its wave form is not particularly critical and it should be provided with means to keep it in step with the line scanning frequency, as is known to be desirable. The transistors 51 and 52 of the triggered output stage of the blocking oscillator circuit could be regarded as constituting a differential amplifier the inputs of which are defined by the base connections of the respective transistors 51 and 52. The input voltage applied to the base connection of transistor 52 is the Zener voltage of the Zener diode 53, thus a constant reference voltage. The operating voltage for the transistors 51 and 52 and for the Zener diode 53 is obtained from the supply voltage UB, which is to say from the rectifier 3. The diode 67 protects the transistor 52, for example at the time of the apparatus being switched on, against damage from an excessively high emitter-base blocking voltage. The capacitor 65 prevents undesired oscillation of the circuit of transistors 51 and 52, which could give rise to undesired disturbances.
At the base of the transistor 51, there is present as input voltage for the circuit a composite voltage that is the sum of three voltages. These are, first, the line scan frequency trigger voltage coupled through the capacitor 63; second, a bias voltage dependent upon the loading of the blocking oscillator stage resulting from the load on the secondary of the transformer 6, but detected by the voltage across the resistor 7 and actually controlled by the load-sensitive control circuit 8, and, third, a regulating voltage applied at the terminal 71 of the resistor 70, which regulating voltage is proportional to the voltage of the secondary winding of the transformer 6 and can accordingly be provided by one or another of the output circuits of the rectifier 14 of FIG. 1 or by a separate winding of the transformer 6 and a separate rectifier element connected in circuit therewith. This regulating voltage and the control voltage provided by the control circuit 8 are applied to the resistor 61 which completes the circuit for both of these bias voltages and their combined effect constitutes the bias voltage for the transistor 51 which determines its working point.
The circuit of the transistors 51 and 52 operates as an overdriven differential amplifier. When the trigger voltage exceeds the threshold determined by the base voltage of the transistor 51, the circuit produces an approximately rectangular output voltage pulse of constant amplitude. Since the trigger voltage is recurrent, the result is a periodic succession of rectangular output voltage pulses, but the duration or pulse width of these pulses depends upon the loading and the output voltage of the stage. The output voltage of the circuit constituted by the transistors 51 and 52 comes from the emitter connection of the transistor 52 and is furnished to the switching transistor 5, preferably through a driver stage 54, such as a transformer or another transistor stage for better matching of the circuit impedances. Of course, the collector circuit of the transistor 5 includes the primary winding of the transformer 6 of FIG. 1.
NORDMENDE Chassis F6TT
Detail viewing:
- Line deflection output + EHT + Tripler
- Line deflection output Transistor BU208 (Telefunken)
- Frame deflection output Unit in place + N/S Raster correction components circuitry.
- Signal processing section - IF + Sound + Synch + Luminance + Chrominance + RGB Amplifiers
CRT Line Output Stage Operation Principle: I'll examine the operation of the line output stage, whose basic job is to generate a sawtooth current in the line scan coils so that the beams are deflected horizontally across the picture tube's screen. The beams are deflected from the left-hand side to the right-hand side to give the forward line scan: this is followed by a rapid, blanked flyback to the left-hand side ready to trace out the next viewed line. Because of the way in which the flyback is achieved, the line output transformer generates various pulse voltages which are rectified to produce the e.h.t. required by the tube and other supplies. The line output stage is not just any sort of amplifier. The active device, almost always a transistor though valves, thyristors and gate -controlled switches have been used in the past, operates as a switch, the inductive components in the stage being mainly responsible for generating the sawtooth current waveform. Tuning is used to generate and control the flyback. The line drive waveform controls the output transistor's on/off switching and thus determines the timing of the cycle of operations, keeping them phase synchronised with the transmitted picture signal.
Basic Operation
Fig. 1 shows in most basic form the main elements in the line output stage, the active device (transistor) being shown as a switch. When the switch is closed, capacitor C and diode D are shorted out and the 150V supply is connected across coil L. Now it's a basic law of inductance that when a d.c. voltage is connected across a coil the current flowing through the coil builds up linearly from zero. Fig. 2(a) shows this as a positive -going ramp that starts at time t 1 , when the switch is closed. After about 26psec (t2), roughly the time required to deflect the beams from screen centre flows via the large -value capacitor CR, charging the tuning capacitor C with the result that the voltage at its 'upper' plate (the one connected to the coil) rises to a relatively high positive value. When all the energy in coil L has been transferred to capacitor C (time t3) the latter begi
ns
to discharge, passing the energy back the other way to L via CR which,
as far as the circuit's a.c. operation is concerned, can be regarded as a
short-circuit. At time t4 the capacitor has discharged, having
transferred the energy back to the coil. This to-and-fro interchange of
energy between L and C, which from the a.c. point of view are in
parallel (CR representing a short-circuit), is the normal action of a
tuned/resonant/oscillatory circuit. The resonant frequency is determined
by the values of L and C. These are selected so that when time t4 is
reached, i.e. after a half cycle of oscillation, the sawtooth current
has passed through zero to a negative point on the ramp and the beams
have been deflected to the left-hand side of the screen ready for the
next active line scan. To complete the oscillatory cycle (the normal
resonant circuit action) the voltage at the upper plate of capacitor C
would have to move negatively with respect to chassis. It can't do so
because of the presence of diode D, which is called the efficiency diode
- we'll explain that in a minute. When the voltage at the cathode of D
tries to swing negatively it conducts, i.e. switches on, providing a
discharge path for the coil. Once again because of the inductance in the
circuit there's a gradual, linear current discharge, the enegery being
returned to the supply's reservoir capacitor CR. During this discharge,
the beams are deflected back towards the centre of the screen (times t4
to t5). At this point the magnetic flux (energy) in L has been
dissipated. C is still in its discharged state, being shorted out by
diode D. So at time t5, with the beams at screen centre (zero
deflection), the switch has to be closed so that the cycle of operation
can be repeated. The action of diode D has, with the inductance in the
circuit, provided half the scan power while in the process returning the
energy (minus inevitable circuit losses) to the reservoir capacitor. No
wonder it's called the efficiency diode. It's important to note that
the beam flyback period t2 to t4 is governed by the time -constant of L
and C, consisting of one half cycle of oscillation. To achieve a flyback
time of 12μsec the duration of one cycle needs to be 24μsec: so the
resonant frequency of L and C works out at 41.67kHz. Fig. 3 illustrates
the four phases in the operation of the line output stage. Now the
voltage developed across an inductor is propor- tional to the rate of
change of the current flowing through it. Thus the voltage across L is
relatively low during the forward scan period but correspondingly high
during the flyback, when the current flow is faster because of the
circuit resonance. The voltage developed at the positive plate of
capacitor C is shown in Fig. 2(b), typically peaking at 1,200V. Both the
line output transistor and the efficiency diode must be capable of
withstanding this high reverse voltage. As we've seen, the circuit
action is highly efficient as the energy stored in L is returned to the
supply during the first half of the forward scan: indeed with 'perfect'
components there would be no net demand on the power supply at all! In
practice because of the resistance of the inductor and the losses in the
diode, switch and capacitor the circuit takes out a little more than it
puts back, while the practice of loading the transformer with rectifier
circuits to provide power for other sections of the set increases the
stage's current demand. To make up for these losses, the line output
transistor is switched on slightly before instead of at the centre of
the forward scan. In a practical circuit L is the primary winding of the
line output transformer and the deflection coils are connected across
it via a d.c. blocking capacitor, CB, as shown in Fig. 4. This coupling
capacitor also provides scan -correction (often referred to as S
-correction). Why is this required? If a linear deflection current was
used to control the scanning with a relatively flat -faced picture tube
the sides of the picture would be stretched out in comparison with the
centre section. Hence S -correction: the value of the coupling capacitor
is chosen so that it resonantes with the inductance of the scan coils
at about 5kHz. This has the effect of adding a sinewave component to the
sawtooth current, as shown in
Fig. 5. Thus the deflection power is tailored to suit the length of the
beam paths as the screen is scanned, correcting the horizontal
linearity of the display. At the line scanning frequency the scan coils
behave as an almost perfect inductor, but their small d.c. resistance is
in series with the fixed voltage that should be present across the
coil. It has the effect of introducing an asymmetric sensitivity loss
during the forward scan. To counteract it a further component is added
in series with the scan coils - an inductor with a saturable magnetic
core, biased by a permanent magnet so that its inductance falls as the
scan current increases. The voltage drop across this inductor, which is
known as the linearity coil, varies in the opposite sense to that
produced by the resistance of the coils, thus providing an equal -but
-opposite cancellation effect. In some TV sets the permanent magnet can
be adjusted to trim the linearity correction, though many modern sets
use components with such tight tolerances that a sealed linearity
-correction coil can be used. With some very small -screen sets the
horizontal non -linearity effect is small enough to be ignored.Practical Line Output Stage
Fig. 6 shows a relatively simple line output stage circuit used with a 90° -deflection tube. Tr5 is the line output transistor, which incorporates the efficiency diode in the same package. The primary winding of the line output trans- former T4 is the section between pins 2 and 10, C95 being the flyback tuning capacitor. Scan coil coupling and S - correction are provided by C94, the line linearity coil L14 being connected in series on the chassis side of the scan current path. L14 is damped by R110 to prevent it ringing when the line flyback pulse occurs - the effect of an undamped linearity coil is velocity modulation of the beams at the beginning of their sweeps, showing up as black -and - white vertical striations at the left-hand side of the screen. C92 is the reservoir capacitor, the h.t. feed being via 8105. 8106 and R109 feed pulses to the second phase -locked loop (APC2) in the sync chip - we dealt with this in last month's instalment. A second pulse feed from the same point goes to the colour decoder chip to provide line blanking, burst gating and PAL switch drive - this particular set doesn't use the sandcastle pulse approach.
Secondary Supplies
So much for the generation and control of the sawtooth scanning current. The rest of the components in this circuit are used to harness the energy in the transformer to provide power supplies for other sections of the receiver. The winding between pins 4 and 8 pulse energises the picture tube's heaters at 6.3V r.m.s. The other supplies make use of the transformer as the heart of a d.c.-to-d.c. converter system, by means of secondary windings that provide pulse feeds to diode/capacitor rectifier circuits. Small -value (0.680) resistors in the 25V and 200V supplies provide surge limiting and protection (by going open -circuit) in the event of a short-circuit in one of these supplies. The most significant supply is obtained from the diode - split winding that starts at pin 9. Although not shown in full detail it consists of several 'cells', each of which consists of an electrically isolated secondary winding, a built-in high - voltage rectifier diode and, as the reservoir capacitor, the carefully contrived capacitance that's present between adjacent, highly -insulated winding layers. These cells are connected in series to form a voltage -multiplier system capable of providing an e.h.t. supply for the tube's final anode of typically 24kV - it may be as high as 30kV in some designs. There's a built-in surge limiter resistor at the output end of the chain of cells. An important part of the e.h.t. multiplier system is the final reservoir capacitor that split chain provides about 8kV to a built-in potential -divider chain that contains two pres
ets:
the one at the top provides the supply for the tube's focus electrode
while the one near the bottom provides its first anode supply of about
800V. The bottom of the diode -split chain (pin 9) is returned to
chassis via a diode/capacitor/resistor network (not shown here). The
voltage developed across this network is proportional to the total beam
current, since this flows from the tube's cathodes via the e.h.t.
connector and the diode -split chain to chassis. Above a certain
threshold the voltage at pin 9 reduces the picture brightness and/or
contrast via the colour decoder/matrixing chip, limiting the beam
current and hence the dissipation in the tube's shadowmask to safe
levels. The winding between pins 10 and 7 of the transformer produces
50-70V pulses that sit on the h.t. voltage present at pin 10. When
rectified by D23 and C100 a 200V supply is provided for the RGB output
stages that drive the tube's cathodes. Secondary winding 4-6 feeds D24
and C99 which provide a 25V supply for the field timebase. In some
designs supplies for the audio output stage and the signal sections of
the receiver are also obtained from the line output transformer: in this
particular chassis they are obtained from the chopper transformer in
the power supply instead. Incidentally there have been one or two
designs, the Ferguson/philco TX10 chassis being a well-known example,
where the e.h.t. is also obtained from the chopper transformer, the line
output transformer then acting mainly as a load for the line output
transistor. In earlier designs a separate diode - capacitor multiplier
unit (tripler) was fed from a single line output transformer overwiding
to provide the e.h.t.Scan Rectification
The e.h.t., focus and 200V supplies d
erived
from the transformer are relatively lightly loaded, i.e. no great
current demand is placed on them. They can therefore be obtained by
rectifying the pulses present during the flyback period (time t2 -t4 in
Fig. 2), which is about twenty per cent of the scan cycle. Where the
current demand is greater, e.g. in a supply for the field timebase or an
audio output stage, the phasing of the relevant transformer winding is
often arranged so that the rectifier diode conducts during the scan
rather than the flyback period. Although the voltage available is much
lower, it's present for a longer period (about eighty per cent of the
scan/duty cycle). As a result the output regulation is much better. The
relatively high peak reverse voltage has to be taken into account in the
rectifier diode's specification.EHT Regulation
The internal impedance of a diode -sp
lit
e.h.t. supply is typically about 1MOhm. Thus with a total beam current
of lmA, present when a bright picture is being displayed on a 22in.
picture tube, the e.h.t. voltage will drop by about 1kV or five per
cent. The result of this is some ballooning, i.e. increase in picture
size. Compensation can be provided by reducing the line scanning power.
Careful choice of the value of the resistor that feeds the line output
transformer - R105 in Fig. 6 - gives automatic compensation in the
horizontal direction, while deriving the supply for the field output
stage from the line output transformer tends to cancel out the
ballooning in the vertical plane. Various 'anti -breathing' arrangements
are used in TV receiver design. Most operate via the diode -modulator
circuit we'll come to shortly. With any line output stage circuit the
picture width and e.h.t. voltage depend on the stage's h.t. supply, so
this must be well regulated and set up correctly. In the circuit shown
in Fig. 6 the h.t. voltage has to be 119V with a 20in. tube and 145V
with a 22in. tube.Pincushion Distortion
The raster produced on an almost -flat faced picture tube by constant -amplitude scan currents has pincushion distortion at all four sides. This is because of the disparity between the image plane and the screen's profile - . As a general rule the deflection yokes used with modern 90° tubes have built-in correction for both NS (vertical) and EW (horizontal) pincushion distortion while 110° tubes (generally above 22in. screen size) have in -yoke correction for NS distortion but cannot fully compensate for the
pincushion
effect at the sides of the screen. Thus with these the line scan
current has to be amplitude -modulated by a parabolic waveform at field
frequency as shown in Fig. 7. With present-day tube designs a modulation
depth of about seven per cent is required. the peak -to -peak scan
current being typically 4.1A at the top and bottom of the screen and
4.4A towards the centre of the screen, where the deflection power is
greatest. Amplitude modulation of the line scan current can be achieved
by including a saturable -reactance transformer in series with the scan
coils, but this is expensive. You could put a suitably -shaped ripple on
the supply to the line output stage, but the parabola would be
superimposed on any secondary supplies derived from the line output
transformer. The most widely used solution is to employ a diode -modu-
lator circuit, since this gives full control of the raster shape and
scan amplitude while providing a constant load current and flyback time.The Diode Modulator
Fig. 8 shows the essence of a diode -modulator arrange- ment. The efficiency diode is split in two, DI and D2, which perform the same clamping action as before. The flyback tuning capacitor is also split in two, Cl and C2: the upper one tunes the transformer and scan coils (L1) as before while the lower one tunes a bridge coil, L2, via C4 to the same flyback frequency of about 42kHz. C3 is the scan coupling capacitor, which corresponds with CB in Fig. 4. Modulation is achieved by using transistor Tr2, whose conduction governs the scan width, to vary the load across C4. When Tr2 is off, the scan energy is shared between the the two series LC combinations C3/L1 and L2/C4. The charge on C3 and C4 is in the ratio of about 7:1, the scan current being reduced in proportion. When Tr2 is fully conductive, C4 is effectively shorted out and acquires no charge. Thus a greater proportion of the energy is present in C3/L1 and the scan current and picture width are increased. By varying the conduction of Tr2 during the forward scan in a parabolic manner, EW pincushion correction is achieved. The basic picture width can be controlled by varying Tr2's standing bias. Choke L3 and the large -value capacitor
C5
filter the line -frequency energy so that it doesn't reach Tr2. And
because both sections of the load (L 1/C1 and L2/C2) are individually
tuned to the flyback frequency the flyback time, and hence the e.h.t.
and the other line output transformer -derived supplies, remain constant
over the field period despite the line scan current variation. There
are several different versions of the diode -modu- lator arrangement.
Some tube/yoke combinations have a scan -geometry characteristic such
that when the line scan current is modulated by a simple parabolic
waveform as described above the raster has inner pincushion distortion
as shown in Fig. 9.
Because
of this. the EW-correction system also has to modulate the S
-correction. Fig. 10 shows, in skeleton circuit form. how this can be
done. There are two coupling/S-correction capacitors. C3 and C3A. C3 is
the usual S -correction capacitor, but C3A has an increasing influence
as the diode modulator begins to have maximum effect towards the centre
of the screen. Critical choice of the value of C3A ensures that the
inner curved verticals shown in Fig. 9 are straightened out to give a
raster completely free from geometric distortion. Although all diode
modulators work on the same basic principle, in some designs a
transformer is used in place of the bridge coil to give better impedance
matching and balance. Fig. 11 shows such an arrangement, used by Bang
and Olufsen. The EW correction waveform is applied to transformer T6.
whose winding 1-2 takes the place of L2 in Figs. 8 and 10. This circuit
also provides inner -pincushion distortion correction as just described,
the supplementary S - correction capacitor being C36.Diode Modulator Drive
The parabolic EW drive waveform required is easily obtained by feeding the field -scan sawtooth waveform to a double integrator. By adding a sawtooth component the shape of the parabolic waveform can be tilted in either direction to give keystone -distortion correction if required - this is not generally necessary with modern tube/yoke designs. These EW correction characteristics are adjustable by preset resistors or, in the case of bus -programmable sets, remote control commands to the deflection processor. Very often the EW modulator is used to correct the previously mentioned picture breathing effect: this is done by feeding to the EW modulator's control circuit a voltage that's proportional to beam current.
BU208(A)
Silicon NPNnpn transistors,pnp transistors,transistors
Category: NPN Transistor, Transistor
MHz: <1 MHz
Amps: 5A
Volts: 1500V
HIGH VOLTAGE CAPABILITY
JEDEC TO-3 METAL CASE.
DESCRIPTION
The BU208A, BU508A and BU508AFI are
manufactured using Multiepitaxial Mesa
technology for cost-effective high performance
and use a Hollow Emitter structure to enhance
switching speeds.
APPLICATIONS:
* HORIZONTAL DEFLECTION FOR COLOUR TV With 110° or even 90° degree of deflection angle.
ABSOLUTE MAXIMUM RATINGS
Symbol Parameter Value Unit
VCES Collector-Emit ter Voltage (VBE = 0) 1500 V
VCEO Collector-Emit ter Voltage (IB = 0) 700 V
VEBO Emitter-Base Voltage (IC = 0) 10 V
IC Collector Current 8 A
ICM Collector Peak Current (tp < 5 ms) 15 A
TO - 3 TO - 218 ISOWATT218
Ptot Total Dissipation at Tc = 25 oC 150 125 50 W
Tstg Storage Temperature -65 to 175 -65 to 150 -65 to 150 oC
Tj Max. Operating Junction Temperature 175 150 150 °C
A Cockcroft-Walton cascade circuit comprises an input voltage source and a pumping and storage circuit with a series array of capacitors with pumping and storage portions of the circuit being interconnected by silicon rectifiers, constructed and arranged so that at least the capacitor nearest the voltage source, and preferably one or more of the next adjacent capacitors in the series array, have lower tendency to internally discharge than the capacitors in the array more remote from the voltage source.
1. An improved voltage multiplying circuit comprising,
2. An improved voltage multiplying circuit in accordance with claim 1 wherein said first pumping capacitor is a self-healing impregnated capacitor which is impregnated with a high voltage impregnant.
3. An improved voltage multiplying circuit in accordance with claim 1 wherein said first pumping capacitor comprises a foil capacitor.
Description:
BACKGROUND OF THE INVENTION
The invention relates in general to Cockcroft-Walton cascade circuits for voltage multiplication and more particularly to such circuits with a pumping circuit and a storage circuit composed of capacitors connected in series, said pumping circuits and storage circuit being linked with one another by a rectifier circuit whose rectifiers are preferably silicon rectifiers, especially for a switching arrangement sensitive to internal discharges of capacitors, and more especially a switching arrangement containing transistors, and especially an image tube switching arrangement.
Voltage multiplication cascades composed of capacitors and rectifiers are used to produce high D.C. voltages from sinusoidal or pulsed alternating voltages. All known voltage multiplication cascades and voltage multipliers are designed to be capacitance-symmetrical, i.e., all capacitors used have the same capacitance. If U for example is the maximum value of an applied alternating voltage, the input capacitor connected directly to the alternating voltage source is charged to a D.C. voltage with a value U, while all other capacitors are charged to the value of 2U. Therefore, a total voltage can be obtained from the series-connected capacitors of a capacitor array.
In voltage multipliers, internal resistance is highly significant. In order to obtain high load currents on the D.C. side, the emphasis in the prior art has been on constructing voltage multipliers with internal resistances that are as low as possible.
Internal resistance of voltage multipliers can be reduced by increasing the capacitances of the individual capacitors by equal amounts. However, the critical significance of size of the assembly in the practical application of a voltage multiplier, limits the extent to which capacitance of the individual capacitors can be increased as a practical matter.
In television sets, especially color television sets, voltage multiplication cascades are required whose internal resistance is generally 400 to 500 kOhms. Thus far, it has been possible to achieve this low internal resistance with small dimensions only by using silicon diodes as rectifiers and metallized film capacitors as the capacitors.
When silicon rectifiers are used to achieve low internal resistance, their low forward resistance produces high peak currents and therefore leads to problems involving the pulse resistance of the capacitors. Metallized film capacitors are used because of space requirements, i.e., in order to ensure that the assembly will have the smallest possible dimensions, and also for cost reasons. These film capacitors have a self-healing effect, in which the damage caused to the capacitor by partial evaporation of the metal coating around the point of puncture (pinhole), which develops as a result of internal spark-overs, is cured again. This selfhealing effect is highly desirable as far as the capacitors themselves are concerned, but is not without its disadvantages as far as the other cirucit components are concerned, especially the silicon rectifiers, the image tubes, and the components which conduct the image tube voltage.
It is therefore an important object of the invention to improve voltage multiplication cascades of the type described above.
It is a further object of the invention to keep the size of the entire assembly small and the internal resistance low.
It is a further object of the invention to increase pulse resistance of the entire circuit.
It is a further object of the invention to avoid the above-described disadvantageous effects on adjacent elements.
It is a further object of the invention to achieve multiples of the foregoing objects and preferably all of them consistent with each other.
SUMMARY OF THE INVENTION
In accordance with the invention, the foregoing objects are met by making at least one of the capacitors in the pumping circuit, preferably including the one which is adjacent to the input voltage source, one which is less prone to internal discharges than any of the individual capacitors in the storage circuit.
The Cockcroft-Walton cascade circuit is not provided with identical capacitors. Instead, the individual capacitors are arranged according to their loads and designed in such a way that a higher pulse resistance is attained only in certain capacitors. It can be shown that the load produced by the voltage in all the capacitors in the multiplication circuit is approximately the same. But the pulse currents of the capacitors as well as their forward flow angles are different. In particular, the capacitors of the pumping circuit are subjected to very high loads in a pulsed mode. In the voltage multiplication cascade according to the invention, these capacitors are arranged so that they exhibit fewer internal discharges than the capacitors in the storage circuit.
The external dimensions of the entire assembly would be unacceptably large if one constructed the entire switching arrangement using such capacitors.
The voltage multiplication cascade according to the invention also makes it possible to construct a reliably operating
arrangement which has no tendency toward spark-overs, consistent with satisfactory internal resistance of the voltage multiplication cascade and small dimensions of the entire assembly. This avoids the above cited disadvantages with respect to the particularly sensitive components in the rest of the circuit and makes it possible to design voltage multiplication cascades with silicon rectifiers, which are characterized by long lifetimes. Hence, a voltage multiplication cascade has been developed particularly for image tube circuits in television sets, especially color television sets, and this cascade satisfies the highest requirements in addition to having an average lifetime which in every case is greater than that of the television set.
A further aspect of the invention is that at least one of the capacitors that are less prone to internal discharges is a capacitor which is impregnated with a high-voltage impregnating substance, especially a high-voltage oil such as polybutene or silicone oil, or mixtures thereof. In contrast to capacitors made of metallized film which have not been impregnated, this allows the discharge frequency due to internal discharges or spark-overs to be reduced by a factor of 10 to 100.
According to a further important aspect of the invention, at least one of the capacitors that are less prone to internal discharges is either a foil capacitor or a self-healing capacitor. In addition, the capacitor in the pumping circuit which is adjacent to the voltage source input can be a foil capacitor which has been impregnated in the manner described above, while the next capacitor in the pumping circuit is a self-healing capacitor impregnated in the same fashion.
Other objects, features and advantages of the invention will be apparent from the following detailed description of preferred embodiments, taken in connection with the accompanying drawing, the single FIGURE of which:
BRIEF DESCRIPTION OF THE DRAWING
is a schematic diagram of a circuit made according to a preferred embodiment of the invention.
DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS
The voltage multiplier comprises capacitors C1 to C5 and rectifiers D1 to D5 connected in a cascade. An alternating voltage source UE is connected to terminals 1 and 2, said voltage source supplying for example a pulsed alternating voltage. Capacitors C1 and C2 form the pumping circuit while capacitors C3, C4 and C5 form the storage circuit.
In the steady state, capacitor C1 is charged to the maximum value of the alternating voltage UE as are the other capacitors C2 to C5. The desired high D.C. voltage UA is picked off at terminals 3 and 4, said D.C. voltage being composed of the D.C. voltages from capacitors C3 to C5. Terminal 3 and terminal 2 are connected to one pole of the alternating voltage source UE feeding the circuit, which can be at ground potential. In the circuit described here, a D.C. voltage UA can be picked off whose voltage value is approximately 3 times the maximum value of the pulsed alternating voltage UE. By using more than five capacitors, a correspondingly higher D.C. voltage can be obtained.
The individual capacitors are discharged by disconnecting D.C. voltage UA. However, they are constantly being recharged by the electrical energy supplied by the alternating voltage source UE, so that the voltage multiplier can be continuously charged on the output side.
According to the invention, in this preferred embodiment, capacitor C1 and/or C2 in the pumping circuit are designed so that they have a lower tendency toward internal discharges than any of the individual capacitors C3, C4 and C5 in the storage circuit.
It is evident that those skilled in the art, once given the benefit of the foregoing disclosure, may now make numerous other uses and modifications of, and departures from the specific embodiments described herein without departing from the inventive concepts. Consequently, the invention is to be construed as embracing each and every novel feature and novel combination of features present in, or possessed by, the apparatus and techniques herein disclosed and limited solely by the scope and spirit of the appended claims.
Inventors:Petrick, Paul (Landshut, DT)
Schwedler, Hans-peter (Landshut, DT)
Holzer, Alfred (Schonbrunn, DT)
ERNST ROEDERSTEIN SPEZIALFABRIK
US Patent References:
3714528 ELECTRICAL CAPACITOR WITH FILM-PAPER DIELECTRIC 1973-01-30 Vail
3699410 SELF-HEALING ELECTRICAL CONDENSER 1972-10-17 Maylandt
3463992 ELECTRICAL CAPACITOR SYSTEMS HAVING LONG-TERM STORAGE CHARACTERISTICS 1969-08-26 Solberg
3457478 WOUND FILM CAPACITORS 1969-07-22 Lehrer
3363156 Capacitor with a polyolefin dielectric 1968-01-09 Cox
2213199 Voltage multiplier 1940-09-03 Bouwers et al.
Chassis F6TT UNITS view.
- Luminance + Chrominance unit 492.045 G8 with TDA2500/3 + TDA2522 (all PHILIPS)
(Note the long luminance delay line)
- Synchronization Unit with TDA2593 (PHILIPS) 492.174 G10
- Sound - Audio Unit with TBA120U (Telefunken) + L130V (Fairchild Semiconductors) 492.044/E3
- RGB Amplifiers with TBA530 (PHILIPS) 943/K7
- Frame deflection unit 492.225/C1
- E/W Correction Unit 492.213/C
TDA2522 PAL TV CHROMA DEMODULATOR COMBINATION
FAIRCHILD LINEAR INTEGRATED CIRCUIT
GENERAL DESCRIPTION- The TDA2522 is a monolithic integrated circuit designed as
a synchronous demodulator for PAL color television receivers. It includes an 8,8 MHz
oscillator and divider to generate two 4.4 MHz reference signals and provides color difference outputs.
PACKAGE OUTLINE 9B
The TDA2522 is Intended to Interface directly with the TDA2560 with a minimum oF external components. The TDA2530 may be added if RGB drive is required. The TDA2522
is constructed using the Fairchild Planar* process.
TDA2520 COLOUR DEMODULATOR COMBINATION
The TDA2520 is an integrated synchronous demodulator combination for colour television
receivers incorporating the following functions :
~ 8, 8 MHZ oscillator followed by a divider giving two 4, 4 MHZ signals used as reference
signals
- keyed burst phase detector for optimum noise behaviour
- a stage to obtain chrominance signal control (a. c. c.) and an a. c. c. reference level
- a colour killer and identification signal detector
- two synchronous demodulators for the (B-Y) and (R-Y) signals
- temperature compensated emitter follower outputs
- PAL switch
- PAL flip-flop
- integrated capacitors in the symmetrical demodulators reduce unwanted carrier-
signals at the outputs.
TDA2593 SYNCHRO AND HORIZONTAL DEFLECTION CONTROL FOR COLOR TV SETDESCRIPTION
The TDA2593 isa circuit intended for the horizontal
deflectionof color TVsets, suppliedwith transistors
or SCR’S.
.LINE OSCILLATOR(two levels switching)
.PHASE COMPARISON BETWEEN SYNCHRO-
PULSE AND OSCILLATOR VOLTAGE
Ø 1, ENABLED BY AN INTERNAL PULSE,
(better parasitic immunity)
.PHASE COMPARISON BETWEEN THE FLYBACK
PULSES AND THE OSCILLATORVOLTAGE
Ø2
.COINCIDENCE DETECTOR PROVIDING A
LARGE HOLD-IN-RANGE .FILTER CHARACTERISTICS AND GATE
SWITCHING FOR VIDEO RECORDER APPLICATION
.NOISE GATED SYNCHRO SEPARATOR
.FRAME PULSE SEPARATOR
.BLANKING AND SAND CASTLE OUTPUT
PULSES
.HORIZONTAL POWER STAGE PHASE LAGGING
CIRCUIT
.SWITCHING OF CONTROL OUTPUT PULSE
WIDTH
.SEPARATED SUPPLY VOLTAGE OUTPUT
STAGE ALLOWING DIRECT DRIVE OF
SCR’S CIRCUIT .SECURITY CIRCUIT MAKES THE OUTPUT
PULSE SUPPRESSED WHEN LOW SUPPLY
VOLTAGE.
NORDMENDE SPECTRA COLOR SC1032 CHASSIS F6TT COLOR AMPLIFIER WITH Constant bandwidth RGB output amplifiers having simultaneous gain and DC output voltage control :
A
color television receiver includes conventional circuitry for
processing and detecting a received color television signal. Three
chrominance-luminance matrices combine detected color difference and
luminance signals forming color red, blue and green video signals.
Emitter follower coupling stages apply the color video signals
individually to each
of three output amplifiers which in turn drive the cathode electrodes
of a unitized gun CRT. Potentiometers couple the emitter electrodes of
the output amplifiers to a source of operating potential providing a
simultaneous signal gain and DC output voltage adjustment for each
amplifier during CRT color temperature setup. A voltage divider controls
the voltage applied to the common screen grid electrode of the CRT
providing a master setup adjustment. 1. In a color televison receiver, for processing and displaying a received television signal bearing modulation components of picture information, having a cathode ray tube including a trio of electron source means producing individual electron beams impinging an image screen to form three substantially overlying images and in which the respective operating points and relative conduction levels of said electron source means determine the color temperature of the reproduced image, the combination comprising:
master conduction means, coupled to said trio of electron source means simultaneously varying said conduction levels;
a plurality of substantially equal bandwidth amplifiers, each coupled to a different one of said electron source means, separately influencing said conduction levels;
low output impedance signal translation means recovering said picture information and supplying it to each of said plurality of amplifiers; and
separate adjusting means individually coupled to at least two of said amplifiers for simultaneously producing predetermined same sense variations in gain and DC output voltage of its associated amplifier while preserving said bandwidths.
2. The combination set forth in claim 1, wherein the transconductance and cutoff voltage of each of said electron source means bear a predetermined relationship and wherein said simultaneous predetermined variations in gain and DC output voltage are determined by said transconductance-cutoff voltage relationship. 3. The combination set forth in claim 2, wherein said plurality of amplifiers each include a gain and DC output voltage determining impedance and wherein each of said separate adjusting means include:
a variable impedance, coupling said gain and DC output voltage determining impedance of said associated amplifier to a source of bias current and forming a shunt path for signals within said amplifier.
4. The combination set forth in claim 3, wherein each of said electron source means include a cathode electrode and wherein each of said amplifiers include:
a transistor having input, common, and output electrodes, said output electrode being coupled to said electron source means cathode.
5. The combination set forth in claim 4, wherein said gain and DC output voltage determining impedance is coupled to said common electrode. 6. The combination set forth in claim 5, wherein said input, common, and output electrodes of said transistors are defined by base, emitter, and collector electrodes, respectively. 7. The combination set forth in claim 6, wherein said gain and DC output voltage determining impedance includes a resistor coupling said emitter electrode to ground and wherein said variable impedance includes:
a resistive control, having a variable resistance, coupling said emitter electrode to a source of operating potential.
8. The combination set forth in claim 7, wherein said three electron source means include control grid and screen grid electrodes common to said three electron guns and wherein variations of cathode electrode voltages permit changes of said relative conduction levels and said respective operating points. 9. The combination set forth in claim 8, wherein said master conduction means includes a variable bias potential source coupled to said common screen grid electrode.
Description:
BACKGROUND OF THE INVENTIONThis invention relates to color television receivers and in particular to cathode ray tubes (CRT) drive systems therefor. Each of the several types of color television cathode ray tubes in current use includes a trio of individual electron sources producing distinct electron beams which are directed toward an image screen formed by areas of colored-light-emitting phosphors deposited on the inner surface of the CRT. The phosphors emit light of a given additive primary color (red, blue or green) when struck by high energy electrons. A "delta" electron gun arrangement, in which the electron sources comprise three electron guns disposed at the vertice
s
of an equilateral triangle, having its base oriented in a horizontal
plane and its apex above or below the base plane, may be used.
Alternatively, the three electron sources may be "in line", that is,
positioned in a horizontal line. In either case, the three beams
produced are subjected to deflection fields and scan the image screen
in both the horizontal and vertical directions thereby forming three
substantially overlying rasters. The phosphor deposits forming the image screen may alternatively comprise round dots, elongated areas, or uninterrupted vertical lines. A parallax barrier or shadow mask, defining apertures generally corresponding to the shape of the phosphor areas, is interposed between the electron guns and the image screen to "shadow" or block each phosphor area from electrons emitted from all but its corresponding electron gun.
A color television signal includes both luminance (monochrome) and chrominance (color) picture components. In the commonly used RGB drive systems the separately processed luminance and chrominance information is matrixed (or combined) before application to the CRT cathodes. Three output amplifiers apply the respective red, blue and green video signals thus produced for controlling the respective electron source currents.
The luminance components have substantially the same effect on all three electron sources whereas the color components are differential in nature, causing relative changes in electron source currents. In the absence of video signals, the combined raster should be a shade of grey. At high gun currents, the grey is very near white and at low settings, it is near black. The "color", commonly called color temperature, of the monochrome raster depends upon the relative contributions of red, blue and green light. At high color temperatures, the raster may appear blue and at low color temperatures it may appear sepia. While the most pleasing color temperature is largely a matter of design preference, ideally the receiver should not change color temperature under high and low brightness nor for high and low frequency picture information.
Generally, the electron sources comprise individual electron guns each including separately adjustable cathode, control grid and screen grid electrodes and a desired color temperature is achieved by adjustment of each electrode voltage during black and white setup. While the exact setup procedure employed varies with the manufacturer and specific CRT configuration, all manufacturers attempt to achieve consistent color temperature throughout the usable range of CRT beam current variations.
A typical color temperature adjustment involves setting the low light color temperature condition of each electron gun by adju
sting
its screen grid electrode voltage to produce the required DC
conditions between electron guns at minimum beam currents. A high light
or dive adjustment at increased CRT beam current is then made to
insure consistent color temperature. In receivers utilizing CRT's with
separately adjustable screen grid electrode voltages, the drive
adjustment may take the form of a minor change in signal gain of the
output amplifiers. The process is, in essence, one of configuring the
operating points of the three electron guns to conform to three
substantially identical output amplifiers. The recently developed economical "unitized gun" type CRT has a combined electron source structure in which three common control grids and three common screen grids are used with the cathodes being the only electrically separate electrodes. The greatly simplified and more economical unitized gun structure, however, imposes some restrictions on the circuitry used to drive the electron sources. Perhaps most significant is the absence of
the
flexibility previously provided by individually adjustable screen
grid electrode voltages. Due in part to the inverse relationship
between electron source transconductance, which may be thought of as
"gain" of the electron source, and cutoff voltage, the typical
individual low level color temperature or equal cutoff adjustment
described above also performs the additional function of establishing
nearly equal transconductances for the three electron sources. As a
result only minor relative changes in electron source currents occur
at higher CRT beam currents. Color temperature adjustment in a receiver with a unitized gun CRT involves a somewhat different process, namely, configuring the drive and bias applied to each of the gun cathodes to accommodate differences in relative electron source characteristics which, without the equalizing effect of separate screen electrode adjustments, may be considerable.
Initially television receivers using unitized gun CRT's utilized a variable DC voltage divider operative upon each output amplifier to provide adjustment of the DC cutoff voltage. Drive, or signal gain, adjustment to accommodate differences in electron source transconductances was generally accomplished by separate individual gain controls operative on each of the output amplifiers.
However, the more recently developed unitized gun systems combine the DC voltage (cutoff) and signal gain (drive) adjustments for each electron source by simultaneously varying the signal gain and DC voltage in the same direction in a predetermined relationship. One such system used three CRT coupling networks each of which includes a variable impedance simultaneously operative on both the amplitude of coupled signal and DC voltage. Another system uses a variable collector load impedance for each of the output amplifiers, making use of the changes in amplifier signal gain and DC output voltage resulting from collector load variations.
While such systems provide an adequate range of adjustment to achieve color temperature setup using a reduced number of controls, they often degrade image quality. Ideally, the luminance portion of the signal is applied uniformly to each of the three electron sources. Although the relative signal amplitudes may be varied to accommodate transconductance differences between electron sources, it is desirable that each applied signal be an otherwise identical replica of the others. The variable impedance elements in the voltage divider networks and variable collector loads of the prior art interact with the capacities inherent in the output amplifiers and electron gun structures to produce unequal bandwidths for the different color video signals, which cause color changes in their high frequency components (which correspond to detailed picture information). The resulting effect upon the displayed image is similar in appearance to the well-known "color fringing" or misconvergence effect.
OBJECTS OF THE INVENTION
It is an object of the present invention to provide an improved color television receiver.
It is a further object of this invention to provide a novel CRT color temperature setup system.
SUMMARY OF THE INVENTION
In a color television receiver, for processing and displaying a received television signal bearing modulation components of picture information, a
cathode
ray tube includes three electron source means producing individual
electron beams which impinge an image screen to form three
substantially overlying images. The respective operating points and
relative conduction levels of the electron source means determine the
color temperature of the reproduced image. Master conduction means,
coupled to the three electron source means, simultaneously vary the
conduction levels and a plurality of substantially equal bandwidth
amplifiers, each coupled to a different one of the electron source
means, separately influence the conduction levels. Low output impedance
signal translation means recover the picture information and supply it
to each of the amplifiers. Separate adjusting means are individually
coupled to at least two of the amplifiers for simultaneously producing
predetermined variations in the gain and DC output voltage of the
amplifiers while preserving the bandwidths. BRIEF DESCRIPTION OF THE DRAWING
The drawing shows a partial-schematic, partial-block diagram representation of a color television receiver constructed in accordance with the present invention.
DESCRIPTION OF THE PREFERRED EMBODIMENT
Referring to the drawing, a signal processor 10 includes conventional circuitry (not shown) for amplifying a received television signal and detecting
the
modulated components of luminance and chrominance information therein.
The output of signal processor 10 is coupled to a luminance amplifier
11 and a chrominance processor 30. Luminance amplifier 11 is
conventional and includes circuitry controlling brightness and contrast
of the luminance signal. The output of luminance amplifier 11 is
coupled to three luminance-chrominance matrices 12, 13 and 14.
Chrominance processor 30 includes conventional chrominance information
detection circuitry for providing three color difference or
color-minus-luminance output signals (R-Y, G-Y and B-Y) which are
individually coupled to luminance-chrominance matrices 12, 13 and 14,
respectively. The signal from luminance amplifier 11 is combined with
the color-minus-luminance signals from chrominance processor 30 to form
the respective red, green and blue video signals which are coupled to
the R, G and B output amplifiers 15, 16 and 17, respectively. The
outputs of amplifiers 15, 16 and 17 are coupled to the cathode
electrodes 23, 24 and 25, respectively, of a CRT 20 having an image
screen 21. A voltage divider, formed by a series combination of
resistors 83 and 84, is coupled between a source of operating potential
+V2 and ground. The junction of resistors 83 and 84 is
connected to a common control grid electrode 28 and to ground by a
filter capacitor 85 which provides a signal bypass. A potentiometer 80
and a resistor 81 are series coupled between a source of operating
potential +V1 and ground, forming another voltage divider.
The junction of potentiometer 80 and resistor 81 is connected to common
screen grid electrode 29 and to ground by a bypass capacitor 82.
Cathode electrodes 23-25, control grid electrode 28 and screen grid
electrode 29 are part of a unitized gun structure in CRT 20 with the
control grid and screen grid being common to each of the three electron
sources defined by the separate cathode electrodes. While luminance-chrominance matrices 12 and 13 are shown in block form, it should be understood that they are identical to the detailed structure of matrix 14. Similarly, red output amplifier 15 and green output amplifier 16 are identical to the detailed structure of blue output amplifier 17. Further, the receiver shown is understood to include conventional circuitry for horizontal and vertical electron beam deflection together with means deriving a CRT high voltage accelerating potential, all of which have, for clarity, been omitted from the drawing.
Luminance-chrominance matrix 14 includes a matrix transistor 40 having an emitter electrode 41 coupled to ground by a resistor 55 and by a series combination of resistors 46 and 47, a base electrode 42 coupled to the output of luminance amplifier 11, and a collector electrode 43 coupled to a source of operating potential +V3 by a resistor 45. The B-Y output of chroma processor 30 is connected to the junction of resistors 46 and 47. An emitte
r-follower
transistor 50 has an emitter electrode 51 coupled to ground by a
resistor 56, a base electrode 52 connected to the collector of matrix
transistor 40, and a collector electrode 53 connected to +V3. Blue amplifier 17 includes an output transistor 60 having an emitter electrode 61 coupled to ground by a series combination of resistors 67 and 68, a base electrode 62 connected to the emitter of transistor 50, and a collector electrode 63 coupled to +V2 by a resistor 66. A series combination of a potentiometer 70 and a resistor 69 couples the junction of resistors 67 and 68 to +V3. Collector 63, which is the output of amplifer 17, is connected to cathode 25 of CRT 20.
During signal reception, the separately processed luminance and B-Y color difference signals are applied to matrix transistor 40. The combi
ned
signal developed at its collector 43 forms the blue video signal which
controls the blue electron beam in CRT 20 and represents the relative
contribution of blue light in the image produced. The blue video signal at collector 43 is coupled via transistor 50 to base 62 of output transistor 60. The low source impedance of emitter follower transistor 50 obviates any detrimental effects upon the blue video signal due to loading at the input to amplifier 17 caused by gain or frequency dependent input impedance variations of amplifier 17. The blue video signal applied to base 62 is amplified by transistor 60 to a level sufficient to control the conduction of its respective electron source.
During color temperature setup, a predetermined setup voltage (corresponding to black) is applied to matrices 12, 13 and 14. The voltage on common screen grid electrode 29 is adjusted, by varying potentiometer 80 which together with resistor 81 and capacitor 82 form master conduction means, to cause a low brightness raster to appear on image screen 21. As will be seen, adjustment of potentiometer 70 and the corresponding potentiometers in amplifiers 15 and 16 establish the correct combination of DC electron source cathode voltages and output amplifier gains to produce the selected color temperature at both low and high CRT beam currents.
Amplifier 17 includes a common emitter transistor stage in which the impedance coupled to emitter electrode 6 is a gain and DC output voltage determining impedance. Signal gain is approximately equal to the ratio of the collector impedance (resistor 66), to this gain and DC voltage determining impedance (ignoring the effects of capacities associated with the tr
ansistor and the electron gun which will be considered later). Because the source of operating potential +V3
coupled to potentiometer 70 forms a good AC or signal ground, the
series combination of resistor 69 and potentiometer 70 are effectively
in parallel with resistor 68 and the total impedance coupling emitter 61
to signal ground comprises resistor 67 in series with this combination
of resistors 68 and 69 and potentiometer 70. Variations in this
impedance caused by adjustment of potentiometer 70 changes the ratio of
collector to emitter impedances and thereby the gain of amplifier 17.
If potentiometer 70 is varied to present increased resistance, gain is
reduced and if varied to present decreased resistance, gain is
increased. The DC voltage at collector 63 of transistor 60 is determined by the product of the collector resistance and quiescent collector current (current in the absence of applied signal) and V2. The voltage at base 62 is established by the emitter voltage of transistor 50. Variations in the resistance of potentiometer 70 cause variations in current flow in the series path including potentiometer 70 and resistors 69 and 68. The voltage developed across resistor 68 is supplied to emitter 61 through resistor 67.
In the absence of signal, the DC voltage at base 62 is constant and the relative voltage between base 62 and emitter 61, which controls the conduction level of transistor 60, is a function of the voltage at emitter 61. Increases in the resistance of potentiometer 70 reduce the emitter voltage, increase the relative base-emitter voltage of transistor 60, and increase collector current. The increased collector current develops a greater voltage drop across collector resistor 66 and reduces the DC voltage at collector 63 (and cathode 25). Conversely, a decrease in the resistance of potentiometer 70 increases the voltage at emitter 61, reducing the relative base-emitter voltage and decreasing collector current. The smaller voltage drop across resistor 66 increases the DC voltage at collector 63 and cathode 25.
Thus, increasing the resistance
of
potentiometer 70 produces proportionate simultaneous reduction of the
DC voltage applied to cathode 25 and the voltage gain of amplifier 17,
whereas decreasing the resistance of potentiometer 70 produces
proportionate simultaneous increase of the DC voltage and signal gain.
As mentioned above, amplifiers 15 and 16 are identical to amplifier 17.
In practice only two of the three output amplifiers require adjustment
to achieve color temperature setup. However, greater flexibility and
optimum use of amplifier signal handling capability is realized if all
three output amplifiers are adjustable. As previously mentioned capacities associated with transistor 60, cathode 25 and corresponding interconnections (such as those used to couple collector 63 to cathode 25) are effectively in parallel with collector load resistor 66 forming a partially reactive "coupling network" which exhibits a frequency characteristic (bandwidth) affecting signals coupled therethrough. In practice, the other coupling networks have identical bandwidths and affect their signals in an equal manner. The setup control adjustments of the present invention do not change the characteristics of these coupling networks and the uniformity of signal coupling for the different color signals is preserved. In contrast, conventional adjustment circuitry (whether variable collector load or voltage divider) place variable impedances within these couplings. The varied adjustments of these impedances to effect color temperature control adjustment disturb the bandwidth characteristics of the coupling networks causing differential variations in the individual color video signals.
What has been shown is an RGB CRT drive system which includes output amplifiers each having a single control which simultaneously achieves changes of the DC
output voltage and signal gain of the amplifier in a predetermined
relationship. The bandwidths of all three output amplifiers and their
associated coupling networks remain substantially undisturbed by these
control adjustments during CRT color temperature setup. While particular embodiments of the invention have been shown and described, it will be obvious to those skilled in the art that changes and modifications may be made without departing from the invention in its broader aspects, and, therefore, the aim in the appended claims is to cover all such changes and modifications as fall within the true spirit and scope of the invention.
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