PHILIPS 26C568 GOYA (PHILIPS K11) CHASSIS K11 INTERNAL VIEW.

The PHILIPS CHASSIS K11 is introducing the InLine PHILIPS 20AX CRT TUBE FAMILY.

This chassis was fitted in many models series phasing out the CRT DELTA TUBES FAMILY of previous era and models.

The chassis structural aspect is similar to CHASSIS K9 dual panel with deflections / power stages on the left big board (Large signal panel) and all signal stages on the smaller board on the right side (small signal board).
The concept and the development are different due to CRT TUBE 20AX TYPE.

This chassis is showing another time the awesome engineering of such tellyes.





This model is equipped with one of the first ULTRASONIC REMOTE CONTROL RECEIVER device in PHILIPS models , to feature remote control capability.

The version shown here in collection is the nD using for first time a DST Transformer.

The PHILIPS K11 was replaced by the PHILIPS K12 MONOCARRIER.




















The chassis K11 is higly sophisticated and complex but it has an unique fashinating structure and design which expands his technology in a way of simplicity which is today, long time, lost and forgotten (forever).

PHILIPS CHASSIS K11 DST High-voltage transformer:

A component holder is arranged on the windings of a diode-split line transformer. On this holder the diodes, the focus bleeder and the high-voltage bleeder can be arranged. The connection strips of the components act as soldering pins for the winding ends. The high voltage is connected via a socket connector which is clamped to the component holder by way of two snap tags. The socket connector is provided with radially extending ribs so as to increase the creepage path. The contact spring of the socket connector has a triangular shape, thus ensuring a large contact surface area.


1. A high-voltage transformer comprising a ferromagnetic core, a primary winding disposed on said core, an insulating layer disposed on said winding, a secondary winding disposed on said layer, a component holder, a socket connector mounted on said holder for the output of the high-voltage to be generated, the secondary winding including a number of winding layers, insulating layers disposed between said layers, a plurality of diodes coupled between an end of the winding layer to a beginning of the subsequent winding layer respectively, all diodes being coupled in the same rectifying sense, the component holder being disposed on the secondary windings, support means being disposed on the component holders for clamping the diodes to prevent rotation about their axis, the connection wires of the diodes comprising connection pins for a part of the windings, and two tag means disposed on the component holder for clamping the connector. 2. A high-voltage transformer as claimed in claim 1, wherein a resistor is clamped on the supports of the component holder. 3. A high-voltage transformer as claimed in claim 1, wherein the socket connector comprises a triangular spring means for the electrical contacting of a pin connector of a high-voltage cable to be connected. 4. A high-voltage transformer as claimed in claim 1, wherein the socket connector can be rotated with respect to the component holder. 5. A television line transformer comprising a high voltage transformer as claimed in claim 1.

Description:

The invention relates to a high-voltage transformer, notably a line transformer for a television receiver, comprising a ferromagnetic core, a primary winding, an insulating layer, a secondary winding, a component holder, and a socket connector for the output of the high voltage to be generated, the secondary winding consisting of a number of winding layers wherebetween insulating layers are inserted, a transition from an end of a winding layer to a beginning of the subsequent winding layer being formed by a diode, all diodes constituting the transitions between the further winding layers being connected in the same rectifying sense. A transformer of this kind is known from British Patent Specification 1,090,995. The component holder of the transformer at the same time acts as the coil holder. On the component holder connection pins are provided whereto the windings of the coils and the diodes can be separately soldered. Consequently, for one diode winding connection two welds are required. Moreover, in this coil holder the position of the windings and the diodes with respect to each other is not fixed, which necessitates, for reasons of safety, a spacing between the diodes themselves and also between the windings and the diodes which is larger than would be required in the case of an ideal situation of the components. As a result, a compact construction is not possible. The high voltage to be generated is output via a high-voltage cable which is soldered either to a terminal rail or to a connection pin on the coil holder. As a result, the high-voltage transformer is no longer a universal circuit component, because it is sometimes desirable to use other types of high-voltage cable, or a cable of the same type but a different length.
The invention has for its object to eliminate the described drawbacks and to provide a high-voltage transformer which has a compact construction which is also more universal.
To this end, the invention is characterized in that the component holder is arranged on the secondary windings, on the component holder supports being formed whereon diodes are clamped such that they cannot rotate about their axis, the connection wires of the diodes serving as connection pins for a part of the windings, the socket connector being clamped by way of two tags formed on the component holder.
It is advantageous to mount bleeder resistors, such as the focus bleeder resistor and the high-voltage bleeder resistor, also on the component holder. The focus and the high-voltage bleeder resistors provide additional loading of the high-voltage transformer, thus causing a reduction of the internal resistance of the high-voltage supply. As a result, the focus voltage and the high voltage are more current-independent. The mounting of the bleeder resistors on the coil holder saves space elsewhere, whilst the dimensions of the high-voltage transformer are only slightly increased. At the same time, connections to bleeder resistors outside the high-voltage transformer are avoided, so that a safer operation of the high-voltage section of the television receiver is achieved.
The invention will be described in detail hereinafter with reference to a drawing showing an embodiment of a high-voltage transformer according to the invention.
FIG. 1 shows a partly broken away front view,
FIG. 2 is a sectional view taken along the line II--II in FIG. 1.
FIG. 3 is a longitudinal sectional view of an embodiment of a socket connector.
FIG. 4 is a cross-sectional view of the socket connector shown in FIG. 3, taken along the line IV--IV.
The high-voltage transformer illustrated in FIG. 1 comprises a core 1 consisting of two parts which are clamped together by means of bolts and nuts 2 (only partly shown). On one leg of the core there are provided a primary winding 3 and a secondary winding 5 which are separated by an insulating layer 4. A component holder 7 is arranged on the winding 5. The component holder 7 is provided with supports 9, which are shaped such that the diodes 11 and the bleeder resistors 13 and 14 clamped therein cannot rotate about their axis. Very good positioning and unambiguous spacing are thus obtained. The welded joint between a winding and the connection wire of a diode and/or of a bleeder resistor should be rounded, because otherwise undesired effects can occur when a high-voltage is generated. The necessary connection pins 15 of the diode 11 and the bleeder resistors 13, 14 project from the moulding compound 24 cast about the high-voltage section of the transformer. The component holder 7 is provided with tags 17 wherebetween a socket connector 19 is secured. The generated high voltage is connected via a socket connector 19.
So as to obtain a better idea of the construction of a high-voltage transformer, a sectional view taken along the lines II--II of FIG. 1 is shown in FIG. 2. The principal components are denoted by references corresponding to those of FIG. 1. It will be obvious that the primary winding 3 can alternatively be partly or completely wound about the other leg of the core. If the primary winding is distributed between the two legs of the core, the part of the primary winding situated underneath the secondary winding is usually referred to as coupling winding. FIG. 3 is a sectional view of the socket connector 19. The Figure shows a pipe 30 wherein a socket 31 is clamped on the one end, the said socket containing a triangular spring 32 for making firm contact with a pin connector (not shown) to be inserted into the socket connector 19.

The high voltage to be generated is connected to the socket 31 via a conductor 29.
Ridges 33 are provided on the pipe 30 so as to increase the creepage path.
For clamping the plug connector to be inserted, the pipe is provided on its other end with tapered claws 34. The claws 34 are pressed towards each other by tightening a coupling nut 35.
FIG. 4 shows a sectional view through one end of the pipe 30, taken along the line IV--IV in FIG. 3. It is to be noted that the output of a high-voltage via a socket connector in combination with a plug connector is known from German Offenlegungsschrift No. 2,052,922. Therein, electrical connection is effected by way of a contact spring having a fork-like shape. The electrical connection between the contact spring and a pin to be contacted thereby consists of only a few contact points.
The embodiment according to the invention is shaped as a triangular spring which offers a larger contact surface area.

The Diode Split LOPT

This  idea of incorporating the e.h.t. rectifier into the line output transformer is not new , it was first patented in 1966 by E. K. Cole Ltd. of Southend. What is new is that the e.h.t. tripler itself has now been integrated into a new type of line output transformer. Extensive testing has indicated that the life expectancy of this unit is excellent.
The new transformer makes use of the interlayer capacitances between a number of secondary windings, thus eliminating the high voltage capacitors necessary in a conventionally constructed voltage tripler. This in itself
leads to greater inherent reliability since these high voltage capacitors are largely responsible for tripler failures. In practical designs, the primary winding and the auxiliary windings - which provide the 1.t., reference flyback pulses, h.t. for the video output stages, etc. - are located on one leg of the core, the secondary windings, with the e.h.t. rectifier diodes and a link winding, being on the other leg. The link winding is connected in parallel with the primary winding and serves to eliminate the high leakage inductance that would otherwise exist between the primary and the secondaries as they are on opposite legs of the core. Fig. 1 shows the circuit diagram of a basic d.s.t. Each of the secondaries has the same number of turns, so each secondary layer will have only a d.c. potential difference between each coil and no a.c. potential difference. This approach makes the interlayer insulation much easier. The diodes are connected as shown in Fig. 2, and a d.c. voltage is obtained whose value is the sum of the rectified a.c. voltages per layer. To obtain an output of about 25kV, four secondary layers and four diodes are used, each carrying a peak flyback voltage of  around 7kV.
Mullard / PHILIPS have designated their first d.s.t. type AT2076/15. This device was specifically designed for use in 20AX receiver designs. A practical line output stage based on the Mullard device is shown in Fig. 3, using the BU208A line output transistor and a diode modulator for EW raster correction.
Unlike most recent designs, the EW modulator is of the high-level type, i.e. linked to the primary instead of to a secondary winding on the line output transformer. Since the 20AX tube's first anodes require a lower voltage than con- ventional tubes the BY184 first anode supply rectifier is fed direct from the collector of the line output transistor, thus reducing the loading on the transformer. As usual, pulses from the line output transformer are rectified to provide the video supply, but in this design the "cold" end of the wind- ing is returned to the main h.t. line, enabling the winding to be reduced in size. Where a lower supply is required the rectifier (BY210-600) can be fed from pin 10 instead of pin 8.
Cost Advantages: Due to the lower manufacturing costs when compared to using a separate line output transformer and e.h.t. tripler, the d.s.t. is likely to offer the setmaker a price advantage. Also, storage and buying problems are reduced by half. Another point in its favour is the reduction in assembly time, again because one is dealing with one rather than two components. Diode split transformers are already in use in a number of continental sets and are currently undergoing evaluation trials with British setmakers for incorporation in the next generation of receivers which we are likely to be meeting later on the future years .

PHILIPS 26C568 CHASSIS K11 E/W CORRECTION Circuit arrangement in an image display apparatus for (horizontal) line deflection

Line deflection circuit in which the deflection coil is east-west modulated. In order to cancel an east-west dependent horizontal linearity defect the inductance value of the linearity correction coil is made independent of the field frequency, for example by means of a compensating current. In an embodiment this current is supplied by the shunt coil of the east-west modulator.


1. Circuit arrangement for use with a line deflection coil, said circuit comprising a generator means adapted to be coupled to said coil for producing a sawtooth line-deflection current through said line deflection coil, said deflection current having a field-frequency component current, a horizontal linearity correction coil adapted to be coupled in series with said deflection coil and including an inductor having a bias-magnetized core, and means for making the inductance value of the linearity correction coil substantially independent of the field frequency component current. 2. Circuit arrangement as claimed in claim 1, wherein said making means includes a current supply source means for producing a compensating line-frequency sawtooth current through a winding of the line

arity correction coil, the amplitude of the compensating current having a field-frequency variation. 3. Circuit arrangement as claimed in claim 2, wherein the direction of curvature of the field-frequency envelope of the compensating current is opposite to the direction of curvature of the field-frequency component current of the line deflection current, whereby the magnetic fields produced in the core of the correction coil by the two currents have the same direction. 4. Circuit arrangement as claimed in claim 2, wherein the direction of curvature of the field-frequency envelope of the compensating current is the same as the direction of curvature of the field-frequency component current of the line deflection current, whereby the magnetic fields produced in the core of the correction coil by the two currents have opposite directions. 5. Circuit arrangement as claimed in claim 2, wherein said correction coil further comprises an additional winding disposed on the core, said additional winding being coupled to said supply source means to receive the compensating current. 6. Circuit arrangement as claimed in claim 5, further comprising modulator means for modulating the line deflection current with said field frequency component, said modulator including a compensation coil coupled in series with said additional winding. 7. Horizontal linearity correction coil comprising a core made of a magnetic material and bias-magnetized by at least one permanent magnet, and an additional winding disposed on the core. 8. Image display apparatus including a circuit arrangement as claimed in claim 1.

Description:

The invention relates to a circuit arrangement in an image display apparatus for (horizontal) line deflection, which apparatus also includes a circuit arrangement for (vertical) field deflection, provided with a generator for generating a sawtooth line-frequency deflecting current through a line deflection coil and with a modulator for field-frequency modulation of this current, the deflection coil being connected in series with a linearity correction coil in the form of an inductor having a bias-magnetized core.
By means of the linearity correction coil the linearity error due to the ohmic resistance of the deflection circuit is corrected. The sign of the bias magnetisation is chosen so that it is cancelled by the deflection current at the beginning of the deflection interval, so that the inductance of the correction coil is a maximum, whereas the voltage drop across the deflection coil then is a minimum. This voltage drop is adjustable by adjustment of the starting inductance of the correction coil. During the deflection interval the core gradually becomes saturated so that the inductance of, and the voltage drop across, the correction coil decrease. Thus the linearity error can be cancelled exactly at the beginning of the interval, that is to say on the left on the screen of the image display tube, and with a certain approximation at other locations.
In image display tubes using a large deflection angle, raster distortion, which generally is pincushion-shaped, of the image displayed occurs. This distortion can be removed in the horizontal direction, the so-called east-west direction, by means of field-frequency modulation of the line deflection current, the envelope in the case of pincushion-shaped distortion being substantially parabolic so that the amplitude of the line deflection current is a maximum at the middle of the field deflection interval.
It was found in practice that the said two corrections are not independent of one another, that is to say the adjustment of the east-west modulation affects horizontal linearity. As long as the modulation depth is not excessive, a satisfactory compromise can be found. However, in display tubes having a deflection angle of 110° and particularly in colour display tubes in which the deflection coils have a converging effect also, it is difficult to find such a compromise. A tube of this type is described in "Philips Research Reports," volume Feb. 14, 1959, pages 65 to 97; the distribution of the deflection field is such that throughout the display screen the landing points of the electron beams coincide without the need for a converging device. Owing to this field distribution, however, the pin-cushion-shaped distortion in the image displayed in the east-west direction is greater than in comparable display tubes of another type. Hence there must be east-west modulation of the line deflection current to a greater depth. It is true that under these conditions horizontal linearity can correctly be adjusted over a given horizontal strip after the east-west modulation has been adjusted correctly, i.e., for a rectangular image, but it is found that in other parts of the display screen a serious linearity error remains. When vertical straight lines are displayed as straight lines in the right-hand part of the screen, they are displayed as curved lines in the left-hand part.
It is an object of the present invention to remove the said defect so that horizontal linearity can satisfactorily be adjusted throughout the screen, and for this purpose the circuit arrangement according to the invention is characterized in that it includes means by which the inductance of the linearity correction coil is made substantially independent of the field frequency.
The invention is based on the recognition that the defect to be removed is due to a field-frequency variation of the said inductance because the latter is current-dependent. According to a further recognition of the invention the circuit arrangement is characterized in that it includes a current supply source for producing a compensating line-frequency sawtooth current through a winding of the linearity correction coil, the amplitude of the current being field-frequency modulated. The circuit arrangement according to the invention may further be characterized in that an additional winding is provided on the core of the linearity correction coil and is traversed by the compensating current. A circuit arrangement in which the modulator for modulating the line deflection current includes a compensation or bridge coil may according to the invention be characterized in that the additional winding is connected in series with the said coil.
The invention also relates to a linearity correction coil for use in a line deflection circuit having a core which is made of a magnetic material and is bias magnetized by at least one permanent magnet, which coil is characterized in that an additional winding is provided on the core.
Embodiments of the invention will now be described by way of example, with reference to the accompanying diagrammatic drawings, in which
FIG. 1 is the circuit diagram of a known circuit arrangement for line deflection in which the line deflection current is east-west modulated,
FIG. 2 shows the distorted image which is displayed on the screen when the circuit arrangement of FIG. 1,
FIG. 3 is a graph explaining the observed defect, and
FIGS. 4 and 7 show embodiments of the circuit arrangement according to the invention by which this defect can be cancelled.
FIG. 1 is a greatl simplified circuit diagram of a line deflection circuit of an image display apparatus, not shown further. The circuit includes the series combination of a line deflection coil L _y , a linearity correction coil L and a trace capacitor C _t , which series combination is traversed by the line deflection current i _y . The collector of an npn switching transistor T _r and one end of a choke coil L ₁ are connected to a junction point A of a diode D, a capacitor C _r and the said series combination. The other end of the choke coil is connected to the positive terminal of a supply voltage source which supplies a substantially constant direct voltage V _b and to the negative terminal of which the emitter of transistor Tr is connected. This negative terminal may be connected to earth. The other junction point B of elements D and C _r and of the series combination of elements C _t , L _y and L is connected to one terminal of a modulation source M for east-west correction which has its other terminal connected to earth. Diode D has the pass direction shown in the FIG.
To the base of transistor Tr line-frequency switching pulses are supplied. In known manner the said series combination is connected to the supply voltage source during the deflection interval (the trace time), diode D and transistor Tr conducting alternately. During the retrace time these elements are both cut off. Under these conditions the current i _y is a sawtooth current. The coil L, which has a saturable ferrite core which is bias-magnetized by means of at least one permanent magnet, serves to correct the linearity of the current i _y during the trace time, whilst the capacitance of the capacitor C _t is chosen so that the currenct i _y is subjected to what is generally referred to as S correction. During the retrace time, at point A pulses are produced the amplitude of which is much higher than that of the voltage V _b and would be constant in the absence of modulation source M. Information from the field deflection circuit, not shown, of the image display apparatus and line retrace pulses, the latter for example by means of a transformer, are supplied in known manner to modulation source M. Amplitude-modulated line retrace pulses having a field-frequency parabolic envelope, as indicated in the FIG., are produced at point B. During the line trace time the voltage at point B is zero. Thus the current i _y is given the desired field-frequency modulated form which is also shown in FIG. 1.
The amplitude of the envelope in point B at the beginning and at the end of the field trace time and the amplitude of this envelope at the middle of the said time can both be adjusted so that the image displayed on the display screen of the display tube (not shown) has the correct substantially rectangular form. If, however, the required modulation depth is comparatively large, a linearity error of the line deflection is produced which cannot be removed by means of the correction coil L.
FIG. 2 shows the image of a pattern of vertical straight lines as it is displayed on the screen with the correction coil L adjusted so that horizontal linearity is satisfactory along and near the central horizontal line. In FIG. 2 the defect is exaggerated. It is found that horizontal linearity is defective in other areas of the screen so that the vertical lines are displayed correctly in the right-hand half of the screen but as curves in the left-hand path, the defect increasing as the line is farther to the left.
This phenomenon can be explained with reference to FIG. 3. In this FIG. the inductance L of the linearity correction coil is plotted as a function of the magnetic field strength H. In the absence of current, H has a value H ₀ owing to the bias magnetization. If an approximately linear sawtooth current i (t) as shown in the bottom left-hand part of FIG. 3 flows through the coil, the field strength H varies proportionally about the value H ₀ , for the mean value of the current is zero. Because the curve of L is not linear, the variation L(t) of L, which is shown in the top right-hand part, is not a linear function of time. The resulting curve may be regarded as composed of a linear component and a substantially parabolic component which is to be taken into account when choosing the capacitance of capacitor C _t .
Because owing to the east-west modulation the amplitude of current i(t) varies, the amplitude of L(t) also varies. This implies a field-frequency variation of L which is non-linear. This variation is undesirable. In the case of a small variation of the amplitude of current i(t) the variation of L(t) can be more or less neglected, but this is no longer possible when the amplitude of current i(t) varies greatly owing to the east-west modulation. L(t) varies according to different curves. FIG. 3 shows two of such curves and also illustrates the fact that the undesirable variation of L(t) is greatest at the beginning of the trace time and smallest at the end thereof.
FIG. 4 shows a circuit arrangement in which the defect described can be corrected. On the core of the correction coil L of the circuit of FIG. 1 an additional winding L ₂ is provided. Winding L ₂ is connected to a current source which produces a compensating current i ₂ which has a line-frequency sawtooth variation and a field-frequency amplitude modulation. The envelope here also is parabolic, however, with a shape opposite to that of deflection current i _y , that is to say having a minimum at the middle of the field trace time. The direction of current i ₂ and the winding sense of winding L ₂ relative to that of coil L are chosen so that the magnetic field produced in the core by winding L ₂ has the same direction as the field produced by coil L. Hence the two field strengths are added. The amplitude of current i ₂ and the turns number of winding L ₂ can be chosen so that current i _y flows through inductances the total value of which is not dependent upon the field frequency. The curve L(t) of FIG. 3 remains substantially unchanged. Consequently the undesirable field-frequency modulation is removed without variation of the bias magnetization, which would have been varied if current i ₂ were a field-frequency current. Obviously the same result can be achieved by a choice such of the direction of current i ₂ and of the winding sense of winding L ₂ that the two field strengths are subtracted one from the other, whilst the curvature of the envelope of current i ₂ has the same direction as that of the envelope of current i _y .
The current source of FIG. 4 may be formed in known manner by means of a modulator in which a line-frequency sawtooth signal is field-frequency modulated, the envelope being parabolic. FIG. 5 shows a circuit arrangement in which current i ₂ is produced by the modulation source which provides the east-west correction. In FIG. 5, the source M of FIG. 1 comprises a diode D', a coil L' and two capacitors C' _r and C' _t , which elements constitute a network of the same structure as the network formed by elements D, L _y , C _r and C _t . The capacitor C' _t is shunted by a modulation source V _m which supplies a field-frequency parabolic voltage having a minimum at the middle of the field trace time.
With the except

ion of the linearity correction means to be described hereinafter, the circuit arrangement of FIG. 5 was described in more detail in U.S. Pat. No. 3,906,305. Hence it will be sufficient to mention that the capacitances of capacitors C _r and C' _r and of a capacitor C ₁ connected between junction point A and earth and the inductance of coil L' are chosen so that the three sawtooth currents flowing through L _y , L' and L ₁ have the same retrace time. The capacitances of capacitors C _t and C' _t , which are large, are ignored. When voltage V _b is constant, current i _y is subjected to the desired east-west modulation having the form shown in FIG. 1.
Coil L _y is connected in series with correction coil L, and winding L ₂ is connected in series with coil L'. FIG. 5 shows that the current flowing through winding L ₂ has the same waveform as the current i ₂ of FIG. 4, for its envelope has the same shape as the voltage supplied by source V _m . By a suitable choice of the number of turns of winding L ₂ it can be ensured that the linearity correction remains the same for every line during the field trace time.
Modified embodiments of the circuit arrangement of FIG. 5 can also be used. FIG. 6 shows such a modified embodiment in which the capacitive voltage divider C _r , C' _r of FIG. 5 is replaced by an inductive voltage divider by means of a tapping on coil L ₁ . A capacitor C ₂ is included between the tapping and the junction point of diodes D and D', whilst capacitor C' _t here forms part of two networks C _t , L _y and C' _t , L' traversed by a sawtooth current. In FIG. 6 modulation source V _m is connected via a choke coil L ₃ to the junction point of D, D', C ₂ and C' _t . One end of winding L ₂ is connected to the junction point of capacitor C' _t and the coil L, whilst the other end is connected to earth via coil L'. The capacitances of capacitors C ₁ and C ₂ and the location of the tapping on coil L ₁ are chosen so that the sawtooth currents flowing through L _y , and L' and L ₁ have the same retrace time, whilst the field-frequency linearity defect of FIg. 2 is cancelled by correctly proportioning winding L ₂ .
Other east-west modulators are known in which the step of FIGS. 5 and 6 can be used. An example is the modulator described in the publication by Philips, Electronic Components and Materials: "110° Colour television receiver with A66-140X standard-neck picture tube and DT 1062 multisection saddle yoke," May 1971, pages 19 and 20, which modulator also comprises two diodes and a compensation coil L', which are arranged in a slightly different manner. In another example the east-west modulator and the line deflection generator are included in a bridge circuit whilst they are decoupled from one another by means of a bridge coil which has the same function as coil L' in FIGS. 5 and 6. In these circuit arrangements coil L and winding L ₂ may be arranged in the same manner as in FIG. 6. The same applies to an east-west modulator using a transductor the operating winding of which is in series with the deflection coil.
In the abovedescribed embodiments of the circuit arrangement according to the invention the compensating current i ₁ is provided by transformer action. In the embodiment of FIG. 7 the current source which supplies the current i ₂ is connected in parallel with correction coil L, i.e., without an auxiliary winding. In this embodiment the east-west modulation is achieved not by means of a modulator, but by means of the fact that the supply voltage V _b is the super-position of a field-frequency parabolic voltage on the direct voltage. In this known manner the supply source also is the modulator.
It will be seen that in the embodiments of FIGS. 4, 5 and 6 current i ₂ counteracts the east-west modulation of deflection current i _y . It was found in practice, however, that this counteraction is slight.

PHILIPS 26C568 CHASSIS K11 (20AX) NORTH SOUTH (NORD / SUD) CORRECTION CIRCUIT ARRANGEMENT FOR CORRECTING THE DEFLECTION OF AT LEAST ONE ELECTRON BEAM IN A TELEVISION PICTURE TUBE BY MEANS OF A TRANSDUCTOR :


A circuit arrangement for raster correction in a television picture tube by means of a transductor whose power winding is connected in parallel with at least a portion of the line deflection coils, the line deflection generator having a low internal impedance. In order to increase this impedance a mainly inductive impedance is connected in series with the generator. In a picture tube employing at least two electron beams the series impedance may include the convergence circuit. As a result the convergence in the corners of the picture screen is also improved. The linearity control circuit may likewise form part of the series impedance.



1. A deflection circuit for a cathode ray tube comprising a transistor horizontal deflection generator; a horizontal deflection coil parallel coupled to said generator; means for pincushion correction of said tube comprising a saturable reactor having a control winding adapted to receive a vertical deflection signal and a power winding parallel coupled to at least a portion of said deflection coil; and means for increasing the effectiveness of said correction means comprising an impedance element external to said generator having a substantially inductive reactance series coupled between said generator and said coil. 2. A circuit as claimed in claim 1 wherein said generator comprises a transformer having a tap and said power winding has a first end coupled to said coil and a second end coupled to said tap. 3. A circuit as claimed in claim 1 wherein said impedance element comprises means for controlling the linearity of the beam deflection. 4. A deflection circuit for a cathode ray tube having at least two electron beams comprising a transistor horizontal deflection generator; a horizontal deflection coil parallel coupled to said generator; means for pincushion correction of said tube comprising a saturable reactor having a control winding adapted to receive a vertical deflection signal and a power winding parallel coupled to at least a portion of said deflection coil; means for increasing the effectiveness of said correction means comprising an Impedance element external to said generator having a substantially inductive reactance series coupled between said generator and said coil; and means for dynamically converging said beams comprising a convergence circuit coupled to said horizontal generator and to said transductor. 5. A circuit as claimed in claim 4 wherein said generator comprises a transformer having a tap and said power winding has a first end coupled to said coil and a second end coupled to said tap. 6. A circuit as claimed in claim 4 wherein said impedance element comprises means for controlling the linearity of the beam deflection.

Description:

The invention relates to a circuit arrangement for correcting the deflection of at least one electron beam (raster correction) in a television picture tube by means of a saturable reactor a power winding of which is connected in parallel with at least a portion of the coils for the horizontal deflection, the current flowing through these coils being supplied by a deflection generator having a low internal impedance.

A circuit arrangement for ras

ter correction with the aid of a transductor is described, for example, in U.S. Pat. No. 3,444,422. In this patent the power winding of a transductor is connected in parallel with the horizontal deflection coils while the control winding receives a signal of field frequency so that the current of line frequency which flows through the deflection coils is modulated at the field-frequency (East-West correction), whereas the vertical deflection current is modulated at the line frequency (North-South correction). However, in this known arrangement there is the difficulty that the transductor can exert little influence on the horizontal deflection current if the internal impedance of the deflection generator is low because the transductor then only constitutes an additional load on the generator. This is the case when the deflection generator includes a valve with feedback -- or a switch formed with one or more transistors. In order to be able to use a transductor arrangement also in such a case the circuit arrangement according to the invention is characterized in that a mainly inductive impedance is connected in series between the said parallel arrangement and the deflection generator.

Due to the step according to the invention the internal impedance of the deflection generator is increased and the different components of the circuit remain mainly inductive so that the deflection current is more or less linear when the voltage provided by the deflection generator during the line scan period is substantially constant. The series impedance may be, for example, a fixed coil. However, the invention is furthermore based on the recognition of the fact that the increase in the internal resistance of the horizontal deflection generator may not only be obtained by a constant impedance, but other arrangements envisaging other improvements of the deflection may be used for this purpose. In that case even special improvements may be obtained as will be apparent hereinafter and possible small non-linearities of the additionally used arrangements have no detrimental results.

It is true that in known convergence circuits in picture tubes employing a plurality of electron beams a satisfactory improvement is obtained for the central horizontal and vertical lines of a picture tube of the shadow mask type. However, it is found that convergence errors may subsist in the corners of the picture. Known circuit arrangements which correct these second-order errors are often complicated and expensive. In the circuit arrangement according to the invention a satisfactory compensation of such convergence errors is possible in a simple manner if the series impedance which is arranged between the horizontal deflection generator and the deflection coils includes the convergence circuit. In this manner the sum of the deflection current and of the current derived for the field correction and modulated by the transductor flows through the convergence circuit so that the desired additional convergence correction in the corners of the written raster is obtained.

In order that the invention may be readily carried into effect a few embodiments thereof will now be described in detail by way of example with reference to the accompanying diagrammatic drawings in which:

FIG. 1 shows a circuit arrangement in which the transductor is connected in parallel with the deflection coils, while in

FIG. 2 the transductor is only fed by part of the voltage applied to the deflection coils.

FIG. 1 shows two line-output transistors 1 and 2 which are arranged in series. The emitter of transistor 2 is connected to ground through a winding 3 while the collector of transistor 1 is connected through a winding 4 and a small series impedance 5, preferably a resistor, to the positive terminal of a supply source V _b whose negative terminal is connected to ground.

Windings 3 and 4 are wound together with an EHT-winding 6 on the same transformer core 7. The ends of windings 3 and 4 remote from each other are connected through the capacitor 10 for the S-correction to the deflection-unit consisting of two windings 8 and 9 arranged, for example, in parallel. The base of transistors 1 and 2 receive pulses of line frequency in a manner not shown in FIG. 1 so that these transistors are cut off during the flyback period. During the scan period, a substantially constant voltage is applied to the deflection unit. Consequently a more or less sawtooth-shaped current flows through windings 8 and 9. The bipartite power winding 11 of a transductor ensuring the raster correction is connected in parallel with this deflection unit 8, 9. The control winding 12 of said transductor, and a converting capacitor 13 in parallel therewith form part of the circuit for the vertical deflection through terminals 14 and 15. An adjustable coil 16 with which the raster correction can be adjusted exactly is connected in series with winding 12.

Windings 3 and 4 have the same number of turns so that pulses of the same amplitude and reversed polarity are produced at the emitter of transistor 2 and at the collector of transistor 1. As a result a disturbing radiation of these pulses is reduced. Furthermore, transistor types are chosen in this Example for transistors 1 and 2 whose collector-base diodes may function as efficiency diodes. All this has been described in U.S. Pat. No. 3,504,224.

According to the invention the convergence circuit 17 is arranged through a separation transformer 20 between the end of winding 3 remote from winding 4 and the horizontal deflection coils 8, 9. Furthermore, this current branch includes the linearity control circuit 21 which comprises the parallel arrangement of a resistor and a coil whose inductance is adjustable, for example, by means of premagnetization of the core of the coil. A current, which is the sum of the current for the deflection coils 8, 9 and of the current for the power winding 11 of the transductor, flows through the primary winding of transformer 20. This primary current is transformed to the secondary circuit of transformer 20 so that a current flows through convergence circuit 17.

In known arrangements the convergence current is only influenced by the deflection current itself. It has been found that in this case the convergence correction is not sufficient in the corners of the picture. At these areas, where the deflection in both directions is at a maximum, a greater intensity of the convergence current is required. This is especially the case in picture tubes having a great deflection angle and according to the invention this is achieved in that the current which is derived from the power winding 11 of the transductor for the raster correction is also applied to the convergence circuit. This current flows from the horizontal deflection generator constituted by windings 3 and 4 through the primary winding of transformer 20 to power winding 11 of the transductor. The transductor current is in fact at a minimum in the center of the picture and increases towards the edges and particularly towards the corners. Thus the convergence current varies in the desired manner. According to the invention the desired improvements of the convergence correction and simultaneously the likewise desired increase in the internal resistance of the horizontal deflection generator is consequently obtained without a considerable increase in the number of required circuit elements and without disturbing the normal operation of the circuit arrangement. Due to transformer 20 a terminal of convergence circuit 17 may be connected to ground so that the convergence can be adjusted safely. If necessary, a suitable impedance transformation may also be obtained with the aid of transformer 20.

The linearity control circuit 21 may alternatively be connected in series with the said branch which includes transformer 20. As a result the internal resistance of the horizontal deflection generator for the line frequency is further increased without the field correction and the convergence correction being disturbingly influenced.

FIG. 2 shows a modification of the circuit arrangement according to the invention in which the deflection current is not changed relative to that of FIG. 1. The end of power winding 11 of the transductor shown on the upper side of FIG. 1 is connected to ground in FIG. 2. In addition convergence circuit 17 is included between winding 3 and ground so that separation transformer 20 may be omitted. If as a first approximation the impedances 5 and 17 are assumed to be negligibly small relative to the other impedance of the circuit arrangement, power winding 11 may be considered to be connected to a tap on the deflection generator 3, 4. Consequently, only approximately half the voltage of the deflection generator is applied to transductor winding 11 which winding must therefore be proportioned in such a manner that it can convey a current which is approximately twice as large as that of FIG. 1. This larger current also flows through convergence circuit 17 which, with the omission of separation transformer 20, is favorable for the convergence in the corners of the picture screen.

In FIG. 2 the emitter of transistor 2 is connected to ground i.e., the said tap on the deflection generator. During the scan period the series arrangement of supply source V _b and windings 3 and 4 FIG. 1 is substantially short-circuited by transistors 1 and 2. In order that these transistors in the circuit arrangement according to FIG. 2 operate under the same circumstances as those in FIG. 1, an additional winding 24 must be wound on core 7 between windings 4 and 6, winding 24 having the same number of turns as winding 3, and the collector of transistor 1 must be connected to the junction of windings 6 and 24.

The end of power winding 11 connected to ground in FIG. 2 may alternatively be connected for the desired adjustment of the corner convergence to a different tap on the transformer, that is to say, on winding 3 or 4.

Resistor 5 serves in known manner mainly as a safety resistor so that in case of an inadmissible load of the EHT, for example, as a result of flash-over in the picture tube, the supply voltage for transistors 1 and 2 is reduced so that overload of these transistors is avoided.

Convergence CORRECTION circuit 20AX PHILIPS CHASSIS K11
PHILIPS 26C568 CHASSIS K11 Colour television display apparatus incorporating a television display tube

Colour television display apparatus provided with a display tube in which the electron beams are generated in one plane. The deflection coils are designed in such known manner that the landing points of the beams on the display screen coincide. Owing to tolerance angular errors of the orientation of the plane relative to one direction of deflection convergence errors are produced in the other direction of deflection. They are corrected by means of a magnetic quadripolar field the polar axes of which substantially coincide with the directions of deflection and the field strength of which is a substantially quadratic function of either deflection current or the sum of both functions.

1. Colour television display apparatus incorporating a television display tube having a display screen and two deflection coils for the deflection in two directions of electron beams which are generated in the tube substantially in one plane, a first direction of deflection being substantially parallel to the said plane whilst the second direction of deflection is substantially at right angles to the first direction, the field generated by the deflection coil for deflection in the first direction having a distribution in which its meridional image plane substantially coincides with the screen whilst the field generated by the deflection coil for deflection in the second direction has a distribution in which its sagittal image plane substantially coincides with the screen, the deflection errors due to comma and anisotropic astigmatism being substantially equal to zero, whilst at least one deflection coil is divided into two substantially equal coil halves, characterized in that in order to correct for tolerance angular errors in the orientation of the plane in which the electron beams are generated relative to the first direction of deflection the split deflection coil generates a magnetic quadripolar field the polar axes of which substantially coincide with the directions of deflection and the field strength of which is a substantially quadratic function of the instantaneous strength of the deflection current flowing through at least one deflection coil, and means for clamping the peak of said quadratic field. 2. Apparatus as claimed in claim 1, characterized in that a substantially parabolic correction current which is adjustable in amplitude and in polarity flows in the same direction as the deflection current in one coil half and in the opposite direction in the other coil half and is zero at the middle of the trace interval of the deflection current. 3. Apparatus as claimed in claim 2, in which one direction of deflection is horizontal and the other is vertical, characterized in that a line-frequency correcting current flows through the coil halves of the deflection coil for horizontal deflection and a field-frequency correction current flows through the coil halves of the deflection coil for vertical deflection. 4. Apparatus as claimed in claim 2, characterized in that a sawtooth current supplied by the deflection current generator which produces the deflection current flows through a potentiometer the setting of the slider on which determines the adjustment of the polarity and of the amplitude of the correcting current. 5. Apparatus as claimed in claim 4, in which the deflection current is of field frequency, characterized in that the setting of the slider on the potentiometer also renders symmetrical the deflection fields generated by the coil halves. 6. A display apparatus as claimed in claim 1 wherein said split coil field strength is substantially the sum of quadratic functions of the current flowing through both coils. 7. A color television deflection system for a television display tube having a display screen, said system comprising two deflection coils for the deflection in two directions of electron beams which are generated in the tube substantially in one plane, said first direction of deflection being substantially parallel to the said plane, the second direction of deflection being substantially at right angles to the first direction, the field generated by the deflection coil for deflection in the first direction having a distribution in which its meridional image plane substantially coincides with the screen, the field generated by the deflection coil for deflection in the second direction having a distribution in which its sagittal image plane substantially coincides with the screen, the deflection errors due to comma and anisotropic astigmatism being substantially equal to zero, at least one deflection coil comprising two substantially equal coil halves, means for correcting for tolernace angular errors in the orientation of the plane in which the electron beams are generated relative to the first direction of deflection comprising means for providing that the split deflection coil generates a magnetic quadripolar field the polar axes of which substantially coincide with the directions of deflection and the field strength of which is a substantially quadratic function of the instantaneous strength of the deflection current flowing through at least one deflection coil, and means for clamping the peak of said quadratic field. 8. A deflection system as claimed in claim 7 wherein said split coil field strength is substantially the sum of quadratic functions of the current flowing through both coils.

Description:

The invention relates to a colour television display apparatus incorporating a television display tube having a display screen and two deflection coils for the deflection in two directions of electron beams which are generated in the tube substantially in one plane, a first direction of deflection being substantially parallel to the said plane whilst the second direction of deflection is substantially at right angles to the first direction, the field generated by the deflection coil for deflection in the first direction having a distribution in which its meridional image plane substantially coincides with the screen whilst the field generated by the deflection coil for deflection in the second direction has a distribution in which its sagittal image plane substantially coincides with the screen, the deflection errors due to comma and anisotropic astigmatism being substantially zero, whilst at least one deflection coil is split into two substantially equal coil halves.

Such an apparatus is described by J. Haantjes and G. J. Lubben in "Philips Research Reports", Volume 14, February 1959, pages 65-97 and in U.S. Pat. No. 2,886,125. In this apparatus the landing points of the electron beams on the display screen coincide everywhere, in other words the various beams, which generally are three in number, which intersect the deflection plane along a straight line are imaged as points on the screen. It is assumed that both the construction of the device or devices which generate the beams, for example three cathodes, and the distribution of the deflection fields exactly satisfy the requirements derived in the said paper. In practice, however errors are produced which are due to tolerances so that the images of the beams on the screen are not points but lines which are substantially parallel to the second direction, i.e. convergence errors, for when a point is referred to what is actually meant is that each electron beam strikes a phosphor dot or stripe on the screen to cause it to luminesce in a given colour, the landing points being associated so as to be perceived as a single point. This is no longer the case if the aforementioned straight line, which is the projection of the plane of the three cathodes in the deflection plane, does not exactly coincide with the first direction of deflection but is at an angle thereto. This error is a tolerance error, i.e. it is small, and may be due to a slight misplacement of the cathodes and/or to a slightly incorrect field distribution within the display tube and hence to tolerances in the construction of the deflection coils.

If the first direction of deflection is horizontal and the second one is vertical, the said error entails a convergence error in the vertical direction. The aforementioned straight line in the deflection plane can be made to coincide with the horizontal direction of deflection by rotation. Attempts have been made to cancel the convergence errors due to this rotation by means of a coil which is axially arranged on the neck of the display tube and through which an adjustable direct current flows. The effect of this coil is comparable to that of a focussing coil; it exerts a force on the travelling electrons which causes their paths to be helical, so that some compensation is obtained. It has been found, however, that this solution has the following disadvantages: the residual errors in the corners are increased; the effect on the horizontal and vertical directions are different, so that satisfactory adjustment in both directions is difficult to realise; depending upon the axial position an undesirable influence may occur at the centre of the screen which in turn can be eliminated by the means, for example permanent magnets, provided for static convergence, requiring iterative and hence time-consuming trimming. Furthermore the coil is an expensive component.

The present invention is based on the recognition that the aforementioned convergence errors due to tolerance errors in the construction of the display tube and/or of the deflection coils can be eliminated by means of simple circuits without the need for additional components to be mounted on the neck of the display tube whilst avoiding the aforementioned disadvantages. For this purpose the apparatus according to the invention is characterized in that to correct for tolerance angular errors in the orientation of the plane in which the electron beams are generated relative to the first direction of deflection the split deflection coil generates a magnetic quadripolar field the polar axes of which substantially coincide with the directions of deflection and the field strength of which is a substantially quadratic function of the instantaneous strength of the deflection current flowing through either deflection coil or the sum of both quadratic functions.

It should be mentioned that it is known to use a split deflection coil to generate a quadripolar field the polar axes of which substantially coincide with the directions of deflection. This is described in U. S. Pat. No. 3,440,483 in which, however, the field strength of the quadripolar field is a function of the product of the values of the two deflection currents so that deflection errors due to anisotropic astigmatism can be corrected. In contradistinction thereto the present application described an apparatus having substantially no anistropic astigmatism whilst the quadripolar field generated according to the invention has a field strength which depends upon the value of either deflection current or upon the sum of the squares of the two deflection currents. For the sake of clarity it should be mentioned that in the apparatus according to the said U.S. Patent, in the absence of the correction quadripolar field described, the image of a beam on the screen is a tilted ellipse, whereas in the present application the corresponding image when not corrected is a vertical line.

The known apparatus has some isotropic astigmatism so that the vertical focal lines, i.e. the Meridional focal lines of the horizontal deflection plane and the sagittal focal lines of the vertical deflection plane, coincide with the display screen. Since the imaginary ribbon-shaped beam produced by the three beams together has substantially no dimension in the vertical direction, its image on the screen is a point. In these circumstances the term "isotropic astigmatism" as used herein in actual fact is to be understood to mean that the coefficients which determine the isotropic astigmatism differ from the desired values. Consequently the cross-sectional area on the screen of the imaginary thick beam of circular cross-section in the deflection plane (see FIG. 2 of the said paper in which, however, the three beams are generated in a vertical plane) does not degenerate into a straight line but takes the form of an ellipse the axes of which are parallel to the directions of deflection. Means for correcting such undesirable isotropic astigmatism is described in U.S. patent application Ser. No. 447,564 filed March 4, 1974. In this means a correcting quadripolar field which varies with the square of the strength of either deflection current is generated in the deflection region. However, the axes of said quadripolar field lie substantially along the diagonal between the axes of the deflection directions and the field is generated by separate windings and not by the deflection coil or coils. It should be noted that the apparatus according to the invention also may be subject to this defect which in this case may be corrected in the manner described in the said U.S. patent application. For the sake of simplicity this will be disregarded hereinafter, that is to say the deflection coil will be assumed to have the correct degree of isotropic astigmatism, causing the landing points of the beams on the screen to coincide in one point everywhere but for the abovementioned tolerance error.

In order that the invention may be more readily understood, embodiments thereof will now be described by way of example with reference to the accompanying diagrammatic drawings, in which:

FIG. 1 is a sectional view of a colour television display tube subject to the defect to be corrected,

FIG. 2 shows schematically the ensuring convergence error on the display screen of the tube,

FIGS. 3, 4, 5 and 7 are circuit diagrams of embodiments of correction circuits, and

FIG. 6 is a wave form obtained in the circuit of FIG. 5.

FIG. 1 is a simplified elevation of a cross-section of a colour television display tube 1 taken on the deflection plane at right angles to the axis of the tube in a direction opposite to the direction of propagation of the electron beams, the deflection coils being omitted for simplicity. Three electron beams L, C and R are generated in one plane, the beam C substantially coinciding with the axis of the tube 1 and the beams L and R being located to the left and to the right respectively of the beam C. If the construction of the devices, for example cathodes, which generate the beams and the field distribution within the tube 1 were exactly as desired, the points of intersection of the beams L, C and R with the deflection plane would be a straight line coinciding for example with the X axis, which coincides with the direction of horizontal deflection, the Y axis coinciding with the direction of vertical deflection. However, owing to tolerances the points of intersection lie on a straight line D which is at an angle α to the X axis which it intersects in C.

The paper mentioned in the second paragraph of this application shows that an imaginary thick beam may be considered the cross-section S of which with the plane of deflection is a circle. The line section LCR of FIG. 1 is a diameter of this circle. If the horizontal deflection field has a distribution in which the meridional image plane substantially coincides with the display screen of the tube 1 whilst the vertical deflection field has a distribution in which the sagittal image plane substantially coincides with the screen, and if moreover the deflection errors due to comma and both anisotropic and isotropic astigmatism are substantially equal to zero, all the points on and within the circle S are imaged on vertical line everwhere on the screen. It is supposed that the correct degree of isotropic astigmatism is actually obtained. Otherwise the image of the circle S would be an ellipse the axes of which are parallel to the X and Y axes, i.e. there would be a horizontal convergence error.

In these circumstances the beams L, C and R of FIG. 3 are imaged on the screen 2 of the tube 1 along vertical lines, some of which are shown (in exaggerated form) in FIG. 2, with the exception of the image at the midpoint of the screen, i.e. without deflection, where they coincide. In the ideal case in which the beams L and R of FIG. 1 would lie on the X axis, i.e. with α = 0, in each triplet L', C', R' in FIG. 2 the points L' and R' would coincide with the point C'. Consequently the error angle α results in a vertical convergence error on the screen. In FIG. 1 the beam L lies above the X axis and the beam R beneath the X axis. Because the beams cross within the tube, the points L' and R' in FIG. 2 always lie beneath and above the point C' respectively.

According to the invention a magnetic correction quadripolar field is generated the polar axes of which substantially coincide with the X and Y axes and four lines of force of which are shown in FIG. 1. The quadripolar field does not influence the beam C which is located at the centre of the deflection plane. The beams L and R are subject to forces F _L and F _R respectively which are superposed on the forces exerted by the deflection fields. FIG. 1 shows that as a result the angle α is effectively reduced to substantially zero so that the convergence error of FIG. 2 is cancelled.

Such a quadripolar field is obtainable by causing an additional current, the difference current, to flow through a deflection coil divided in two coil halves in a manner such that the said current is added to the deflection current in one coil half and subtracted from it in the other coil half. FIG. 2 shows that the convergence errors on the left-hand and right-hand halves of the screen 2 have the same sign and that they have the same sign in the upper and lower halves. Hence it is desirable for the value of the difference current to vary substantially as the square of each deflection. Because initially the value and polarity of the angle α are unknown, the current must be adjustable both in amplitude and in polarity. At the middle of the line and field trace intervals the angle must be zero. For this purpose either one or both deflection coils may be used.

Because the images L', C', R' in FIG. 2 are vertical, i.e. are not tilted, the convergence error to be corrected is to be considered as an isotropic astigmatic deflection error. Hence the line-frequency component of the difference current must be a function of horizontal deflection only and its field-frequency component must be a function of vertical deflection only. Thus it is simpler, but not necessary, to cause the line-frequency component of the difference current to flow through the split deflection coil for horizontal deflection and its field-frequency component to flow through the split deflection coil for vertical deflection.

FIG. 3 shows a simple circuit for generating a line-frequency difference current which satisfies the said requirements. A line deflection current generator 3 at one terminal supplies a line-frequency sawtooth current i _H to line deflection coil halves 4' and 4", which in this embodiment are connected in parallel for the current i _H . Adjustable coils 5' and 5" of low inductance are connected in series with the coil halves 4' and 4" respectively. The coils 5' and 5" may be adjusted jointly and oppositely so as to eliminate in knwon manner any asymmetry of the deflection fields generated by coil halves 4' and 4". The ends of the coils 5' and 5" not connected to the coil halves 4' and 4" respectively are connected to one another via a potentiometer 6 the slider on which is connected to the other terminal of the generator 3. The resistance of, for example, 4.7 ohms of the potentiometer 6 is low compared with the impedance of the coil halves 4' and 4" for the line repetition frequency. Thus a sawtooth voltage the polarity and amplitude of which depend upon the position of the slider is produced across the potentiometer 6. As a first approximation this voltage may be considered as being produced by a voltage source of low internal impedance. The coil halves 4' and 4" pass a current which is proportioned to the integral of the voltage across the potentiometer 6 and consequently is the required parabolic correction difference current i _KH . In one coil half it flows in the same direction as the current i _H /2 and in the other coil half it flows in the opposite direction. For this purpose it is required that the position of the slider on the potentiometer 6 should differ from the electric midpoint thereof.

The potentiometer 6 is shunted by the series combination of two resistors 7' and 7" the junction point of which is connected to the anode of a diode 8 the cathode of which is connected to the slider on the potentiometer 6. The diode 8 and the resistors 7' and 7" ensure that the peak of the parabola will be at zero. In actual fact the diode 8 produces a direct current which compensates for the sagging of the parabola, provided that the resistances of the resistors 7' and 7" are equal and have the correct value, for example 8.2 ohms. This direct current also is a difference current and since the diode 8 is connected to the slider on the potentiometer 6 the reversal of its polarity is automatically effected together with that of the parabolic component.

A disadvantage of the circuit of FIG. 3 may be that the obtainable amplitude of the current i _KH is limited because the permissible value of the potentiometer 6 is limited, for a comparatively large value of this potentiometer will increase dissipation and give rise to a linearity error of the deflection current whilst the current i _KH will no longer be parabolic but will also include higher-order components. The amplitude i _KH may be increased without increasing the resistance of the potentiometer 6 by coupling the latter to the remainder of the circuit by means of a transformer. This may be achieved by an autotransformer, as is shown in FIG. 4. Two windings 19' and 19" which are bifilarly wound on the same core and have the polarities shown are connected in series between the ends of the coils 5' and 5" not connected to the coil halves 4' and 4" respectively. The potentiometer 6 is connected between two tappings on the windings 19' and 19" which are symmetrical with respect to the junction point thereof and the potentiometer slider is connected to said junction point via the series combination of the diode 8 and a resistor 7.

In the circuit shown in FIG. 4 the operation of the balancing coils 5' and 5" is not disturbed, provided that the overall inductance value of the windings 19' and 19" between the junction point of the winding 19' and the coil 5' and the junction point between the winding 19" and the coil 5" is small compared with the inductance value of the coil halves 4' and 4" and the coils 5' and 5" measured between the same points. In a practical embodiment of the circuit of FIG. 4 the latter value is 3.55 mH and the former value is 1.25 mH. This means that the effect of the balancing coils 5' and 5" is reduced by only about one third. The tappings on the windings 19' and 19" are provided at the midpoints thereof, the value of the resistor 6 is about 3.3 ohms and that of the resistor 7 about 0.5 ohm. It should be noted that the resistance of the windings 19' and 19" should not be too small, for otherwise the direct component of the difference current would be short-circuited.

FIG. 5 shows a simple circuit for producing a field-frequency difference current in field-frequency deflection coil halves 9' and 9". Since these coil halves are predominantly resistive for the field repetition frequency, the circuits shown in FIGS. 3 and 4 cannot be used. A field deflection current generator 10 supplies a fieldfrequency sawtooth current i _V to coil halves 9' and 9" which are connected in series in this embodiment. The series combination of a diode 11', a potentiometer 12' and a second diode 13' and the series combination of a third diode 11", a potentiometer 12" and a fourth diode 13" are connected in parallel with the series combination of the said coil halves, the said four diodes having the polarities shown in FIG. 5. An isolating resistor 14' is connected between the slider on the potentiometer 12' and the junction point of the coil halves 9' and 9", and an isolating resistor 14" is connected between the slider of the potentiometer 12" and the said junction point, the values of the isolating resistors being high relative to the impedance of the coil halves, for example about 100 ohms.

During one half of the field trace interval the current i _V flows in the direction shown. Diodes 11' and 13' are conducting whereas diodes 11" and 13" are cutoff. Across the potentiometer 12' a sawtooth voltage is produced so that, if the position of the slider of the potentiometer 12' is different from the electric midpoint of the potentiometer, a sawtooth correction difference current i' _KV flows through the coil half 9', for example in a direction opposite to that of the current i _V , whilst the coil half 9" passes a sawtooth correction difference current i" _KV in the same direction as the current i _V , the currents i' _KV and i" _KV being substantially equal. It should be noted that a difference current, in this embodiment i" _KV , flows through a diode, in this embodiment 13', from the cathode to the anode. However, because the elements 9', 9", 11', 12', 13' and 14' form a Wheatstone bridge comprising resistors, the diodes cannot be cut off.

During the other half of the trace interval current i _V flows in the other direction. The diodes 11" and 13" are conducting and the diodes 11' and 13' are cut off. Sawtooth difference currents are produced which are derived from the slider on potentiometer 12". In FIG. 6 the variation of the extreme value i _KVmax of the difference currents is shown as a function of time, T denoting the field trace interval. At the middle of the interval T these currents are zero, because the current i _V and hence the voltage across the potentiometer 12' or 12" respectively are zero. Owing to the voltage drop across the diode the difference currents are zero for a certain time before and after the middle of the interval T. The resulting curves may be regarded as approximate parabolas, for practice has shown that the residual convergence error is negligibly small. Because the difference currents produced are sawtooth currents, the potentiometers 12' and 12" ensure also that the deflection fields generated by the coil halves 9' and 9" are symmetrical. An advantage of the circuit of FIG. 5 is that the adjustments of the upper half and of the lower half are independent of one another, which conduces to clarity. In the embodiment described both potentiometers have a resistance of about 330 ohms.

It will be appreciated that the quadripolar field generated will only be capable of correcting for the vertical convergence error if the angle α is very small. The error introduced by the incorrect position of the line D is compensated for by the quadripolar field according to the invention, it is true, however, at large values of the angle α this field in turn introduces new errors, especially in the corners of the screen. Practice has shown that an angle of from 2° to 3° still can be corrected.

Hereinbefore no statement has been made about the construction of the deflection coils. If they are in the form of saddle coils, no special steps are required. If, however, they are wound toroidally, a step as described in U.S. patent application Ser. No. 390,701 filed August 23, 1973 must be used which consists in the introduction of the difference currents into the deflection coil halves via tappings. In this case the simple circuits of FIGS. 3, 4 and 5 are to be replaced by circuits in which the parabolic difference currents are generated in a different manner, for example by separate generators.

In the embodiments described the coil halves 4' and 4" for horizontal deflection are connected in parallel for the line deflection current i _H , whereas the coil halves 9' and 9" for vertical deflection are connected in series for the field deflection current i _V . Obviously this is not of importance for the invention and the coil halves may be connected in a different manner. FIG. 7 shows an embodiment in which the coil halves 4' and 4" are connected in series for the current i _H . In this embodiment two diodes 8' and 8" are required. It will further be appreciated that the invention may also be applied if the electron beams are generated in a plane of substantially vertical orientation, in which case the convergence error to be corrected is horizontal.

PHILIPS 26C568 CHASSIS K11 ULTRASONIC REMOTE CONTROL RECEIVER PHILIPS CHASSIS K11.




Includes on separated PCB the Ultrasonic Remote Control Receiver Devices Assy's With Amplifier and Discriminator and LOGIC.





(NOTE that PHILIPS has developed his own circuit, and these descriptions are for educational purphoses)
Based on ASIC Circuit AMI DD7661 and DD7662 A
Icludes Basic commands: Tuning search mode FAST Tuning search mode Fine/Slow.
Contrast Setting, Store Channel, band selection, sound tone controls, muting, bright setting, volume, ON/OFF, OSD !!
and a led wich indicates perfect in tuning channel mode (AFC).





An ultrasonic remote control receiver wherein an incoming ultrasonic signal is converted to square wave pulses of the same frequency by a Schmitt trigger circuit; digital circuits are thereafter used to count pulses resulting from the incoming signal over a predetermined period of time; a decoder activates one of a plurality of outputs in dependance to the number of pulses counted, provision is made to prevent interference signals from producing undesired control outputs.



1. An ultrasonic remote control receiver for applying a control signal to a selected one of a plurality of control channels in response to and dependent on the frequency of a received ultrasonic signal comprising:

2. An ultrasonic remote control receiver comprising:

3. An ultrasonic remote control receiver comprising:

4. The ultrasonic remote control receiver as defined in claim 3, wherein said means producing square pulses is a Schmitt trigger circuit and said means providing a signal input to said sequence controller is a retriggerable monostable multivibrator.

5. An ultrasonic remote control receiver comprising:

6. An ultrasonic remote control receiver comprising:

7. An ultrasonic remote control receiver as defined in claim 6 further comprising a monostable multivibrator between the output of said Schmitt trigger circuit and the remaining elements of said receiver.

8. An ultrasonic remote control receiver as defined in claim 6 further comprising a bistable multivibrator between the output of said Schmitt trigger circuit and the remaing elements of said receiver.

9. The ultrasonic remote control receiver as defined in claim 7 wherein the hold period of said monostable multivibrator is slightly less than one half the period of said square wave pulses from said Schmitt trigger circuit.

Description:

The invention relates to an ultrasonic remote control receiver for receiving signals having different useful frequencies each associated with a channel, comprising a plurality of outputs which are ach associated with one of the channels and from which a control signal is emitted on receipt of a signal having the corresponding useful frequency.

To obtain the simplest possible transmitter construction in ultrasonic remote control, modulation of the emitted ultrasonic frequencies is not employed; to control different operations different frequencies are emitted which must be recognized in the receiver and evaluated for carrying out the different functions associated therewith. Presently, to recognize the different frequencies, use is made of resonant circuits, each of which contains one or more coils tuned in each case together with a capacitor to one of the useful frequencies.

These hitherto known receivers have numerous disadvantages. Thus, for example, before starting operation of the receiver a time-consuming alignment procedure must be carried out with which the resonant frequencies of the individual resonant circuits are set. Since it is inevitable that with time the resonant circuits become detuned, it may be necessary to repeat the alignment procedure.

A further disadvantage is that the known receivers cannot be made by integrated techniques because the coils used therein are not suitable for such techniques.

The problem underlying the invention is thus to provide an ultrasonic remote control receiver of the type mentioned at above which is extremely simple to set and in addition can be made by integrated techniques.

To solve this problem, according to the invention an ultrasonic remote control receiver of the type mentioned above contains a counter for counting the useful frequency oscillations received during a fixed measuring time, a sequence control device which determines the measuring time and which is started on receipt of a useful frequency, and a decoder comprising several outputs which is connected to the outputs of the counter, said decoder emitting a control signal at the output associated with the count reached at the end of the measuring time.

In the receiver constructed according to the invention the frequency emitted by the transmitter is identified by counting the oscillations received during a measuring time. The evaluation of the count reached at the end of the measuring time takes place in a decoder which emits a control signal at a certain output according to the count. The measuring time is fixed by a sequence control device which is set in operation on receipt of useful frequency signals.

In such a receiver the only quantity which has to be exactly fixed is the measuring time; it is therefore no longer necessary to align components to certain frequencies. Since no coils are required, the novel receiver can also be made up of integrated circuits.

A further development of the invention resides in that an interference identifying device is provided which on receipt of interference frequencies differing from the useful frequencies interrupts the operation of the sequence control device.

Hitherto known ultrasonic remote control receivers respond to any oscillation received if the frequency thereof has a value which excites a resonant circuit in the receiver. There is no way of distinguishing between oscillations received from the remote control transmitter and from interference sources.

Interfering ultrasonic oscillations may be due to many different causes. For example, noises such as hand clapping, rattling of short keys such as safety keys, operating cigarette lighters, rattling of crockery and the like cover a frequency spectrum reaching from the audio frequency range far into the ultrasonic region. The ultrasonic components may have the effect of simulating a useful frequency and cause an erroneous function in the receiver.

The interference identifying device according to the further development is constructed in such a manner that it recognizes oscillations having frequencies deviating from the useful frequencies and as a result of this recognition switches off the sequence control device. This switching off prevents the counter state reached from being passed to the decoder and consequently the latter cannot emit an erroneous control signal.

With this further development of the ultrasonic remote control receiver the operation of equipment such as radio and television sets is made extremely reliable and interference-free. During the operation of such a set it is no longer possible for the remote control to become operative, triggered by interference noises, eliminating for example the possibility of unintentional program or volume changes.

Examples of embodiment of the invention are illustrated in the drawings, wherein:

FIG. 1 shows a block circuit diagram of a remote control receiver according to the invention;

FIG. 2 is a diagram explaining the mode of operation of the circuit according to FIG. 1;

FIG. 3 shows another embodiment of the invention;

FIG. 4 is a diagram explaining the mode of operation of the circuit according to FIG. 3;

FIG. 5 is a diagram illustrating interference frequency identification in the circuit according to FIG. 3;

FIG. 6 shows a block circuit diagram of another embodiment of part of the circuit according to FIG. 3;

FIG. 7 is a diagram explaining the mode of operation of the embodiment according to FIG. 6;

FIG. 8 is a block circuit diagram of a further embodiment of a part of the circuit according to FIG. and, an

FIG. 9 is a diagram explaining the mode of operation of the embodiment according to FIG. 8.

The ultrasonic remote control receiver shown in FIG. 1 comprises an input 1 which is connected to an ultrasonic microphone intended to receive ultrasonic signals coming from a remote control transmitter. For each function to be performed by the receiver the remote control transmitter emits one of several unmodulated different useful frequencies which are spaced from each other a constant channel spacing Δ f and which all lie within a useful frequency band.

To obtain a signal which is as free as possible from noise at the input 1, a band filter and a limiting amplifier are preferably incorporated between the ultrasonic microphone and the input 1. The band filter may be made up of two active filters whose resonant frequencies are offset with respect to each other so that a pass band curve in the useful frequency band is obtained which is as flat as possible.

The input 1 leads to a Schmitt trigger 2 which converts the electrical signal applied thereto with the frequency of the ultrasonic signal to a sequence of rectangular pulses. The output 3 of the Schmitt trigger 2 is connected to the input 6 of a frequency divider 7 which is in operation for the duration of a control pulse applied to its control input 8 and divides the recurrence frequency of the pulses supplied thereto at the input 6 thereof in a constant division ratio. The output 9 of the frequency divider 7 is connected to the input 10 of a counter 11 which counts the pulses coming from the frequency divider 7. The counter 11 is a four-stage binary counter whose stage outputs are connected to the inputs of a store (register) 12 which is so constructed that on application of a control pulse to the input 12 thereof it takes on the counter state in the counter 11 and stores said counter state until the next pulse at the input 13. The stage outputs of the store 12 are fed to the inputs of a decoder 14 which decodes the counter state contained in the store 12 in such a manner that a control signal is emitted at that one of its outputs D0 to D9 which is associated with the decoded counter state.

The output 3 of the Schmitt trigger 2 is also connected to the input 4 of a monoflop 5 which is brought into its operating state by each pulse at the output 3 of the Schmitt trigger. It returns from this operating state to its quiescent state after expiration of a hold time depending on its intrinsic time constant if it does not receive a new pulse prior to expiration of this hold time. It is held in the operating state by each pulse received during the hold time until it finally flops back into the quiescent state when the interval between two successive pulses is greater than its hold time.

The output 15 of the monoflop circuit 5 is connected to the input 16 of a sequence control device 17 which is set in operation by the signal emitted in the operating state of the monoflop 5. Supplied to the sequence control device by 17 via a Schmitt trigger 18 at a control input 19 are pulses having a recurrence frequency derived from oscillations of the same frequency, for example, twice the mains frequency of 100 c/s, applied to the input 20. The sequence control device 17 is so constructed that in a cyclically recurring sequence in time with the pulses supplied to it at the input 19 it emits pulses at the outputs 21, 22 and 23 whose duration is equal to the period of the oscillation applied to the input 20. The output 21 of the sequence control device 17 is connected to the control input 8 of the frequency divider 7, the output 22 is connected to the control input 13 of the store 12 and the output 23 thereof is connected to the reset input 24 of the counter 11.

The mode of operation of the circuit of FIG. 1 will now be explained with the aid of the diagram of FIG. 2 which shows the variation with time of the signals at the output 3 of the Schmitt trigger 2 and at the inputs 16 and 19 as well as the outputs 21, 22 and 23 of the sequence control device 17.

It will be assumed that a useful frequency oscillation is being received at the input 1. The Schmitt trigger 2 then emits at the output 3 rectangular pulses whose recurrence frequency is equal to the frequency of said useful frequency oscillation. The first pulse emitted by the Schmitt trigger 2 puts the monoflop 5 into its operating state. The hold time of the monoflop 5 is so dimensioned that for all useful frequencies occurring it is longer than the recurrence period of the rectangular pulses emitted at the output 3. The monoflop 5 therefore remains in its operating state for as long as the useful frequency oscillation is applied to the input 1 and supplies to the control input 16 of the sequence control device 17 a control signal throughout this time.

Due to the control signal applied to the input 16 the sequence control device 17 emits at its outputs 21, 22 and 23 in time with the pulses supplied to it via the Schmitt trigger 18 at the input 19 mutually offset control pulse sequences, the duration of the control pulses being equal to the time interval of the leading edges of the pulses supplied at the input 19 and thus equal to the period of the oscillation applied to the input 20 and the pulse sequences being offset with respect to each other by one pulse duration. The control pulses emitted by the sequence control device 17 perform the following functions:

a. The first control pulse appearing at the output 21 sets in operation for its duration via the input 8 the frequency divider 7 so that the latter divides the recurrence frequency of the pulses supplied thereto from the Schmitt trigger 2 and thus the frequency of the useful frequency oscillations received in a constant ratio and passes counting pulses to the input 10 of the counter 11 with a correspondingly reduced recurrence frequency.

b. Via the input 13 the second pulse occurring at the output 22 causes the store 12 to take on and to store the count of the counter 11 reached at the end of the first control pulse.

c. The third control pulse appearing at the output 23 resets the counter 11 via the reset input 24.

COntrol pulse sequences continue to be emitted for as long as the monoflop 5 remains in its operating state.

Since the stage outputs of the store 12 are permanently connected to the inputs of the decoder 14, the store content is continuously being decoded. The decoder 14 therefore emits a control signal at the output which is associated with the count contained in the store.

During each group of three offset control pulses of the three control pulse sequences emitted by the sequence control device 17, the counter 11 receives counting pulses from the frequency divider 8 only for the duration of the control pulse of the first control pulse sequence emitted at the output 21. The duration of this control pulse thus determines the measuring time during which the oscillations of the useful frequency signal received are counted. Since the duration of the control pulses emitted by the sequence control device 17 is however equal to the period of the oscillation applied to the input 20, the measuring time is fixed by the period of said oscillation.

The frequency divider 7 is connected in front of the counter 11 so that a small capacity of the counter 11 is sufficient to obtain a clear indication of the received frequency even when the measuring time is so long that a large number of periods of the useful frequency oscillation is received during the measuring time. This is for example, the case when the oscillation supplied to the input 20 has twice the mains frequency. Since the frequency divider 7 divides the frequency of the useful frequency oscillations received in the constant ratio k, the counter 11 need count only the oscillations having a correspondingly reduced frequency. If the division ratio k of the divider 7 is so set that it is equal to the product of the measuring time t and channel spacing Δ f, only a frequency which differs by at least the channel spacing Δ f from a previously received frequency will change the count of the counter 11.

The purpose of the monoflop 5 is to prevent interference frequencies supplied to the input 1 from producing at one of the outputs D0 to D9 of the decoder 14 a control signal which could lead to an erroneous function of the equipment being controlled. The interference sources usually encountered emit a frequency spectrum whose components lie predominantly in the audio region, i.e., below the ultrasonic region. If the hold time of the monoflop 5 is set to a value slightly greater than the period of the smallest useful frequency but smaller than the period of the highest interference frequency occurring, the monoflop 5 returns to its quiescent state before the end of the period of an interference frequency. Since in this state no signal is supplied to the control input 16 of the sequence control device 17, the latter is put out of operation and consequently the received signal cannot be evaluated because the count of the counter 11 is not transferred to the store 12 and thus no decoding takes place.

To facilitate understanding of the invention, the function of the circuit of FIG. 1 will now be explained numerically by way of example. The channel spacing Δ f will be taken as 1,200 c/s so that for a frequency of 100 c/s of the oscillation applied to the input 20 and thus a measuring time of 10 ms a division ratio of the frequency divider 7 of k = t ^. Δf = 12 results. It will further be assumed that ten different channel frequencies are to be evaluated; the counter 11 is therefore so connected that it has a capacity of 10. With these values, during the measuring time the counter 11 runs through several count cycles. This means that for the received frequency during the measuring time the counter 11 reaches its maximum count several times and then starts counting again from the beginning. The count reached at the end of the measuring time is however still a clear indication of the received useful frequency provided the number of useful frequencies having a channel spacing Δf is at the most equal to the counter capacity Z. The relationship between the useful frequency f received and the count reached at the end of each measuring time t while this useful frequency is being received is expressed by the following equation:

f = (k/t) ^. (n ^. Z + m + 0.5)

wherein

f = useful frequency received in c/s

t = measuring time in seconds

k = division ratio of the frequency divider 7

Z = capacity of the counter 11

n = number of count cycles passed through (integral)

m = count

The term 0.5 in brackets is a correction factor which ensures that a new count is reached whenever the received frequency differs at least by half the channel spacing Δf from the channel center frequency of the neighboring channel. With a channel spacing Δ of 1,200 c/s, a measuring time t of 10 ms, a division ratio k of the frequency divider 7 of 12, a capacity Z of the counter 11 of 10 and an input frequency f of 33 k c/s, the count 7 is for example reached after two complete count cycles. This is because the input frequency of 33 k c/s is first divided by 12 by the frequency divider 7 so that pulses having a recurrence frequency of 2.750 k c/s reach the input 10 of the counter 11. Since the frequency divider 7 emits counting pulses only during the measuring time of 10 ms, during said time only 27.5 pulses reach the input 10 of the counter 11. For this number of pulses the counter thus runs through two complete cycles and finally stops at the count 7. Similarly, for an input frequency of 39 k c/s the counter stops at the count 2 after passing through three complete counter cycles. With the numerical values given up to 10 different frequencies may be received without any ambiguity occurring in the evaluation.

FIG. 3 illustrates a further embodiment of an ultrasonic remote control receiver which differs from the embodiment described above primarily in that to fix the measuring time it is not necessary to supply a reference frequency. In the illustration of FIG. 3 the same reference numerals as in FIG. 1 are used for identical circuit components. The part of the circuit enclosed in the dashed line represents the sequence control device 17' which emits at its outputs 21', 22', 23' control signals which have substantially the same functions as the control signals emitted at the outputs 21, 22 and 23 of the sequence control device 17 of FIG. 1.

The useful frequency signal received is again supplied to the input 1. The input 1 is connected to the input of the Schmitt trigger 2 which again converts the input useful frequency oscillations into a sequence of pulses whose recurrence frequency is equal to the input useful frequency. The output 3 of the Schmitt trigger 2 is connected to the input B1 of a monoflop 25 which is contained in the sequence control device 17' and which is so constructed that it is switched to its operating state by a pulse received at the input B1 but during its hold time cannot be tripped again by any further pulse. The output 3 of the Schmitt trigger 2 is also connected to the input 26 of an AND gate 27 whose other input 28 is connected to that output 21' of the sequence control device 17' which is directly connected to the output Q1 of the monoflop 25. The output Q1 of the monoflop 25 which emits the signal complementary to the signal at the output Q1 is connected to the input B2 of a further monoflop 29 whose output Q2 is connected to the input A1 of the monoflop 25. The input 10 of the counter 11 is connected to the output of the AND gate 27. The stage outputs of the counter 11 are connected to the inputs of a gate circuit 30 which on receipt of a control pulse at its input 31 transfers the count contained in the counter 11 to the decoder 14 connected to its outputs. In the decoder 14 the count is then decoded in the manner already explained in conjunction with FIG. 1 so that a control signal is emitted at the output corresponding to the transferred count.

The output 3 of the Schmitt trigger 2 is further connected to the input 32 of an AND gate 33 which is contained in the sequence control circuit 17' and the other input 34 of which is connected to the output of a NOR gate 35. The output Q1 of the monoflop 25 is directly connected to one input 36 of the NOR gate 35 and is connected to the other input 37 via a delay member 38 and an inverter 39.

The output of the AND gate 33 represents the output 22' of the sequence control circuit 17' which is directly connected to the control input 31 of the gate circuit 30. In addition, the output of the AND gate 33 is directly connected to one input 40 of a NOR gate 41 and to the other input 42 thereof via a delay member 43 and an inverter 44. The output of the NOR gate 41 represents the output 23' of the sequence control circuit 17', to which output the reset input 24 of the counter 11 is connected.

The mode of operation of the circuit of FIG. 3 is explained in FIG. 4. Since the measuring time in the arrangement of FIG. 3 is substantially shorter than in the arrangement of FIG. 1, the time scale in FIG. 4 has been enlarged compared with FIG. 2 in order to clarify the illustration. When useful frequency oscillations are supplied to the input 1 of the receiver, pulses whose recurrence frequency is equal to the useful frequency appear at the output 3 of the Schmitt trigger 2. It will be assumed that the presence of a pulse corresponds to the logical signal value 1 whereas a pulse space represents the logical signal value 0. The leading edge of the first pulse at the output 3 puts the monoflop 25 into its operating state in which it emits the signal value 1 for the duration of its hold time at its output Q1, resulting in the control pulse at the output 21', which passes to the input 28 of the AND gate 27. Since the other input 26 of the AND gate 27 is directly connected to the output 3 of the Schmitt trigger 2, for the duration of each pulse at the output 3 the signal value 1 is also applied to the input 26 of the AND gate 27. Thus, the pulses occurring at the output 3 of the Schmitt trigger 2 are transferred for the duration of the control pulse at the output 21', i.e. during the hold time of the monoflop 25, as count pulses to the counter 11 and counted by the latter. The hold time of the monoflop 25 thus determines the measuring time; the capacity of the counter 11 must be greater than the number of pulses received during the measuring time for the greatest useful frequency. The count of the counter 11 reached at the end of the measuring time is then a clear indication of the received useful frequency.

When the monoflop 25 flops back into the quiescent state at the end of its hold time, it applies the signal value 0 via its output Q1 to the input 28 of the AND gate 27 so that no further count pulses can enter the counter 11. At the same time there appears at the output Q1 of the monoflop 25 the signal value 1 which at the input B2 puts the monoflop 29 into the operating state. In this state the monoflop 29 emits at its output Q2 the signal value 1 which blocks the monoflop 25 via the input A1 for the duration of the hold time of the monoflop 29 in such a manner that it cannot be switched into the operating state by pulses at the input B1. This is necessary to enable the sequence control device 17' to have sufficient time to generate the control pulses appearing at the outputs 22' and 23' for the transfer of the count or resetting of the counter.

With the return of the monoflop 25 to its quiescent state, the signal value 0 passes to the input 26 of the NOR gate 35 directly connected to the output Q1. During the operating state of the monoflop 25 the signal value 0 is applied with a delay determined by the delay member 38 via the inverter 39 to the input 37 of the NOR gate 35, said signal value 0 being replaced by the signal value 1 only after the delay time of the delay member 38 and not simultaneously with the flop back of the monoflop 25. Thus, for the duration of this delay time the signal value 0 is applied to both inputs 36 and 37 of the NOR gate 35 and consequently for this period of time the signal value 1 appears at the output of the NOR gate 35. The circuits 35, 38, 39 thus effect the generation of a short pulse which immediately follows the return of the monoflop 25 and the duration of which is determined by the delay of the delay member 38. This pulse is applied to the input 34 of the AND gate 33 (FIG. 4). The same effect could obviously alternatively be obtained with a monoflop which is tripped by the signal at the output Q1 changing from the value 1 to the value 0.

Now, if during this time a pulse is emitted at the output 3 of the Schmitt trigger 2, i.e., a signal value 1 is at the input 32 of the AND gate 33, said gate supplies to the control input 31 of the gate circuit 30 a control pulse for the duration of the delay of the delay member 38. This control pulse opens the gate circuit so that it allows the count reached at the end of the hold time of the monoflop 25 to pass to the decoder 14. The latter then emits a control signal at the output associated with this count. The signal value 1 present at the output of the AND gate 33 during the delay of the delay member 38 also passes directly to the input 40 of the NOR gate 41, at the other input 42 of which the signal value 0 is applied for the duration of the same pulse but with a delay determined by the delay member 43. Thus, in a manner similar to the circuits 35, 38, 39 the circuits 41, 43, 44 produce a short pulse which immediately follows the end of the output pulse of the AND gate 33 and appears at the output 23' of the sequence control circuit and is applied to the reset input 24 of the counter 11 (FIG. 4). This pulse resets the counter 11.

The hold time of the monoflop 29 is so set that it flops back into its quiescent state again only when the transfer process from the counter to the decoder via the gate circuit and the resetting of the counter has been effected. When the monoflop 29 returns to its quiescent state, it emits at its output Q2 the signal value 0 which brings the monoflop 25 via the input A1 thereof into such a condition that it can again be brought into its operating state by a pulse at the output 3 of the Schmitt trigger 2. In this manner the measuring and evaluating periods can be repeated for as long as useful frequency oscillations are supplied to the input 1.

In the circuit according to FIG. 3, interference frequencies are suppressed by setting a certain hold time of the monoflop 25. It is apparent from the above description of the function that the transfer of the count of the counter 11 to the decoder 14 takes place immediately following the end of the hold time of the monoflop 25, i.e., immediately following the end of the measuring time. However, a control signal initiating the transfer can be applied by the AND gate 33 to the control input 31 of the gate circuit 30 only when simultaneously with the end of the measuring time a pulse, i.e., the signal value 1, is present at the output 3 of the Schmitt trigger 2. Now, if the hold time of the monoflop 25 is made equal to the reciprocal of the channel spacing Δf, this coincidence at the AND gate 33 at the end of the measuring time occurs only when quite definite frequencies are applied to the input 1; these frequencies lie only within frequency bands which in the example described here, in which the output pulses of the Schmitt trigger 2 have a pulse duty factor of 1:2, have the width of half a channel spacing. These frequency bands each contain one of the useful frequencies. Between these frequency bands there are gaps having the width of half the channel frequency and frequencies falling in these gaps do not produce coincidence at the AND gate 33 and consequently cannot be evaluated by transfer of the count of the counter 11 to the decoder 14. Thus, frequency windows are formed over the entire frequency range which can occur at the input 1 and only frequencies lying within these windows are treated by the circuit according to FIG. 3 as useful frequencies. All intermediate frequencies are recognized as interference frequencies and excluded from evaluation.

If the measuring time is made exactly equal to the reciprocal of the channel spacing the frequency bands in which evaluation takes place are such that the rated frequencies of the signals transmitted by the transmitter are disposed at the lower end of the frequency bands. Thus, in this case only frequencies starting from a rated frequency in each case and extending up to the frequency in the center between two channels would be evaluated as useful frequencies. Since the frequency of the signals emitted by the transmitter can however also fluctuate below the rated frequency, it is desirable to place the frequency bands in which evaluation takes place so that the rated frequencies lie substantially in the center of the bands. To achieve this, the hold time of the monoflop 25 and thus the measuring time is lengthened by a quarter of the reciprocal of the maximum rated frequency. Although with this setting only the maximum rated frequency lies exactly in the center of the corresponding frequency band, the other rated frequencies still lie within the corresponding frequency bands and consequently the frequencies of the useful signals can also deviate from the rated frequency downwardly without preventing evaluation. The frequency gaps including the frequencies treated as interference frequencies then lie in each case substantially in the center between two rated frequencies.

To facilitate understanding of the type of interference identification just outlined attention is drawn to FIG. 5; the latter shows at Q1 the output signal of the monoflop 25 determining the measuring time, at 3-F1, 3-F2, 3-F3 the pulse sequences appearing at the output 3 of the Schmitt trigger 2 for three different useful frequencies F1, F2, F3 and at 3-FS the pulse sequence which appears at the output 3 when an interference frequency FS is received which lies between the useful frequencies F2 and F3. It is apparent from this diagram that at the end of the measuring time a pulse is present at the output 3 of the Schmitt trigger only when useful frequencies are being received and that when an interference frequency is applied there is a pulse space at the end of the measuring time. Thus, at the AND gate 33 the presence of a pulse at the end of the measuring time is employed as criterion for the receipt of a useful frequency. It is also apparent from FIG. 5 that with the useful frequency F1 the counter 11 counts 4 pulses, with the useful frequency F2 up to 5 pulses and with the useful frequency F3 6 pulses.

Isolated short interference pulses which could reach the input 1 of the circuit of FIG. 3 between two useful pulses and undesirably increase the count may be made ineffective by inserting a flip-flop circuit 45 between the output 3 of the Schmitt trigger 2 and the rest of the circuit as illustrated in FIG. 6. The mode of operation of this flip-flop circuit 45 will be explained with the aid of FIG. 7, which shows the signals at the output 3 of the Schmitt trigger 2 and at the output 3a of the flip-flop circuit 45 firstly without interference and secondly with interference. The flip-flop circuit 45 is tripped by the leading edge of each output pulse of the Schmitt trigger 2. If a short interference pulse is received, the flip-flop circuit 45 supplies at its output 3a the signal value 0 for example on receipt of the useful pulse preceding the interference pulse, the signal value 1 on receipt of the interference pulse and the signal value 0 on receipt of the next useful pulse. If no interference pulse had occurred, the flip-flop circuit would not have been switched to the signal value 1 at the output until receipt of the next useful pulse. The flip-flop circuit thus effects on receipt of an interference pulse (and in general on receipt of an odd number of interference pulses) between two useful pulses a reversal of the signal values so that at the end of the measuring time coincidence is not reached at the gate 33 although a useful frequency was received. Without the flip-flop circuit 45 the count would be transferred, although because of the interference pulse received it would not correspond to the useful frequency received.

The embodiment of FIG. 3 differs from the embodiment of FIG. 1 also in that instead of the store (register) 12 the gate circuit 30 is used that allow the count to be evaluated to pass briefly only once in a measuring and evaluating time. Thus, at the output of the decoder 14, instead of a uniform signal as in the case of the embodiment of FIG. 1, a series of pulses appears with the spacing of the control signals at the input 31 of the gate circuit 30. The use of a gate circuit instead of a store is suitable in applications where the equipment to be controlled must be actuated with control pulses and not with a uniform signal.

The immunity to interference may be further increased if in accordance with FIG. 8 a further monoflop 46 which cannot be triggered again during its hold time is inserted between the output 3 of the Schmitt trigger 2 (or the output 3a of the flip-flop circuit 45 of FIG. 6) and the remainder of the circuit. This hold time is set to half the period of the highest useful frequency. This modification is effective against a particular type of interferences, i.e., cases where an amplitude break occurs within an oscillation at the input 1 of the Schmitt trigger 2; this break would lead at the output 3 of the Schmitt trigger to the emission of two pulses instead of the single pulse per oscillation emitted in the normal case. These two pulses give the same effect as the receipt of a frequency which is twice as high and consequently without the additional monoflop 46 erroneous evaluations could arise. However, the monoflop 46 prevents the two pulses from becoming separately effective because it always emits pulses having the duration of its hold time; short double pulses which can arise due to amplitude breaks in the received signal thus cannot have any effect. FIG. 9 shows the action of the monoflop 46 when an amplitude break occurs at the input 1 of the Schmitt trigger 2 which produces a double pulse at the output 3 of the Schmitt trigger. As is apparent, the pulses at the output 3b of the monoflop 46 are not affected by this double pulse.

One embodiment of the remote control receiver may also reside in that a sequence control counter fed by the pulses at the output of the Schmitt trigger 18 is used for the sequence control device 17 of FIG. 1; the stage outputs of said counter are connected to a decoder which is so designed that it activates one after the other one of its outputs for each count. Thus, for example, this decoder may have 10 outputs which are activated successively in each counting period of the sequence control counter. Since in accordance with the description of the example of embodiment of FIG. 1 a total of three control signals are required for the evaluation of the frequency received, the output signals at the fourth, fifth and seventh outputs may be used respectively for activating the frequency divider 7, opening the store 12 and resetting the counter 11. Since in this case the evaluation of the received frequency by the control pulses emitted from the output of the decoder of the sequence control device does not begin until the decoder emits a signal at its fourth output, there is an evaluation delay which has the advantage that short interference pulses produce no response in the receiver.

The advantageous formation of frequency band windows are used in the embodiment of FIG. 3 can also be applied in the embodiment of FIG. 1 if instead of the retriggerable monoflop 5 a monoflop is used which has no dead time and which is not retriggerable again during its hold time which as in the monoflop 35 of FIG. 3 is made equal to the reciprocal of the channel spacing Δ f. This monoflop thus always flops back into its quiescent state when there is a pulse pause at its input at the end of its hold time whereas it is returned to its operating state practically without dead time by a pulse applied to its input at the end of the hold time. Since a pulse at the input of the monoflop at the end of its hold time however occurs only for frequencies lying within the frequency bands mentioned in connection with the description of FIG. 3, only frequencies which lie within the frequency bands can be treated as useful frequencies. For all intermediate frequencies, the monoflop returns to its quiescent state in which it interrupts the sequence control device and thus prevents evaluation of said frequencies. For the same reasons as in the circuit of FIG. 3, in this case as well the hold time of the monoflop should be lengthened by a quarter of the reciprocal of the highest useful frequency.

The ultrasonic remote control receiver described above can be used not only to control television sets, radio sets and the like but is particularly suitable also for industrial use in which high immunity to interference is very important. It may, for example, be used for remote control of cranes on large building sites, where there are a great number of different interference sources. The ultrasonic remote control receiver according to the above description is so immune to interference that it operates satisfactorily even under the difficult conditions encountered in the aforementioned use.

The following table provides examples of integrated circuits from Texas Instruments Incorporated which may be used in the foregoing invention.

______________________________________ Schmitt-triggers 2 and 18 SNX 49713 Monoflops 25, 29 and 46 SN 74121 Monoflop 5 SN 74122 Frequency divider 7 SN 7492 Counter 11 SN 7490 Store 12 SN 7475 Control 17 SN 7476 Gate 30 SN 7432 Decoder 14 SN 7442 ______________________________________




Receiver tuning circuit PHILIPS CHASSIS K11

A receiver tuning circuit in which without operation of extra switches a change-over can be made from tuning by means of a continuously varying tuning voltage to tuning by means of one of a number of adjusted tuning voltages by using a capacitor controlled by an automatic tuning correction current source circuit for obtaining said voltage, and an automatic switch for applying the desired tuning voltages to this capacitor.



1. A receiver tuning circuit comprising a tuning section having a tuning input, a capacitor means coupled to said tuning input for applying a tuning voltage thereto, a controllable current source coupled to said capacitor, a tuning correction signal detector means coupled between said tuning section and said current source for applying an automatic tuning correction signal to said capacitor means through said current source, and means for immediately tuning said tuning section to a selected frequency independently of the previous voltage on said capacitor comprising a first switch coupled to said capacitor and an operating device means for controlling said switch for an operating period, said operating device including a memory means for storing the last adjusted state of said operating device, at least one potentiometer and a generator means for effecting that a signal from said potentiometer is applied to said capacitor through said switch upon operation of said operating device, and said first switch including a first time constant circuit means coupled to said generator for maintaining said switch in an on position for a selected period of time independent of said operating period.

2. A receiver tuning circuit as claimed in claim 1, wherein said switch comprises a current source which can be influenced by an operating signal, said source being coupled to two parallel branches the first of which includes a transistor having an emitter coupled to said current source, a base coupled to an input of the switch, and a collector, a current mirror circuit having an input coupled to said collector and an output, the second branch including a pair of series connected diodes coupled to the current source and to said output of the current mirror circuit, and output of the switch being coupled to the pair of diodes.

3. A circuit as claimed in claim 1, further comprising a manually operable second switch means for obtaining a continuous coupling between said potentiometer means and said capacitor.

4. A circuit as claimed in claim 1 further comprising a supply circuit means for obtaining a desired tuning voltage, said memory means being independent of said supply circuit, said first switch further comprising a second time constant circuit means coupled to said supply circuit means for temporarily applying a tuning voltage determined by the potentiometer to said capacitor when the supply voltage is switched on.

5. A receiver tuning circuit comprising a tuning section having a tuning input, a capacitor means coupled to said tuning input for applying a tuning voltage thereto, a controllable current source coupled to said capacitor, a tuning correction signal detector means coupled between said tuning section and said current source for applying an automatic tuning correction signal to said capacitor through said current source, means coupled to said capacitor for immediately tuning said tuning section to a selected frequency independently of the previous voltage on said capacitor, a memory means coupled to said immediate tuning means for storing a tuning voltage corresponding to a selected frequency and signal amplitude detector means coupled to said immediate tuning means for effecting that said tuning voltage stored in said memory means is applied to said capacitor through said immediate tuning means when said signal amplitude goes below a selected value.

Description:

The invention relates to a receiver tuning circuit having a tuning section tunable by means of a tuning voltage obtained from a capacitor whose charge can be changed by means of a current source circuit controllable by at least an automatic tuning correction signal, while furthermore a desired tuning voltage can temporarily be applied to said capacitor with the aid of a switch controllable by an operating device so as to make it possible to immediately tune to a desired frequency independently of the previous charge condition of said capacitor.
A tuning circuit of the kind described above is known from German Offenlegungsschrift No. 2,025,369 in which the said capacitor is optionally connected to a tuning potentiometer by means of a push-button switch for applying a voltage determined by said potentiometer to said capacitor as long as the push-button switch is operated, whereafter a tuning frequency thus selected is corrected with the automatic tuning correction signal through the current source circuit and the charge of said capacitor.
It is an object of the invention to enhance the comfort of operation of such a tuning circuit.
To this end a tuning circuit of the kind described in the preamble is characterized in that the operating device includes a memory for storing the last adjusted state of said operating device, and a signal generator which upon operation of the operating device applies a signal to an output thereof, which output is coupled to a time-constant circuit coupled to said switch for maintaining said switch switched on for a period determined by the time-constant circuit independently of the operating duration of the operating device.
Due to the step according to the invention it is possible at any moment to ascertain, by means of the state of the memory, the last operating action of the operating device, maintaining the advantage of a temporary tuning voltage supply to the capacitor so that subsequently other functions such as, for example, a tuning correction device or a search tuning device can become active on said capacitor through the current source circuit.
The invention will now be described with reference to the drawing.
FIG. 1 shows by way of a block-schematic diagram a receiver tuning circuit according to the invention;
FIG. 2 shows by way of a principle circuit diagram a possible embodiment of part of the receiver tuning circuit according to the invention.
In FIG. 1 a tuning section 1 has an input 3 to which a received RF signal is applied and an output 5 from which an IF signal is obtained. This IF signal is applied to an input 7 of an IF amplifier 9 and derived in an amplified form from an output 11 thereof and applied to an input 13 of a tuning correction signal detector 15 and an input 17 of a signal amplitude detector 19.
Furthermore, the tuning section 1 has an input 21 which receives a tuning voltage from a capacitor 23. The charge of the capacitor 23 can be changed with the aid of a current source circuit 25 for which purpose an output 27 thereof is connected to the capacitor 23 whose other end is connected to ground.
An input 29 of the current source circuit 25 is controlled by a tuning correction signal originating from an output 31 of the tuning correction signal detector 15. This correction signal can be rendered inactive with the aid of a switch-off device 33 incorporated in the connection between the output 31 and the input 29, and with the aid of signals applied to an input 35 or 37 thereof.
For this purpose the input 35 of the switch-off device 33 is connected to an output 39 of a station finder 41 two outputs 43, 45 of which are connected to inputs 47, 49 of the current source circuit 25. Thus, the station finder 41 can continuously bring about a charge or discharge of the capacitor 23 when the automatic tuning correction is switched off so that the tuning section 1 is continuously detuned. When a station is found, a signal is produced at the output 31 of the tuning correction signal detector 15, which signal causes stop signal at an input 55 of the station finder through a polarity correction circuit 51 and a delay circuit 53, and this for a certain period, for example, 1.5 seconds so that station finding is temporarily discontinued and the automatic tuning correction is activated. As a result, tuning is effected immediately and correctly at the frequency of the received station. If this station is not desired, further station finding can be continued after 1.5 seconds.
The capacitor 23 providing the tuning voltage for the tuning section 1 may be controlled not only by the current source circuit 25, but also by an output 57 of a switch 59 an input 61 of which is connected to an output 63 of an operating device 65.
A voltage originating from one of a plurality of tuning potentiometers 67, 69, 71 can be temporarily applied to the capacitor 23 with the aid of the operating device 65. When the device 65 is operated a signal is obtained to that end from a signal generator 73. This signal is applied through an output 75 of the operating device to an input 77 of a time-constant circuit 79. The time constant circuit 79 is coupled to the switch 59 and closes it for a certain time so that the capacitor 23 assumes the desired voltage of a selected potentiometer 67, 69 or 71.
The operating device 65 has a memory which is symbolically shown in the figure as a block 81. This memory 81 ensures that it can always be seen which potentiometers 67, 69 or 71 is interconnected to the output 63 of the operating device 65, while due to the action of the time-constant circuit 79 the voltage originating from this potentiometer is not continuously present at the capacitor 23. The said memory 81 may be either a mechanical or an electrical memory. When using a mechanical memory, the signal generator 73 may be an AFC switch which is present on many operating devices. When using an electrical memory, as is common practice with touch controls in the operating device 65, any change of state of this memory may be converted in a simple manner into a signal applied to the output 75.
The switch 59 has an output 83 which applies a signal to the input 37 of the switch-off device 33. This signal renders the automatic tuning correction inactive as long as the switch 59 is closed, as is the case when a tuning voltage is applied to the capacitor 23 with the aid of the operating device 65. The tuning correction is active again immediately when the switch 59 is open so that tuning is effected immediately and correctly when a selected station is received.
To be able to adjust the potentiometers 67, 69 or 71, easily, a switch 85, which can be operated manually, is connected to a further input 87 of the switch 59 which can be maintained closed with the aid of the manually operated switch 85 as long as is desired for adjustment.
Coupled to the switch 59 is a further time-constant circuit 89 which has an input 91 connected to an output 93 of a supply circuit 95. Thus, whenever the receiver is switched on, the switch 59 is maintained closed for some time so that firstly the station to which the operating device 65 is adjusted is tuned to, even if the station finder 41 were switched on. In that case the operating device 65 must have, for example, a mechanical memory 81, which is independent of the supply voltage, in order to maintain its adjustment also when the supply voltage is switched off.
Furthermore, the switch 59 has an input 97 which is connected to an output 99 of the signal amplitude detector 19. When the signal received by the receiver becomes too weak, the switch 59 can be closed via this path so that tuning to a frequency selected by the operating device 65 is maintained and is stil present when the received signal becomes stronger again. A further possibility, which may be particularly attractive for motorcar radios, is to incorporate a switch which can be operated in this manner between the capacitor and an output of a memory which can be coupled to that capacitor. When the field strength is sufficient, this memory may be written in with the voltage on the capacitor and when the field strength is insufficient, an output of this memory may be coupled to the capacitor for transferring the memory voltage to the capacitor. This memory may be, for example, a motor adjusting a potentiometer and operated with the aid of a control system. When the supply voltage drops out, the last adjusted state of the potentiometer is maintained.
The described tuning circuit may immediately change over from, for example, a search tuning state to a state tuned to a desired station without operating extra switches and only by operating the relevant operating members.
It will be evident that the switch tuning may be omitted, if desired.
FIG. 2 shows a possible embodiment of the switch 59 and, coupled thereto, the time-constant circuits 79 and 89 of the receiver tuning circuit of FIG. 1. The inputs and outputs have the same reference numerals as the corresponding inputs and outputs in FIG. 1.
The input 61 of the switch 59 is connected to the base of a npn transistor 201. The emitter of this transistor 201 is connected through a diode 203 to the collector of an npn transistor 205 arranged as a current source whose emitter is connected to the output 83 and is furthermore connected to ground through a resistor 207.
The collector of the transistor 201 is connected through a diode 209 to the input 91 to which the supply voltage is applied. The diode 209 shunts the base-emitter path of a pnp transistor 211 which together with the diode 209 constitutes a current mirror circuit. The collector of the transistor 211 allows a current to flow through a series arrangement of two diodes 215, 217, which current has substantially the same intensity as the current flowing through the diode 203. Furthermore, the diode 217 is connected to the collector of the transistor 205, while the junction of the collector of the transistor 205 and the diode 215 is connected to the output 57.
The base of the transistor 205 is connected to a tap on a potential divider 219, 221 between the supply voltage and ground. This potential divider will raise the voltage at the base of the transistor 205 to such an extent that it produces a current, which is further determined by the emitter resistor 207, equally distributed over the collector branches with the diode 203 and the transistor 201 and with the diodes 217 and 215, respectively. When the circuit is designed in a integrated form, it can be achieved in a simple manner that the output 57 will always assume the same voltage as the input 61. Since the output 57 is connected to the capacitor 23, both a discharge and a charge of this capacitor 23 is possible. Charging is effected through the transistor 211 and discharging is effected through the diodes 215, 217. The circuit is independent of temperature influences. The diode 203 and consequently the diode 217 are provided to prevent a too large voltage difference at the base-emitter junction of the transistor 201.
The current source 205 can be turned off by connecting the base of transistor 205 to ground with the aid of a npn transistor 223 connected across the resistor 221. This is effected when the base of this transistor receives a voltage from a potential divider comprising three resistors 225, 227, 229. However, when the base of the transistor 223 receives a low voltage through the input 87 or the input 97, the transistors 223 is cut off and the transistor 205 conducts so that the switch 59 is closed.
The voltage at the base of the transistor 223 remains low for some time after switching on the supply voltage because a capacitor 231, which is connected to the junction between the resistors 225 and 227, must firstly be charged. Thus, the switch 59 is closed during that period.
Furthermore, the voltage at the base of the transistor 223 may be decreased by discharging the capacitor 231 through a resistor 233 to the input 77 when this input is earthed for a moment during operating device 65. The voltage at the capacitor 231 will subsequently increase in accordance with a certain time constant and after a certain time the transistor 223 conducts again and the switch 59, which was closed when the transistor 223 was cut off, will be open again.
The input 97 is interconnected to the input 87 so that the transistor 223 is also cut off and the switch 59 starts to conduct when the voltage at the input 97 becomes low upon a drop-out of a transmitter signal.
The switch 59 in this embodiment also acts as an amplifier so that the adjustments of the tuning potentiometer 67, 69 or 71 do not have any influence on the rate at which the charge of the capacitor 23 is changed.
PHILIPS 26C568 CHASSIS K11 UNITS VIEW.

CHASSIS PHILIPS K11 VIEWING OF ALL UNITS.

- DEMOD 4822 212 20062 TDA2661

- CROM LUM 4822 212 20079 TBA560B

- REF COMB 4822 212 20081

- LINE CONTR 4822 212 20084 TBA240B TBA720A

- IF SOUND 4822 212 20252 TBA750A

- IF AMPL 4822 212 20083

- FRAME CONTR 4822 212 20085 TBA730 TBA760





PHILIPS 26C568 CHASSIS K11 Circuit arrangement for generating a control signal for the field output stage in a television receiver PHILIPS CHASSIS K11 (20AX) FRAME CONTROL / LINE CONTROL RELATION THEORY

A circuit arrangement for generating a control signal for the field output stage in a television receiver provided with a frequency divider circuit with which the double line frequency is divided either by a number which is equal to the number of lines per image or by a number which deviates from this number. The choice between the two divisors is made by a storage element dependent on the on-phase state or the off-phase state of the received field synchronizing pulses and the divider pulses. Resetting of the frequency divider circuit is therefore realised by the store and not by the received pulses, which results in less sensitivity to interference. Since there are two states, a small brightly lit image no longer occurs. In the case of the nominal divisor, the divider pulse is extended. When receiving non-standard signals (for example, from video recorders) direct synchronisation is used with the divisor being larger than the nominal divisor.

1. A circuit arrangement for generating a control signal for the field output stage in a television receiver, said circuit comprising means for the reception of line and field synchronising pulses in which a number of fields constitutes an image raster, a generator means coupled to said reception means for generating a signal of the line frequency or an integer multiple thereof, a frequency divider circuit coupled to said generator means and having a first state wherein the divisor equals the number of lines per image and a second state wherein the divisor deviates from said number, a comparator stage means for continuously comparing said received field synchronization pulses with an output signal from said frequency divider having a first input means coupled to said reception means for receiving said field synchronization pulses, a second input coupled to said divider, and an output means for supplying a signal which is dependent on the phase difference between the compared pulses, a memory element means coupled to said output means for bringing and maintaining the frequency divider circuit in its first state when the compared pulses at least partly coincide and brings it to the second state when the compared pulses have not coincided for a given period, and a gate coupled between said memory and divider. 2. A circuit arrangement as claimed in claim 1, further comprising an adjusting gate means coupled to said divider and said memory element for adjusting the frequency divider circuit every time at the commencement of each raster, said adjusting gate having an input means for adjusting the frequency divider circuit by means of the memory element. 3. A circuit arrangement as claimed in claim 1, wherein the memory element comprises a bistable element means for receiving reset pulses when the compared pulses coincide at least partly and and for receiving set pulses when the compared pulses have not coincided for a given period. 4. A circuit arrangement as claimed in claim 3, wherein the divisor in the divider in the second state is larger than that in the first state, and further comprising means coupled to the bistable element in the memory element for preventing reset pulses from being received by said bistable element while the frequency divider circuit is being adjusted by means of received field synchronizing pulses. 5. A circuit arrangement as claimed in claim 1, further comprising a pulse shaper having an input coupled to said memory and an output means for applying a pulsatory output signal to the comparator stage, the memory element bringing the pulse shaper to a first state when it brings the frequency divider circuit to its first state, the duration of the output pulse from the pulse shaper in its first state being longer than in a second state to which the pulse shaper is brought by the memory element when it brings the frequency divider circuit to its second state. 6. A circuit arrangement as claimed in claim 5, wherein the frequency divider circuit comprises a plurality of serially coupled bistable elements and the pulse shaper includes a keyed gate having a first state in which it receives the output signal from a divider bistable element and a second state in which it receives the output signal from another divider bistable element, the period of the former output signal being longer than the period of the latter, the output signal from the keyed gate being the first half period of the relevant output signal after the instant of resetting the frequency divider circuit. 7. A circuit arrangement as claimed in claim 1 wherein the frequency divider circuit comprises a plurality of serially coupled bistable elements, and further comprising an adjusting gate having a plurality of inputs, the outputs of a plurality of the divider bistable elements being coupled to inputs of said adjusting gate for adjusting the frequency divider circuit at the commencement of each raster, and a controllable switch coupled between one of said plurality of inputs and one of said plurality of outputs. 8. A circuit arrangement as claimed in claim 7, wherein the difference between the two divisors is a combination of powers of 2. 9. A circuit arrangement as claimed in claim 7 wherein said memory element is coupled to said controllable switch. 10. A method comprising generating a television line frequency signal or an integral multiple thereof from a received line synchronization signal, said method comprising obtaining a field synchronization signal by frequency dividing said generated signal by a first divisor, continuously detecting lack of synchronization between said obtained field frequency signal and a received field frequency signal, changing said divisor in said dividing step to a second divisor upon detecting said lack of synchronization, and changing said divisor back to said first divisor upon detecting synchronization between said signals.

Description:

The invention relates to a circuit arrangement for generating a control signal for the field output stage in a television receiver suitable for the reception of line and field synchronising pulses in which a number of fields constitutes a picture, provided with a generator for generating a signal of the double line frequency, a frequency divider circuit and means for applying received field synchronising pulses to a comparator stage for comparing the phase between these pulses and the pulses generated by the frequency divider circuit, the comparator stage being capable of applying a signal to a gate which signal is dependent on the phase difference between the compared pulses.

A circuit arrangement of this kind is described in U.S. Pat. No. 3,708,621. Since in this known circuit arrangement the control signal is derived by frequency division from the line synchronising signal, its frequency is correct as soon as the line synchronising circuit has pulled in in frequency which is usually effected at a comparatively fast rate. The comparator stage which may be formed as a coincidence stage and an integrator ensure the correct phase of the received field control signal relative to the field synchronising pulses originating from the transmitter and received by the television receiver. In the off-phase condition the comparator stage provides a pulse during the occurrence of a pulse originating from the frequency divider circuit. When the integrator, which may be a counter, has received a given number of these pulses, it in turn applies a signal which enables the gate. The frequency divider circuit, which consists of a plurality of bistable elements, is then reset in that one of the received field synchronising pulses is passed by the gate. The phase is then correct, the comparator stage no longer supplies any pulse and the received synchronising pulses can in principle no longer reach the divider circuit, at least not as long as the signal generated by the circuit arrangement maintains the same frequency and the same phase as the received pulses.

The known circuit arrangement has the following drawbacks. Firstly, at the instant when the frequency divider circuit is reset, the vertical deflection discontinues and subsequently commences again which means that one vertically directed deflection lasts shorter than the others. When this deflection is very short or when, in contrast, is almost as long as a normal deflection, i.e. 20 ms in a television system using 50 fields per second, this is not a very great drawback. When, however, the shorter deflection lasts, for example, 10 ms, the mean level of the sawtooth current flowing through the field deflection coil is shifted considerably which may result in the transistors of the field output stage providing the said current being cut off for a given period. A brightly lit horizontal line then appears on the display screen of the receiver which is disturbing for the observer and which may be harmful for the screen.

Secondly, in the case when the received signal is weak, the coincidence stage not only receives the useful field synchronising pulses originating from the transmitter but also noise and interferences. It may then occur that the coincidence stage receives too little information during the occurrence of the synchronising pulses which may be observed as the off-phase condition. The gate may therefore be enabled at any arbitrary instant so that interferences can directly influence the frequency divider circuit and cause a wrong phase. Also the divider circuit may at any instant be reset so that the vertical deflection can commence and end at any instant. The height of the image displayed then continuously varies and may be very small if interferences succeed each other at a fast rate. This has the same disturbing effect as that described above.

An object of the invention is to obviate the said drawbacks and, to this end, the circuit arrangement according to the invention is characterized in that the frequency divider circuit, dependent on the output signal from the gate is switchable under the control of a memory element between two states having different divisors, to wit a first state whereby the divisor is equal to the number of lines per image and a second state whereby the divisor deviates from said number, whereby the memory element brings and maintains the frequency divider circuit in its first state when the compared pulses at least partly coincide and brings it to the second state when the compared pulses have not coincided for a given period.

The invention is based on the recognition of the fact that the said drawbacks of the known circuit arrangement are caused in that the received field synchronising pulses can be directly applied to the frequency divider circuit. According to the invention, no received signal and hence no interference can directly reach the divider circuit. This may be compared with the known circuit arrangements in which a field oscillator to be synchronised is used instead of a divider circuit (or a counter). In these circuit arrangements, the oscillator initially receives the received field synchronising pulses (direct synchronisation) until the frequency and the phase of the generated signal are correct. Subsequently, the direct path is completely or partly cut off while the frequency and the phase are always recontrolled (indirect synchronisation) unless the off-phase condition occurs again for some reason or other. In the known circuit arrangement referred to in the U.S. patent mentioned hereinbefore a direct synchronisation is used only once for resetting for the case where the off-phase condition (= non-coincidence) has taken longer than a given period whereafter the frequency divider circuit operating as a field generator does not receive anything anymore unless, as has been explained, the received signal is weak. In the circuit arrangement according to the invention which is also provided with a frequency divider circuit, this divider circuit is not adjusted by an external signal. This again is an indirect synchronisation, but one which is only active in the off-phase condition and is thereafter no longer active. The circuit arrangement according to the invention therefore has the advantage of the circuit arrangements employing indirect synchronisation, i.e. the greater insensitivity to interference, as well as the advantage of generating the field frequency by means of frequency division, i.e. obtaining substantially immediately the exact frequency of the control signal applied to the field output stage.

Moreover, since the frequency divider circuit in the circuit arrangement according to the invention has two states with different dividers which are dependent on the output voltage of the gate, that is to say, on the fact whether the received and generated signals of field frequency are either in phase or not in phase (= coincidence), the above described interfering phenomena cannot occur. In fact, it is obvious that the second divider will in practice be chosen sufficiently closely to the first one in order that switching over from one to the other divider cannot result in an image having a small height and a strong brightness.

A further aspect of the invention is that the relevant circuit arrangement is alternatively suitable for the reception of "non-standard signals", i.e. signals in which the number of lines per image deviates from the number prescribed for the relevant television system. Such signals are generated by some test signal generators or may be produced when using video recorders. An embodiment of the circuit arrangement according to the invention makes synchronisation in such a case possible as well, whereby the direct synchronisation is used. To this end, the circuit arrangement according to the invention is characterized in that the memory element comprises a bistable element receiving reset pulses when the compared pulses at least partly coincide and receiving set pulses when the compared pulses have not coincided for a given period, while the divider in its second state is larger than that in its first state, means being provided by which the bistable element in the memory element cannot receive reset pulses, while the frequency divider circuit is adjustable by means of received field synchronising pulses.

The invention will be described in greater detail by way of example with reference to the accompanying Figures, in which,

FIG. 1 shows a block schematic diagram of a television receiver provided with a circuit arrangement according to the invention,

FIG. 2 shows part of the circuit arrangement according to the invention,

FIGS. 3, 4, 5, 6 and 7 show waveforms which occur in the circuit arrangement according to the invention,

FIG. 8 shows a further part of the circuit arrangement according to the invention and

FIG. 9 shows part of a second embodiment of the circuit arrangement according to the invention.

In FIG. 1, 1 denotes an aerial with which a television signal can be received. This signal is applied to an RF and detection section 2. The detected signal subsequently reaches at one end the sound section 3 of the television receiver and at the other end a video amplifier 4 at whose output a complete video signal, possibly with a chrominance signal in the case of colour television, is available. This signal is applied to a section 5 in which it is processed whereafter a picture display tube 6 is driven as well as a synchronising separator 7. The output voltage thereof comprises line synchronising pulses which are applied to a phase detector 8 whose output voltage can influence an oscillator 11 through a fly-wheel filter 9 and a reactance circuit 10. Oscillator 11 generates a voltage of double the line frequency 2f _H , i.e. 31250 Hz in case of reception of a signal in accordance with a television system employing 625 lines per complete image, 2 interlaced rasters per image and 50 fields per second. Another possibility is that oscillator 11 generates a voltage of the line frequency f _H whose frequency is subsequently doubled. The voltage of the frequency 2f _H controls a frequency divider circuit 12 in which its frequency is divided by two and the signal thus obtained is applied through a pulse shaper 13 to the line output stage 14 which provides the line deflection current for the deflection coil (not shown) for the horizontal deflection of the electron beam(s) in tube 6.

The voltage available at the output of oscillator 11 also controls a frequency divider circuit 15 in which its frequency is divided by the divisor 625 in a first state and by another divisor in the second state. When oscillator 11 has the correct frequency, i.e. after frequency pull-in of the circuit 8, 9, 10, 11 for the indirect line synchronisation, the frequency of the signal generated by the divider circuit 15 is correct, that is to say, it is equal to the field frequency, in case of the given standard of 50 Hz, if the divider circuit is in its first state. An adjusting gate 16 ensures that the divider has the correct value. A pulse shaper 17 receives the signal generated by the divider circuit 15 and controls the field output stage 18 which applies the field deflection current to the deflection coil (not shown) for the vertical deflection of the electron beam(s) in tube 6. The two pulse shapers 13 and 17 ensure that the line and the field control signal acquire the waveform for stages 14 and 18, respectively.

The output voltage from synchronising separator 7 also comprises field synchronising pulses which are separately obtained by means of a field synchronising separator 19 whereafter they are applied to an input of a coincidence stage 20. The divider pulses originating from the output of pulse shaper 17 are present at a second input of said stage. In the on-phase state, that is to say, in the case where a field synchronising pulse originating from the separator 19 and a divider pulse at least partly coincide, stage 20 does not provide a signal. In the offphase state, it provides a signal, namely the divider pulse to an integrator 21 which is followed by a level detector 22. When this state lasts at least approximately 0.4 s, which corresponds to approximately 20 pulses, the detected level exceeds a given threshold level so that a gate 23 formed as control switch starts to conduct. The field synchronising pulses at the output of separator 19 are also applied to a trigger 24 which, for example, by means of differentiation, generates pulses whose leading edges coincide with those of the synchronising pulses. When switch 23 conducts, some of these pulses are passed and they reach according to the invention a memory element 25. Memory element 25 influences adjusting gate 16 through a gate 26 in a manner which will be described hereinafter with the result that the on-phase state occurs. Coincidence stage 20 then does not provide a pulse so that switch 23 is cut off. Element 25 influences also pulse shaper 17 as will be described hereinafter.

FIG. 2 shows in greater detail the parts 15, 16, 25 and 26 of the circuit arrangement according to the invention. In this embodiment, frequency divider circuit 15 consists of ten bistable elements, in this case flipflops 15 ₁ , 15 ₂ , . . . 15 ₁₀ which are formed in known manner and each of which divides by two. In order that divider circuit 15 is reset after the 625th period of the signal with the frequency 2f _H to the initial position, i.e. the position at the commencement of the first period, the outputs of oscillator 11 and of flipflops 15 ₅ , 15 ₆ , 15 ₇ and 15 ₁₀ are connected to five inputs of the adjusting gate 16 formed as a NAND gate, while gate 26 is connected to a sixth input 27 thereof.

FIG. 3 shows the operation of divider circuit 15 and adjusting gate 16 in which input 27 is initially left out of consideration, FIG. 3 shows the output signal S11 from oscillator 11 as well as the output signals Q15 ₁ , Q15 ₂ , . . . Q15 ₁₀ from flipflops 15 ₁ , 15 ₂ , . . . 15 ₁₀ and the signal S28 at the reset line 28 of the flipflops, which line connects the reset terminals (S ₂ ) of all flipflops of divider circuit 15 and which is connected to the output of gate 16. The numbers T ₁ , T ₂ , . . . T ₆₂₃ , T ₆₂₄ , T ₆₂₅ indicate the periods of signal S11, while H ₁ , H ₂ , . . . H ₃₁₁ , H ₃₁₂ indicate the corresponding line periods. For the raster commencing after period T ₆₂₅ the notations T' ₁ , T' ₂ , . . . and H' ₁ apply. Each flipflop reverses when a leading edge occurs in the output signal from the previous flipflop or from signal S11.

At the commencement of period T ₁ , signals S11, Q15 ₁ , Q15 ₂ , . . . Q15 ₁₀ are "high" which may be indicated by the binary number 0. At the commencement of period T ₂ , flipflop 15 ₁ reverses, signal Q15 ₁ becomes "low" which corresponds to the number 1. FIG. 3 shows that for the first six periods the flipflops indicate the following:

T ₁ : 0000000000

t ₂ : 0000000001

t ₃ : 0000000010

t ₄ : 0000000011

t ₅ : 0000000100

and

T ₆ : 0000000101

in which signals Q15 ₁ Q15 ₂ , . . . Q15 ₁₀ are written from the right to the left. These are the numbers 0, 1, 2, 3, 4 and 5 in the decimal system, that is to say, the number of the corresponding period reduced by 1. For the periods T ₆₂₂ , T ₆₂₃ , T ₆₂₄ and T ₆₂₅ the flipflops indicate the following:

T ₆₂₂ : 1001101101

t ₆₂₃ : 1001101110

t ₆₂₄ : 1001101111

and

T ₆₂₅ : 1001110000.

these are the numbers 621, 622, 623 and 624.

It is evident that gate 16 has received at least one 0 at one of its input up to the first half of period T ₆₂₅ . FIG. 3 shows that during the second half of period T ₆₂₅ signals S11, Q15 ₅ Q15 ₆ , Q15 ₇ and Q15 ₁₀ are simultaneously equal to 1 so that signal S28 becomes equal to 0. The pulse thus generated is applied as a reset pulse through line 28 to all flipflops of divider circuit 15. The flipflops which were not in the high state, i.e. flipflops 15 ₅ , 15 ₆ , 15 ₇ and 15 ₁₀ are brought to that state while the other flipflops do not change their state. At the commencement of the next period, period T' ₁ , all flipflops indicate 0 and a new raster commences.

Memory element 25 includes a flipflop 29, an output 30 of which is connected to an input of gate 26 while another input of gate 26 is connected to the output of flipflop 15 ₄ . The outputs of an OR gate 32 and of an OR gate 33 are connected to the set (S ₁ ) and the reset terminal (S ₂ ), respectively, of flipflop 29. The output signal Q from flipflop 29 is present at output 30 and the other output signal Q is present at the other output 31 thereof. The output signals from switch 23 and pulse shaper 17 are applied to gate 33 while the output signals from switch 23 and an inverter stage 34 are applied to gate 32, while stage 34 reverses the output signal from pulse shaper 17 in its polarity.

In the off-phase state, received pulses are passed by switch 23 (FIG. 4a) which do not coincide with the divider pulses (FIG. 4b) originating from pulse shaper 17. Consequently, gate 33 does not provide a signal (FIG. 4d). However, gate 32 receives the pulses of FIG. 4b and the pulses of FIG. 4c which are inverted relative thereto so that this gate is enabled. The first of the output pulses from gate 32 (FIG. 4e) sets flipflop 29 whose output signals consequently become Q = 1 and Q = 0. If flipflop 29 were already in this state, nothing would be changed.

When the on-phase state has been achieved after some time, one pulse from switch 23 (FIG. 5a) and one divider pulse (FIG. 5b) coincide for at least a part. Gate 33 is enabled (FIG. 5d) while gate 32 which receives the pulse of FIG. 5a and the pulse of FIG. 5c which is inverted relative to that of FIG. 5b does not provide a signal (FIG. 5e). Flipflop 29 is reset by the pulse of FIG. 5d, that is to say, the signals Q = 0 and Q = 1 are present at terminals 30 and 31, respectively. Gate 26 is a controlled switch and does not conduct under these circumstances. The operation of divider circuit 15 therefore remains the same as has been extensively described hereinbefore. If due to the inertia of integrator 21 other pulses might be passed by switch 23, the state will not change.

In the off-phase state for which Q = 1, gate 26 conducts so that the output signal Q15 ₄ from flipflop 15 ₄ is present at the input 27. FIG. 3 shows that Q15 ₄ = 0 during period T ₆₂₅ . Signal S28 consequently remains equal to 1 and divider circuit 15 is not reset. FIG. 6 shows the further variation. This Figure shows that only during the second half of period T ₆₃₃ , i.e. 8 periods later than in the on-phase state, the signals S11, Q15 ₄ , Q15 ₅ , Q15 ₆ , Q15 ₇ and Q15 ₁₀ applied to gate 16 are all equal to 1 so that a reset pulse S28 = 0 is generated. The next period is thus the first period T' _i of a new raster.

The foregoing shows that memory element 25 ensures that, dependent on the fact whether the on-phase or off-phase state has occurred, the frequency divider circuit divides the frequency 2f _H from signal S11 by the divisors 625 and 633 respectively. The following cases may occur:

1. on-phase state, with Q = 0 : the previous divisor was 625, Q remains 0, the on-phase state is maintained and neither the memory element nor the divider circuit are influenced;

2. on-phase state with Q = 1 : the previous divisor was 633, Q becomes 0 and the divisor becomes 625, which is the case as under 1;

3. off-phase state (longer than approximately 0.4 s) with Q = 0 : the previous divisor was 625, Q becomes 1 so that the divisor becomes 633; the pulses in FIGS. 5a and 5b have different repetition frequencies and are displaced relative to each other; after a given pull-in period, the on-phase state is reached which is the case as under 2;

4. off-phase state (longer than approximately 0.4 s) with Q = 1 : the previous divisor was 633 and remains so because Q remains equal to 1 which is the case as under 3.

The case under item 1 is thus always the final state in which neither any received signal nor any noise or interferences can reach the divider circuit. If the received field synchronising pulses drop out after this state has been reached the divider circuit continues to divide by 625 due to the action of the memory element so that the image displaced on the display screen of tube 6 remains in place. This is also the case with the known circuit arrangement which in fact has no other divisor than 625 but which, as already stated, is more sensitive to noise and interference with the attendant drawbacks. When the received signal is so weak that substantially no distinction can be made between interferences and field synchronising pulses or when these pulses drop out before the on-phase state is reached, the divisor 633 is used for division. The image "rolls over" in the vertical direction which is less disturbing to the observer and is less harmful for the screen than the brightly lit narrow images which can be displayed with the known circuit arrangement.

FIG. 7 shows how the pull-in process is effected, i.e. when Q = 1. FIG. 7a shows the output pulses from pulse shaper 17 and FIG. 7b shows the pulses originating from trigger 24. Since the frequency of the latter pulses (=2f _H /625) is higher than that of the former pulses (=2f _H /633),

they are displaced relative thereto to the left until coincidence takes place in gates 32 and 33 : Q becomes 0; coincidence also takes place in the coincidence stage 20 and switch 23 is open. For each period of the frequency divider circuit the relative time difference Δ t between two received pulses 1 and 2 is equal to 8 times one period of signal S11, i.e. approximately 8 × 32 μs = 256 μs. One period of the frrquency divider circuit corresponds to 633 × 32/8 × 32 = 79 times Δ t. In the most unfavourable case in which the process commences with the pulse in the position 3 in FIG. 7b, this process will therefore take approximately 75 field periods, i.e. 1.5 s. In this extreme case, which is very improbable, it takes consequently approximately 0.4 + 1.5 = 1.9 s before the displayed image comes to a standstill.

It may occur that just before the end of the pull-in process the pulse of FIG. 7b occupies such a position, indicated by 4, that at the next position 5 after a period of time Δ t the leading edge of the pulse occurs just before the trailing edge of the pulse of FIG. 7a. As a result, the divider circuit is brought to the state with the divisor 625. Since the coincidence period is then very short, it is however evident that this situation is very sensitive to interference. This drawback is obviated according to an aspect of the invention in that the duration of the pulse in FIG. 7a is made longer at the instant when the 625-state occurs. Information therefor may be obtained from memory element 25 which thus fulfils a second function, which information is applied to pulse shaper 17.

FIG. 8 shows pulse shaper 17 in greater detail. The outputs of flipflop 15 ₄ and flipflop 15 ₅ are connected to the inputs of controlled switches 35 and 36, respectively. The switches 35 and 36 are controlled by the signals at the outputs 30 and 31, respectively, of memory element 25. The outputs of switches 35 and 36 are connected to two inputs of an adder stage 37. The output signal thereof is applied to a keyed gate 38 which is keyed by the output signal Q15 ₁₀ from flipflop 15 ₁₀ .

A pulse shaper is required in any case. The output signal Q15 ₁₀ from the frequency divider circuit has a natural frequency of 2f _H /2 ¹⁰ which corresponds to a natural period of approximately 33 ms. Circuit 15 is reset approximately 20 ms after the commencement of the period, i.e. 20 - 33/2 ≉ 3.5 ms after reversal in the middle of the natural period. Signal Q15 ₁₀ therefore has a flyback period of approcimately 3.5 ms and is thus not usable as a field control signal. According to the invention, the pulse shaper has also two states. In the state with divisor 633 of the divider circuit for which Q = 1 and Q = 0, switch 35 conducts while switch 36 is cut off. The output signal Q15 ₄ from flipflop 15 ₄ whose natural period is equal to 2 ⁴ /2f _h ≉ 512 μs is applied to gate 38. This gate is keyed in such a manner that only the first positive half period of signal Q15 ₄ is passed, that is to say, its output signal lasts until the first trailing edge. This may be obtained in known manner with the aid of bistable elements. This is the pulse in FIG. 7a, it lasts approximately 256 μs from the instant of resetting divider circuit 15. Since Δt ≉ 256 μs, this is exactly the duration which is at least required. As soon as the state with divisor 625 is achieved, there applies that Q = 0 and Q = 1. Switch 36 then conducts while switch 35 is cut off so that the output signal from gate 38 is the first positive half period of signal Q15 ₅ . This is the pulse in FIG. 7c and it lasts approximately 512 μs from the instant of resetting divider circuit 15. As a result, it is insured that the pulse in FIG. 7b in any case completely coincides therewith while the output signal from gate 38, hence from pulse shaper 17, is always suitable for controlling the field output stage 18. The flyback period thereof is in fact shorter than approximately 1 ms.

FIG. 7b shows that the largest possible time difference between the leading edges of the pulses is slightly shorter than approximately 256 μs, i.e. 0.256/20 ≉ 1.3 percent of a field period. This slight deviation is maintained as long as the occurred on-phase state lasts and produces a deviation in the vertical position of the image. It may be noted that this value, as well as the maximum duration of the pull-in process, emanates from the difference between the two divisors 633 and 625, i.e. 8 = 2 ³ . A value different from 633 for the divider in the off-phase state may, however, be chosen. Instead of applying signal Q15 ₄ to adjusting gate 16, the supply of signal Q15 ₅ to this gate may be interrupted so that the divisor becomes 625 - 2 ⁴ = 609. In this case, the time difference Δt in FIG. 7b lasts approximately 16 × 32 μs = 512 μs, so that the pull-in time has been reduced by 50 percent relative to the described case with a divisor of 633 while the minimum duration of the pulse in FIG. 7a must also be 512 μs. The extended pulse of FIG. 7c then lasts at least approximately 700 μs which is still just suitable as a field control signal. The largest possible vertical deviation is then, however, doubled.

Divisors other than 633 and 609 may be obtained by applying or not applying one or more output signals from the flipflops of divider circuit 15 to adjusting gate 16. For 633 the difference from the nominal divisor 625 is equal to +2 ³ and for 609 it is -2 ⁴ . The divisor 613, for example, corresponds to -2 ⁴ +2 ² = -12 and may be realised by connecting for Q = 1 the output of flipflop 15 ₃ to an input of gate 16 and by interrupting the connection between the output of flipflop 15 ₅ and the relevant input of gate 16. Divisors other than 625 are obtained in that the connection between at least one of the flip-flops 15 ₅ , 15 ₆ , 15 ₇ , 15 ₁₀ to gate 16 is interrupted and/or in that at least one of the other flipflops is connected to gate 16. The considerations regarding divisor 609 have, however, shown that the choice of the second divisor cannot be limited in practice, while it is evident that in practice the second divisor should not deviate too much from the nominal divisor 625. Moreover, divisors 633 and 609 can be obtained in the easiest manner.

It is to be noted that it may occur that coincidence takes place in stage 20 but not in memory element 25 due to the shorter duration of the pulse generated by trigger 24. However, in such a case the divisor deviates from the value 625 so that the process described with reference to FIG. 7 is effected.

FIG. 9 shows another embodiment of the circuit arrangement according to the invention in which a higher value than the nominal value 625, namely 633, is chosen for the divisor in the off-phase state. This embodiment is based on the following recognition. There are some test signal generators in which the number of lines per image is not 625 but, for example, 624, so that the displayed image is not interlaced. They are used, for example, for adjusting the convergence in colour television receivers. When using video recorders, it may occur that, for example, when displaying a stationary image, the number of line synchronising pulses per image slightly deviates from 625. Field synchronisation with the known circuit arrangement is impossible when such "non-standard signals" are received. The object of the embodiment of FIG. 9 is to establish synchronisation in such a case and for this purpose the direct synchronisation is used while the frequency divider circuit is set to the state with divisor 633. Since this state corresponds to a lower frequency than the nominal one, the direct synchronisation is possible while the drawbacks thereof do not in most cases apply because the received signal often comprises little noise and interferences.

The embodiment of FIG. 9 comprises parts which also occur in those of the previous Figures and which have the same reference numerals. In FIG. 9, 39 denotes a switch which may be, for example, manually operated and which is closed when the above-mentioned non-standard signals are received. As a result, the delay introduced by integrator 21 is reduced, for example, by making a time constant associated with this integrator shorter or by switching off integrator 21 completely. Coincidence stage 20 thus does not have any effect any longer. Also, the closure of switch 39 renders the input voltage of an amplifier 40 "high" so that its output voltage likewise becomes high (= 0). Under these circumstances, a controlled switch 41 which is connected to the output of trigger 24 starts to conduct so that the pulses originating from the trigger are passed and reach an input of an AND gate 42. Another input thereof is connected to the output of gate 16 and the output thereof is connected to the reset line 28 of the ten flipflops of divider circuit 15. The output voltage from amplifier 40 is also applied to an inverter stage 43 whose output voltage is low (= 1) when switch 39 is closed and is applied to an input of an OR gate 44. Another input of gate 44 is connected to the output of integrator 21 and its output is connected to the input of level detector 22. Furthermore, the output of inverter stage 43 is also connected to an extra input of OR gate 33.

Under the circumstances described, gates 44 and 33 do not provide any signal (= 1). The output signal from level detector 22 becomes 0 so that the controlled switch 23 conducts. Synchronising pulses originating from trigger 24 are passed on to OR gate 32. If they do not already have the values Q = 1 and Q = 0 the output signals from flipflop 29 consequently become Q = 1 and Q = 0. This is the state as described with reference to FIG. 6 for which frequency divider circuit 15 should be reset after the 633rd period of oscillator 11. In fact, the output signal from gate 16 then is 0 so that signal S28 is likewise 0. Before the 633rd period, however, a received field synchronising pulse is present through switch 41 at the relevant input of gate 42 (= 0) so that S28 = 0. Divider circuit 15 is thereby reset.

When standard signals are received (that is to say with 625 lines per image) switch 39 is open so that integrator 21 has the original time constant whilst switch 41 is cut off. Received field synchronising pulses can no longer reach gate 42. The output signal from inverter stage 43 is 0 so that those from OR gates 44 and 33 only depend on the signals from integrator 21 and switch 23, respectively. The circuit arrangement of FIG. 9 operates in the same way as those of FIGS. 1 and 2. It may be noted that the extension of the duration of the divider pulse after the occurrence of the on-phase state in case of reception of non-standard signals is not effected in the embodiment of FIG. 9 because signal Q remains equal to 1. This is no drawback because in the most cases little noise and interference are received.

The so-called negative logic is employed in the foregoing. It is evident that this choice is not important for the essence of the invention. For the positive logic only the denomination of the logical gates shown in the Fig. would have to be changed in known manner.

Elements 10 to 13, 15 to 17, 20 to 26 and 39 to 44 of the described circuit arrangements, with the exception of a capacitor possibly associated with integrator 21, may be advantageously integrated in a semi-conductor body. In view of the large number of components thereof, it is evident that a non-integrated embodiment would not be economical. It may be noted that the described frequency divider circuit and the memory element consist of binary elements. Embodiments of the same scope as described in the present patent application are, however, feasible, using elements of different types.

A television system employing 625 lines per image, two interlaced rasters per image and 50 fields per second has been taken as an example in the foregoing. It will be evident that modifications of the circuit arrangement according to the invention without an essential difference are possible for the reception of television signals in accordance with a different system.

PHILIPS 26C568 CHASSIS K11 Circuit arrangement for generating a control signal for the field output stage in a television receiver PHILIPS CHASSIS K11 (20AX) LINE CONTROL AND FRAME CONTROL UNITS
A circuit arrangement for generating a control signal for the field output stage in a television receiver, provided with a frequency divider circuit by which the double line frequency is divided by a number equal to the number of lines per image. An automatic selection circuit insures that direct synchronization is used in the off-phase state and in case of reception of non-standard signals (for example, from video recorders).



A circuit arrangement for generating a control signal for the field output stage in a television receiver suitable for the reception of line and field synchronizing pulses, in which a plurality of fields constitutes an image, said circuit arrangement comprising a generator for generating a signal of the line frequency or an integer multiple thereof, a field frequency generator coupled to said line frequency generator including a frequency divider circuit having an initial state, means for applying received field synchronizing pulses to a comparison stage for comparing the phase between these pulses and the pulses generated by the frequency divider circuit, the comparison stage applying a signal to a gate associated with said comparison stage, which signal is dependent on the phase difference between the compared pulses, the field generator having direct and indirect synchronization states, a gating pulse generator generating gating pulses, namely a first gating pulse during the occurrence of which the frequency divider circuit is reset to said initial state in the off-phase state of the compared pulses and a second gating pulse which is applied to an automatic selection circuit coupled to said field frequency generator, which selection circuit switches the field generator to the indirect synchronization state when a received field synchronizing pulse and a pulse generated by the frequency divider circuit at least partly occur simultaneously, and switches the field generator to the direct synchronization state when said pulses do not coincide during the occurrence of the second gating pulse. 2. A circuit arrangement as claimed in claim 1, wherein the gating pulses and the frequency divider pulses each have a repetition frequency, the repetition frequency of the gating pulses is equal to the repetition frequency of the pulses generated by the frequency divider circuit divided by an integer n and that each second gating pulse occurs a number of field periods after a first gating pulse. 3. A circuit arrangement as claimed in claim 2, wherein the pulse duration of the two gating pulses is at least approximately one field period. 4. A circuit arrangement as claimed in claim 1, wherein the automatic selection circuit includes a gating device and a bistable element, the field synchronizing pulses, the pulses generated by the frequency divider circuit and the second gating pulses being applied to said gating device, the output signal from the gating device being applied to the bistable element an output signal of which is a second input signal for the gate associated with the comparison stage. 5. A circuit arrangement as claimed in claim 4, wherein said bistable element has a state corresponding to direct synchronization and further comprising a second gating device receiving field synchronizing pulses through a controlled switch rendered conducting by the gate associated with the comparison stage during the off-phase state of the compared pulses, the output signal from the second gating device resetting the frequency divider circuit during the occurrence of the first gating pulse and bringing the bistable element into the state corresponding to the direct synchronization. 6. A circuit arrangement as claimed in claim 1, further comprising a free-running oscillator which is synchronized by the pulses originating from the frequency divider circuit in case of indirect synchronization and by the received field synchronizing pulses in case of direct synchronization. 7. A circuit arrangement as claimed in claim 1, wherein at least the frequency divider circuit, the gating pulse generator, the automatic selection circuit and the gating devices are integrated in a semiconductor body. 8. A circuit arrangement as claimed in claim 2, wherein the gating pulse generator includes an auxiliary frequency divider circuit for dividing the repetition frequency of the pulses generated by the frequency divider circuit. 9. A circuit arrangement as claimed in claim 8, wherein the integer n is determined by the condition ##EQU2## wherein N is the number of lines per image in the television system for which the television receiver is suitable and d is chosen as a function of the number of lines per image of the received television signal. 10. A circuit arrangement for generating a control signal for the field output stage in a television receiver suitable for the reception of line and field synchronizing pulses, in which a plurality of fields constitutes an image, said circuit comprising a generator means for generating a signal of the line frequency or an integer multiple thereof, a field frequency generator circuit coupled to said line frequency generator and having direct and indirect synchronization states and including a frequency divider circuit having an initial state, a comparison stage means coupled to said divider, means for applying received field synchronizing pulses to said comparison stage means for comparing the phase between these pulses and the pulses generated by the frequency divider circuit, a gate, the comparison stage means comprising means for applying a signal to said gate which signal is dependent on the phase difference between the compared pulses, an automatic selection circuit coupled to said gate, a gating pulse generator means coupled to said selection circuit for generating gating pulses, said pulses comprising a first gating pulse during the occurrence of which the frequency divider circuit is reset to said initial state in the off-phase state of the compared pulses and a second gating pulse which is applied to said automatic selection circuit which selection circuit switches the field generator circuit to indirect synchronization when a received field synchronizing pulse and a pulse generated by the frequency divider circuit at least partly occur simultaneously, and switches the field generator circuit to direct synchronization when said pulses do not coincide during the occurrence of the second gating pulse.

Description:

The invention relates to a circuit arrangement for generating a control signal for the field output stage in a television receiver suitable for the reception of line and field synchronizing pulses, in which a number of fields constitutes an image, provided with a generator for generating a signal of the line frequency of an integer multiple thereof, a frequency divider circuit and means for applying received field synchronizing pulses to a comparison stage for comparing the phase between these pulses and the pulses generated by the frequency divider circuit, the comparison stage being capable of applying a signal to a gate which signal is dependent on the phase difference between the compared pulses, the circuit arrangement being switchable between direct and indirect synchronization.

Such a circuit arrangement is described in U.S. Pat. No. 3,708,621. Since in this known circuit arrangement the control signal is derived from the line synchronizing signal by means of frequency division, the frequency thereof is correct as soon as the line synchronizing circuit has synchronized in frequency with respect to the received line synchronization signal, which is generally effected fairly quickly. The correct phase of the field control signal obtained relative to the field synchronizing pulses originating from the transmitter and received by the television receiver is insured by the comparison stage which may be formed as a coincidence gate, and an integrator. In the off-phase state the comparison stage provides a plurality of pulses so that the integrator supplies a signal after a given period which enables the gate. The frequency divider circuit is then reset: this is direct synchronization in which the generated control signal is directly influenced by the received synchronizing pulses. The phase is then correct, the comparison stage no longer provides any pulse and the received synchronizing pulses can in principle no longer reach the divider circuit, at least not as long as the signal generated by the circuit arrangement maintains the same frequency and same phase as the received pulses: this is indirect synchronization in which the received synchronizing pulses cannot directly influence the generated control signal.

It is an object of the invention to provide a circuit arrangement which is also suitable for the reception of "non-standard signals" which are signals in which the number of lines per image deviates from the number prescribed in the relevant television system. Such signals are generated by some test signal generators in which the displayed image is not interlaced and which are used, for example, for adjusting the convergence in colour television receivers or may be produced when using video recorders, for example, for the display of still pictures. Field synchronization with the known circuit arrangement is impossible when such signals are received, for the frequency of the received field synchronizing pulses deviates from the frequency of the pulses obtained by division so that a vertical roll-over is obtained. Some received synchronizing pulses reset, however, the frequency divider circuit so that the image occasionally jumps in the vertical direction.

The circuit arrangement according to the invention is characterized in that it further comprises a gating pulse generator generating gating pulses, namely a first gating pulse during the occurrence of which the frequency divider circuit is reset in the off-phase state of the compared pulses and a second gating pulse which is applied to an automatic selection circuit which selection circuit switches the circuit arrangement to indirect synchronization when a received field synchronizing pulse and a pulse generated by the frequency divider circuit at least partly coincide, and switches the circuit arrangement to direct synchronization when said pulses do not coincide during the occurrence of the second gating pulse.

The invention will be described in greater detail by way of example with reference to the accompanying Figures in which

FIG. 1 shows a block-schematic diagram of a television receiver provided with the circuit arrangement according to the invention,

FIG. 2 shows details of the circuit arrangement according to the invention,

FIGS. 3, 4 and 5 show waveforms which occur in the circuit arrangement according to the invention.

In FIG. 1, 1 is an aerial by which a television signal can be received. This signal is applied to an RF and detection section 2. The detected signal subsequently reaches the audio section 3 of the television receiver and a video amplifier 4 at the output of which a complete video signal, possibly with a chrominance signal in the case of colour television is available. This signal is applied to a section 5 in which it is processed whereafter a picture display tube 6 is controlled, and to a sync. separator 7. The output voltage thereof includes line synchronizing pulses which are applied to a phase detector 8 whose output voltage can influence an oscillator 11 through a flywheel filter 9 and a reactance circuit 10. Oscillator 11 generates a voltage of the double line frequency 2f _H , i.e., 31250 Hz upon reception of a signal in accordance with the television system using 625 lines per complete image, 2 interlaced fields per image and 50 fields per second. Another possibility is that oscillator 11 generates a voltage of the line frequency f _H whose frequency is subsequently doubled. The voltage of the frequency 2f _H controls a frequency divider circuit 12 in which its frequency is divided by two and the signal thus obtained is applied through a pulse shaper 13 to the line output stage 14 which provides the line deflection current for the deflection coil (not shown) for the horizontal deflection of the electron beam(s) in tube 6.

The voltage available at the output of oscillator 11 is also applied to a generator 15 of field frequency signals in which its frequency is divided by the divisor 625 and is further processed. When oscillator 11 has the correct frequency after in frequency synchronization of the circuit 8, 9, 10, 11 for the indirect line synchronization, the frequency of the signal generated by generator 15 is also correct, that is to say, it is equal to the field frequency at the mentioned standard of 50 Hz. A pulse shaper 17 receives the signal generated by generator 15 and controls the field output stage 18 which applies the field deflection current to the deflection coil (not shown) for the vertical deflection of the electron beam(s) in tube 6. Both the line and the field control signals have the waveform required for stages 14 and 18 due to the two pulse shapers 13 and 17, respectively. If the output signal of divider circuit 12 or generator 15 has already this shape, pulse shapers 13 and 17, respectively, may be omitted.

The output voltage of sync. separator 7 also includes field synchronizing pulses which are separately obtained by means of a field sync. separator 19 and are subsequently applied to an input of a coincidence gate 20. The divider pulses originating from an output of generator 15 are present at a second input of gate 20. In the on-phase state that is to say, in the case where a field synchronizing pulse originating from separator 19 and a divider pulse coincide at least partly, stage 20 does not provide a signal. In the off-phase state it provides a signal namely the divider pulse for an integrator 21 which is followed by a level detector 22. When this state lasts at least approximately 0.4 s which corresponds to approximately 20 pulses, the detected level exceeds a given threshold value so that a signal is applied to an input of an AND gate 16. The field synchronizing pulses at the output of separator 19 reach an input of an OR-gate 25 through a controlled switch 23 which can be rendered conducting by the output signal from gate 16. The same output signal is also applied to generator 15.

A further input of gate 16 is connected to an output of an automatic selection circuit 26 while the output signal from gate 25 is applied in a manner to be described hereinafter to generator 15 and to circuit 26. The output signal from pulse shaper 17 is applied to a pulse generator 27 having two outputs one of which is connected to an input of gate 25 and one is connected to an input of circuit 26. The divider pulses which are applied to coincidence stage 20 are also applied to an input of selection circuit 26 while another input thereof as well as another input of generator 15 receive the pulses originating from separator 19.

A control signal for the field output stage is generated by generator 15 with the aid of pulse generator 27 and selection circuit 26. This control signal always has the correct frequency and the correct phase after a short pull-in period irrespective of whether the field synchronizing pulses received from separator 19 are standard signals or not. This will be explained with reference to FIG. 2 in which elements 15, 26 and 27 of FIG. 1 are shown in greater detail.

Generator 15 includes a frequency divider circuit 29 which in known manner, for example, by means of bistable elements, divides the frequency 2f _H of the signal generated by oscillator 11 by 625. As is known ten bistable elements must be present so that the output signal from circuit 29 has a natural frequency of (2f _H /2 ¹⁰ ) which corresponds to a natural period of approximately 33ms. Circuit 29 is internally reset after a field period i.e., approximately 20 ms after the commencement of the period, i.e., 20 - 33/2 ≉ 3.5 ms after reversal in the middle of the natural period. A pulse shaper 30 reduces this flyback pulse to approximately 300 μs which is slightly longer than the field synchronizing pulse provided by separator 19 and whose duration is approximately 200 μs. These pulses are compared in coincidence stage 20. The output signal from pulse shaper 30 also reaches through a first controlled switch 31 a NOR-gate 32. A further input of gate 32 receives through a second controlled switch 33 the output signal from separator 19. The output signal from gate 32 serves as a trigger signal for an oscillator 34 formed in known manner which provides the output signal of generator 15. Switch 33 can be rendered conducting by the output signal from gate 16 while the same signal can render switch 31 conducting through an inverter stage 35. Finally the reset terminal (S ₂ ) of divider circuit 29 is connected to the output of gate 25.

Pulse generator 27 includes an auxiliary frequency divider circuit 36 which may be a counter and by which the repetition frequency in this example 50 Hz of its input signal (FIG. 3a) is divided by an integer n. In this example n is equal to 16, so that the output signal from circuit 36 has a period of n × 20 = 16 × 20 = 320 ms and has the shape as is shown in FIG. 3b. This signal is applied to a gating pulse shaper 37 which generates in known manner two series of gating pulses of the same repetition frequency as that of the signal in FIG. 3b. The first gating pulse (FIG. 3c) is applied to gate 25 and has a duration of approximately 20 ms, that is to say approximately one field period. It is generated for example by a monostable element which is responsive to a trailing edge of the signal in FIG. 3b. The second gating pulse (FIG. 3d) has approximately the same duration as the first and occurs a given number of field periods later, in this example n - 1 = 15, so that its final edge coincides with the leading edge of the next first gating pulse. The second gating pulse is applied to an input of an OR gate 38 forming part of the automatic selection circuit 26. In FIG. 3a the field frequency pulses are shown to be very narrow. In practice they have a given duration so that every time one of these pulses coincides with one of the pulses of FIG. 3c and FIG. 3d. The pulses of FIG. 3c and 3d may alternatively be shifted in such a manner that they commence and end in the period located between two pulses of FIG. 3a.

The divider pulses from pulse shaper 30 and the synchronizing pulses from separator 19 are applied to other inputs of gate 38. The outputs of gate 38 and of gate 25 are connected to the set (S ₁ ) and reset terminals (S ₂ ), respectively, of a flipflop 39 whose Q-output is connected to an output of gate 16.

Oscillator 34 is a free-running oscillator, for example an astable multivibrator, which receives trigger pulses through a gate 32. FIG. 2 shows that these pulses originate from either separator 19 (direct synchronization), or from frequency divider circuit 29 (indirect synchronization), which will now be described in greater detail.

In the on-phase state level detector 22 does not apply a signal to gate 16 which may be indicated by the binary digit 1. During the period of the second gate pulse all input signals from gate 38 coincide at least partly which corresponds to the digit 0 for each input. Under these circumstances the output signal from gate 38 is also 0, that is to say, a set pulse is applied to terminal S ₁ of flipflop 39 so that the output signal Q thereof is 1. The output signal from gate 16 is therefore 1 with the result that switches 23 and 33 are cut off when switch 31 is conducting. The divider pulses are applied through gate 32 to oscillator 34. One of the inputs of gate 25 conveys the signal 1, the output signal thereof is consequently 1: neither divider circuit 29 nor flipflop 39 can be reset. As long as the on-phase state prevails, which means that coincidence occurs every second gating pulse at gate 38 and that level detector 22 provides the signal 1, the situation shown is maintained while the generated control signal cannot be influenced by the received synchronizing pulses.

When the off-phase state occurs, level detector 22 provides after approximately 0.4 s a signal which is equal to 0 for gate 16. The output signal thereof becomes 0 so that switches 23 and 33 conduct while switch 31 is cut off. The synchronizing pulses received from separator 19 are applied through gate 32 to oscillator 34 while the divider pulses cannot influence this oscillator (direct synchronization). The generated control signal is then synchronous with the received signal, but as is shown the duration of this state must be short due to the higher sensitivity to interference of the circuit at least when receiving broadcasting television signals. This is effected as follows. Since the first gating pulse from generator 27 takes approximately one field period a synchronizing pulse occurs fairly quickly simultaneously with a first gating pulse. Both inputs of gate 25 are therefore equal to 0 so that the output thereof is also 0. Frequency divider circuit 29 is reset. Since there is no coincidence in gate 38 the output signal thereof is equal to 1. This is the signal at the terminal S ₁ of flipflop 39 while terminal S ₂ receives a 0. Flipflop 39 is thus reset: Q becomes 0, but circuit 26 has no influence on the rest of the circuit because level detector 22 provides a signal which is equal to 0 for gate 16.

When the incoming signals are standard signals, the above-described situation remains while the divider pulses and the synchronizing pulses are always in phase, this until the occurrence of the next second gating pulse from pulse generator 27. During this occurrence the three input signals of gate 38 are equal to 0 so that a reset pulse 0 is applied to the terminal S ₁ of flipflop 39: Q becomes 1. The input signals from coincidence stage 20 coincide since the beginning of the direct synchronization so that both input signals of gate 16 become equal to 1 at the instant when Q = 1. The output signal from gate 16 thus becomes 1 so that switches 23 and 33 are cut off and switch 31 conducts. Consequently the received signal does not reach the oscillator 34 while the divider pulses are applied to this oscillator (indirect synchronization). Nothing is changed in the state of divider circuit 29 because the output signal of switch 23 and consequently that of gate 25 becomes equal to 1 so that circuit 29 and flipflop 39 are not reset.

After the second gating pulse the output signal from gate 38 becomes equal to 0, but this does not change the state of flipflop 39. During the next second gating pulse a set pulse is applied to flipflop 39 but the output signal Q thereof was already 1 and thus does not change. The foregoing shows that the time elapsing until indirect synchronization occurs is as long as the time interval between both gating pulses, i.e., in the described example (n-1) × 20 = 15 × 20 = 300 ms after the off-phase state is established, i.e., approximately 0.4 s after its occurrence, increased by the time which is necessary for coincidence of a synchronizing pulse with a first gating pulse. Since the divisor corresponding to the incoming synchronizing pulses will not deviate in practice very much from 625 the latter period will last not more than a period of the first gating pulse, i.e., 320 ms. This is the reason why both gating pulses have a duration of approximately one field period. When this duration is shorter it is possible that no coincidence is effected even when receiving standard signals so that the circuit never pulls in. On the other hand a too long duration might render the circuit more sensitive to interference. FIG. 3 shows that a duration of approximately one field period can simply be realized.

When the incoming signals are not standard signals the circuit behaves in a different manner. Since divider circuit 29 is reset during the occurrence of the first gating pulse the two input signals from coincidence stage 20 coincide at least once. It is however uncertain whether this happens more than once and what is the output signal from level detector 22. This, however, has no influence on the manner of synchronization: in fact the divider pulse and the synchronizing pulse do not occur simultaneously during the occurrence of the second gating pulse so that the output signal Q from flipflop 39 remains equal to 0 and hence those of gates 16 and 25 also remain 0 independent of the situation in stage 20. As long as non-standard signals are received the synchronization of oscillator 34 thus remains direct which is no drawback because the signal generated by test signal generators and video recorders generally includes little noise and interference.

Divider circuit 29 is reset at each first gating pulse. When the incoming signal is a standard signal, the divider pulses and the synchronizing pulses are in phase. At the next second gating pulse coincidence is effected in gate 38 so that the circuit is immediately switched over to indirect synchronization. Otherwise a new cycle of n field periods will start. Another function of resetting divider switch 29 every time is the following. When receiving nonstandard signal the time difference between the divider and the synchronizing pulse would increase without this step, with the risk that coincidence might take place in gate 38 at an arbitrary instant so that an unwanted indirect synchronization might be the result.

After the divider pulse and the synchronizing pulse have coincided during the occurrence of the first gating pulse a time difference increasing each period is produced between these pulses upon reception of non-standard signals. Since the period of the signal generated by oscillator 11 is 1/2f _H ≉ 32 μs, this difference after one field period is equal in μs to 32 × (625 - d) in which d is the divisor of the incoming signal deviating from 625. The number n must be chosen to be such that the time difference (n-1) × 32 × (625-d) can be observed by gate 38 after n-1 periods. FIG. 4a shows a synchronizing pulse and FIG. 4b shows a divider pulse which pulses have the stated duration of approximately 200 μs and in an extreme case of the on-phase state this is the state at which the two leading edges coincide. FIG. 4c shows the extreme case of the off-phase state which might occur subsequently and which is the state at which the leading edge of the synchronizing pulse coincides with the trailing edge of the divider pulse. FIG. 5a, 5b and 5c show the opposite situation. FIG. 4a, 4b, 4c, and 5a, 5b and 5c show that the above-mentioned time difference must be in the order of 300 μs. The number n is thus determined by the condition ##EQU1## which proves that the more the divisor d deviates from 625 the less n may be. When for the sake of security the smallest possible difference 625-d = ± 1 is chosen, i.e., d is 624 or 626, a value is found for n which is at least equal to 11. Auxiliary frequency divider circuit 36 might in principle divide the field frequency by 11, but it is simpler to divide by 16, for example, by means of four binary elements, for example, flipflops. As a result the pull-in time is slightly extended relative to the case where n = 11, namely 300 ms instead of 10 × 20 = 200 ms, but it is still acceptable while also the reliability and the insensitivity to interference are increased.

The foregoing shows that the output signal Q from automatic selection circuit 26 is equal to 0 in the off-phase state and becomes 1 after some time when standard signals are received so that synchronization is firstly effected directly and then indirectly. When receiving nonstandard signals Q remains equal to 0 so that direct synchronization is maintained.

It will be noted that the gating pulses in FIGS. 3c and 3d succeed each other after n-1 field periods so that the final edge of one pulse coincide with the leading edge of the other. It will be evident that this is not of essential importance, that is to say, a given time may elapse between these edges. Neither is it necessary for auxiliary frequency divider circuit 36 to divide the frequency of the signal from pulse shaper 17 and not, for example, that from oscillator 34 or from divider circuit 29. Oscillator 34 may be omitted in the case where the output signal from gate 32 has the correct waveform to control pulse shaper 17. In the embodiment of FIG. 2 the gating pulses are obtained by means of auxiliary frequency divider circuit 36. A different method is alternatively possible, namely the integration of the pulses of FIG. 3a. By means of suitable pulse shapers gating pulses can then be obtained whose repetition frequency is not necessarily equal to the field frequency divided by an integer such as is the case with circuit 36.

The so-called negative logic is used in the foregoing, that is to say, the logic in which 0 means "signal" and 1 means "not signal." It is obvious that this choice is not important for the essence of the invention. With the positive logic only the terms for the logical gates shown in FIGS. 1 and 2 would have to be changed in known manner.

Elements 10 to 13, 15 to 17 and 20 to 27 of the described circuit, except for a capacitor optionally associated with integrator 21 may advantageously be integrated in a semiconductor body. In view of the large number of components thereof it is obvious that a non-integrated embodiment would not be economical. It may be noted that the described embodiment includes binary elements. Embodiments of the same scope as in the present application are, however, feasible in which different elements may be used.

A television system using 625 lines per image, 2 interlaced fields per image and 50 fields per second has been used hereinbefore as an example. It will be evident that modifications of the circuit according to the invention are possible without essential difference for the reception of television signals in accordance with a different system.


Philips TBA SERIES SINCE the last part in this series Philips have released details of a PAL -D decoder developed in their laboratories in which most of the circuitry has been integrated into four i.c.s a TBA560Q which undertakes the luminance and chrominance signal processing, a TBA540Q which provides the reference signal channel, a TBA990Q which provides synchronous demodulation of the colour -difference signals, G -Y signal matrixing and PAL V switching, and a TBA530Q which matrixes the colour -difference signals and the luminance signal to obtain the R, G and B signals which after amplification by single -transistor output stages drive the cathodes of the shadowmask tube.

The TBA540Q and TBA560Q and also the TBA500Q and TBA510Q which provide an alternative luminance and chrominance signal processing arrangement will be covered this time.
The internal circuits of the TBA530Q and TBA520Q (predecessor to the TBA990Q which shows how fast things are moving at present) were shown in Part 6 in order to give an idea of the type of circuitry used in these linear colour receiver i.c.s. The internal circuitry is not however of great importance to the user or service engineer: all we need to know about a particular i.c. are the functions it performs, the inputs and outputs it requires and provides and the external connections necessary. The i.c.s we shall deal with in this instalment are highly complex internally the TBA560Q for example contains some 67 integrated transistor elements alone. This time therefore we shall just show the immediate external circuitry in conjunction with a block diagram to indicate the functions performed within the i.c.

TBA540Q Reference Signal Channel:

A block diagram with external connections for this i.c. is shown in Fig. 1. In addition to providing the reference signal required for synchronous demodulation of the colour difference signals this i.c. incorporates automatic phase and amplitude control of the reference oscillator and a half line frequency synchronous demodulator which compares the phases and amplitudes of the burst ripple and the square waveform from the PAL V switch circuit in order to generate a.c.c., colour killer and ident outputs. The use of a synchronous demodulator for these functions provides a high standard of noise immunity in the decoder. The internal reference oscillator operates in conjunction with an external 4.43MHz crystal connected between pins 1 and 15. The nominal load capacitance of the crystal is 20pF. The reference oscillator output, in correct phase for feeding to the V signal synchronous demodulator, is taken from pin 4 at a nominal amplitude of 1.5V peak -to -peak. This is a low -impedance output and no d.c. load to earth is required here. The bifilar inductor Ll provides the antiphase signal necessary for push-pull reference signal drive to the burst detector circuit, the antiphase input being at pin 6. The U subcarrier is obtained from the junction of a 900 phase shift network (R1, C1) connected across Ll. The oscillator is controlled by the output at pin 2. This pin is fed internally with a sinewave derived from the reference signal and controlled in amplitude by the internal reactance control circuit. The phase of the feedback from pin 2 to the crystal via C2 is such that the value of C2 is effectively increased. Pin 2 is held internally at a very low impedance. Thus the tuning of the crystal is automatically controlled by the amplitude of the feedback waveform and its influence on the effective value of C2. The burst signal is fed in at pin 5. A burst waveform amplitude of 1V peak -to -peak is required (the minimum threshold is 0.7V) and this is a.c. coupled. The a.p.c. loop phase detector (burst detector) loads and filter (R2, C4, C5 and C6) are connected to pins 13 and 14. A synchronously -generated a.c.c. potential is produced at pin 9. The voltage at this pin is set by R3 to 4V with zero burst input. The synchronous demodu- lator producing this output is fed with the burst signal and the PAL half line frequency squarewave which is a.c. coupled at pin 8 at 2.5V peak -to -peak. If the phase of the squarewave is correct the potential at pin 9 will fall and normal a.c.c. action will commence. If the phase of the squarewave is incorrect the voltage at pin 9 will rise, providing the ident action as this rise will make the PAL switch miss a count thereby correcting its phase. A colour -killer output is provided at pin 7 from an internal switching transistor. If the ident conditions are incorrect this transistor is saturated and the output at pin 7 is about 250mV. When the ident conditions are correct (voltage at pin 9 below 2.5V) the transistor is cut off providing a positive -going turn -on bias at pin 7. The network between pins 10 and 12 provides filtering and a.c.c. level (R3) setting. The control connected to pin 11 is set so that in conjunction with the rest of the decoder circuitry the level of the burst signal at pin 5 under a.c.c. control is correct. The positive d.c. supply required is applied to pin 3 and the chassis connection is pin 16.

TBA560Q Chroma-Luminance IC:

A block diagram with external connections for this i.c. is shown in Fig. 2. The i.c. incorporates the circuits required to process the luminance and chrominance signals, providing a luminance output for the RGB matrix and a chrominance output for the PAL delay line circuit.
The luminance input is a.c. coupled from the luminance delay line terminating resistor at pin 3. This pin also requires a d.c. bias current which is obtained via the 22kI resistor shown. The brightness control is connected to pin 6: variation from OV to 1 2V at this pin gives a variation in the black level of the luminance output at pin 5 of from OV to 3V, which is a greater range than is needed in practice. The contrast control is connected to pin 2 and the potential applied here controls the gain of both the luminance and the chrominance channels so that the two signals track together correctly. Picture tube beam current limiting can be applied at either pin 6 or pin 2 (by taking the earthy side of one of the controls to a beam limiter network). To maintain correct picture black level it is preferable to apply the beam limiting facility to reduce the contrast. A positive going pulse timed to coincide with the back porch period is fed in at pin 10 to provide burst gating and to operate the black -level clamp in the luminance channel: the black -level clamp requires a charge storage capacitor which is connected to pin 4. The luminance output is obtained from an internal emitter follower at pin 5, an external load resistor of not less than 2kS2 being required here. The output has a nominal black level of 1.6V and 1V black -to -white amplitude. The chrominance signal is applied in push-pull to pins 1 and 15. A.c.c. is applied at pin 14, a negative going potential giving a 26dB control range starting at 1V and giving maximum gain reduction at 200mV. The saturation control is connected to pin 13 and the colour -killer potential is also applied to this pin : the chrominance channel is muted when the voltage at this pin falls below IV. The chrominance output, at an amplitude of about 2V peak -to -peak, is obtained at pin 9: an external network is required which provides d.c. negative feedback in the chrominance channel via pin 12. The burst output, at about 1V peak -to -peak, is obtained at pin 7. A network connected to this pin also provides d.c. feedback to the chrominance input transformer (connected between pins 1 and 15) to give good d.c. stability. Line and field blanking pulses are fed in at pin 8 to the luminance and chrominance channels : these negative -going pulses should not exceed -5V in amplitude. The d.c. supply is applied to pin 11 and pin 16 is the chassis connection.

TBA500Q Luminance IC:

A block diagram with external connections for this i.c. is shown in Fig. 3. This i.c. provides a colour receiver luminance channel incorporating luminance delay -line matching stages, a black -level clamp and a d.c. contrast control which maintains a constant black level over its range of operation. A beam current limiting facility which first reduces picture ,contrast and then picture brightness is provided and line and field flyback blanking can be applied. A video input signal of 2V peak -to -peak with negative -going sync pulses is required at pin 2, a.c. coupled. A clamp potential obtained from pin 13 via a smoothing circuit is fed to pin 2 to regulate the black level of the signal at pin 2 to about 10-4V. The smoothing network for the black -level control potential should have a time -constant which is less than the time constant of the video signal coupling network. The 3V peak -to -peak composite video output with positive -going sync pulses obtained at pin 3 from an emitter -follower can be used as a source of chroma signal: in Fig. 3 it is used as a source of sync pulses for the black -level clamp, fed in at pin 15. This pin requires positive -going sync pulses of 2V amplitude or greater for sync -cancelling the black -level clamp. The other input to the clamp consists of negative going back porch pulses fed in at pin 1 to operate the clamp. The timing of these pulses is not critical provided the pulse does not encroach on the sync pulse period and that it dwells for at least Zus on any part of the back porch-clamp pulse overlap into the picture line period is unimportant. A low-pass filter capacitor for the clamp is connected at pin 14 to prevent the operation of the clamp being affected by the bursts or h.f. noise. The contrast control is connected to pin 5 and is linked to the saturation control so that the two track together. A variation of from 2 to 4V at pin 5 gives a control range of at least 40dB, the relationship between the video at pin 4 and the potential at pin 5 being linear. An output to drive the luminance delay line is provided at pin 4. This is a low -impedance source and a luminance delay line with a characteristic impedance of 1-2.7161 can be used. The delayed luminance signal is fed back into the i.c. at pin 8. Line and field flyback banking pulses and the brightness control are also connected to this pin. The gain of the luminance channel is determined by the value of the resistor connected to pin 9. The luminance output is taken from an emitter -follower at pin 10, an external load resistor being required. The voltage output range available is from 0.7V to 5-5V. The potential of the black level of the output signal is normally set to 1.5V by appropriate setting of the potential at pin 8. A luminance signal output amplitude of 2.8V black to white at maximum contrast is produced : superimposed on this is the blanking waveform which remains of constant amplitude independently of the contrast and brightness control settings. A beam current limiting input is provided at pin 6. A rising positive potential at this pin will start to reduce the contrast at about 2V. Further increase in the voltage at this pin will continue to reduce the contrast until a threshold is reached, determined by the potential applied to pin 7, when the d.c. level of the video signal is reduced giving reduction in picture brightness. The d.c. supply is connected to pin 12 and pin 16 is the chassis connection.

TBA510Q Chrominance IC:

A block diagram with external connections for this i.c. is shown in Fig. 4. It provides a colour receiver chrominance signal processing channel with a variable gain a.c.c. chroma amplifier circuit, d.c. control of chroma saturation which can be ganged to the opera- tion of the contrast control, chroma blanking and burst gating, a burst output stage, colour -killer circuit and PAL delay line driver stage. The chroma signal is a.c. coupled to pin 4, the a.c.c. control potential being applied at pin 2. The non - signal side of the differential amplifier used for the a.c.c. system is taken to pin 3 where a decoupling capacitor should be connected. A resistor can be connected between pins 2 and 3 to reduce the control sensitivity of the a.c.c. system to any desired level. The saturation control is connected to pin 15, the d.c. control voltage range required here being 1.5-4-5V. For chrominance blanking a negative -going line flyback pulse of amplitude not greater than 5V is fed in at pin 14. A series network is connected to pin 6 to decouple the emitter of one of the amplifying stages in the i.c.: the value of the resistor in this network influences the gain of both the burst and the chroma channels in the i.c. The chrominance signal outputs are obtained at pin 8 (collector) to drive the chroma delay line and pin 9 (emitter) to feed the chrominance signal matrix (undelayed signal). A resistive path to earth is essen- tial at pin 9. The colour -killer turn -on bias is applied to pin 5 : colour is "on" at 2.3V, "off" at 1.9V. Chroma signal suppression when killed is greater than 50dB. The burst signal output is at pin 11 (collector) or 12 (emitter). If a low -impedance output is required pin 11 is connected direct to the 12V supply rail and the output is taken from pin 12. An external load of 2kn connected to chassis is required here. The burst gating pulse is fed in at pin 13, a negative -going pulse of not greater than 5V amplitude being required. Pins 7 and 10 are connected to an internal screen whose purpose is to prevent unwanted burst and chroma outputs : the pins must be linked together and taken via a direct path to earth. Pin 1 is the d.c.
supply pin and pin 16 the chassis connection.

A TBA510 as example  is used in the Grundig 1500/3010 series and also the YR 1972 Grundig colour chassis (5010 / 5050  series) introduced in the70's. Grundig continue in these models to favour colour -difference tube drive. The 5010 series uses a TBA510 together with a TAA630 colour demodulator i.c. in the chrominance section and a TBA970 luminance i.c. which drives a single BF458 luminance output transistor operated from a 280V rail. As this series has been appearing more and more i.c.s have come to be used in television receivers, both monochrome and colour, and more and more i.c.s designed for television set use have been announced. Some of these have been mentioned in recent argumentations here in this Web Museum. There seems little doubt that a major increase in the use of integrated circuits in television receivers is about to occur in the future. Fully integrated i.f. and vision detector sections are already in use (PHILIPS K9-K11) and this is the likely area, together with the decoder in colour sets, in which integration will most rapidly spread. Elsewhere integrated line and field oscillators using circuits without inductors have been developed and a field output stage in integrated form is now feasible. Line output stages consisting of hybrid i.c. and thick film circuits (PHILIPS K12) have been built and there is a programme of work directed to the integration of the r.f. tuner, using digital frequency synthesisers to provide local oscillator action controlled by signals from a remote point.

We seem to have reached the position where the only part of the set which does not attract the i.c. manufacturers is the picture tube itself !

PHILIPS CHASSIS K11 COLOR BURST CIRCUIT WITH A.G.C.A color television receiver has at least partially separate color information and burst signal paths. A passive burst subcarrier regenerator is located within said burst signal path. In order to supply a constant amplitude regenerated subcarrier without effecting the amplitude of the color information signal, an amplitude detector is coupled to the output of the regenerator. The detected signal goes through a high-pass filter and is used to control the gain of an amplifier located exclusively within the burst signal path.


1. A circuit comprising: means for receiving a color television signal having amplitude varying color information and burst signal components, means coupled to said receiving means for separating said components from said television signal, a burst signal path coupled to said separating means for receiving only said burst signal, said path comprising the serial coupling of means for passively regenerating a subcarrier reference signal from said burst signal, means having a control terminal for controlling the amplitude of said subcarrier reference signal within said burst signal path without effecting the amplitude of said color information signals, means for detecting the amplitude of the output of said amplitude controlling means and a high pass filter coupled between said detecting means and said control terminal; whereby said reference signal is kept at a substantially constant amplitude regardless of the rapidity of said variations. 2. A circuit as claimed in claim 1 further comprising a chrominance amplifying means for amplifying both said color information and burst signal components and means for controlling the gain of said chrominance amplifier coupled to the output of said detecting means. 3. A circuit as claimed in claim 2 wherein said chrominance signal amplifier-controlling means comprises a low-pass filter having a higher cutoff frequency than said high-pass filter.

Description:

The invention relates to a color television receiver having a color information signal path and a burst signal path, which burst signal path includes a color subcarrier regenerator of a passive type (passive integrator) so that a burst signal obtained from a received color television signal can be converted into a color subcarrier reference signal, the burst signal path including a detection circuit for obtaining at an output thereof an automatic gain control signal from the color subcarrier reference signal.

In known receivers of the above-mentioned type the drawback occurs that particularly upon reception of weak signals the color subcarrier reference signal may show great fluctuations as a result of the burst signal decreasing in amplitude sometimes during a number of successive line periods. This reference signal is used for synchronously demodulating the color difference signals which are passed through the color information signal. Upon variation in the amplitude of the reference signal this may give rise to color errors upon this demodulation. Consequently, to prevent this phenomenon a limiter stage is generally included after the passive integrator. However, this limiter does not operate at a slight amplitude of the integrated burst signal. The operation of this limiter may be rendered more effective by using more amplification stage for this limiter. From an economic point of view this is, however, not particularly interesting. An object of the invention is to avoid as must as possible the occurrence of color errors upon reception of weak signals.

According to the invention a color television receiver of the type described in the preamble is characterized in that the said output of the detection circuit is connected through a low cutoff filter to a gain-control input of an amplifier included in the burst signal path outside the color information signal path.

As a result an automatic gain control is obtained which acts upon comparatively rapid variation in the amplitude of the regenerated color subcarrier reference signal and which tends to maintain the amplitude of this reference signal constant. By the step according to the invention this rapid automatic gain control is not effective in the color information signal path. This is based on the recognition of the fact that due to the short duration of the burst signals amplitude variation often occur in the individual bursts, which variation are not representative of the amplitude variations which occur in the associated line periods in the color information signal. Any automatic gain control for the color information signal path obtained from the burst signal path must therefore not be influenced by accidental fluctuations of the burst signal amplitude, such as occur particularly upon reception of weak signals.

In order that the invention may be readily carried into effect it will now be described in detail, by way of example, with reference to the accompanying diagrammatic drawing which shows a color television receiver according to the invention in a block diagram.

Details which are not important for the understanding of the invention have been omitted as much as possible for the sake of clarity.

In the Figure, a section of the receiver is indicated by 1 in which a color television signal receiver through an input 3 is amplified and converted into a brightness signal Y, a chrominance signal 1 Chr and a synchronization signal S. These signals occur at the outputs 5, 7 and 9, respectively, of the section 1.

The output 5 of the section 1 is connected to an input 11 of a picture display section 13. The brightness signal Y is applied through this line to the picture display section 13.

The output 7 of the section 1 is connected to an input 15 of a chrominance amplifier 17. An output 19 of the chrominance amplifier 17 is connected to an input 21 of a separator stage 23. The separator stage 23 further has an input 25 which is connected to an output 27 of a time base state 29, through which line it is possible to apply a switching signal to the separator stage 23.

The time base stage 29 receives a synchronization signal S from an input 31 connected to the output 9 of the section 1 and supplies time base currents to the picture display section 13 through an output 33 which is connected to an input 35 of the picture display section 13.

The chrominance signal Chr becoming available at the output 7 of the section 1 comprises a color information signal and a burst signal. The color information signal is applied from the output 7 through the chrominance amplifier 17 and the separator stage 23 to an output 37 thereof and the burst signal is applied from the output 7 through the chrominance amplifier 17 and the separator stage 23 to an output 39 of this stage. To this end a time selection is applied on the chrominance signal in the separator stage 23 with the aid of a switching signal applied to the input 25.

The output 37 is connected to an input 41 of a color information signal amplifier 43. An output 45 thereof is connected to an input 47 of a demodulator and matrix circuit 49. The demodulator and matrix circuit 49 has three outputs 51, 53 and 55 which are connected to inputs 57, 59 and 61, respectively, of the picture display section 13.

The signal path leading from the output 7 of the section 1 through the chrominance amplifier 17, the separator stage 23, the output 37 of this separator stage, the color information amplifier 43 to the input 47 of the demodulator and matrix circuit 49 belongs to the color information signal path. The color information signal is applied through this path to the demodulator and matrix circuit 49.

The output 39 of the separator stage 23 is connected to an input 63 of a burst signal amplifier 65. An output 67 of the burst signal amplifier 65 is connected to an input 69 of a passive integrator circuit 71 and to an input 73 of a phase detection circuit 75. The passive regenerator is a high-Q crystal circuit. This circuit along with the phase detector and their operation are described in "Proceedings of the I.R.E.," Jan. 1954, vol. 42, pp. 111--112.

The color subcarrier burst are integrated to form a continuous reference signal with the aid of the passive integrator circuit 71. This reference signal becomes available at the output 77. The output 77 is connected to an input 79 of a reference signal amplifier 81. A reference signal which is applied to an input 85 of the demodulator and matrix circuit 49 becomes available at an output 83 of this amplifier.

The reference signal is further applied to an input 87 of the phase detection circuit 75. The phase of the burst signal applied through the input 73 is compared in the phase detection circuit 75 with that of the integrated burst signal (reference signal) applied through the input 87. A voltage which is a measure of the phase deviation between these two signals is obtained at an output 89. The output 89 is connected to the input 91 of the passive integrator circuit 71. A phase deviation possibly produced in the integrator circuit 71 is corrected with the aid of the voltage applied through this line, so that the phase of the reference signal obtained at the output 83 is maintained as much as possible the same as that of the burst signal applied to the input 69.

The reference signal obtained at the output 83 is further applied to a detection circuit having a diode 92, a capacitor 93 and a resistor 95. A voltage dependent on the amplitude of the reference signal is obtained from an output 97 of the detection circuit 92, 93, 95. This voltage is applied through a low-pass filter serving as a high cutoff filter including a resistor 99 and a capacitor 101 to a gain control input 103 of the chrominance amplifier 17. The gain of the chrominance amplifier 17 is thus dependent on the average amplitude of the burst signal. This average amplitude is thus maintained substantially constant. The average amplitude of the burst signal is a measure of the amplitude of the color information signal. Hence the color information signal appears with a automatically corrected amplitude at the output 19 of the chrominance amplifier 17. The saturation of a picture obtained with the aid of the color information signal will thus be substantially independent of variations in the transmission of the transmission path of the color television signal.

The above described trajectory from the output 7 of the section 1 through the chrominance amplifier 17, the output 39 of the separator stage, the burst signal amplifier 65, the passive integrator circuit 71, the reference signal amplifier 81 and the detection circuit 92, 93 95 belongs to the burst signal path. Part of the burst signal path, namely the chrominance amplifier 17, coincides with part of the color information signal path.

According to the invention the output 97 of the detection circuit 92, 93, 95 provided in the burst signal path is connected through a high-pass filter serving as a low cutoff filter, including a capacitor 105 and a resistor 107, to a gain control input 109 of an amplifier 81 included outside the color information signal path. Rapid variations in the output signal of the reference signal amplifier 81 will be readjusted by the automatic gain control circuit thus formed without exerting influence on the color information signal path.

According to the invention this rapid automatic gain control, which is effected outside the color information signal path, is based on the recognition of the fact that rapid variations in the amplitude of the burst signal such as occur, for example, upon reception of weak signals or during the frame flyback period, are no measure of the variations in the color information signal and hence must not exert influence on a possible automatic gain control in the color information signal path.

By the step according to the invention a very constant reference signal voltage amplitude is obtained at the input 85 of the demodulator and matrix circuit 49 so that it will substantially be impossible for color errors to occur due to the demodulation of the color information signal, even with unfavorable conditions of reception.

The lower limit frequency of the high-pass filter 105, 107 is preferably chosen to be such that it is higher than the upper limit frequency of the low-pass filter 99--101.

It will be evident that the rapid automatic gain control according to the invention can be used in color television receivers for both the NTSC-system and the PAL-system.

Although the described embodiment includes a control voltage from the output 97 of the detection circuit 92, 93, 95 to the input 103 of the chrominance amplifier 17. It is readily evident that this voltage is not essential for using the step according to the invention. However, to ensure a satisfactory operation of the color difference signal demodulators it is generally desirable to apply this control voltage to the input 103.

In the embodiment described the feedback of the rapid automatic gain control is effected in the reference signal amplifier 81 following the passive integrator circuit 71. The feedback may in principle also be effected in an amplifier, for example, preceding the passive integrator circuit or, at will, preceding as following it.

PHILIPS PAL CHROMA DELAY LINE:An improved ultrasonic delay line comprising a solid glass body having one or more slits in the side walls extending inwardly from the outer edge faces of the body. The slits are arranged in the path of the propagating ultrasonic energy so as to effectively increase the number of energy transmission paths in the body by acting as additional energy reflecting surfaces. The slits extend the effective length of the delay line. The slits also operate to reduce undesired cross-coupling between the input and output transducers.

1. An ultrasonic delay line comprising a solid body having a plurality of energy reflecting edge walls and composed of ultrasonic wave energy transmitting material, said edge walls being arranged to provide a first point for introducing ultrasonic energy and a second point for extracting said energy from the body and further providing a plurality of multiply reflected internal transmission paths for delaying said ultrasonic energy, and an energy reflecting surface positioned in the desired energy transmission path and formed by a wall slit arranged to block the passage therethrough of impinging ultrasonic energy and extending inwards from an outer edge wall of the solid body and positioned so as to provide substantially complete reflection of the desired energy from opposite faces thereof to the edge walls thereby to redirect said energy through 2. A delay line as claimed in claim 1 wherein the wall slit is arranged relative to one or more reflecting edge walls of the body so as to produce an odd number of said paths between a reflection from one face of the slit and a subsequent reflection from the same or opposite face of said slit. 3. A delay line as claimed in claim 2 further comprising first and second electromechanical transducers coupled to said body at said first and 4. A delay line as claimed in claim 1 further comprising first and second electromechanical transducers coupled to said body at said first and 5. A delay line as claimed in claim 1 wherein the wall slit is located in the plane of symmetry of the body, said delay line further comprising first and second electromechanical transducers coupled to said body at 6. A delay line as claimed in claim 1 wherein said body includes a second wall slit extending inwards from an outer edge wall and positioned so that the ultrasonic energy is reflected off of opposite faces of the second 7. A delay line as claimed in claim 1 wherein the wall slit is arranged in the body relative to one or more reflecting edge walls thereof so as to multiply reflect the desired energy from said wall slit to produce an odd number of said paths between a first reflection from one face of the slit 8. An ultrasonic delay line comprising a solid body having a plurality of energy reflecting edge walls and composed of ultrasonic wave energy transmitting material, said edge walls being arranged to provide a first point for introducing ultrasonic energy and a second point for extracting said energy from the body and further providing a plurality of multiply reflected internal transmission paths for delaying said ultrasonic energy, and an energy reflecting surface positioned in the desired energy transmission path and formed by a wall slit arranged to block the passage therethrough of impinging ultrasonic energy and extending inwards from an outer edge wall of the solid body to reflect the desired energy to the edge walls thereby to redirect said energy through the body, said wall slit being arranged in the body so as to intercept ultrasonic energy propagating along given undesired transmission paths between said first and second energy points of the body thereby to reduce any direct coupling of scattered secondary ultrasonic energy between said first and second 9. An ultrasonic delay line comprising a solid body having at least five energy reflecting edge walls and composed of ultrasonic wave energy transmitting material, two of said edge walls being parallel to each other and orthogonal to a third edge wall, the fourth and fifth edge walls each being at an angle of approximately 135° to a respective one of said parallel edge walls, said edge walls being arranged to provide a first point for introducing ultrasonic energy and a second point for extracting said energy from the body and further providing a plurality of multiply reflected internal transmission paths for delaying said ultrasonic energy, and an energy reflecting surface positioned in the desired energy transmission path and formed by a wall slit extending centrally inwards into the body orthogonal to said third edge wall and arranged to block the passage therethrough of impinging ultrasonic energy thereby to redirect 10. A delay line as claimed in claim 9 wherein said wall slit extends 11. A delay line as claimed in claim 9 wherein said fourth and fifth edge walls intersect one another and said wall slit extends inwardly from the 12. A delay line is claimed in claim 1 wherein said body has a generally rectangular shape and a second wall slit extending inwards from an outer edge wall of the body so as to reflect the ultrasonic energy, the first and second wall slits extending inwards from opposite parallel edge walls 13. A delay line as claimed in claim 9 further comprising first and second electromechanical transducers coupled to said body at one or more edge 14. A delay line as claimed in claim 11 further comprising first and second electromechanical transducers coupled to said body at said fourth and fifth edge walls, respectively, thereby to reduce direct coupling of 15. An ultrasonic delay line comprising a solid body having a plurality of energy reflecting edge walls and composed of ultrasonic wave energy transmitting material, said edge walls being arranged to provide a first point for introducing ultrasonic energy and a second point for extracting said energy from the body and further providing a plurality of multiply reflected internal transmission paths for delaying said ultrasonic energy, an energy reflecting surface positioned in the desired energy transmission path and formed by a wall slit arranged to block the passage therethrough of impinging ultrasonic energy and extending inwards from an outer edge wall of the solid body thereby to redirect said energy through the body, and a second wall slit extending inwards from an outer edge wall of the body so as to reflect the ultrasonic energy, the first and second wall slits extending inwards from opposite parallel edge walls of the body, and wherein said body has a parallelogram cross-section and said first and second wall slits extend orthogonally inwards from the longer pair of 16. A delay line as claimed in claim 12 further comprising first and second electromechanical transducers coupled to said body at said first and second points which are located on edge walls other than the edge walls 17. An ultrasonic delay line comprising a solid body having at least five energy reflecting edge walls and composed of ultrasonic wave energy transmitting material, two of said edge walls being parallel to each other and two other edge walls being at right angles thereto, a fifth edge wall being at an angle of approximately 135° to each of two adjacent edge walls, said edge walls being arranged to provide a first point for introducing ultrasonic energy and a second point for extracting said energy from the body and further providing a plurality of multiply reflected internal transmission paths for delaying said ultrasonic energy, and an energy reflecting surface positioned in the desired energy transmission path and formed by a wall slit extending into the body centrally of and orthogonal to one of the edge walls located opposite to the fifth edge wall and arranged to block the passage therethrough of impinging ultrasonic energy thereby to redirect said energy through the 18. A delay line as claimed in claim 17 further comprising first and second electromechanical transducers coupled to said body at said first and 19. A delay line as claimed in claim 18 wherein said first and second energy points are located on the fifth edge wall.

Description:

This invention relates to ultrasonic delay lines of the type using a solid medium such as quartz or glass through which an acoustic signal wave is made to travel to provide a time delay between the application of the wave and its extraction. In such delay lines it is known to shape the solid medium so as to provide internal peripheral reflective surfaces for the ultrasonic wave in order to fold the wave over a plurality of legs to increase the length of the transmission path through the medium and thus increase the wave delay with a minimum mass of solid medium.

It is also known to increase the length of the transmission path of an ultrasonic wave by including specially shaped openings in the solid medium to provide additional reflective surfaces. In this case such openings have to be very accurately positioned and dimensioned to ensure proper operation.

In connection with such delay lines there arises a number of problems. Some of these concern the solid medium itself and its thermal properties. Delay lines using wavelengths equivalent to several Megahertz require very accurate dimensioning to reduce internal energy scatter and give an accurate source of extraction. This requires a solid medium having a very low temperature coefficient. A special glass having such properties is available but it is relatively costly for use in mass production so that any design steps that will allow an overall reduction in the mass of the delay medium will not only in itself reduce thermal problems but will also reduce overall costs.

In certain color television receiver systems a prescribed signal delay is required so that the delay line has to provide stable operation and yet lend itself to mass production at a very low cost.

Another problem which confronts the designer of such delay lines is the prevention of direct signal coupling between the application and extraction points of the signal which can result in the desired delayed signal being masked by a strong undelayed signal arriving at the extraction point. A further problem is the suppression of alternative signal paths which contribute a train of secondary spurious signals each having a different delay and which make extraction of the wanted delayed signal difficult.

The purpose of this invention is to provide a simple delay line construction in which the overall mass of the delay line medium is reduced in a manner which will also allow greater freedom from expensive manufacturing processes as well as providing enhanced electro-acoustical performance.

According to this invention there is provided an ultrasonic delay line using a solid medium through which an ultrasonic signal wave is made to travel and which is reflected over a plurality of paths to increase the time delay between the application point of the ultrasonic signal and its point of extraction, wherein the path followed by the ultrasonic waves includes at least one reflecting surface constituted by the side wall or face of a slit extending inwards from an edge face of the solid medium.

In order to make maximum utilization of a given delay line mass, the delay line may include several slits arranged so that both side walls of the slits can be used as reflective surfaces. Furthermore, if the geometrical pattern of the reflected signal legs or path is so arranged that an odd number of legs exists between reflections on the same or associated slit wall, this gives the advantage that the angular orientation of the slit is non-critical and it displays self-cancelling properties for minor errors.

Furthermore, the use of slits to provide reflective walls also has the advantage of reducing spurious secondary signals in that a greater control can be exercised over the required signal path by the very high damping barrier provided by the absence of any delay line medium forming the slit. This reduces any signal transference across the slit to a value far below the minimum requirements.

It should be noted that the use of notches introduced in the edge surfaces of a solid medium for a delay line to reduce secondary waves from reaching the output transducer is known per se. However, these notches do not constitute reflecting walls for the desired signal.

Examples of this invention will now be described with reference to the accompanying drawings in which FIG. 1 is a plan view of a substantially rectangularly shaped delay line showing a simplified embodiment of applicant's invention.

FIG. 2 is a plan view of a substantially rectangularly shaped delay line showing two slits for further increasing the length of the delay line of FIG. 1.

FIG. 3 is a plan view of a delay line having five reflecting faces for further increasing the length of the delay line of FIG. 1.

FIG. 4 is a plan view of a delay line shaped as a parallelogram having four slits.

FIG. 5 is a plan view of a delay line having five edges and a central slit.

FIGS. 1 to 5 show five different embodiments of delay lines according to this invention. Each Figure has certain design features which will be discussed below.

FIG. 1 shows a solid body 1 made, for example, of glass and having a substantially rectangular cross-section. Two corners of the body 1 are beveled and transducers A and B are arranged on the surfaces 14 and 15, respectively. The surfaces 14 and 15 are at respective angles of 135° to the surfaces 17, 18 and 18, 19 of the body 1. The input transducer A has an electric signal applied to it which is converted by the transducer into an acoustic ultrasonic signal. This acoustic signal propagates in the form of a wave through the body 1 and after a number of reflections it reaches the transducer B which reconverts it into an electric signal. The time required for the acoustic ultrasonic wave to cover the entire path (shown in dotted lines) from the transducer A to the transducer B determines the delay time between the application of the electric input signal at the transducer A and the electric output signal recovered at the transducer B. Use is preferably made of piezo-electric transducers which are so polarized that shear mode vibrations are produced so that the overall reflection at each of the reflective surfaces occurs without energy conversion of the shear vibrations into longitudinal vibrations.

According to this invention, a slit 2, in the form of a saw-cut having plane parallel walls, is provided at the plane of symmetry in the body 1 so that the waves originating from the transducer A first reflect at the left-hand wall of the slit 2 and then at the rectangular walls 16, 17, 18, 19, and 20 of the body 1, whereupon they are reflected from the right-hand wall of the slit 2 and finally strike the transducer B. The energy path from transducer A to transducer B is made up of eight reflected signal legs shown by dashed lines with arrowheads. It will be apparent from FIG. 1 that an increased path length for the ultrasonic wave is thus obtained in a simple manner. Moreover, secondary waves are suppressed by the slit 2. The angle at which the ultrasonic wave strikes the various reflective surfaces is always 45°. However, in this embodiment the angle 3 of 90° between the slit 2 and the surfaces 16 and 20 must be very accurately defined in order that the waves may follow the path indicated.

In the delay line of FIG. 2, the signal paths (shown in dotted lines) are obtained by providing two slits 2 and 4 at suitably chosen areas at right-angles to the long surfaces 21 and 22 of the delay line medium 1. In this embodiment the ultrasonic waves also strike the reflective surfaces at angles of 45°. However, after reflection at one wall of the slit 2, an odd number of signal legs (five) occurs before reflection at the other wall of the slit 2. As a result, the orientation of the angles 5 and 6 of 90° is not critical and the angular errors introduced into the reflected signals are cancelled automatically. In this construction, the slits 2 and 4 also cause a reduction of secondary (spurious) signals, and moreover the formation of any direct or secondary transmission path between the input transducer A and the output transducer B is prevented.

The delay line construction of FIG. 3 provides an increased length of the transmission path while retaining the advantages of the delay line constructions shown in FIGS. 1 and 2. In this case, the body 1 has a square cross-section (a corner of the square being denoted by x--x) and the opposite corner of the square is removed so that an additional wall 31 is formed on the body 1 which is at an angle of 135° to the walls 32 and 33. The transducers A and B are arranged side by side on the wall 31, while a slit 8 is provided at right angles to and approximately centrally of a wall 34 of the body 1 and extends approximately as far as half the length x into the body 1. The ultrasonic waves again follow the path indicated by dotted lines.

Either the transducer A or the transducer B may be used as input or output. Since the number of signal legs between the reflections at one wall and those at the other wall of the slit 8 is odd (five), the orientation of the angle 7 of 90° between the slit 8 and the surface 34 is not critical because the angular error introduced into the signal wave is automatically canceled. This self-canceling effect is illustrated in FIG. 3, in which the slit 8 is purposely slightly tilted. A practical embodiment of a glass delay line of this construction for use in a PAL color television receiver system has the following approximate dimensions:

x = 33 mm, y = 15 mm, and z = 6 mm.

The width of the slit 8 is approximately 1 mm and this slit extends over approximately 15 mm into the delay line 1. The electric characteristics give a delay of one line period, i.e., approximately 64 μ sec, at a band center frequency of 4.4 Mc/s.

FIG. 4 shows a body 1 in the form of a rectangular prism having a cross-section in the form of a parallelogram whose sides 41, 42 and 43, 44 respectively are at angles of 45° to each other. Slits 8, 10 and 9, 11, respectively, are provided at right angles to the side faces 42 and 44. In this delay line, only one side wall of each of the slits 8, 9, 10, and 11 is used at a time. An input transducer A is arranged for injecting an ultrasonic signal which follows the path shown in dotted lines and which is extracted by the output transducer B. In this construction, any angular displacements of the slits are not automatically canceled and the angles are therefore critical, but the remote positioning and interspersion of the slits between the input transducer A and the output transducer B provides a high degree of decoupling for spurious (secondary) signals when compared with known delay lines.

The surface of the delay line of FIG. 5 has a cross-section in the form of a pentagon having two parallel sides 51 and 52 and a third side 53 at right angles to the sides 51 and 52, while the fourth and fifth sides 54 and 55 are at angles of 135° to the sides 51 and 52, respectively. The latter sides 54 and 55 support the transducers A and B, respectively. According to the invention, a slit 56 is positioned at the intersection of the sides 54 and 55 and extends into the body 1 parallel to the sides 51 and 52 over a distance approximately equal to half the length of the sides 51 and 52. The path followed by the ultrasonic waves is shown in dotted lines. Small angular displacements of the surfaces 51 and 52 again substantially do not influence the overall delay time and the direction in which the waves strike the output transducer B. Also, the slit 56 prevents the direct coupling of scattered radiation from the input transducer A to the output transducer B.

It will be evident from the foregoing that delay lines constructed in accordance with this invention can be easily and economically mass produced. A comparatively long rod of delay line medium may be profiled, for example, in the desired shape, while the slits may be accurately arranged throughout its length. The method of manufacturing separate delay lines then merely resides in parting off portions of the rod to the desired thickness. This results in a high reproducibility of components of individual delay lines.

The invention is not limited to the delay line described consisting of a single layer, but the advantages of this invention may also be obtained in delay lines consisting of several layers, the path followed by the signal in one layer then being reflected at a suitable point to a further layer so that it can pass on through this further layer before it is extracted.

PHILIPS CHASSIS K11 GLASS TECHNOLOGY FOR PHILIPS PAL CHROMA DELAY LINE / ACOUSTIC DELAY LINE:A glass for an acoustic delay line which consists of SiO₂, Al₂ O₃, B₂ O₃ and an oxide of a bivalent metal and satisfies the requirement that -5×10^-6 <Σ_i α_i x₁ < +5×10^-6 where α_i is the temperature coefficient of the rate of propagation in the range of 20°-70° C for the oxide component i and x_i is the molar fraction of that component.

1. In an acoustic delay line of the type having signal converting elements on the surface of a glass body for converting an input electric signal into an acoustic signal and an output acoustic signal into an electrical signal, the improvement comprising that said body of glass consist of the following compositions in wt. percent: 2. In an acoustic delay line of the type having signal converting elements on the surface of a glass body for converting an input electric signal into an acoustic signal and an output acoustic signal into an electric signal, the improvement comprising that said body of glass consist of the following composition in wt. percent:

Description:

The invention relates to an acoustic delay line in which the delay medium is glass.

Such delay lines are known per se for electronic uses in which delays of electric signals in the order 0.01-1 millisecond are to be obtained with bandwidths of a few tens of mc/s. The delay is produced in that an electric signal is converted, by means of a piezo-electric element, into an ultrasonic mechanical vibration, preferably a shear vibration, and after said acoustic signal has traversed the delay medium this is likewise converted again into an electric signal by a piezo-electric element, said signal having experienced the desired delay with respect to the original signal. The rate of propagation of the acoustic shear waves in a solid is approximately 10 ⁵ times smaller than that of electro-magnetic waves so that a comparatively large delay can be obtained over a comparatively small distance.

Delay lines are used inter alia in electronic computers, in radar technology and in television technology. In two color television systems delay lines are used for combining the color information of adjacent lines of a frame. The delay time required for this purpose is approximately 64 μsec. with 625 lines and a frequency of 50 c/s. At the frequency to be considered of 4.43 mc/s and the required bandwidth of approximately 2 mc/s, glass is a suitable delay medium.

A known glass which is excellency suitable for this purpose has the following composition in mol. percent:

SiO ₂ 70-78 PbO 15-30, of which maximally 5 mol. percent may be replaced by one or more of the oxides MgO, BaO, CaO and SrO, Na ₂ O + K ₂ O 0-7 Na ₂ O ≤0.5 SB ₂ O ₃ + As ₂ O ₃ ≤ 0.5

this glass is distinguished by the quality of various properties which are of importance for the end in view. Taking into account the temperature variations of ±30° C occurring in practice, the delay times does not vary more than 0.02 μsec. This means that the temperature coefficient of the delay time dτ/(τdτ) of these glasses is smaller than 10 × 10 ^-6 per ° C and in some cases even smaller than 1 × 10 ^-6 per ° C.

The damping of the acoustic vibrations in delay lines of this class is not too large. The mechanical attenuation of said glass is not more than 9 × 10 ^-3 dB/μs. Mc/s which is amply sufficient for delay lines in television receivers.

A further advantage of this glass consists in that it is very slightly sensitive to the previous thermal history of the glass which means that it has substantially no influence on the temperature coefficient of the delay time, whether the glass has been cooled relatively rapidly or slowly from temperatures in the proximity of the annealing temperature. Large variations in the treatment which consists of a heating for approximately 10 minutes at a temperature which lies approximately 50° C above the annealing temperature succeeded by cooling at a rate of approximately 1.5° C per minute, do substantially not influence the reproducibility.

Finally, a hysteresis effect is not present in this glass to any inconvenient extent, in contrast with some other known glasses. This hysteresis effect manifests itself in the delay time when the glass is heated from room temperature to a temperature between 60° and 80° C, is kept at said temperature for more than 1 hour, and is then cooled to room temperature again. The delay time at room temperature may be increased 1 to 10 ⁴ , said increase disappearing again gradually in the course of a few days. In the above-mentioned glasses said variation is at most 3 to 10 ⁵ at the temperature cycle described.

The rate of propagation for shear waves in these glasses is comparatively low and varies only slightly with the composition (2,400-2,600 m/sec.).

A difficulty in manufacturing the glass compositions required for delay lines is associated with the fact that small variations in the composition of a chosen glass may cause variations in the acoustic properties, notably in the temperature coefficient of the delay time. This is most undesirable, particularly when used in delay lines for color television. So this involves the necessity of keeping the content of the components of the glass constant between narrow limits. The known glasses have a high content of lead monoxide. However, lead monoxide has the property of partly evaporating at the surface of the glass melt so that there the PbO-content is considerably reduced. If such a glass, originating from the surface layer of the melt, forms part of the delay body, the good operation as a delay medium may be disturbed.

Possibilities are known, it is true, to restrict said evaporation of PbO. However, these requires special precautionary measures.

The invention provides a class of glasses of which the drawback of evaporation of one or more of the components with the resulting adverse influence on the acoustic properties of the glass is considerably smaller while the above-mentioned advantageous properties of the known glass are maintained therein.

According to the invention the acoustic delay line, the delay body of which consists of glass which contains the components SiO ₂ , K ₂ O and oxide of bivalent metal, is characterized in that the glass has the following composition in percent by weight:

SiO ₂ 50-75 K ₂ O + Na ₂ O 0-8 Na ₂ O ≤0.5 Sb ₂ O ₃ + As ₂ O ₃ ≤ 1.5 B ₂ O ₃ < 5 Al ₂ O ₃ < 15 PbO 0-10 CaO 0-20 BaO 0-40 MgO 0-10 ZnO 0-25 totally 20-50 CdO 0-35 SrO 0-30 Bi ₂ O ₃ 0-30

on the understanding, however, that the requirement is also satisfied, that -5 × 10 ^-6 <Σ _i α _i x _i < +5 × 10 ^-6 , where α _i is the factor for the temperature coefficient of the rate of propagation in the range of 20° to 70° C for the oxidic component i and x _i is the molar fraction in which said component is present in the glass.

During the experiments which led to the invention it was found that the temperature coefficient of the rate of propagation of acoustic shear waves is an additive quantity with respect to said quantity for the free oxidic components. In order that the temperature coefficient of the delay line be substantially zero, the above condition should be fulfilled. Within the above-mentioned range of compositions, only those glasses may be used as a delay medium in ultrasonic delay lines for the above-mentioned purposes in which the said condition is fulfilled without having to use additive ancillary means which have for their object to improve a delay line the temperature coefficient of which is not equal to zero, for example, by the combination with an electric transit time line the temperature coefficient of which is equal to but opposite to that of the glass delay line.

In the following Table I the values of the factors α _i are listed for the oxides to be considered.

TABLE I

Oxide i α _i + 10 ⁶ SiO ₂ - 100 B ₂ O ₃ - 90 Al ₂ O ₃ + 180 ZnO +165 PbO +285 CaO +340 BaO +350 MgO +325 CdO +210 Bi ₂ O ₃ + 350 SrO + 350 K ₂ O +300

as ₂ O ₃ and Sb ₂ O ₃ may be neglected in the calculation. The accuracy of the value of the temperature coefficient calculated by means of the formula is such that for glasses which have been cooled at a rate of approximately 1° C per minute from the highest annealing temperature or 50° C above said temperature said value does not differ from the experimentally determined value of the temperature coefficient more than ±5 × 10 ^-6 /° C over the temperature range of 20° - 70° C. With a desired greater accuracy a quantity of one or more components, starting from a previously chosen composition, may be varied until the desired value of the temperature coefficient has been reached. As a rule the desired value for glasses which are used as an acoustic medium will be equal to or substantially equal to zero but in some cases a value differing slightly from zero is desirable in order to obtain an optimum action of the delay line in a temperature range other than the said range of 20° to 70° C or to compensate for the temperature coefficient of the transducers and/or other components of the associated electric circuit. Alternatively, a different manner of cooling may result in a slightly differing value of the temperature coefficient.

The glasses according to the invention for the present use and a good stability, that is to say that the above-mentioned hysteresis effects do not occur to any inconvenient extent also after prolonged use.

Whereas for most of the known glasses the delay time τ in accordance with temperature has an approximately parabolic variation:

(Δτ)/τ = c ^. (T - T _o ) ²

in the temperature range in which │T-T _o │ ≤50° C and in which c is approximately +0.04 × 10 ^-6 /(° C) ² , the value of c for a large number of glasses according to the invention is only +0.02 × 10 ^-6 /(° C) ² , so that the constancy of the delay time as a function of the temperature for these glasses is still larger than for the known types of glass.

The rate of propagation of acoustic shear waves varies for the glasses with compositions within the range according to the invention from 2,800 to 3,500 m/sec. These values are somewhat higher than the above-mentioned known glasses (2,400-2,600 m/sec.) which means that for the same delay time a proportionally larger length of the acoustic beam is necessary. For delay lines having a small delay time of, for example, 64 μsec., however, that is no objection.

A preferred range of compositions is determined by the following limits (also in percent by weight).

SiO ₂ 60-70 K ₂ O+Na ₂ O 2-6 Na ₂ O ≤0.5 Sb ₂ O ₃ +As ₂ O ₃ ≤ 1.5 B ₂ O ₃ < 5 Al ₂ O ₃ < 15 PbO 0-5 CaO 0-10 BaO 0-25 MgO 0-5 together 25-38 i.e., the remainder not less than 25 ZnO 0-15 CdO 0-20 SrO 0-15 Bi ₂ O ₃ 0-20

a few examples of glass types which are used according to the invention as a delay medium in an acoustic delay line are the following which are stated in mol. percent and in wt. percent. Stated are the following properties: the average temperature coefficient TC = (Δτ)/(TΔT) in the temperature range of 20° - 70° C in 10 ^-6 per ° C, the variation (ΔTC) at 20° C of the temperature coefficient in 10 ^-6 per ° C after a cooling treatment in which the glass is heated from room temperature to 50° C above the annealing temperature of the glass and is then cooled to room temperature at a rate of 1 1/2° C per minute compared with that of the glass in which it is cooled at a rate of approximately 100° C per minute and the value of the constant c from the above formula in 10 ^-8 per (° C) ² . ------------------------------------------------------------ --------------- TABLE II

1 2 3 4 Mol Wt. Mol Wt. Mol Wt. Mol Wt. % % % % % % % % ____________________________________________________________ ______________ SiO ₂ 63.7 69.2 54.3 67.0 62.0 72.9 60.7 B ₂ O ₃ 3.0 2.7 3.0 3.2 Al ₂ O ₃ 5.0 6.7 5.0 7.9 K ₂ O2.5 3.4 2.5 3.1 2.5 3.6 2.5 3.3 PbO CaO 7.9 6.4 5.0 4.3 5.0 3.9 BaO 7.7 16.9 12.1 24.2 6.5 13.8 ZnO 7.7 9.0 8.0 8.5 12.3 15.3 7.9 8.9 MgO 5.0 3.1 CdO 5.0 8.9 As ₂ O ₃ 0.2 0.6 0.2 0.5 0.2 0.6 0.2 0.5 ____________________________________________________________ ______________ TC 0 ± 1 0 ± 1 0 ± 1 0 ± 1 ΔTC 4 3 6 6 c. 3 3 4 3 ____________________________________________________________ ______________ ____________________________________________________________ ______________ 5 6 7 8 Mol Wt. Mol Wt. Mol Wt. Mol Wt. % % % % % % % % ____________________________________________________________ ______________ SiO ₂ 53.9 73.3 58.5 70.1 60.7 72.6 62.5 B ₂ O ₅ 5.0 5.1 Al ₂ O ₃ 5.0 7.4 K ₂ O2.5 2.8 2.5 3.1 2.5 3.4 2.5 3.4 PbO CaO 5.0 3.3 5.0 4.0 7.0 5.6 BaO 5.5 9.9 11.0 22.4 7.2 15.9 7.0 15.4 ZnO 8.0 8.6 10.7 12.5 MgO 5.0 2.3 5.0 2.9 SrO 5.0 6.9 Bi ₂ O ₃ 5.0 27.3 As ₂ O ₃ 0.2 0.5 0.2 0.5 0.2 0.6 0.2 0.6 ____________________________________________________________ ______________ TC 0 ± 1 0 ± 1 0 ± 1 0 ± 1 ΔTC 6 3 3 5 c. 2 3 2 3 ____________________________________________________________ ______________

PHILIPS 26C568 CHASSIS K11 POWER SUPPLY:


























































































Control circuit for a switched-mode power supply, particularly for a television receiver: PHILIPS CHASSIS K11 SWITCH MODE POWER SUPPLY.

The Control UNIT 4822 212 20077 is developed around the PHILIPS TDA2640.
"Television Switched-Mode Power Supply Using the TDA2640", Mullard Technical Communications, L. M. White, pp. 258-279, Jul. 1975.

A switched-mode power supply provided with a control stage and a switching stage coupled by means of a transformer. The collector of an additional transistor is connected to the transformer. In this manner the ratio of the collector current to the base current of the switching transistor can assume a predetermined value, for example a constant value whatever the value of the mains voltage applied to the power supply.

What is claimed is:

1. A control circuit for a switched-mode power supply, said power supply comprising a non-regulated rectified DC voltage source, a driver transistor, a first transformer having primary and secondary windings, an end of said primary being coupled to the collector-emitter path of said driver transistor, a switching transistor having a base coupled to said secondary, a second transformer having a primary winding coupled in series with said switching transistor, and a plurality of secondary windings, said control circuit comprising a first additional transistor having a collector coupled to the remaining end of the primary winding of the first transformer not connected to the driver-transistor and an emitter coupled to the non-regulated direct voltage source.

2. A control circuit as claimed in claim 1, further comprising a constant voltage source coupled to the base of the additional transistor.

3. A control circuit as claimed in claim 1, further comprising a constant current source, and a resistor coupled between the emitter of the additional transistor and the constant current source.

4. A control circuit as claimed in claim 3, wherein the constant current source comprises a second additional transistor, the two additional transistors being of complementary conductivity and their emitters being connected with each other through said resistor, the collector of the second additional transistor being coupled to the non-regulated rectified direct voltage source and the collector of the first additional transistor being coupled to the end of the primary winding of the first transformer not connected to the driver transistor.

5. A control circuit as claimed in claim 4, further comprising a resistor coupled in series with the collector circuit of said second additional transistor and the non-regulated rectified direct voltage source.

6. A control circuit as claimed in claim 5, further comprising a zener diode coupled between the base of the second additional transistor and the non-regulated voltage source.

7. A control circuit as claimed in claim 6, further comprising a resistance bridge coupled to the base of the first additional transistor and arranged between the two electrodes of the zener diode.

8. A control circuit as claimed in claim 7, wherein the driver transistor and the switching transistor do not conduct simultaneously, and the voltage between the two electrodes of the zener diode as well as the values of the resistors arranged between the said electrodes and of the resistor arranged between the emitters of the two additional transistors are chosen so that the first additional transistor is in the saturated state at the lowest value of the non-regulated voltage while it operates in the linear state at a higher value of said non-regulated voltage.

Description:

The present invention relates to a control circuit for a switched-mode power supply, particularly in a television receiver, said power supply comprising a rectified, non-regulated rectified DC voltage source, a driver transistor whose collector-emitter path is arranged in series with a primary winding of a first transformer, a secondary winding of the latter being coupled to the base of a switching transistor which is arranged in series with a primary winding of a second transformer having a plurality of secondary windings.
This type of switched-mode power supply is used more and more because of the numerous advantages it presents as regards energy efficiency, reliability, compactness, etc. However, as for the majority of the other types of power supplies, its operation on mains supplies of different voltages imposes the use of either a transformer with taps or switch-over from full wave rectification at the highest mains voltage to a voltage doubler rectification for the lowest mains voltage.
It is known that the specific qualities of a switched-mode power supply depend for a large part on the switching speed of the switching transistor at the moment at which the latter passes periodically from the conductive state to the blocking state; this speed is at its maximum when the switching transistor presents, at the turn-off moment, a certain ratio between the collector current and the base current IC/IB: if this ratio is too low, the delay in the recombination of the charges stored in the base increases the switching time; if it is too high there is the risk that the transistor is brought out of saturation before it is blocked, which results in its substantially immediate destruction. For the known switched-mode power supplies it is not possible to maintain a suitable IC/IB ratio in the presence of large variations of the non-regulated rectified DC voltage which result from the connection to the nominal mains voltages of, for example, 110 or 220 V; actually, if the variations in IB are substantially proportional to the variations in the non-regulated voltage, the same does not happen for those of the IC whose amplitude is less.
However, the importance of having a power supply which can operate without any switching on mains supplies of 110 or 220 V is evident: for the manufacturer it is cheaper to produce and the reliability is increased; while the user does not run the risk of incorrect manipulations, particularly when the power supply is destined for use in portable television sets.
One of the objects of the invention is to realize a control circuit which permits the switched-mode power supply to operate without switching in conditions which are substantially optimum and in the presence of mains voltage variations in the range of 90 to 250 Volts.
A further object of the invention is to ensure that said IC/IB ratio of the switching transistor has a predetermined and, more particularly a constant value at the turn-off moment whatever the value of the mains voltage applied to the power supply.
The control circuit according to the invention is characterized in that the end of the primary winding of the first transformer not connected to the driver transistor is connected to the collector of an additional transistor whose emitter is coupled with the non-regulated direct voltage source. Advantageously it is characterized in that the emitter of the additional transistor is connected to one end of a resistor, the other end of this resistor being connected to a constant current source, and that the constant current source is constituted by a second additional transistor, the two additional transistors being of complementary conductivity and their emitters being connected with each other through a resistor, whilst the collector of the second additional transistor is connected to one of the poles of the non-regulated rectified direct voltage source and the collector of the first additional transistor is connected to the end of the primary winding of the first transformer not connected to the driver transistor.
Whilst combining the action of a ballast transistor with that of a variable current generator, the circuit according to the invention thus maintains automatically a desired IC/IB ratio of the switching transistor whatever the value of the mains voltage applied to the power supply.





List of sets known to have the K11 chassis (made from approximately 1975-1978)

= means that models are most likely the same or very similar, but the styling can be

different in some cases. Information was amongst others taken from the Philips model

number survey 2003, 3122 785 14570.

A side note for those who have noticed the K10 chassis is missing from the line up. Rumour has, that this was a K9 variant with another tube, probably Trinitron, that didn’t make it beyond the prototype stage. Instead, Philips decided to use the 20AX tube and named the chassis K11. This chassis was designated K9i in some countries, most notable Germany. The differences between the K9 and K11 chassis were probably thought of as minor as the K11 chassis was basically an improved version of the K9 chassis with some minor (evolutionary) updates, another tube and as a result less complicated convergence circuits.

General models
22C545
22C549
26C364
26C466
26C555
26C556
26C557
26C560
26C561
26C564
26C565
26C566
26C567
26C568
26C569
26C655
26C657
26C663
26C667
26C677
26C750
26C752
26C753
26C762
26C764
26C768
26C770
26C782
26C840

Germany
Factory location Krefeld (KR)

It seems very strange that only one German model is mentioned. Quite possibly the person who compiled the official Philips model number survey got confused by the K9i nomenclature. As a result of that, the D26C865 mentioned in the K9 overview might actually be a K11 set. Other German K11 sets probably exist.

D26C662
D26C865??

Sweden
Factory location Norrköping (NF)
SK22C462
SK26C464
SK26C466
SK26C467
SK26C468
SK26C476
SK26C477
SK26C478
SK26C764
SK26C765
SK26C773
SK26C776
SK26C777
SK26C778
SK26C865

South Africa
Factory location Martinsville
V26k606
V26K609

Other brands (Erres, possibly Schneider (F), ..)
Erres branded sets mostly used the prefix RS
The suffix KSK instead of K might indicate a Swedish model. I haven’t actually seen it on a set in person.

22264KSK
22545K = 22C545
26555K
26557K
26565K
26566K
26568K
26655K
26756K
26764KSK
26768K
26965KSK
26966KSK
263637K
263737K

Other Brands
As a rule, the model number below is prefixed by letters indicating the brand name as

follows (not all brands may be used, others may exist):

AR = Aristona
SA = Siera
RA = Radiola
DX = Dux
CT = Conserton?

The infix KSK instead of K might indicate a Swedish model. I haven’t actually seen it on a set in person.

56KSK264
56K545 = 22C545
56K549 = 22C549
26K0624 (?)
66KSK364
66KSK365
66KSK366
66KSK375
66KSK376
66K466
66K555
66K557
66K565
66K566
66K568
66K655
66K756
66KSK764
66K768
66K4627
66K4727
66K5520
66K5522
66K5624

PROGRESSIVE BY YEAR LIST OF COLOR TELEVISION SETS WITH PHILIPS CHASSIS K11 20AX CRT TUBE.
26C466 CHASSIS K11 YEAR 1974
26565K CHASSIS K11 YEAR 1975
26566K CHASSIS K11 YEAR 1975
26C567 CHASSIS K11 YEAR 1975
66K565 CHASSIS K11 YEAR 1975
66K566 CHASSIS K11 YEAR 1975
22264KSK CHASSIS K11 YEAR 1976
22545K CHASSIS K11 YEAR 1976
22C545 CHASSIS K11 YEAR 1976
22C549 CHASSIS K11 YEAR 1976
26555K CHASSIS K11 YEAR 1976
26557K CHASSIS K11 YEAR 1976
26568K CHASSIS K11 YEAR 1976
26655K CHASSIS K11 YEAR 1976
26756K CHASSIS K11 YEAR 1976
26764KSK CHASSIS K11 YEAR 1976
26966KSK CHASSIS K11 YEAR 1976
26C555 CHASSIS K11 YEAR 1976
26C557 CHASSIS K11 YEAR 1976
26C565 CHASSIS K11 YEAR 1976
26C566 CHASSIS K11 YEAR 1976
26C568 CHASSIS K11 YEAR 1976
26C569 CHASSIS K11 YEAR 1976
26C655 CHASSIS K11 YEAR 1976
56K545 CHASSIS K11 YEAR 1976
56K549 CHASSIS K11 YEAR 1976
56KSK264 CHASSIS K11 YEAR 1976
66K555 CHASSIS K11 YEAR 1976
66K557 CHASSIS K11 YEAR 1976
66K568 CHASSIS K11 YEAR 1976
66K655 CHASSIS K11 YEAR 1976
66K756 CHASSIS K11 YEAR 1976
66KSK365 CHASSIS K11 YEAR 1976
66KSK366 CHASSIS K11 YEAR 1976
SK22C462 CHASSIS K11 YEAR 1976
SK26C464 CHASSIS K11 YEAR 1976
SK26C466 CHASSIS K11 YEAR 1976
SK26C467 CHASSIS K11 YEAR 1976
SK26C765 CHASSIS K11 YEAR 1976
sk26c865 CHASSIS K11 YEAR 1976
V26K606 CHASSIS K11 YEAR 1976
V26K609 CHASSIS K11 YEAR 1976
263637K CHASSIS K11 YEAR 1977
26768K CHASSIS K11 YEAR 1977
26C364 CHASSIS K11 YEAR 1977
26C556 CHASSIS K11 YEAR 1977
26C564 CHASSIS K11 YEAR 1977
26C657 CHASSIS K11 YEAR 1977
26C667 CHASSIS K11 YEAR 1977
26C677 CHASSIS K11 YEAR 1977
26C762 CHASSIS K11 YEAR 1977
26C764 CHASSIS K11 YEAR 1977
26C768 CHASSIS K11 YEAR 1977
26C770 CHASSIS K11 YEAR 1977
26C782 CHASSIS K11 YEAR 1977
56K0624 CHASSIS K11 YEAR 1977
66K4627 CHASSIS K11 YEAR 1977
66K5624 CHASSIS K11-TRIPLER YEAR 1977
66K768 CHASSIS K11 YEAR 1977
66KSK375 CHASSIS K11 YEAR 1977
66KSK376 CHASSIS K11 YEAR 1977
66KSK764 CHASSIS K11 YEAR 1977
SK26C468 CHASSIS K11 YEAR 1977
SK26C476 CHASSIS K11 YEAR 1977
SK26C478 CHASSIS K11 YEAR 1977
SK26C764 CHASSIS K11 YEAR 1977
SK26C773 CHASSIS K11 YEAR 1977
SK26C776 CHASSIS K11 YEAR 1977
SK26C777 CHASSIS K11 YEAR 1977
263737K CHASSIS K11 YEAR 1978
26965KSK CHASSIS K11 YEAR 1978
26C560 CHASSIS K11 YEAR 1978
26C561 CHASSIS K11 YEAR 1978
26C663 CHASSIS K11 YEAR 1978
26C750 CHASSIS K11 YEAR 1978
26C752 CHASSIS K11 YEAR 1978
26C753 CHASSIS K11 YEAR 1978
26C840 CHASSIS K11 YEAR 1978
66K466 CHASSIS K11 YEAR 1978
66K4727 CHASSIS K11 YEAR 1978
66K5520 CHASSIS K11 YEAR 1978
66K5522 CHASSIS K11 YEAR 1978
66KSK364 CHASSIS K11 YEAR 1978
D26C662 CHASSIS K11 YEAR 1978
SK26C477 CHASSIS K11 YEAR 1978
SK26C778 CHASSIS K11 YEAR 1978