NORDMENDE SPECTRA COLOR L2UT YEAR 1972.

The NORDMENDE SPECTRA COLOR L2UT  is a Color television wood console with 26 inches (66cm) color screen and 8 programs preselection mechanical push button keyboard / Nixie fluorescent display, first time ultrasonic remote control  and potentiometric tuning search system.

The mechanical turret approach to television tuning has been used almost exclusively for the past 60 years. Even though replete with the inherent disadvantages of mechanical complexity, unreliability and cost, such apparatus has been technically capable of performing its intended function and as a result the consumer has had to bear the burdens associated with the device. However, with the " recent "  Broadcast demands for parity of tuning for UHF and VHF channels, the increasing number of UHF and cable TV stations have imposed new tuning performance requirements which severely tax the capability of the mechanical turret tuner. Consequently, attempts are now being made to provide all electronic tuning to meet the new requirements.

 One " " new " " tuning system currently being incorporated in some television receivers uses a varactor tuner which overcomes some of the disadvantages of mechanical turret tuner by accomplishing tuning electronically. As the name indicates, the heart of such a tuner is a varactor diode which is used as a capacitive tuning element in the RF and local oscillator sections. In this system, channel selection is made by applying a given reverse bias voltage to the varactor to change its electrical capacitance. The channel selection biasing can be performed by mechanically or electrically switching approximately 5 or many more preset potentiometers. The problem with such arrangement is that it quite seriously limits the number of channels available to the consumer. Additionally, it suffers from the drawback that all potentiometers require adjusting for the desired channels. The VHF channels are usually factory adjusted while the six UHF channels require on-location adjustment. Moreover, using this arrangement, the only indication--during adjustment--of which channel is selected is by station identification.

This invention relates to a wireless remote control system which is particularly useful for the control of television receivers.
Where such apparatus as television receivers are to be controlled from a viewer location as to channel, volume, brightness, etc., the remote control systems usually are made up of a hand held transmitter which transmits an ultrasonic signal to a receiver connected to or built within the television receiver. The depression of buttons on the transmitter causes a variety of signals or signal frequencies to be transmitted, whereby channel change, volume change, etc. is responsively obtained.
However such systems have individually suffered from one or more problems, such as inability to have direct access to the desired channel, slow access, insufficient noise immunity making it often possible to operate the system with the jingling of a key chain or an ultrasonic sound originating from a dishwasher etc., unreliable control due to the absence of means to detect and suppress transmission errors resulting from echoes, interfering signal sources, etc. Also some control systems are not suitable for continuous analog commands such as volume, brightness, etc. Existing systems also often require the need for bandpass filters and accurate crystal oscillators which make them costly. Many systems are not very suitable for integration into custom integrated circuits. To obtain the simplest possible transmitter construction in ultrasonic remote control, modulation of the emitted ultrasonic frequencies is not employed; to control different operations different frequencies are emitted which must be recognized in the receiver and evaluated for carrying out the different functions associated therewith. Presently, to recognize the different frequencies, use is made of resonant circuits, each of which contains one or more coils tuned in each case together with a capacitor to one of the useful frequencies.

These hitherto known receivers have numerous disadvantages. Thus, for example, before starting operation of the receiver a time-consuming alignment procedure must be carried out with which the resonant frequencies of the individual resonant circuits are set. Since it is inevitable that with time the resonant circuits become detuned, it may be necessary to repeat the alignment procedure.

A further disadvantage is that the known receivers cannot be made by integrated techniques because the coils used therein are not suitable for such techniques.

The potentiometers knobs are hidden under the programs push knobs.

The "Nixie Tube" is a tube which utilizes a transparent envelope that contains an anode electrode and a plurality of cathode glow indicator electrodes aligned in a stack one above the other. Such tubes require substantial thickness because the electrode indicator units are placed one above the other and a number of electrodes are used and are relatively expensive to manufacture. Such tubes are difficult to manufacture and are illegible unless the observer is directly in front of the indicator tube.

In the end of the 60's  increasingly attention was focused on the varicap diode tuner as the latest, sophisticated means of television receiver frontend tuning in both colour and black and white sets.
 The main purpose of this article is to investigate the servicing problems associated with this comparatively new method of tuning.

First however let's briefly recap on the principles involved in this tuning system:

 The tuners use variable capacitance (or "varicap") diodes as the variable tuning elements: the effective capacitance of the diodes is controlled by the reverse bias applied across them, tuning being achieved by varying this voltage. As the reverse bias across a varicap diode is increased so its junction depletion region widens thus reducing its capacitance.
A VHF/ UHF television tuner is constructed in accordance with the present invention includes a preselector tuned circuit having a solid state voltage controlled capacitor as its tunable element, a radio frequency amplifier coupled to the preselector circuit and alsoother circuit to perfect the signal receiving capability and the application the like.

Considering the Mechanical Tuner Problems:

To get the servicing problems in perspective let us next consider the tuning arrangements previously used.
 The earliest of these, employed on v.h.f., was the switched tuner which was either of the turret or incremental type.
 The turret tuner substituted a coil bearing "biscuit" mounted on the rotating drum or turret when channels were changed. Twelve positions were normally provided, with a fine tuning knob to adjust the local oscillator frequency. As its name suggests the incremental tuner simply added more inductance to the tuned circuits at every downward channel movement: thus the highest inductance was present on channel one and the least on channel 12 (which normally covered 13 as well with manipulation of the fine tuner).
The movement towards u.h.f. TV working, initially with dual standard sets and later with single standard ones, brought about the need for u.h.f. tuners. In the earliest u.h.f. receivers valve tuners which were not particularly efficient were used.

The drive mechanism was usually a dual  speed rotary system calibrated from channels 21 to 68. Experience in the field indicated that 625 line television was in many cases considered by the viewer to be inferior to 405 -line reception, on account of the poor signal to noise ratio achieved by the valve tuners. Many viewers were not prepared to use external u.h.f. aerials of course, having achieved satisfactory reception on v.h.f. with an indoor aerial: this aggravated the situation even more.
Another aspect which caused difficulty was the care needed to tune in a u.h.f. channel using a rotary tuner covering the whole of Bands IV and V. Many viewers simply could not tune in BBC 2  or ZDF or ORF or any channel correctly with such a tuning mechanism, finding that they had passed right over the channel they wanted before realising what they had done.
The advent of transistor tuners rapidly improved the quality of u.h.f. reception but use of a rotary mechanism was continued by many manufacturers. Thus while potential reception was improved the same tuning difficulties remained and viewers continued to gravitate towards 405 line viewing using the "old faithful" switched tuner. The operational breakthrough came with the introduction of the push-button u.h.f. channel change. 

The mechanism is basically simple. Adjustable push buttons press down on a lever bar which in turn rotates the tuner's variable capacitors to the appropriate position. Each button is capable of tuning over the entire u.h.f. bands and this leads to customer confusion at times when after some adjustments which were too heavy handed they find themselves receiving ITV on a BBC button or a ORF and ZDF broadcasting or any channel possible !

Mechanical Faults:

 Mechanical tuning obviously has its snags. There are for example contact springs which earth the tuning capacitor and go intermittent. This gives rise to the most random tuning defects, capable of driving the. most patient viewer to a state of total exasperation. It is also possible for the rotation mechanism to hang up and jam intermittently, or just become sticky, so that the reset accuracy of the mechanism is impaired and the receiver has to be retuned every time the channel is changed.
The vanes in the tuning capacitor can also short out at different settings, thereby eliminating some channels. The  Varicap Tuner It will be seen then that mechanical defects can cause very irritating fault symptoms. If one thinks along the lines that anything mechanical is nasty, then the elimination of mechanical parts can only be to the good.

The logic of this is splendid provided the electronic replacement for the mechanical system is more reliable! Otherwise we are leaping out of the frying pan into the fire! In the light of experience gained with mechanical tuning devices it seems great that with the varicap tuner we have at last dispensed with the dreaded rotary tuning capacitor, replacing it instead with a variable voltage to the tuner. 
Let us think about this however since things are never quite as simple as they first appear. The tuning voltage has to be variable in order to tune the receiver. Obviously then a means of varying the voltage has to be provided to act as the tuning control.
As it is a voltage that has to be varied the tuning control takes the form of a potentiometer., Now we have returned to a mechanical system again, though in a less complex form.
A potentiometer is required for each channel, selected by pressing the appropriate channel button.

We have lost a tuning capacitor and its rotating mechanism and gained a set of pots and selector switches therefore. Provided the pots and switches are mechanically more reliable than the tuning capacitor we should be better off-or should we? 

Need for Voltage Stabilisation.
 The voltage selected by the pots cannot be allowed to drift otherwise the receiver will go off -tune. The voltage supply to the potentiometers has to be stabilised therefore and a stabilising zener diode or integrated circuit (TAA550) .is needed for this purpose.

Any failure in this part of the circuit will give rise to tuning drift or worse, a total loss of reception. A short-circuit TAA550 for example will completely remove the tuning voltage while if it is open circuit the tuning can vary with picture brightness. Likewise any intermittency in the potentiometers or associated switching and/or resistors can also cause problems.

Relative Reliability of Tuners:

 It  will be seen then that in order to lose our troublesome mechanical arrangement we have had to introduce considerably more electronics which we trust are going to be more reliable. In addition we have not so far considered the relative reliability of the varicap tuner itself compared with the mechanical type. Since two r.f. transistors are generally used to compensate for the reduced Q of the varicap tuned circuits we immediately have twice the likelihood of an r.f. stage breaking down! 
And being semiconductors the varicap diodes themselves are more likely to fail than the sections of a ganged tuning capacitor. It is reasonable then to conclude that if mechanical faults are the most prevalent the use of varicap tuners will make life easier. Mechanical faults are generally not too difficult to sort out however and the field engineer can often cope with them in the home. 
Can the same be said of the varicap tuner? It seems that this type of tuner does not need so much attention as its mechanical counterpart but is likely to throw up some much more difficult faults when it does, resulting in bench repairs being needed. So far my own experience has indicated that varicap tuning faults nearly always need servicing on the bench.
Generally speaking it seems true to say that varicap tuners themselves are adequately reliable: the snags result from the tuning system and stabilised power supply.

Tuning Drift with Varicap Tuners:

 If a varicap tuned receiver is constantly drifting off tune the +30V supply should be the number one suspect. It is best to connect an Avometer permanently to the supply so that it can be precisely monitored-if necessary write down the exact voltage measured.
 If the receiver drifts, check the voltage. If it has changed, even slightly, this may well be enough to be the cause of the fault. To pinpoint and confirm the diagnosis aerosol freezer should be applied to the stabiliser i.c. or zener. If the voltage returns to normal or changes wildly for the worse the stabiliser is almost certainly the cause of the trouble and should be replaced.
A prolonged soak test should then be carried out. Another point concerning varicap tuners arises with their use in colour receivers.

 There were  makers of the most expensive colour receiver on the market still didn't use a varicap tuner but instead use a mechanical one. The makers' claim is that the signal to-noise ratio of the varicap tuner is inadequate for their colour standards. Undoubtedly the results obtained on the receiver seem to confirm this. Interestingly, the same manufacturers use varicap tuners in their black -and -white receivers, and the tuning button system is often full of troublesome intermittent contacts. The varicap tuner has its advantages and disadvantages then. Probably the simplest comment would be to say that when it is good it is very very good but when it is bad it is horrid!

The NORDMENDE SPECTRA COLOR L2UT was first NORDMENDE COLOR TV sporting a 110 degrees delta CRT. More deflection power was needed and convergence circuitry was mostly active using transistors instead of passive as before. 

The chassis of the NORDMENDE SPECTRA COLOR L2UT  has still 4 tubes and is a hybrid type with many complex and sophisticated discretes components based signal circuits.NO IC's are featured, all parts are exclusively discretes based.

 In 1972 came the 110° tube. As yet it was still in the form of a delta -gun thick -neck device, needing prodigious deflection power and elaborate high -power convergence and raster -correction arrangements. Some European setmakers introduced 110° models immediately, others sat on the fence, as did the public and the dealers - not surprising, in view of the considerable complexity of the sets, which often produced a picture no better and in some respects (like purity, focus and convergence) sometimes markedly poorer than their contemporary 90° counterparts. All this for 3 in. or so off the cabinet depth! Some customers did not even realise at the time of purchase that they had bought or rented a 110° set; certainly the sales staff did not make it a big selling point - perhaps they were unaware too! The arrival of 110° deflection did not make a big impact in my part of the world. Probably the most elaborate wide angle set was the 3400 model from Bang and Olufsen. This receiver boasted ten valves, 104 transistors and 89 diodes, with two PL509s providing line deflection/e.h.t. Other features were dynamic (line -rate) focusing and a separate class B push-pull output stage for each set of horizontal dynamic convergence coils.

The chassis of the NORDMENDE SPECTRA COLOR L2UT  was even featuring first time a EHT Multiplier, The invention relates to a circuit arrangement for supplying EHT to the acceleration anode of a picture display tube in a picture display apparatus, comprising a line output transformer, an EHT winding wound on the core of the transformer, a multiplier circuit constituted by capacitors and rectifiers, which multiplier circuit is connected to a terminal of the EHT winding, a diode and a parallel arrangement of a resistor and a capacitor being connected to terminals of the EHT winding.  

It has a  picture tube apparatus employing the so-called delta gun  shadow mask  The present invention relates to a color television picture tube  type color cathode-ray tube which is provided with three electron guns positioned respectively at the vertices of an equilateral triangle, " the so-called delta gun  shadow mask ".

To obtain a fine color picture on the screen of the cathode-ray tube of this type, the following requirements should be satisfied that the electron beam is emitted from each electron gun onto the center of each corresponding phosphor dot on the screen of the cathode-ray tube, the purity of colors is high and the electron beams are converged onto a group of phosphor dots.
These requirements can be comparatively easily satisfied in the case of the cathode-ray tube with a spherically formed screen, whereas it is difficult to satisfy the requirements in the case of a narrow-necked, wide-angle deflection cathode-ray tube. The wide angle deflection cathode-ray tube is advantageous in practical use because the distance between the electron guns and the screen is small and the screen is almost flat with a curvature approximate to that of a flat surface; however, it is necessary to control the electron beams so that the electron beams may be emitted exactly onto the 3-color phosphor dots on the screen because the incident angle and distance of the electron beams which reach the phosphor dots on the screen have the values proper to respective phospher dots.

The total power consumption is 350 watts due to the higher deflection currents in comparison with 90° deflection colour tv sets. The set, which was sold in Germany, has such a high total power consumption because of large amount of circuits.

Some characteristics:

Nordmende Spectra-Color L2UT 4.570.D/telec.;146 Dioden, 2 Thyristoren, Hochspannungskaskade, Programmanzeige mit Leuchtziffernröhre, ACS - Buchse, Kopfhörer-TB oder HiFi Anlage, VHF/UHF-Tuner nach CCIR B/G, Fernbedienung Telecontrol 1.

 Net weight (2.2 lb = 1 kg) 66.1 kg / 145 lb 9.5 oz (145.595 lb) 

(To see the Internal Chassis Just click on Older Post Button on bottom page, that's simple !)



Nordmende was a manufacturer of entertainment electronics based in Bremen, Germany.
The original company, Radio H. Mende & Co, was founded in 1923 by Otto Hermann Mende (1885-1940) in Dresden. Following the destruction of the plant during the bombing raids in 1945, Martin Mende (the founder's son) created a new company in Bremen in 1947, in a former Focke-Wulf plant, under the name North German Mende Broadcast GmbH. The name was subsequently changed to Nordmende: subsequently the company became one of the prominent German manufacturers of radios, televisions, tape recorders and record players in the 1950s and 1960s.

In the 1970s, Nordmende televisions were renowned for their innovative chassis, and for the rigorous testing and quality control of their finished products. Both created high costs, however, which soon proved a competitive disadvantage when the price of colour televisions began to plunge.

In 1969, Mende's sons took over the company, and in 1977 a majority shareholding was sold to the French Thomson Brandt company and the chassis remains the original NordMende until CHASSIS F9. The following year, the family sold their remaining shares to Thomson. In the 1980s, the factories in Bremen were closed, Nordmende becoming purely a Thomson trademark (Starting from chassis F10 F11 they're all THOMSON).








In the 1990s, the name Nordmende was used with decreasing frequency, and it eventually disappeared in favour of the Thomson name. In 2005 Videocon Group acquired all cathode ray tube activities from Thomson. This led to the creation of VDC Technologies, which manufactures TV sets using the Nordmende brand under licence from Thomson.
The Nordmende brand name was relaunched in Ireland in September 2008 by the KAL Group. Although Nordmende was well known for its televisions throughout Ireland during the 1970s and 1980s, the company bought the rights to the name and launched a range of white goods including fridges, freezers, washing machines, and dishwashers, alongside a revamped range of flat-screen TVs and stereos.

     
NORDMENDE HISTORY IN GERMAN:

Die Vorkriegsgeschichte findet sich unter Mende. Nach dem Totalverlust in Dresden gründet Martin Mende (30.12.1898-1982) unter Mitwirkung von Hermann Weber am 26. August 1947 [FT5901] in Bremen-Hemelingen die Norddeutsche Mende-Rundfunk GmbH.

Die ersten Gehäuse liefert ein Tischler in Achim gegen Kompensation von fünf Gehäusen zu einem Rundfunkgerät. Der frühere Mende-Konstrukteur, Obering. Heer zeichnet wieder für die Geräte verantwortlich [FT49??].

Ab 27. Juli 1948 liefert die neue, zuerst 18 und bald 60 Personen umfassende Firma auf Grund von Währungsreform, Krediten und Zulieferverträgen die neue Radioproduktion.

Das Regime in Ostdeutschland lässt den Namen Mende nicht zu, so dass Martin Mende mit grafischen Konstruktionen im Zusammenhang mit «Nord» an seinen Vorkriegserfolg anschliesst.

Die Hallen der ehemaligen Focke-Wulf AG beim Bahnhof Seebaldsbrück dienen als Werkstätten. 1950 beschäftigt das Unternehmen 700, 1959 schon 3500 und im Zenit 6300 Personen.

1950 beginnt die Firma mit UKW-, 1953 mit Fernseh- und 1954 mit Mess- und Prüfgeräten. Gegen Ende der 50er Jahre heisst die Firma Norddeutsche Mende Rundfunk KG [RP7901].

Nachdem sich Nordmende bislang nicht mit Magnettongeräten befasst hat, bringt das Werk 1958 das erste deutsche Heim-Tonbandgerät mit drei Motoren auf den Markt. Allerdings dominieren auf diesem Sektor eindeutig andere Firmen wie AEG/Telefunken und Grundig. Von Nordmende kommen jeweils nur ein bis zwei Geräte (1960 keines) in die Kataloge. Dafür hat die Firma Erfolg mit einem anderen Neueinstieg:


1958 stellt Nordmende mit «Mambo» ihr erstes Reisegerät vor - aber nicht «das erste deutsche, serienmässig hergestellte und volltransistorisierte Koffergerät», wie man aus einer Quelle nachlesen kann. Danach wird Nordmende in Deutschland auf dem Sektor Reisegeräte besonders stark, obwohl sie keine Röhren-Koffer baute. Immerhin kosten die in «Mambo» verwendeten 8 Halbleiter dann im Einzelhandel DM 98.70, während für die vier D-Röhren der 90er-Serie - auch zum Katalogpreis - etwa DM 35.- auszugeben wären. Preis des ganzen Gerätes: DM 189.- plus zwei Flachbatterien von 4,5 V.

Bis 1969 gibt es ca. 92 Modelle der tragbaren Radios (Koffer- bzw. «Handradios», d.h. «Hand held radios»). Beispielsweise finden sich im Katalog 1961/62 [448] je 11 Tischradios und Radiomöbel sowie 8 Modelle von Reiseradios. 17 verschiedene Fernsehmodelle zeigen dagegen, wo in jener Zeit der Erfolg zu holen war.

Gemäss [FT7901] liegt Nordmende während kurzer Zeit mit der sogenannten «Tippomatik-Bedienung» sogar technisch vorne. Siehe auch Philips etc.

Auch Konzertschränke scheinen Ende der 50er bis Anfang 60er Jahre eine tragende Säule für Nordmende zu sein. Dabei verwendet die Firma immer wieder gleiche Namen wie «Cabinet», «Caruso», «Casino», «Cosima» sowie «Arabella» und «Isobella» mit wechselnden Zusatz-Nummern oder den Zusatz «Stereo», z.B. in den Jahren 1959 und 1960/61.

Im März 1967 nimmt das Werk die Produktion von Farbfernsehgeräten auf. Zum Firmenjubiläum erscheint eine Gerätereihe mit der Bezeichnung 'Goldene 20'. 1969 übernehmen die Mende-Söhne Karl und Hermann die Geschäftsführung.

1977 führt der verschärfte Wettbewerb zum Verkauf der Mehrheit an den französischen Konzern Thomson-Brandt; die Familie Mende zieht sich anschliessend ganz aus dem Unternehmen zurück. Martin Mende stirbt 1982.

Weblinks:

 Commons: Nordmende – Sammlung von Bildern, Videos und Audiodateien
Tote Marke NORDMENDE – Verblasster Stolz. Artikel im Manager-Magazin
Ein neues Programm. Artikel in der Zeit
Fernseher: Inder produzieren neue Nordmende. Artikel bei itespresso.de
Videocon produziert Plasmaschirme für Nordmende. Artikel im pressetext.de

Einzelnachweise

Spectra Color Studio und Spectra SK2 Color de Luxe Studio auf radiomuseum.org
Marke Nordmende mit Digital- und Internetradios zurück, teltarif.de, Artikel vom 2. September 2017.


Company profile. Phillar, archiviert vom Original am 6. März 2008; abgerufen am 26. April 2013 (englisch).


Broschüre – Sprachmanager24. Abgerufen am 5. Januar 2015.

NORDMENDE SPECTRA COLOR L2UT CHASSIS FFS 772.535.A 772.536.A (4.570.D) INTERNAL VIEW.

The  NORDMENDE SPECTRA COLOR L2UT CHASSIS FFS 772.535.A 772.536.A (4.570.D) was first tv color chassis from NORDMENDE sporting a 110° degree A66-140X CRT TUBE. 

It's an hybrifd chassis with semiconductors and tubes.

The Tubes are used in Line deflection and frame deflection and horizontal oscillator too.

All other part are based on sophisticated - complex discretes circuits.

The chassis features ultrasonic remote control.

Was first NORDMENDE TV COLOR CHASSIS featuring a EHT Multiplier unit.

NORDMENDE SPECTRA COLOR L2UT CHASSIS FFS 772.535.A 772.536.A (4.570.D) power supply CONSTANT-VOLTAGE CONVERTER EMPLOYING THYRISTOR:

A constant voltage converter having a rectifier for rectifying AC power and with a thyristor connected between the rectifier and a filter for selectively passing therethrough a rectified output to an output terminal. There is a wave generator connected to the output of the rectifier for producing a first signal and an intergrator circuit connected to the output of the wave generator for producing an integral output in response to this first signal. In addition there is a detector circuit for detecting a fluctuation of the rectified output power and for producing second signal. A comparison circuit is connected between the intergrator circuit and the detector circuit for producing third signal in accordance with the comparison. A trigger circuit is connected between the comparison circuit and the control gate of the thyristor for supplying a phase control signal to the thyristor to thereby obtain a constant voltage output regardless of the fluctuation of the rectified output.

1. A constant voltage converter comprising an input of a power supply means, an output terminal, filter means, rectifier means connected to said input for rectifying a.c. power and for supplying output thereof to said output terminal, thyristor means connected between said rectifier means and said filter means for selectively passing therethrough a rectified output to the output terminal by way of said filter means, saw-tooth wave generator means connected between the output of said rectifier means and at least one integrator circuit means for producing an integral output in response to a saw-tooth wave produced, a first transistor in said saw-tooth wave generator, the input of said integrator circuit means being connected to a collector of said first transistor, detector circuit means connected to said output terminal for detecting a fluctuation of the rectified output power and for producing an output signal, said detector circuit means having a second transistor, pulse generator circuit means connected between said saw-tooth wave generator means and said detector circuit means for producing a trigger pulse to said thyristor through a trigger means, a third transistor in said pulse circuit generator means, the base of said third transistor being connected to the output of said integrator circuit means, the emitter thereof being connected to the emitter of said second transistor in said detector circuit means, and the collector thereof being connected to the gate of the thyristor means so as to supply a phase control signal thereto, thereby obtaining a constant voltage output regardless of the fluctuation of the rectified output.

Description:

This invention relates to constant-voltage converters and more particularly to a constant-voltage converter employing a thyristor.

Conventional constant-voltage converters of the type employing a thyristor are arranged to phase shift and full-wave-rectify an input a.c. power applied thereto and to maintain the output voltages constant by regulating the firing angle of the thyristor in comparison of the output voltages with the phase-shifted and rectified input a.c. power. When, however, these converters are connected to a common a.c. source having a relatively high internal impedance, the waveform of the phase-shifted and rectified a.c. input power is distorted thereby causing undesired operations of the converters.

It is therefore an object of the present invention to provide a constant-voltage converter which correctly operates notwithstanding the distortion of the input a.c. voltage.

Another object of the invention is to provide a constant-voltage converter which effectively suppress an undesired rush current.

Another object of the invention is to provide a constant-voltage converter having an improved feed-back circuit of a substantially constant loop gain .

In the drawings:

FIG. 1 is a schematic view of a converter according to the present invention;

FIG. 2 is a diagram showing a circuit arrangement of the converter of FIG. 1;

FIG. 3 is a diagram showing various waveforms of signals appearing in the circuit of FIG. 2;

FIG. 4 is a diagram showing various waveforms appearing in the circuit of FIG. 2 when an a.c. power is supplied to the circuit;

FIG. 5 is a diagram showing another circuit arrangement of the converter of FIG. 1;

FIG. 6 is a diagram showing waveforms of signals appearing in the circuit of FIG. 5; and

FIG. 7 is a diagram showing further another circuit arrangement of generator the of FIG. 1.

Referring now to FIG. 1, a constant-voltage converter 10 according to the present invention comprises a rectifier 11 having two input terminals 12 and 13 through which an a.c. power is supplied. The rectifier 11 is preferably a full-wave rectifier although a half-wave rectifier may be employed. An output 14 of the rectifier 11 is connected through a line 15 to an anode of a thyristor 16. The thyristor 16 passes therethrough the rectified a.c. power in only one direction from its anode to cathode when triggered by a trigger pulse through its gate. The cathode of the thyristor 16 is connected through a line 17 to an input of a smoothing filter 18. The smoothing filter 18 smoothes the power from the thyristor 16. An output of the smoothing filter 18 is connected through a line 19 to an output terminal 20. The output 14 of the rectifier 11 is also connected through a line 21 to a saw-tooth wave generator 22 which generates a saw-tooth wave signal having the same repetition period as the rectified input a.c. power. An output of the saw-tooth wave generator 22 is connected through a line 23 to one input of a trigger pulse generator 24. The other input of the trigger pulse generator 24 is connected through a line 25 to the line 19. An output of the trigger pulse generator 24 is connected through a line 26 to the gate of the thyristor 16. The trigger pulse generator 24 produces a trigger pulse on its output when the voltage of the saw-tooth wave signal reaches a level which is varied in response to the output voltage on the terminal 20. The trigger pulse generator 24 may be variously arranged and in this case arranged to comprise rectangular generator 27 having one input connected through the line 23 to the saw-tooth wave generator 22 and the other input connected through a line 28 to an output voltage detector 29. The detector 29 produces a reference signal representing the output voltage on the terminal 20. The pulse generator 27 is adapted to produces a rectangular pulse when the saw-tooth wave signal to the one input reaches a level which defined is in accordance with the reference signal. An output of the rectangular pulse generator 27 is connected through a line 30 to an input of a trigger circuit 31. The trigger circuit 31 is adapted to convert the rectangular pulse into a spike pulse. An output of the trigger circuit 31 is connected through the line 26 to the gate of the thyristor 16.

FIG. 2 illustrates a preferred circuit arrangement of the converter shown in FIG. 1 which comprises a rectifier 11 of a full-wave rectifier consisting of rectifiers 40, 41, 42 and 43. Inputs of the rectifier are connected to terminals 12 and 13 through which an a.c. power is applied. The output 14 of the rectifier 11 is connected through a line 15 to an anode of a thyristor 16. A cathode of the thyristor 16 is connected through a line 17 to a smoothing filter 18 which includes a capacitor C4 having one terminal connected to the line 17 and the other terminal grounded. The output of the smoothing filter 18 is connected through a line 19 to an output terminal 20.

The saw-tooth wave generator 22 includes a resistor R ₁ having one terminal connected to the line 21 and the terminal connected through a junction J ₁ to one terminal of a resistor R ₂ . The other terminal of the resistor R ₂ is grounded. The junction J ₁ is connected through a coupling capacitor C ₁ to a base of a transistor T ₁ of PNP type. An emitter of the transistor T ₁ is connected through a resistor R ₃ to the line 21. A resistor R ₄ is provided between the emitter and the base of the transistor T ₁ so as to apply a bias potential to the base. A collector of the transistor T ₁ is grounded through a parallel connection of a resistor R ₅ and capacitor C ₂ . To the emitter is connected a capacitor C ₃ which is in turn grounded and passes therethrough only a.c. signals to the ground.

The rectangular pulse generator 27 comprises a transistor T ₂ of PNP type having a base connected through a resistor R ₆ to the collector of the transistor T ₁ . An emitter of the transistor T ₂ is connected through a resistor R ₇ to the emitter of the transistor T ₁ . A collector of the transistor T ₂ is grounded through a resistor R ₈ and connected through the line 30 to one terminal of a capacitor C ₄ of the trigger circuit 31. The other terminal of the capacitor C ₄ is connected through a line 26 to the gate of the thyristor 16.

The output voltage detector 29 includes a transistor T ₃ of NPN type having an emitter grounded through a zener diode ZD. A collector of the transistor T ₃ is connected through a line 28 to the emitter of the transistor T ₂ and, on the other hand, connected through a capacitor C ₅ to the grounded. A base of the transistor T ₃ is connected to a tap of an adjustable resistor R ₉ connected through a resistor R ₁₀ and a line 25 to the line 19 and connected, in turn, to the ground through a resistor R ₁₁ .

When, in operation, an a.c. electric power is applied through the input terminals 12 and 13 of the rectifier 11, a full-wave rectified power as shown in FIG. 3 (a) appears on the output 14. The rectified power is applied through the line 15 to the anode of the thyristor 16. The thyristor 16 passes therethrough the rectified power while its firing angle is regulated by the trigger signal applied to the gate. The rectified power passed through the thyristor 16 is applied through the line 17 to the smoothing filter 18. The smoothing filter smoothes the power by removing the ripple component in the power. The smoothed power appears on the line 19 which is to be supplied to a load through the output terminal 20. The smoothed power on the line 19 is, on the other hand, delivered through the line 25 to the resistor R ₁₀ of the output voltage detector 29. The resistor R ₁₀ constitutes a voltage divider in cooperation with the resistors R ₉ and R ₁₁ . The output of the voltage divider is applied through the tap of the resistor R ₉ to the base of the transistor T ₃ . When the potential of the base of the transistor T ₃ exceeds the zener voltage of the zener diode ZD, a base current flows through the transistor T ₃ so as to render the transistor T ₃ conductive. The potential of the collector of the transistor T ₃ then varies in accordance with the voltage of the smoothed output power on the line 19. The potential variation at the collector of the transistor T ₃ is then applied through the line 28 to the trigger pulse generator 27 and utilized to regulate the triggering timing of the thyristor 16.

The full-wave rectified power is, on the other hand, applied through the line 21 to the saw-tooth wave generator 22. Since the resistors R ₁ and R ₂ consistute a voltage divider to reduce the voltage of the full-wave rectified power to a potential at the junction J ₁ , a charging current to the capacitor C ₁ flows from the emitter to the base of the transistor T ₁ whereby the transistor T ₁ repeats ON-OFF operation in accordance with the voltage of the rectified power. If the transistor T ₁ is conductive when the voltage of the full-wave rectified power is lower than a threshold voltage v ₁ as shown in FIG. 3(a), then the potential at the collector of the transistor T ₁ is varied as shown in FIG. 3 (b) due to the charge and discharge of the capacitor C ₂ . The variation of the potential at the collector of the transistor T ₁ is supplied through the line 23 to the resistor R ₆ of the trigger pulse generator 27.

As long as the voltage of the smoothed power on the line 19 equals to the rated output voltage, the transistor T ₂ is adapted to become conductive when the voltage of the saw-tooth wave signal falls below a threshold value v ₃ shown in FIG. 3(b). Therefore, a potential at the collector of the transistor T ₂ varies as shown in FIG. 3(c). The potential variation, that is, a pulse signal at the collector of the transistor T ₂ is supplied through the line 30 to the capacitor C ₄ of the trigger circuit trigger 31. The trigger circuit 31 converts the pulse signal into a spike pulse or a trigger pulse shown in FIG. 3(d) which is then applied through the line 25 to the gate of the thyristor 16. Upon receiving the spike pulse, the thyristor 16 becomes conductive until the voltage of the rectified power on the line 15 falls below the cut-off voltage of the thyristor 16.

When the voltage of the smoothed power on the line 19 exceeds the rated output voltage, the collector current of the transistor T ₃ increases with the result that the current flowing through the resistor R ₇ increases. The threshold voltage of the transistor T ₂ therefore reduces to a voltage v ₂ as shown in FIG. 3(b). At this instant, leading edge of the pulse signal delays as shown by dot-and-dash lines in FIG. 3(c), so that each trigger pulse delays as shown by dot-and-dash line in FIG. 3(d). When on the contrary, the voltage of the smoothed signal on the line 19 lowers below the rated output voltage, the collector current of the transistor T ₃ decreases whereby the threshold voltage rises to a voltage v ₄ in FIG. 3(b). Each leading edge of the signal pulse now leads as shown by dotted line in FIG. 3(d). Being apparent from the above description, the appearance timing of each trigger pulse is regulated in accordance with the voltage of the smoothed power on the line 19 so that the voltage of the output voltage at the terminal 20 is held substantially constant.

Referring now to FIG. 4, start operation of the converter 10 is discussed hereinbelow in conjunction with FIG. 2. When an a.c. voltage is applied to the input terminals 12 and 13, the capacitor C ₃ begins to be charged by the voltage on the line 15, and the capacitor C ₅ also begins to be charged through the resistors R ₃ and R ₇ . It is important that the time constant of power supply circuit constituted by the resistor R ₃ and the capacitor C ₃ is selected to be much larger than that of the time constant of another power supply circuit constituted by the resistor R ₇ and the capacitor C ₅ . Thus, the emitter potential of the transistor T ₁ is built up more quickly than that of the transistor T ₂ . Upon completion of the charging of the capacitor C ₃ , the saw-tooth wave generator 22 begins to generate saw-tooth wave signal as shown in FIG. 4(b). Since the capacitor C ₅ is, on the other hand, slowly charged, the emitter voltage of the transistor T ₂ slowly rises as shown in FIG. 4(c), so that, the threshold voltage of the transistor T ₂ gradually rises as shown by a dotted line in FIG. 4 (b). Accordingly, the trigger pulses is produced on the gate of the thyristor 16 as shown in FIG. 4(d), whereby the firing angle of the thyristor 16 is gradually reduced as shown in FIG. 4(a) which illustrates the voltage at the output terminal 14 of the rectifier 11. The output voltage on the output terminal 20 therefore gradually rise up as shown in FIG. 4(e). It is to be understood that since the output voltage of the converter 10 starts to gradually rise up as shown in FIG. 4(e), an undesired rush current is effectively suppressed.

FIG. 5 illustrates another form of the converter 10 which is arranged identically to the circuit arrangement of FIG. 1 except that an integrator 50 is interposed between the output of the saw-tooth wave generator 22 and the input of the trigger pulse generator 27. The integrator 50 includes a resistor R ₁₂ having one terminal connected to the output of the saw-tooth wave generator 22 and the other terminal connected to the input of the rectangular pulse generator 27, and a capacitor C ₇ having one terminal connected to the other terminal of the resistor R ₁₂ and the other terminal grounded.

In operation, the saw-tooth wave generator 22 produces on its ouput a saw-tooth wave signal having decreasing exponential wave form portion as shown in FIG. 6 (a), although the saw-tooth wave signal ideally is illustrated in FIG. 3. This saw-tooth wave signal is converted by the integrator 50 into another form of saw-tooth wave having a increasing exponential wave form portion as shown in FIG. 6(b).

It should be noted that the saw-tooth wave signal of FIG. 6(a) has a smaller inclination near 180°. Hence, when the integrator 50 is omitted and the saw-tooth wave signal as shown in FIG. 6(a) is applied to the trigger pulse generator 27, the rate of change of the output voltage of the converter 10 become larger at a firing angle near to 180°. On the other hand, it is apparent from FIG. 6(c) that the rate of change the output voltage of the thyristor 16 with respect to the firing angle become large at a firing angle near to 180°. Therefore, the loop gain of the trigger pulse generator 24 increases when the firing angle of the thyristor 16 is near to 180°. It is apparent through a similar discussion that the loop gain of the trigger pulse generator 24 decreases when the firing angle is near to 90°. Such non-uniformity of the loop gain of the trigger pulse generator invites a difficulty of the regulation of the output voltage of the converter. It is to be noted that the saw-tooth wave signal shown in FIG. 6(b) has a large inclination at an angle near 180°. Therefore, when the saw-tooth wave signal of FIG. 6(b) is applied to the trigger pulse generator 24, the loop gain of the trigger pulse generator 24 is held substantially constant, whereby the output voltage of the converter is effectively held constant.

It is to be understood that the integrator 50 may be substituted for by a miller integrator and a bootstrap integrator. Furthermore, a plurality of integrator may be employed, if desired.

FIG. 7 illustrates another circuit arrangement of the converter according to the present invention, which is arranged identically to the circuit of FIG. 2 except for the trigger circuit 31 and the smoothing circuit 18.
The trigger circuit 31 of FIG. 7 comprises a transformer TR with primary and secondary coils. One terminal of the primary coil is connected to the resistor R ₇ of the pulse generator 27. The other terminal of the primary coil is connected to a collector of a transistor T ₄ of NPN type. The secondary coil has terminals respectively connected to the gate and cathode of the thyristor 16. An emitter of the transistor T ₄ is grounded through a resistor R ₁₃ . A base of the transistor T ₄ is grounded through a resistor R ₁₄ and connected through a capacitor C ₈ to the collector of the transistor T ₂ of the pulse generator 27.

The smoothing filter 18 of FIG. 7 comprises a choke coil CH connected to the lines 17 and 19, and to capacitors C ₉ and C ₁₀ which are in turn grounded. The circuit of FIG. 7 operates in the same manner as the circuit of FIG. 2.

Obviously many modifications and variations of the present invention are possible in the light of the above teachings. It is therefore to be understood that within the scope of the appended claims the invention may be practiced otherwise than as specifically described.



The CRT TUBE IS a VALVO PHILIPS A66-140X.






NORDMENDE SPECTRA COLOR L2UT CHASSIS FFS 772.535.A 772.536.A (4.570.D) NORD SOUTH (NORD/SUD) CORRECTION CIRCUIT ARRANGEMENT FOR CORRECTING THE DEFLECTION OF AT LEAST ONE ELECTRON BEAM IN A TELEVISION PICTURE TUBE BY MEANS OF A TRANSDUCTOR :



A circuit arrangement for raster correction in a television picture tube by means of a transductor whose power winding is connected in parallel with at least a portion of the line deflection coils, the line deflection generator having a low internal impedance. In order to increase this impedance a mainly inductive impedance is connected in series with the generator. In a picture tube employing at least two electron beams the series impedance may include the convergence circuit. As a result the convergence in the corners of the picture screen is also improved. The linearity control circuit may likewise form part of the series impedance.



1. A deflection circuit for a cathode ray tube comprising a transistor horizontal deflection generator; a horizontal deflection coil parallel coupled to said generator; means for pincushion correction of said tube comprising a saturable reactor having a control winding adapted to receive a vertical deflection signal and a power winding parallel coupled to at least a portion of said deflection coil; and means for increasing the effectiveness of said correction means comprising an impedance element external to said generator having a substantially inductive reactance series coupled between said generator and said coil. 2. A circuit as claimed in claim 1 wherein said generator comprises a transformer having a tap and said power winding has a first end coupled to said coil and a second end coupled to said tap. 3. A circuit as claimed in claim 1 wherein said impedance element comprises means for controlling the linearity of the beam deflection. 4. A deflection circuit for a cathode ray tube having at least two electron beams comprising a transistor horizontal deflection generator; a horizontal deflection coil parallel coupled to said generator; means for pincushion correction of said tube comprising a saturable reactor having a control winding adapted to receive a vertical deflection signal and a power winding parallel coupled to at least a portion of said deflection coil; means for increasing the effectiveness of said correction means comprising an Impedance element external to said generator having a substantially inductive reactance series coupled between said generator and said coil; and means for dynamically converging said beams comprising a convergence circuit coupled to said horizontal generator and to said transductor. 5. A circuit as claimed in claim 4 wherein said generator comprises a transformer having a tap and said power winding has a first end coupled to said coil and a second end coupled to said tap. 6. A circuit as claimed in claim 4 wherein said impedance element comprises means for controlling the linearity of the beam deflection.

Description:

The invention relates to a circuit arrangement for correcting the deflection of at least one electron beam (raster correction) in a television picture tube by means of a saturable reactor a power winding of which is connected in parallel with at least a portion of the coils for the horizontal deflection, the current flowing through these coils being supplied by a deflection generator having a low internal impedance.

A circuit arrangement for raster correction with the aid of a transductor is described, for example, in U.S. Pat. No. 3,444,422. In this patent the power winding of a transductor is connected in parallel with the horizontal deflection coils while the control winding receives a signal of field frequency so that the current of line frequency which flows through the deflection coils is modulated at the field

-frequency (East-West correction), whereas the vertical deflection current is modulated at the line frequency (North-South correction). However, in this known arrangement there is the difficulty that the transductor can exert little influence on the horizontal deflection current if the internal impedance of the deflection generator is low because the transductor then only constitutes an additional load on the generator. This is the case when the deflection generator includes a valve with feedback -- or a switch formed with one or more transistors. In order to be able to use a transductor arrangement also in such a case the circuit arrangement according to the invention is characterized in that a mainly inductive impedance is connected in series between the said parallel arrangement and the deflection generator.

Due to the step according to the invention the internal impedance of the deflection generator is increased and the different components of the circuit remain mainly inductive so that the deflection current is more or less linear when the voltage provided by the deflection generator during the line scan period is substantially constant. The series impedance may be, for example, a fixed coil. However, the invention is furthermore based on the recognition of the fact that the increase in the internal resistance of the horizontal deflection generator may not only be obtained by a constant impedance, but other arrangements envisaging other improvements of the deflection may be used for this purpose. In that case even special improvements may be obtained as will be apparent hereinafter and possible small non-linearities of the additionally used arrangements have no detrimental results.

It is true that in known convergence circuits in picture tubes employing a plurality of electron beams a satisfactory improvement is obtained for the central horizontal and vertical lines of a picture tube of the shadow mask type. However, it is found that convergence errors may subsist in the corners of the picture. Known circuit arrangements which correct these second-order errors are often complicated and expensive. In the circuit arrangement according to the invention a satisfactory compensation of such convergence errors is possible in a simple manner if the series impedance which is arranged between the horizontal deflection generator and the deflection coils includes the convergence circuit. In this manner the sum of the deflection current and of the current derived for the field correction and modulated by the transductor flows through the convergence circuit so that the desired additional convergence correction in the corners of the written raster is obtained.

In order that the invention may be readily carried into effect a few embodiments thereof will now be described in detail by way of example with reference to the accompanying diagrammatic drawings in which:

FIG. 1 shows a circuit arrangement in which the transductor is connected in parallel with the deflection coils, while in

FIG. 2 the transductor is only fed by part of the voltage applied to the deflection coils.

FIG. 1 shows two line-output transistors 1 and 2 which are arranged in series. The emitter of transistor 2 is connected to ground through a winding 3 while the collector of transistor 1 is connected through a winding 4 and a small series impedance 5, preferably a resistor, to the positive terminal of a supply source V _b whose negative terminal is connected to ground.

Windings 3 and 4 are wound together with an EHT-winding 6 on the same transformer core 7. The ends of windings 3 and 4 remote from each other are connected through the capacitor 10 for the S-correction to the deflection-unit consisting of two windings 8 and 9 arranged, for example, in parallel. The base of transistors 1 and 2 receive pulses of line frequency in a manner not shown in FIG. 1 so that these transistors are cut off during the flyback period. During the scan period, a substantially constant voltage is applied to the deflection unit. Consequently a more or less sawtooth-shaped current flows through windings 8 and 9. The bipartite power winding 11 of a transductor ensuring the raster correction is connected in parallel with this deflection unit 8, 9. The control winding 12 of said transductor, and a converting capacitor 13 in parallel therewith form part of the circuit for the vertical deflection through terminals 14 and 15. An adjustable coil 16 with which the raster correction can be adjusted exactly is connected in series with winding 12.

Windings 3 and 4 have the same number of turns so that pulses of the same amplitude and reversed polarity are produced at the emitter of transistor 2 and at the collector of transistor 1. As a result a disturbing radiation of these pulses is reduced. Furthermore, transistor types are chosen in this Example for transistors 1 and 2 whose collector-base diodes may function as efficiency diodes. All this has been described in U.S. Pat. No. 3,504,224.

According to the invention the convergence circuit 17 is arranged through a separation transformer 20 between the end of winding 3 remote from winding 4 and the horizontal deflection coils 8, 9. Furthermore, this current branch includes the linearity control circuit 21 which comprises the parallel arrangement of a resistor and a coil whose inductance is adjustable, for example, by means of premagnetization of the core of the coil. A current, which is the sum of the current for the deflection coils 8, 9 and of the current for the power winding 11 of the transductor, flows through the primary winding of transformer 20. This primary current is transformed to the secondary circuit of transformer 20 so that a current flows through convergence circuit 17.

In known arrangements the con

vergence current is only influenced by the deflection current itself. It has been found that in this case the convergence correction is not sufficient in the corners of the picture. At these areas, where the deflection in both directions is at a maximum, a greater intensity of the convergence current is required. This is especially the case in picture tubes having a great deflection angle and according to the invention this is achieved in that the current which is derived from the power winding 11 of the transductor for the raster correction is also applied to the convergence circuit. This current flows from the horizontal deflection generator constituted by windings 3 and 4 through the primary winding of transformer 20 to power winding 11 of the transductor. The transductor current is in fact at a minimum in the center of the picture and increases towards the edges and particularly towards the corners. Thus the convergence current varies in the desired manner. According to the invention the desired improvements of the convergence correction and simultaneously the likewise desired increase in the internal resistance of the horizontal deflection generator is consequently obtained without a considerable increase in the number of required circuit elements and without disturbing the normal operation of the circuit arrangement. Due to transformer 20 a terminal of convergence circuit 17 may be connected to ground so that the convergence can be adjusted safely. If necessary, a suitable impedance transformation may also be obtained with the aid of transformer 20.

The linearity control circuit 21 may alternatively be connected in series with the said branch which includes transformer 20. As a result the internal resistance of the horizontal deflection generator for the line frequency is further increased without the field correction and the convergence correction being disturbingly influenced.

FIG. 2 shows a modification of the circuit arrangement according to the invention in which the deflection current is not changed relative to that of FIG. 1. The end of power winding 11 of the transductor shown on the upper side of FIG. 1 is connected to ground in FIG. 2. In addition convergence circuit 17 is included between winding 3 and ground so that separation transformer 20 may be omitted. If as a first approximation the impedances 5 and 17 are assumed to be negligibly small relative to the other impedance of the circuit arrangement, power winding 11 may be considered to be connected to a tap on the deflection generator 3, 4. Consequently, only approximately half the voltage of the deflection generator is applied to transductor winding 11 which winding must therefore be proportioned in such a manner that it can convey a current which is approximately twice as large as that of FIG. 1. This larger current also flows through convergence circuit 17 which, with the omission of separation transformer 20, is favorable for the convergence in the corners of the picture screen.

In FIG. 2 the emitter of transistor 2 is connected to ground i.e., the said tap on the deflection generator. During the scan period the series arrangement of supply source V _b and windings 3 and 4 FIG. 1 is substantially short-circuited by transistors 1 and 2. In order that these transistors in the circuit arrangement according to FIG. 2 operate under the same circumstances as those in FIG. 1, an additional winding 24 must be wound on core 7 between windings 4 and 6, winding 24 having the same number of turns as winding 3, and the collector of transistor 1 must be connected to the junction of windings 6 and 24.

The end of power winding 11 connected to ground in FIG. 2 may alternatively be connected for the desired adjustment of the corner convergence to a different tap on the transformer, that is to say, on winding 3 or 4.

Resistor 5 serves in known manner mainly as a safety resistor so that in case of an inadmissible load of the EHT, for example, as a result of flash-over in the picture tube, the supply voltage for transistors 1 and 2 is reduced so that overload of these transistors is avoided.

NORDMENDE SPECTRA COLOR L2UT CHASSIS FFS 772.535.A 772.536.A (4.570.D) E/W CORRECTION Circuit arrangement in an image display apparatus for (horizontal) line deflection:

Line deflection circuit in which the deflection coil is east-west modulated. In order to cancel an east-west dependent horizontal linearity defect the inductance value of the linearity correction coil is made independent of the field frequency, for example by means of a compensating current. In an embodiment this current is supplied by the shunt coil of the east-west modulator.



1. Circuit arrang

ement for use with a line deflection coil, said circuit comprising a generator means adapted to be coupled to said coil for producing a sawtooth line-deflection current through said line deflection coil, said deflection current having a field-frequency component current, a horizontal linearity correction coil adapted to be coupled in series with said deflection coil and including an inductor having a bias-magnetized core, and means for making the inductance value of the linearity correction coil substantially independent of the field frequency component current. 2. Circuit arrangement as claimed in claim 1, wherein said making means includes a current supply source means for producing a compensating line-frequency sawtooth current through a winding of the linearity correction coil, the amplitude of the compensating current having a field-frequency variation. 3. Circuit arrangement as claimed in claim 2, wherein the direction of curvature of the field-frequency envelope of the compensating current is opposite to the direction of curvature of the field-frequency component current of the line deflection current, whereby the magnetic fields produced in the core of the correction coil by the two currents have the same direction. 4. Circuit arrangement as claimed in claim 2, wherein the direction of curvature of the field-frequency envelope of the compensating current is the same as the direction of curvature of the field-frequency component current of the line deflection current, whereby the magnetic fields produced in the core of the correction coil by the two currents have opposite directions. 5. Circuit arrangement as claimed in claim 2, wherein said correction coil further comprises an additional winding disposed on the core, said additional winding being coupled to said supply source means to receive the compensating current. 6. Circuit arrangement as claimed in claim 5, further comprising modulator means for modulating the line deflection current with said field frequency component, said modulator including a compensation coil coupled in series with said additional winding. 7. Horizontal linearity correction coil comprising a core made of a magnetic material and bias-magnetized by at least one permanent magnet, and an additional winding disposed on the core. 8. Image display apparatus including a circuit arrangement as claimed in claim 1.

Description:

The invention relates to a circuit arrangement in an image display apparatus for (horizontal) line deflection, which apparatus also includes a circuit arrangement for (vertical) field deflection, provided with a generator for generating a sawtooth line-frequency deflecting current through a line deflection coil and with a modulator for field-frequency modulation of this current, the deflection coil being connected in series with a linearity correction coil in the form of an inductor having a bias-magnetized core.

By means of the linearit

y correction coil the linearity error due to the ohmic resistance of the deflection circuit is corrected. The sign of the bias magnetisation is chosen so that it is cancelled by the deflection current at the beginning of the deflection interval, so that the inductance of the correction coil is a maximum, whereas the voltage drop across the deflection coil then is a minimum. This voltage drop is adjustable by adjustment of the starting inductance of the correction coil. During the deflection interval the core gradually becomes saturated so that the inductance of, and the voltage drop across, the correction coil decrease. Thus the linearity error can be cancelled exactly at the beginning of the interval, that is to say on the left on the screen of the image display tube, and with a certain approximation at other locations.

In image display tubes using a large deflection angle, raster distortion, which generally is pincushion-shaped, of the image displayed occurs. This distortion can be removed in the horizontal direction, the so-called east-west direction, by means of field-frequency modulation of the line deflection current, the envelope in the case of pincushion-shaped distortion being substantially parabolic so that the amplitude of the line deflection current is a maximum at the middle of the field deflection interval.

It

was found in practice that the said two corrections are not independent of one another, that is to say the adjustment of the east-west modulation affects horizontal linearity. As long as the modulation depth is not excessive, a satisfactory compromise can be found. However, in display tubes having a deflection angle of 110° and particularly in colour display tubes in which the deflection coils have a converging effect also, it is difficult to find such a compromise. A tube of this type is described in "Philips Research Reports," volume Feb. 14, 1959, pages 65 to 97; the distribution of the deflection field is such that throughout the display screen the landing points of the electron beams coincide without the need for a converging device. Owing to this field distribution, however, the pin-cushion-shaped distortion in the image displayed in the east-west direction is greater than in comparable display tubes of another type. Hence there must be east-west modulation of the line deflection current to a greater depth. It is true that under these conditions horizontal linearity can correctly be adjusted over a given horizontal strip after the east-west modulation has been adjusted correctly, i.e., for a rectangular image, but it is found that in other parts of the display screen a serious linearity error remains. When vertical straight lines are displayed as straight lines in the right-hand part of the screen, they are displayed as curved lines in the left-hand part.

It is an object of the present invention to remove the said defect so that horizontal linearity can satisfactorily be adjusted throughout the screen, and for this purpose the circuit arrangement according to the invention is characterized in that it includes means by which the inductance of the linearity correction coil is made substantially independent of the field frequency.

The invention is based on the recognition that the defect to be removed is due to a field-frequency variation of the said inductance because the latter is current-dependent. According to a further recognition of the invention the circuit arrangement is characterized in that it includes a current supply source for producing a compensating line-frequency sawtooth current through a winding of the linearity correction coil, the amplitude of the current being field-frequency modulated. The circuit arrangement according to the invention may further be characterized in that an additional winding is provided on the core of the linearity correction coil and is traversed by the compensating current. A circuit arrangement in which the modulator for modulating the line deflection current includes a compensation or bridge coil may according to the invention be characterized in that the additional winding is connected in series with the said coil.

The invention also relates to a linearity correction coil for use in a line deflection circuit having a core which is made of a magnetic material and is bias magnetized by at least one permanent magnet, which coil is characterized in that an additional winding is provided on the core.

Embodiments of the invention will now be described by way of example, with reference to the accompanying diagrammatic drawings, in which

FIG. 1 is the circuit diagram of a known circuit arrangement for line deflection in which the line deflection current is east-west modulated,

FIG. 2 shows the distorted image which is displayed on the screen when the circuit arrangement of FIG. 1,

FIG. 3 is a graph explaining the observed defect, and

FIGS. 4 and 7 show embodiments of the circuit arrangement according to the invention by which this defect can be cancelled.

FIG. 1 is a greatl simplified circuit diagram of a line deflection circuit of an image display apparatus, not shown further. The circuit includes the series combination of a line deflection coil L _y , a linearity correction coil L and a trace capacitor C _t , which series combination is traversed by the line deflection current i _y . The collector of an npn switching transistor T _r and one end of a choke coil L ₁ are connected to a junction point A of a diode D, a capacitor C _r and the said series combination. The other end of the choke coil is connected to the positive terminal of a supply voltage source which supplies a substantially constant direct voltage V _b and to the negative terminal of which the emitter of transistor Tr is connected. This negative terminal may be connected to earth. The other junction point B of elements D and C _r and of the series combination of elements C _t , L _y and L is connected to one terminal of a modulation source M for east-west correction which has its other terminal connected to earth. Diode D has the pass direction shown in the FIG.

To the base of transistor Tr line-frequency switching pulses are supplied. In known manner the said series combination is connected to the supply voltage source during the deflection interval (the trace time), diode D and transistor Tr conducting alternately. During the retrace time these elements are both cut off. Under these conditions the current i _y is a sawtooth current. The coil L, which has a saturable ferrite core which is bias-magnetized by means of at least one permanent magnet, serves to correct the linearity of the current i _y during the trace time, whilst the capacitance of the capacitor C _t is chosen so that the currenct i _y is subjected to what is generally referred to as S correction. During the retrace time, at point A pulses are produced the amplitude of which is much higher than that of the voltage V _b and would be constant in the absence of modulation source M. Information from the field deflection circuit, not shown, of the image display apparatus and line retrace pulses, the latter for example by means of a transformer, are supplied in known manner to modulation source M. Amplitude-modulated line retrace pulses having a field-frequency parabolic envelope, as indicated in the FIG., are produced at point B. During the line trace time the voltage at point B is zero. Thus the current i _y is given the desired field-frequency modulated form which is also shown in FIG. 1.

The amplitude of the envelope in point B at the beginning and at the end of the field trace time and the amplitude of this envelope at the middle of the said time can both be adjusted so that the image displayed on the display screen of the display tube (not shown) has the correct substantially rectangular form. If, however, the required modulation depth is comparatively large, a linearity error of the line deflection is produced which cannot be removed by means of the correction coil L.

FIG. 2 shows the image of a pattern of vertical straight lines as it is displayed on the screen with the correction coil L adjusted so that horizontal linearity is satisfactory along and near the central horizontal line. In FIG. 2 the defect is exaggerated. It is found that horizontal linearity is defective in other areas of the screen so that the vertical lines are displayed correctly in the right-hand half of the screen but as curves in the left-hand path, the defect increasing as the line is farther to the left.

This phenomenon can be explained with reference to FIG. 3. In this FIG. the inductance L of the linearity correction coil is plotted as a function of the magnetic field strength H. In the absence of current, H has a value H ₀ owing to the bias magnetization. If an approximately linear sawtooth current i (t) as shown in the bottom left-hand part of FIG. 3 flows through the coil, the field strength H varies proportionally about the value H ₀ , for the mean value of the current is zero. Because the curve of L is not linear, the variation L(t) of L, which is shown in the top right-hand part, is not a linear function of time. The resulting curve may be regarded as composed of a linear component and a substantially parabolic component which is to be taken into account when choosing the capacitance of capacitor C _t .

Because owing to the east-west modulation the amplitude of current i(t) varies, the amplitude of L(t) also varies. This implies a field-frequency variation of L which is non-linear. This variation is undesirable. In the case of a small variation of the amplitude of current i(t) the variation of L(t) can be more or less neglected, but this is no longer possible when the amplitude of current i(t) varies greatly owing to the east-west modulation. L(t) varies according to different curves. FIG. 3 shows two of such curves and also illustrates the fact that the undesirable variation of L(t) is greatest at the beginning of the trace time and smallest at the end thereof.

FIG. 4 shows a circuit arrangement in which the defect described can be corrected. On the core of the correction coil L of the circuit of FIG. 1 an additional winding L ₂ is provided. Winding L ₂ is connected to a current source which produces a compensating current i ₂ which has a line-frequency sawtooth variation and a field-frequency amplitude modulation. The envelope here also is parabolic, however, with a shape opposite to that of deflection current i _y , that is to say having a minimum at the middle of the field trace time. The direction of current i ₂ and the winding sense of winding L ₂ relative to that of coil L are chosen so that the magnetic field produced in the core by winding L ₂ has the same direction as the field produced by coil L. Hence the two field strengths are added

. The amplitude of current i ₂ and the turns number of winding L ₂ can be chosen so that current i _y flows through inductances the total value of which is not dependent upon the field frequency. The curve L(t) of FIG. 3 remains substantially unchanged. Consequently the undesirable field-frequency modulation is removed without variation of the bias magnetization, which would have been varied if current i ₂ were a field-frequency current. Obviously the same result can be achieved by a choice such of the direction of current i ₂ and of the winding sense of winding L ₂ that the two field strengths are subtracted one from the other, whilst the curvature of the envelope of current i ₂ has the same direction as that of the envelope of current i _y .

The current source of FIG. 4 may be formed in known manner by means of a modulator in which a line-frequency sawtooth signal is field-frequency modulated, the envelope being parabolic. FIG. 5 shows a circuit arrangement in which current i ₂ is produced by the modulation source which provides the east-west correction. In FIG. 5, the source M of FIG. 1 comprises a diode D', a coil L' and two capacitors C' _r and C' _t , which elements constitute a network of the same structure as the network formed by elements D, L _y , C _r and C _t . The capacitor C' _t is shunted by a modulation source V _m which supplies a field-frequency parabolic voltage having a minimum at the middle of the field trace time.

With the exception of the linearity correction means to be described hereinafter, the circuit arrangement of FIG. 5 was described in more detail in U.S. Pat. No. 3,906,305. Hence it will be sufficient to mention that the capacitances of capacitors C _r and C' _r and of a capacitor C ₁ connected between junction point A and earth and the inductance of coil L' are chosen so that the three sawtooth currents flowing through L _y , L' and L ₁ have the same retrace time. The capacitances of capacitors C _t and C' _t , which are large, are ignored. When voltage V _b is constant, current i _y is subjected to the desired east-west modulation having the form shown in FIG. 1.

Coil L _y is connected in series with correction coil L, and winding L ₂ is connected in series with coil L'. FIG. 5 shows that the current flowing through winding L ₂ has the same waveform as the current i ₂ of FIG. 4, for its envelope has the same shape as the voltage supplied by source V _m . By a suitable choice of the number of turns of winding L ₂ it can be ensured that the linearity correction remains the same for every line during the field trace time.

Modified embodiments of the circuit arrangement of FIG. 5 can also be used. FIG. 6 shows such a modified embodiment in which the capacitive voltage divider C _r , C' _r of FIG. 5 is replaced by an inductive voltage divider by means of a tapping on coil L ₁ . A capacitor C ₂ is included between the tapping and the junction point of diodes D and D', whilst capacitor C' _t here forms part of two networks C _t , L _y and C' _t , L' traversed by a sawtooth current. In FIG. 6 modulation source V _m is connected via a choke coil L ₃ to the junction point of D, D', C ₂ and C' _t . One end of winding L ₂ is connected to the junction point of capacitor C' _t and the coil L, whilst the other end is connected to earth via coil L'. The capacitances of capacitors C ₁ and C ₂ and the location of the tapping on coil L ₁ are chosen so that the sawtooth currents flowing through L _y , and L' and L ₁ have the same retrace time, whilst the field-frequency linearity defect of FIg. 2 is cancelled by correctly proportioning winding L ₂ .

Other east-west modulators are known in which the step of FIGS. 5 and 6 can be used. An example is the modulator described in the publication by Philips, Electronic Components and Materials: "110° Colour television receiver with A66-140X standard-neck picture tube and DT 1062 multisection saddle yoke," May 1971, pages 19 and 20, which modulator also comprises two diodes and a compensation coil L', which are arranged in a slightly different manner. In another example the east-west modulator and the line deflection generator are included in a bridge circuit whilst they are decoupled from one another by means of a bridge coil which has the same function as coil L' in FIGS. 5 and 6. In these circuit arrangements coil L and winding L ₂ may be arranged in the same manner as in FIG. 6. The same applies to an east-west modulator using a transductor the operating winding of which is in series with the deflection coil.

In the abovedescribed embodiments of the circuit arrangement according to the invention the compensating current i ₁ is provided by transformer action. In the embodiment of FIG. 7 the current source which supplies the current i ₂ is connected in parallel with correction coil L, i.e., without an auxiliary winding. In this embodiment the east-west modulation is achieved not by means of a modulator, but by means of the fact that the supply voltage V _b is the super-position of a field-frequency parabolic voltage on the direct voltage. In this known manner the supply source also is the modulator.

It will be seen that in the embodiments of FIGS. 4, 5 and 6 current i ₂ counteracts the east-west modulation of deflection current i _y . It was found in practice, however, that this counteraction is slight.

NORDMENDE SPECTRA COLOR L2UT CHASSIS FFS 772.535.A 772.536.A (4.570.D) CHROMA-BURST SEPARATOR AND AMPLIFIER CIRCUIT :

A combined separator/amplifier for deriving chroma and burst signals comprises a differential amplifier having a pair of differentially acting transistors coupled to a common current source. The current source is formed by a transistor driven by unseparated chroma and burst information from a composite color television signal. Bias networks force one differential transistor to be normally conductive and the other differential transistor to be normally nonconductive. An amplified chroma signal is available at the collector of the normally conductive transistor. During retrace, a single flyback pulse drives the differential transistors into their opposite conduction states, causing an amplified burst signal to be available at the collector of the normally nonconductive transistor. The circuit includes automatic chroma control and color killer action.

1. In a color television receiver for receiving a composite color television signal including a color reference burst signal and a chroma information signal, said burst signal and said chroma signal occurring at different points in time, a circuit for separating and amplifying both said burst signal and said chroma signal, comprising: 2. The circuit of claim 1 wherein said common means comprises a third amplifying means having a first electrode, a second electrode, and an output electrode, means coupling said output electrode of said third amplifying means to said commonly connected first electrodes of said first amplifying means and said second amplifying means, means coupling o

ne of said first and second electrodes of said third amplifying means to a reference potential, and means coupling the other of said first and second electrodes of said third amplifying means to a source of said burst signal and said chroma signal, whereby said common means forms a common current source for said first and second amplifying means. 3. The circuit of claim 2 including ACC means for developing a control signal for automatic chroma control of the color television receiver, and means coupling said control signal to said third amplifying means to control the current flow therethrough in proportion to said control signal. 4. The circuit of claim 2 wherein said first amplifying means and said second amplifying means each comprise a transistor having emitter, base, and collector electrodes corresponding to said first, second, and output electrodes, respectively, said common connecting means and said bias means causing said transistors to form a common emitter driven, differential operating amplifier. 5. The circuit of claim 4 wherein said third amplifying means comprises a transistor having emitter, base and collector electrodes corresponding to said first, second and output electrodes, respectively, whereby the collector electrode of said third amplifying means drives the emitter electrodes of said first and second amplifying means. 6. The circuit of claim 1 including a source of color killer signal generated when the color television receiver is receiving a black-and-white transmission, and said bias means includes means responsive to said color killer signal for biasing the differential amplifying means to cause said second amplifying means to be substantially nonconductive. 7. The circuit of claim 6 wherein said second amplifying means includes a semiconductor junction, and said color killer signal responsive means couples said color killer signal to the semiconductor junction with a polarity to back bias the semiconductor junction. 8. The circuit of claim 1 including deflection and high voltage means in said color television receiver for generating a flyback pulse occurring when said color reference burst signal is present, and said control means couples the flyback pulse to one of the first and second amplifying means to cause said differential amplifier to switch conduction states, said flyback pulse corresponding to said control signal. 9. The circuit of claim 8 wherein said first amplifying means includes a semiconductor junction, and said control means couples said flyback pulse to the semiconductor junction of said first amplifying means with a polarity to forward bias said semiconductor junction.

Description:

BACKGROUND OF THE INVENTION

This invention relates to a combined separator and amplifier circuit used in a color television receiver for deriving separate, amplified burst and chroma signals.

In a color television receiver, a separator and amplifier circuit is necessary to derive burst and chroma signals from a composite color television signal. Circuits are known which combine the function of a separator and an amplifier into a single stage. Typically, such circuits require a pair of flyback pulses to separately and alternately enable a burst channel and a chroma channel. For example, it has been known to drive a split-pentode vacuum tube with a pair of opposite going flyback pulses in order to alternately enable and disable chroma and burst channels connected to the pair of plates of the pentode.

Prior combined separator/amplifier circuits for deriving chroma and burst signals have a number of disadvantages. Some circuits require two flyback pulses of different polarity. Also such prior circuits have not been suitable for incorporation into linear integrated circuits. In addition, these circuits have been relatively complex, and not readily adapted for use with automatic chroma control and color killer action.

SUMMARY OF THE INVENTION

In accordance with the present invention, an improved separator/amplifier circuit uses a single differential amplifier to derive separate, amplified burst and chroma signals. Only a single flyback pulse is required to operate the circuit, and automatic chroma control and color killer action can easily be added with no increase in components or complexity. The circuit is readily adapted to linear integrated circuit techniques, and is of simple design and straightforward operation.

One object of this invention is to provide an improved chrominance and burst separating and amplifying circuit which operates as a differential amplifier.

Further objects and features of the invention will be apparent from the following description, and from the drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of a color television receiver incorporating a novel chroma and burst separator and amplifier; and

FIG. 2 is a schematic diagram of the chroma and burst separator and amplifier shown in block form in FIG. 1.

DESCRIPTION OF THE PREFERRED EMBODIMENT

While an illustrative embodiment of the invention is shown in the drawings and will be described in detail herein, the invention is susceptible of embodiment in many different forms and it should be understood that the

present disclosure is to be considered as an exemplification of the principles of the invention and is not intended to limit the invention to the embodiment illustrated.

Turning to FIG. 1, a color television receiver is illustrated in which an incoming composite color television signal is received by an antenna 10 and coupled to conventional RF and IF amplifying stages 12. The amplified IF signal is coupled to a video detector 13 in order to reproduce the modulating video information which includes a luminance or Y signal, a chrominance or chroma signal modulated on a 3.58 megahertz carrier, and a 3.58 megahertz burst signal which is transmitted during the blanking interval for each scanning line.

A video amplifier 15 amplifies the luminance or Y signal and couples it to a tri-color cathode ray tube or CRT 17 through a delay line 18. A deflection and high voltage circuit 20, responsive to the output of video amplifier 15, derives the horizontal and vertical scanning signals for CRT 17. During the retrace time period, a flyback pulse for blanking the video display is generated from the horizontal output transformer in circuit 20, and appears on a line 21.

The chroma information signal modulated on the 3.58 megahertz carrier, and the 3.58 megahertz burst signal, is coupled through a chroma take-off circuit 22, such as a chroma bandpass filter, and via output line 23 to the applicant's novel combined chroma and burst separator/amplifier 25, shown in detail in FIG. 2. Circuit 25 provides, on a chroma output line 27, a separated and amplified chroma signal which is coupled to a color demodulator and matrix 30 in order to derive three color difference signals R-Y, B-Y, and G-Y for driving the CRT 17. Circuit 25 also has a burst output line 32 on which an amplified burst reference signal is coupled to a conventional injection locked oscillator 34 which generates oscillatory signals coupled to the color demodulator and matrix 30 for the purpose of demodulating the chroma signal.

The injection locked oscillator 34 also generates, during reception of a black-and-white transmission, a color killer signal which is coupled to a color killer amplifier 36. Amplifier 36 has an output line 37 which couples a color killer voltage to the circuit 25. In addition, oscillator 34 further generates an automatic chroma control or ACC voltage, on an output line 39, which is coupled to circuit 25. While the color killer and ACC signals have been illustrated as being derived from an injection locked oscillator, it will be appreciated that any conventional circuit may be used to derive these signals. By way of reference, a suitable injection locked oscillator which derives color killer and ACC voltages is shown in U.S. Pat. No. 2,982,812, issued May 2, 1961 to R. N. Rhodes et al.

In the block diagram of the color television receiver, certain additional circuits of known construction have not been illustrated, as they are not necessary for an understanding of the present invention. Other conventional arrangements for a color television receiver can be utilized, as desired. For example, the chroma take-off circuit 22 may include cascaded video amplifiers having an output directly coupled to the circuit 25. In such an event, the necessary bandpass filters would be added to the circuit 25, rather than being located in block 22.

In FIG. 2, the novel combined chroma and burst separator/amplifier circuit 25 is illustrated in detail. The circuit comprises a single differential amplifier having a pair of NPN transistors 50 and 51 coupled to a common current source formed by a third NPN transistor 52. The emitter electrodes of both transistors 50 and 51 are tied together and are in common with the collector electrode of transistor 52. The collector electrode of transistor 50 is coupled through a tuned tank consisting in parallel of an inductor 55, a capacitor 56, and a resistor 57 located between the collector electrode and a source of B+ voltage, such as 35 volts DC. The junction between the tank and the collector electrode of transistor 50 forms the burst output line 32. The collector electrode of transistor 51 is connected to a similar tuned tank consisting in parallel of an inductor 60, capacitor 61, a resistor 62 located between the collector electrode and the same source of B+. The chroma output line 27 is located between the tank and the collector electrode of transistor 51.

In order to bias the pair of transistors 50 and 51 in a differential or alternate manner, the base electrode of transistor 50 is connected through a coupling capacitor 67 to the flyback pulse line 21 which has, during retrace time, a positive going flyback pulse 69 thereon having a peak amplitude of 10 volts. The base electrode of transistor 50 is also coupled through a resistor 70 to a source of reference potential or ground 72. The base electrode of transistor 51 is coupled to ground 72 through the parallel combination of a resistor 75 and a capacitor 76. The base electrode is also directly coupled to the color killer amplifier output line 37.

Common current source transistor 52 has its emitter electrode coupled to ground 72 through a parallel resistor 80 and capacitor 81. The base electrode of transistor 52 is similarly shunted to ground 72 through a resistor 83, and is coupled to the chroma and burst input line 23 through a coupling capacitor 85. The ACC output line 39 is directly connected to the base electrode of transistor 52.

In operation, the bias voltages are selected to cause transistor 51 to be normally conductive and thereby amplify the chroma information signal. When the positive going flyback pulse 69 is applied to the base of transistor 50, it drives transistor 50 into conduction. Since transistors 50 and 51 operate as a differential pair, the conduction of transistor 50 drives transistor 51 to cut-off, thereby terminating the chroma output signal on the chroma output line 27. At the same time, the signal from the current source 52, which now consists of burst information, is amplified by the conducting transistor 50 and appears on the burst output line 32.

The differential amplifier including current source 52 is very suitable for incorporation into a linear integrated circuit. By using a simple differential amplifier, the burst is separated from the chroma, and both signals are separately amplified. In one embodiment which was constructed, the gain of the chroma channel including transistor 51 was approximately 13, and the gain of the burst channel including transistor 50 was approximately 16.

The gains of transistors 50 and 51, and therefore the resulting collector currents, can be varied by controlling the base bias of transistor 52. Therefore, automatic chroma control (ACC) can readily be provided by applying to the base of transistor 52, via ACC output line 39, a voltage proportional to the burst amplitude. Since the burst amplitude is also varied, a closed loop ACC circuit is formed.

Color killer action is provided by coupling a negative cut-off or back bias to the base-emitter semiconductor junction of transistor 51, in the absence of burst. Such a negative cut-off voltage is available on the killer output line 37 from the color killer amplifier.

If closed loop ACC was not desired, the connection of output line 39 to the base of transistor 52 can be replaced with a resistor (not illustrated) coupled to a B+ source. If the B+ source had a DC voltage of 35 volts, for example, then the replacement resistor could have a value of 12 kilohms, and the resistor 83 could have a value of 560 ohms. If color killer action was not desired, the output line 37 coupled to the base of transistor 51 can be replaced with a resistor (not illustrated) coupled to the same B+ source. Again, if the B+ source had a DC value of 35 volts, then the replacement resistor could have a value of 220 kilohms, and the resistor 75 could have a value of 33 kilohms. The last named resistors form a voltage divider which bias transistor 51 normally into conduction. This in turn drives transistor 50, in which resistor 70 could have a value of 33 kilohms, into nonconduction in the absence of a flyback pulse. When color killer and ACC are to be incorporated in the circuit 25, then the color killer amplifier and the source of the ACC signal, respectively, should be construed to provide the same biasing as described above.

Circuit 25 can be modified in various ways without departing from the present invention. For example, the circuit could be connected so that the flyback pulse was coupled to transistor 51 in order to drive it nonconductive, rather than the illustrated circuit in which the flyback pulse is coupled to transistor 50 in order to drive it conductive. Similarly, the flyback pulse can be coupled to either the base or emitter of transistors 50 and 51, with a polarity to either forward bias or reverse bias, respectively, the base-emitter semiconductor junction in each transistor 50 and 51. Other changes will be apparent to those skilled in the art.

NORDMENDE SPECTRA COLOR L2UT CHASSIS FFS 772.535.A 772.536.A (4.570.D)  PAL-TYPE COLOR SIGNAL PROCESSING

Burst components of PAL-type encoded signal are retained with modulated subcarrier components as they are processed in 1H delay line assembly and delivered to respective demodulators. Reference oscillation phase to which R-Y demodulator responds is effectively reversed every other line, in response to PAL switch apparatus, in order to provide desired R-Y output in successive lines. Reference oscillation phase to which B-Y demodulator responds is alternated by quadrature switch apparatus between B-Y phase (applied throughout each line interval) and R-Y phase (applied during each inter-line blanking interval). A first gating circuit, coupled to the output of the B-Y demodulator, selects that portion of the B-Y demodulator output developed during the burst interval for passage to integrating and amplifying means in order to develop an AFPC voltage for phase control of the local reference oscillator. A second gating circuit, coupled to the output of the R-Y demodulator, selects that portion of the R-Y demodulator output developed during the burst interval for passage to ACC and color killer circuitry. During color operation (enabled state of bandpass chrominance amplifier) the ACC circuiry develops a control current from the second gating circuit output that adjusts the chrominance amplifier gain in a direction appropriate to maintaining burst amplitude substantially constant at a level set by a manual chroma control. The color killer enables the chrominance amplifier for color operation only when the gated R-Y output indicates by its amplitude the presence of a burst in the received signal and by its polarity the correct switching mode for the PAL switch. Unless such circumstances are present, the color killer disables the chrominance amplifier during each line interval; the killer is keyed, however, to enable the chrominance amplifier during each burst interval so that recovery from the disable state may be effected when appropriate. The color killer circuitry also passes a reset pulse to the PAL switch in the absence of a correct mode indication in the gated R-Y output. The color killer circuitry further serves to control the effectiveness of a subcarrier trap for the receiver's luminance channel, removing the trap during line intervals of monochrome operation.

1. In apparatus for processing PAL-type encoded color television signals, the combination comprising: 2. Apparatus in accordance with claim 1, also including: 3. Apparatus in accordance with claim 2, also including: 4. Apparatus in accordance with claim 2, also including 5. Apparatus in accordance with claim 2, wherein said second reference oscillation supplying means includes means for reversing the phase of the supplied reference oscillation in alternate line intervals, and wherein said apparatus also includes: 6. Apparatus in accordance with claim 5, also including a source of line rate triggering pulses; and 7. Apparatus in accordance with claim 6, also including:

Description:

This invention relates generally to color television signal processing systems, and, particularly, to novel and improved systems for processing color television signals of the PAL type.

In a color television receiver responding to a PAL transmission, the video signal output of the receiver's video detector includes, in addition to a wideband luminance component, a chrominance component in the form of a modulated subcarrier, and representing the summation of (a) the sideband products of the modulation of a subcarrier wave of fixed frequency and a first given phase by blue color-difference (B-Y) signals, and (b) the sideband products of the modulation of a subcarrier wave of the same fixed frequency, but with a quadrature phase relation to the first given phase, by red color difference (R-Y) signals, the second phase, however, being shifted by 180° in successive line intervals. The video signal, moreover, includes a color synchronizing burst component occurring during the inter-line blanking interval, incorporated in the transmission with a fixed amplitude and fixed (subcarrier) frequency, but alternating in phase in successive blanking intervals ±45° about a -(B-Y) phase (thereby corresponding to the summation of a fixed amplitude, constant-phase -(B-Y) burst component and a line-by-line phase reversing R-Y burst component of comparable fixed amplitude).

In a widely used approach to the processing of such detector PAL signals, the following functions are performed: A bandpass chrominance channel provides frequency selective amplification of the subcarrier sideband components, to the exclusion of low frequency luminance signals. The selectively amplified signals are applied to a 1H delay line assembly to develop two outputs respectively corresponding to an additive combination of undelayed and delayed signals, and a subtractive combination of undelayed and delayed signals. One output (in which the B-Y components for successive line intervals reinforce, whereas the R-Y components for successive line intervals mutually cancel) is supplied to a B-Y demodulator, while the other output (in which the R-Y components for successive line intervals reinforce, whereas the B-Y components for successive line intervals mutually cancel) is supplied to a R-Y demodulator. Each demodulator functions as a synchronous detector, controlled by the application of the appropriate phase of subcarrier frequency oscillations of fixed amplitude from a local reference oscillator. The reference phase applied to the B-Y demodulator is constant line-to-line, whereas the reference phase applied to the R-Y demodulator is shifted by 180° in successive line intervals. A takeoff for the burst component of the received signal is provided at a point in the chrominance channel prior to the delay line assembly, with appropriately gated apparatus extracting the burst component alone for amplification and delivery to a phase detector for comparison with an output of the local reference oscillator. An AFPC control voltage derived from the phase detector serves to lock the oscillator in a fixed phase relationship to the average phase of the "swinging" burst. Information derived from the separated burst is also used in performance of color killer and automatic chroma control (ACC) functions (determining the enabling or disabling of the chrominace channel, and the relative gain thereof when enabled). The burst component is eliminated from the chrominance signal delivered to the delay line assembly.

In accordance with the principles of the present invention, novel approaches to PAL color signal processing are contemplated which depart, in many regards, from the above-described widely used approach. Pursuant to the principles of the present invention, burst separation prior to delay is not effected, a separate burst amplifying channel and separate AFPC phase detector are not employed, and burst suppression is not effected for the signal delivered to the 1H delay line assembly. Rather, the burst is retained in the signal delivered to the 1H delay line assembly, and the respective B-Y and R-Y components of the burst pass to the respective demodulators. The B-Y demodulator then serves a dual function: as the B-Y demodulator during line intervals, and as an AFPC Phase detector during interline burst intervals. The phase of reference oscillations supplied to the B-Y demodulator is switched from its normal B-Y phase to an R-Y phase between line intervals, so that the polarity of the demodulator output during a burst interval is indicative of the direction of departure from correct phase relationship between local oscillator and incoming signal. A gating circuit, coupled to the output of the B-Y demodulator, selects that portion of the B-Y demodulator output developed during the burst interval for passage to an integrating and amplifying means in order to develop an AFPC voltage to control the local reference oscillator.

In accordance with further aspects of the present invention, the R-Y demodulator also serves a dual function: as the R-Y demodulator during line intervals, and as a synchronous in-phase detector of burst amplitude during the inter-line burst intervals. A second gating circuit, coupled to the output of the R-Y demodulator, selects that portion of the R-Y demodulator output developed during the burst interval for passage to automatic chroma control (ACC) and color killer circuitry. During color operation (enabled state of bandpass chrominance amplifier) the ACC circuitry develops a control current from the second gating circuit output that adjusts the chrominance amplifier gain in a direction appropriate to maintaining burst amplitude substantially constant at a level set by a manual chroma control. The color killer enables the chrominance amplifier for color operation only when the gated R-Y output indicates by its amplitude the presence of a burst in the received signal and by its polarity the correct switching mode for the PAL switch (i.e., for the reference phase reversing switch associated with the R-Y demodulator). Unless such circumstances are present, the color killer disables the chrominance amplifier during each line interval; the killer is keyed, however, to enable the chrominance amplifier during each inter-line interval so that recovery from the disabled state may be effected when appropriate.

In accordance with still further aspects of the present invention, the color killer circuitry may serve several additional functions, viz.: (a) passing a reset pulse to the PAL switch apparatus, in the absence of a correct mode indication in the gated R-Y output (so that PAL switching mode synchronization may be realized; and (b) controlling the effectiveness of a subcarrier trap for the receiver's luminance channel, removing the trap during line intervals of monochrome operation.

An object of the present invention is to provide novel and improved signal processing apparatus for PAL-type color television signals.

Other objects and advantages of the present invention will be readily apparent to those skilled in the art upon a reading of the following detailed description and an inspection of the accompanying drawings in which:

FIG. 1 is a block diagram illustration of a portion of a color television receiver incorporating color signal processing apparatus embodying the principles of the present invention;

FIG. 2 depicts schematically illustrative apparatus for performing the AFPC function in the system of FIG. 1;

FIG. 3 depicts schematically illustrative apparatus for performing the ACC function in the system of FIG. 1; and

FIG. 4 depicts schematically illustrative apparatus for performing the color killer (and associated PAL switch resetting, and color subcarrier trap switching) functions in the system of FIG. 1.

In FIG. 1, a portion of a PAL color television receiver, incorporating an embodiment of the present invention, is illustrated. The video detector 11 recovers a PAL encoded signal from the output of the receiver's intermediate frequency amplifier (not illustrated). The detector output is applied to a video amplifier 15 via a manual contrast control 13, which is bypassed by a burst circuit 14.

The manual contrast control 13 provides a facility for adjustment of the peak-to-peak magnitude of the video signals delivered to amplifier 15; however, the bypass circuit 14 permits the color synchronizing burst component to pass to amplifier 15 without being affected by contrast control adjustment. This arrangement ensures that contrast control adjustment does not introduce an undesired change in saturation of the image colors; i.e., the contrast control provides concomitant adjustments of the luminance and chrominance components, but does not disturb the burst component amplitude (to which subsequent ACC circuitry is responsive).

The output of video amplifier 15 is applied to a wideband luminance channel, including a luminance amplifier (not illustrated), and also, via chroma takeoff circuitry 17, to a chrominance channel, including a gain controlled bandpass amplifier 19. The chroma takeoff circuitry 17 provides a frequency selective input for the chrominance channel, passing the color subcarrier sideband components, to the substantial exclusion of low frequency luminance components; the chroma takeoff circuitry 17 also functions as a subcarrier trap for the luminance channel, significantly reducing the response of the luminance channel to signal frequencies in the vicinity of the color subcarrier. Desirably, the effectiveness of the trapping function is controlled as a function of whether the signal received is a monochrome or color transmission, with trapping eliminated in the former instance; the manner in which such trapping control is effected with be subsequently described.

The output of bandpass amplifier 19 is supplied to a 1H delay line assembly 21, which provides a pair of outputs representing additive and subtractive combinations of delayed and undelayed signals. At output terminal U of the delay line assembly 21, a combination is provided in which the B-Y components of succesive lines reinforce, whereas the shifting R-Y components tend to cancel; this output is supplied to an input terminal (35) of a B-Y demodulator 30. At a second output terminal (V) of the delay line assembly 21, a signal combination is provided in which the R-Y components of successive lines reinforce, whereas the B-Y components tend to cancel; this output is supplied to an input terminal (45) of an R-Y demodulator 40.

Each of the demodulators 30 and 40 function as a synchronous detector, heterodyning the respective delay line assembly output with unmodulated reference oscillations, of subcarrier frequency and respectively appropriate phase. Illustratively, each demodulator is of a type having (1) a pair of output terminals at which appear respective opposite polarity versions of the color-difference signal product of demodulation, and (2) a pair of reference oscillation input terminals with opposing effects on the polarity of the demodulator outputs.

The source of reference oscillations for the demodulators is reference oscillator 65, operating at the subcarrier frequency (e.g., 4.43 MHz.) and subject to phase control in a manner to be described. An output of oscillator 65 is applied to a quadrature switch 67, controlled by a horizontal blanking pulse input, the switch serving to alternately deliver (a) reference oscillations in a B-Y phase (during each line interval to reference input terminal 31 of demodulator 30, and (b) reference oscillations in a R-Y phase (during each inter-line blanking interval) to reference input terminal 33 of demodulator 30.

The B-Y component output of delay line assembly 21 is thus subject to in-phase synchronous detection during each line interval to a provide a B-Y color-difference signal output at terminal 37, and a -(B-Y) color-difference signal output at terminal 39.

At this point, it is appropriate to note that the color synchronizing burst portion of the video signal amplified in video amplifier 15 has been retained with the line interval subcarrier sideband components throughout the chrominance channel (17, 19, 21). The constant phase -(B-Y) component of the swinging burst thus appears in the signal output at delay line assembly terminal U. This component, accordingly, is subject to quadrature synchronous detection in demodulator 30, in view of the delivery by quadrature switch 67 of reference oscillations in the R-Y phase to the (inverting) reference input terminal 33.

B-Y demodulator 30 thereby conveniently serves as the equivalent of the burst phase detector employed in the usual AFPC arrangement. A B-Y burst interval gate 61, activated by an appropriately timed burst gate pulse, is coupled to output terminal 37, and serves to pass the portion of the demodulator output developed during the burst interval, i.e., the result of phase detection of the -(B-Y) burst component, to an AFPC amplifier 63. An integrated and amplified version of the gated output, with amplitude and polarity respectively indicative of degree and direction of departure from correct phase relationship between oscillator and received signal, is supplied by amplifier 63 to a suitable phase control element of oscillator

Reference oscillations in the R-Y phase are delivered in a linewise alternating fashion from the PAL switch apparatus 69, controlled by a horizontal blanking pulse input, to the respective reference input terminals (noninverting terminal 41 and inverting terminal 43) of R-Y demodulator 40. If the switching mode of the PAL switch 69 is the correct one, the alternating polarity line interval R-Y component at terminal V of delay line assembly 21 will be subject to in-phase detection by demodulator 40 in the desired fashion, developing a R-Y color-difference signal at output terminal 47, and a -(R-Y) color-difference signal at output terminal 49. The latter output signal is supplied, along with the -(B-Y) output of demodulator 30, to a matrix circuit 50, for development of a third (G-Y) color-difference signal.

An R-Y burst component also appears in the signal input to terminal 45 of the R-Y demodulator 40, and is subject to in-phase synchronous detection when the correct switching mode is in effect. An R-Y burst interval gate 71, coupled to output terminal 47 of demodulator 40, is gated by a suitably timed burst gate pulse to pass that portion of the R-Y demodulator output developed during the burst interval to a pair of circuits (ACC amplifier circuit 73 and keyed color killer circuit 77).

The ACC (automatic chroma control) circuitry 73 functions to integrate and amplify the gated R-Y demodulator output in order to develop a control current for controlling the gain of bandpass amplifier 19. The gain control is effected in a direction to oppose spurious variations in the amplitude of the R-Y burst component (which is transmitted with fixed amplitude), thereby to minimize spurious variations in the chrominance signal amplitude that may result in incorrect saturation (chroma) of the displayed image colors. A facility for manual adjustment of the saturation of the image colors is provided in the form of a manual chroma control 75, which supplies an adjustable reference potential to ACC amplifier 73 for comparison with the gated R-Y demodulator output from gate 71 to determine the control current magnitude.

The keyed color killer circuit 77 controls the enabling and disabling of the bandpass amplifier 19, responding to the amplitude and polarity of the gated R-Y demodulator output from gate 71. The amplifier 19 is enabled, permitting amplification thereby of the line interval subcarrier sideband components, when the gate 71 output amplitude indicates presence of a color transmission with a burst of adequate amplitude for synchronization, and when gate 71 output polarity indicates operation of the PAL switch in the correct switching mode. In the absence of such circumstances, the color killer circuit 77 holds the amplifier in a disabled state; the color killer circuit is, however, keyed in response to a horizontal blanking pulse input in a manner enabling operation of the amplifier 19 during the burst interval to ensure the ability of the system to recover from the disabled state when appropriate. Alteration of the PAL switch operation to a correct mode is also facilitated by the keyed color killer circuit 77, which permits passage of a reset pulse to the PAL switch apparatus, when circuit 77 holds amplifier 19 in a disabled state.

The keyed color killer circuit 77 also serves the previously mentioned trap switching function, causing circuit 17 to be effective as a subcarrier trap for the luminance channel when amplifier 19 is enabled, and to be ineffective as a subcarrier trap when amplifier 19 is disabled.

FIG. 2 provides, in schematic detail, an illustration of particular circuit arrangements that may advantageously be employed for portions of the FIG. 1 system (and in particular, those portions associated with oscillator synchronization: B-Y demodulator 30, B-Y burst interval gate 61, AFPC amplifier 63, reference oscillator 65, and quadrature switch 67).

The B-Y demodulator 30 in FIG. 2 employs six transistors (301, 302, 303, 304, 305 and 306 conveniently realized in integrated form on a common monolithic integrated circuit chip 300) arranged in a cross-coupled differential amplifier pair configuration. In the circuit arrangement, the emitters of transistors 301 and 302 are joined directly and returned to a bias supply (e.g., - 15 volts) via the collector-emitter path of transistor 303 and emitter resistor 310; likewise, the emitters of transistors 304 and 305 are joined directly and returned to the bias supply via the collector-emitter path of transistor 306 and the common emitter resistor 310.

The base of transistor 301 serves as the non-inverting reference input terminal 31 of the demodulator; the base (terminal 31') of transistor 304 is directly linked thereto. The base of transistor 302 serves as the inverting reference input terminal 33 of the demodulator the base (terminal 33') of transistor 305 is directly linked thereto. The collector of transistor 301 serves as the B-Y color-difference signal output terminal 37 of the demodulator; the collector (terminal 37') of transistor 305 is directly linked thereto. The collector of transistor 302 serves as the -(B-Y) color-difference signal output terminal 39 of the demodulator; the collector (terminal 39') of transistor 304 is directly linked thereto.

The base of transistor 303 serves as the modulated subcarrier input terminal 35 of the demodulator, receiving the signals appearing at terminal U of the delay line assembly 21 (FIG. 1). The base of transistor 306 is effectively held at AC ground potential by suitable bypassing.

The signal output appearing at terminal 37, free of subcarrier frequency components due to cancellation effects from the contributing transistors (301, 305), is applied to emitter follower transistor 307. A B-Y color-difference signal output is available at the emitter of transistor 307 for combination with a luminance component in the matrix and display portion of the receiver (not illustrated).

The emitter of transistor 307 is also linked by a path including resistor 613 and capacitor 614 to the junction (J) of oppositely poled electrodes of a pair of diodes 611 and 612. The collector-emitter path of a gate transistor 610 short circuits junction J to ground throughout each line interval. During each burst interval, however, the short circuit is removed, as transistor 610 is cut off by the positive-going pulse portion b of a gating waveform applied to its base. The cutoff of transistor 610 during each burst interval permits conduction by one of the diodes (611 or 612, depending upon the polarity of the burst interval output of demodulator 30) to charge the respectively associated capacitor (615 or 616) to a level dependent upon the magnitude of the burst interval output of demodulator 30. Transistor 610 and associated circuitry thus performs the function of the B-Y burst interval gate 61 of the FIG. 1 system.

AFPC amplifier 63 includes a pair of transistors 631 and 633 disposed in a differential amplifier configuration, with the base of input transistor 631 coupled to respond to the potential across the charged capacitor (615 or 616). The integrated output of amplifier 63 appears across capacitor 635, coupled between the collector of output transistor 633 and ground.

Reference oscillator 65 employs a transistor 651 associated with reactive circuit elements in a Colpitts configuration, with the inductive circuit branch including a frequency determining crystal 653 in series with a variable capacitance diode 652. A resistor links the collector of AFPC amplifier output transistor 633 to the junction of crystal 653 and diode 652, whereby the reverse bias on diode (and hence its capacitance) is subject to variation in accordance with the integrated output of amplifier 63 in order to effect the desired frequency and phase synchronization.

The output of reference oscillator 65 is derived from the collector of transistor 651 and applied via an emitter follower transistor 655 to a reference oscillation feed point R. Quadrature switch apparatus 67 controls the application of reference oscillations from feed point R to respective reference input terminals of the B-Y demodulator 30.

Quadrature switch 67 employs a pair of switching transistors 675 and 676. Switching transistor 676 is normally conducting, but is cut off during each inter-line blanking interval by the neagive-going pulse portion n of a gating waveform applied to its base. In complementary fashion, switching transistor 675 is rendered conducting only during the inter-line blanking interval by the positive going pulse portion p of a gating waveform applied to its base.

The collector-emitter path of switching transistor 676 is connected between the demodulator reference input terminal 33 and ground, while the collector-emitter path of switching transistor 675 is connected between the demodulator reference input terminal 31 and ground. A resistor 674 links feed point R to reference input terminal 33. A resistor 671 in series with a coil 672 links feed point R to reference input terminal 31. A capacitor 673 is connected between reference input terminal 31 and ground, and is adjusted for series resonance with coil 672 at the reference oscillation frequency.

during each line interval, the conduction of switching transistor 676 short circuits reference input terminal 33 to ground, precluding the feeding of reference oscillations to that terminal. Switching transistor 675, however, is nonconducting each line interval, permitting the feeding of reference oscillations to terminal 31. Circuit elements 672 and 673 introduce a phase shift of 90° from the R-Y phase to which the oscillator output is held, so that the reference oscillations delivered during line intervals are at the B-Y phase.

During each inter-line blanking interval, the conduction of switching transistor 675 short circuits reference input terminal 31 to ground, precluding the feeding of reference oscillations to that terminal. Switching transistor 676, however, is nonconducting during each inter-line blanking interval, permitting the feeding of reference oscillations to terminal 33 in the R-Y phase.

FIG. 3 provides, in schematic detail, an illustration of particular circuit arrangements that may advantageously be employed for additional portions of the FIG. 1 system (particularly, those portions associated with automatic chroma control: R-Y demodulator 40, R-Y burst interval gate 71, ACC amplifier 73, manual chroma control 75, video amlifier 15, chroma takeoff 17, and bandpass amplifier 19).

The R-Y demodulator 40 employs six transistors (401, 402, 403, 404, 405 and 406) disposed on a monolithic integrated circuit chip 400, and arranged in a cross-coupled differential amplifier configuration identical to that previously explained for the B-Y demodulator 30.

The base of transistor 401 serves as the non-inverting reference input terminal 41 of the demodulator, the base (terminal 41') of transistor 404 is directly linked thereto. The base of transistor 402 serves as the inverting reference input terminal 43 of the demodulator; the base (terminal 43') of transistor 405 is directly linked thereto. The collector of transistor 401 serves as the R-Y color-difference signal output terminal 47 of the demodulator; the collector (terminal 47') of transistor 405 is directly linked thereto. The collector of transistor 402 serves as the -(B-Y) color-difference signal output terminal 49 of the demodulator; the collector (terminal 49') of transistor 404 is directly linked thereto.

The base of transistor 403 serves as the modulated subcarrier input terminal 45 of the demodulator, receiving the signals appearing at terminal V of delay line assembly 21 (FIG. 1). The base of transistor 406 is effectively held at AC ground potential by suitable bypassing.

The signal output appearing at terminal 47, free of subcarrier frequency components, is applied to emitter follower transistor 407. An R-Y color-difference signal output is derived from the emitter of transistor 407. A path, including, in series, a resistor 713, capacitor 714 and resistor 715 is also provided between the emitter of transistor 407 and the base of an additional emitter follower transistor 711. The emitter-collector path of a gating transistor 710 is connected between ground and the junction of capacitor 714 and resistor 715; the junction is short circuited to ground throughout each line interval by the conducting gate transistor 710. During each burst interval, however, the short circuit is removed, as transistor 710 is cut off by the positive-going pulse portion b of a gating waveform applied to its base. The cutoff of transistor 710 during each burst interval permits emitter follower transistor 711 to respond to the burst interval portion of the output of demodulator 40. Transistor 710 and associated circuitry thus performs the function of the R-Y burst interval gate 71 of the FIG. 1 system.

An output of emitter follower transistor 711 is applied to the keyed color killer circuit 77 (for which a detailed showing will appear in the subsequently described FIG. 4). ACC amplifier 73 responds to another output of emitter follower transistor 711 in a manner to be now described.

ACC amplifier 73 includes a pair of cascaded amplifier stages incorporating transistors 730 and 731. The emitter of the ACC input transistor is connected to the adjustable tap of a potentiometer 750, the end terminals of which are connected to respective bias supply terminals of opposite polarity (e.g., -15 volts and + 15 volts). The base of ACC input transistor 730 is connected to the emitter of emitter follower transistor 711 by an isolating diode 712, rendered conducting only during each burst interval by the positive-going pulse portion of a gating waveform applied to the transistor 730 base. The degree of conduction, if any, by transistor 730 during the gating interval (i.e., the burst interval) is dependent upon a comparison of the magnitude and polarity of the gated R-Y demodulator output with the magnitude and polarity of the emitter bias selected by adjustment of potentiometer 750 (which, as will be shown, performs the function of the manual chroma control 75 of the FIG. 1 system). Capacitive feedback between collector and base of transistor 730 reduces high frequency response, to prevent high frequency noise in the gated demodulator output from affecting the ACC voltage to be developed.

When the gated R-Y demodulator output is more positive than the selected emitter bias potential, conduction by ACC input transistor 730 in turn drives the (complementary type) ACC output transistor 731 into conduction, charging filter capacitor 732 in its collector circuit. The voltage developed across capacitor 732, representing an integration of successive output pulses of transistor 731, causes a current to flow via the series combination of resistor 735, diode 733, resistor 736 and diode 192 into the base of the amplifier transistor 190 of the bandpass amplifier 19 (to be described in detail subsequently).

When the difference between the gated demodulator output and the selected emitter bias potential is sufficiently small, the voltage across the filter capacitor 732 will be sufficiently small that diode 733 will be reverse biased, permitting no ACC control current flow into the transistor 190 base, leaving transistor 190 in its maximum gain condition determined by fixed biasing parameters. When the burst component delivered to the R-Y demodulator is large enough to increase the gated demodulator output above the aforementioned level at which diode 733 is cut off, a control current will flow into the base of transistor to reduce its gain appropriately.

The above-described ACC action requires the condition that the switching mode of the PAL switch 69 (FIG. 1) controlling the feeding of reference oscillations to demodulator 40 is the correct one, so that the polarity of the gated demodulator output is correct (positive). Also required is that the keyed color killer circuit 77 has placed amplifier 19 in its enabled state for color operation. While a more detailed explanation of keyed color killer circuit 77 will be presented subsequently in connection with FIG. 4, a portion of the killer circuit (comprising transistor 790, which is held cut off when conditions are correct for color operation, and which is conducting during line intervals when conditions are otherwise) has been illustrated in FIG. 3 to permit a full showing of bandpass amplifier 19.

Bandpass amplifier 19 receives signals from an output of video amplifier 15, the latter incorporating an amplifier transistor 150, disposed in grounded base configuration and receiving at its emitter video signals from contrast control 13 and burst bypass circuit 14 (FIG. 1). An output lead from the collector of transistor 150 couples signals therefrom to suitable luminance amplifier circuitry (not illustrated).

The collector of transistor 150 is also connected, by means of the series combination of capacitor 170, coil 171 and the previously mentioned diode 192, to the base of the bandpass amplifier transistor 190. Coil 171 is adjusted for series resonance with capacitor 170 at the subcarrier frequency. A pair of resistors 194 and 195 are connected in series across diode 192, and the emitter-collector path of color killer transistor 790 is connected between negative supply terminal (e.g., -15 volts) and the junction of resistors 194 and 195.

A diode 791 is shunted across the base-emitter path of bandpass amplifier transistor 190, with poling opposite to that of the base-emitter diode. A tuned load is provided for amplifier transistor 190, the primary winding of bandpass transformer 191 being connected in the collector circuit of transistor 190; the secondary winding of transformer 190 couples the amplfier output to the delay line assembly 21 of the FIG. 1 system. DC feedback resistor 193 is coupled between a point in the collector circuit of transistor 190 and the junction of coil 171 and diode 192.

During color operation (when killer transistor 790 is cut off), diode 192 and the base-emitter diode of transistor 190 are forward biased and provide a low impedance return to ground for the series resonant circuit 170, 171. The latter then functions as a frequency selective input circuit for amplifier 19, and also as a subcarrier trap for the circuitry feeding signals to the luminance amplifier (thereby performing the functions of the chroma takeoff and subcarrier trap apparatus 17 of FIG. 1 system). Under these color operation conditions, shunt diode 791 is biased off, and the conductive state of diode 192 permits the feeding of a variable control current from ACC amplifier 73 to the transistor 190 base when appropriate.

When color killer transistor 790 is conducting, however, a substantial change in the biasing conditions for transistor 190 and associated components is brought about. Conduction of killer transistor 790 brings the junction of resistors 194 and 195 to a negative potential. reverse biasing diode 192 and forward biasing shunt diode 791. The reverse biasing of diode 192 blocks the passage of signals to transistor 190, and the conduction of diode 791 holds transistor 190 in a cutoff condition. No low impedance return to AC ground is provided for the series resonant circuit 170, 171, whereby its effectiveness as a subcarrier trap for the luminance channel is eliminated. Diode 734 is rendered conducting under the altered biasing conditions to preclude the ACC filter capacitor 732 from changing to a negative potential.

FIG. 4 provides, in schematic detail, an illustration of particular circuit arrangements that may advantageously be employed for further portions of the FIG. 1 system, particularly including the keyed color killer circuit 77 and the PAL switch apparatus 69. Also repeated in FIG. 4 are illustrative circuit arrangements for system components 15, 19 and 71 to aid in an explanation of the color killer operation.

As previously explained, the keying of gate transistor 710 into cutoff during each burst interval permits emitter follower transistor 711 to respond only to the burst interval portion of the output of the R-Y demodulator 40 (FIGS. 1 and 3). The emitter of transistor 711 is linked not only to the previously described ACC amplifier circuitry (FIG. 3) but also, via a path including compensating diode 770, to the base of feedback amplifier transistor 771.

The collector of amplifier transistor 771 is coupled by means of the series combination of storage capacitor 773 and diode 774 to the base of a succeeding amplifier transistor 776. The emitter-collector path of a gating transistor 772 is connected between ground and the junction of capacitor 773 and diode 774. Gating transistor 772 is rendered conducting during the burst interval only by the positive-going pulse portion b of the gating waveform applied to its base. The conduction of gating transistor short circuits one terminal of storage capacitor 773 to ground during the burst interval, so that the burst interval output of R-Y demodulator 40 is integrated by capacitor 773. During the succeeding line interval, when gating transistor 772 is cutoff, the voltage developed across capacitor 773 (charge reduction caused by the detected burst integration) is transferred via diode 774 to capacitor 775, connected between ground and the base of transistor 776.

Transistor 776 is disposed in a differential amplifier configuration with an additional amplifier transistor 777, the emitters of transistors 776 and 777 being returned to a negative bias supply terminal (e.g., -15 volts) via a common emitter resistor. The collector of transistor 776 is connected to a positive bias supply terminal (e.g., -15 volts) by means of a collector resistor 778. The collector of transistor 766 is also cross-coupled to the base of transistor 777 by means of resistor 779. Resistor 780 is connected between the base of transistor 777 and ground.

Due to the presence of cross coupling resistor 779, the differential amplifier has only two stable states. In the absence of a signal input to the base of transistor 776, transistor 777 is in saturation and transistor 776 is cutoff. However, when the gated R-Y demodulator output is such that a positive potential appears across capacitor 775 with adequate magnitude relative to a threshold determined by the divider 778, 779, 780, the differential amplifier switches to its other stable state in which transistor 776 is in saturation and transistor 777 is cutoff. The latter condition is established only when the received signal includes synchronizing bursts of adequate amplitude, reference oscillator 65 is properly synchronized in phase, and PAL switch 69 is operating in the correct mode.

A resistor 781 links the collector of transistor 777 to the base of transistor 783 (complementary in type to transistor 777); the base of the previously mentioned kiler transistor 790 (similar in type to transistor 777) is connected to a point in the collector circuit of transistor 783. When transistor 777 is cutoff (i.e., when conditions are correct for color operation, as indicated by the R-Y demodulator output during the burst interval). the other transistors of the complementary cascade chain (783, 790) are likewise driven to cutoff. As previously noted, the result of cutoff of transistor 790 is the forward biasing of diode 192 and the base-emitter path of band pass amplifier transistor 190, with the consequence that bandpass amplifier 19 is fully enabled and responds to signals selectively passed by chroma takeoff circuit elements 170, 171 and conducting diode 192; elements 170, 171 are also effective as a subcarrier trap for the luminance channel under these conditions.

When transistor 777 is in saturation, however, in the absence of an indication of correct operating conditions by the gated R-Y demodulator output, the other transistors of the complementary cascade chain (783,790) are also in saturation. The effects of conduction by killer transistor 790 have been previously described: cutoff of diode 192 to bar signal passage to the transistor 190 base and to eliminate the effectiveness of elements 170, 171 as a subcarrier trap, and forward biasing of diode 791 to hold transistor 190 in cutoff.

When killer transistor 790 is conducting to establish the disabled state for bandpass amplifier 19, thereby barring color operation, means must be provided to permit the system to recover from the disabled state when appropriate. For this purpose, a gating waveform, having a positive-going pulse portion p occurring during each inter-line blanking interval, is applied to the base of transistor 783 via a resistor 784, forward biasing the diode 782 (coupled across the base-emitter path of transistor 783 with opposite poling to that of base-emitter diode) during the blanking interval. The pulse application ensures that transistors 783 and 790 are cut off during each interline blanking interval, independent of the conducting state of transistor 777, whereby bandpass amplifier 19 is always in the enabled state for the burst component of a received signal (to be fed on to the demodulators to permit resumption of color operation when appropriate).

A negative-going blanking pulse waveform is developed in the collector circuit of transistor 783 (under color-off conditions) in response to the aforementioned pulse application. This waveform is passed by isolating diode 785 to the series combination of capacitor 786 and resistor 787, the junction of which elements is directly linked to the collector of transistor 776 (cut off during color-off conditions). A differentiated version of the negative-going pulse appears at the junction; the positive-going spike portion of the differentiated waveform, occurring at the end of the inter-line blanking interval, is passed via sterring diodes 696 and 697 to the PAL switch 69 as a reset pulse.

During color-on operation, the saturated state of transistor 783 precludes the inverted blanking pulse development. Additionally, the conduction of transistor 776 reverse biases the sterring diodes 696 and 697 to protect the PAL switch from spurious output variations in the collector circuit of transistor 783, should they occur.

The PAL switch apparatus 69 includes a bistable multivibrator, incorporating transistors 690 and 691 with conventional cross-coupling from collector to base. A triggering waveform, having a positive-going pulse portion p occurring during each inter-line blanking interval, is applied to a differentiating circuit formed by the series combination of capacitor 680 and resistor 681. The differentiated waveform appearing at the junction of elements 680, 681 includes positive-going spikes, occurring at the beginning of each inter-line blanking interval, which are passed by steering diodes 694 and 695 to the bases of the multivibrator transistors 690, 691 to effect triggering of the multivibrator between its stable states.

When the multivibrator is in one of its stable states, transistor 690 is heavily conducting while transistor 691 is cut off; in this state, switching transistor 692, complementary in type to transistor 690 and having its base coupled to a point in the collector circuit of transistor 690, is driven into conduction, while switching transistor 693, complementary in type to transistor 691 and having its base coupled to a point in the collector circuit of transistor 691, is driven into cutoff. The collector-emitter path of switching transistor 692 is directly connected between the noninverting reference input terminal 41 of R-Y demodulator 40 and ground, while the collector-emitter path of switching transistor 693 is directly connected between the inverting reference input terminal 43 of R-Y demodulator 40 and ground. Thus in the noted state of the multivibrator, conduction by switching transistor 692 precludes the feeding of R-Y phase reference oscillations in from feed point R to noninverting reference input terminal 41, whereas cutoff of switching transistor 693 permits the feeding of R-Y phase reference oscillations from feed point R to the inverting reference input terminal 43.

When the multivibrator is triggered to its other stable state, transistor 690 (and switching transistor 692) is dirven into cutoff, while transistor 691 (and switching transistor 693) is driven into conduction. In this state, R-Y phase reference oscillations are permitted to feed noninverting reference input terminal 41, but precluded from feeding inverting reference input terminal 43.

In the absence of reset pulse application from transistor 783, the trigger pulse application via diodes 694, 695 effects a line-by-line reversal of the effective angle of demodulation employed in the R-Y demodulator. When this line-by-line reversal is carried out in the incorrect mode, the reset pulse application permits alteration to the correct mode. It will be noted that when a monochrome signal, lacking a burst component, is received, continued reset pulse application ensures, with the consequence that the phase reversing effect will be overcome during successive line intervals to reduce the possibility of undesired "Hanover bar" type disturbances of the displayed monochrome image.

While specific circuit arrangements have been illustrated for the various components of the FIG. 1 system, it will be appreciated that these are given by way of example, and a variety of other specific circuit arrangements may be substituted therefor in carrying out the principles of the invention. It will also be appreciated that various portions of the system of FIG. 1 may be advantageously employed, with different techniques than those described employed in performing the remaining functions.

NORDMENDE SPECTRA COLOR L2UT CHASSIS FFS 772.535.A 772.536.A (4.570.D)  AUTOMATIC CHROMA GAIN CONTROL SYSTEM:

Cascaded first and second gain-controlled amplifiers are used in the chrominance channel of a color television receiver. The gain of the first amplifier is controlled by an ACC loop employing a noise-immune detector to detect color burst information. The gain of the second amplifier is controlled by the output of a peak detector which detects picture-interval information at the output of the second amplifier. An ACC system with improved performance during the reception of noisy signals results.

1. In a color television receiver an automatic chroma gain control system for processing input chroma signals having burst information and picture-interval information components, said system comprising, 2. In a color television receiver an automatic chroma gain control system as claimed in claim 1 wherein said peak detector is characterized by being: 3. In a color television receiver an automatic chroma gain control system as claimed in claim 1 wherein said threshold peak detector comprises: 4. In a color television receiver an automatic gain control system as claimed in claim 3 wherein said resistance is chosen large enough that the return to said quiescent charge condition when said semiconductor means is no longer biased into conduction requires a time longer than said period. 5. In a color television receiver an automatic chroma gain control system as claimed in claim 1 wherein, 6. In a color television receiver an automatic chroma gain control system as claimed in claim 1 including: 7. In a color television receiver an automatic gain control system as claimed in claim 1 including: 8. In a color television receiver including 1. a first chroma amplifier providing intermediate chroma signals at its output terminal and having its gain controlled by an automatic chroma gain control responsive to the burst information component of said intermediate chroma signals and 2. chroma demodulators having their input terminal adapted to receive signals to be demodulated, which signals if their peak excursions were excessively large could cause oversaturation to occur in said receiver, in combination therewith the improvement comprising: 9. In a color television receiver the improvement claimed in claim 8 wherein: 10. In a color television receiver the improvement as claimed in claim 9 wherein said peak detector and said manual chroma gain control means are embodied in circuitry comprising:

Description:

The present invention relates to color television receivers and more particularly to circuitry for providing improved automatic chroma control (ACC).

Automatic chroma control (ACC) is an automatic gain control system applied to a chroma amplifier in a color television receiver. The control system is conventionally responsive to burst information appearing in the horizontal blanking intervals of the chroma signal and acts to maintain the amplitude of the burst information in the output circuit of the chroma amplifier more nearly constant than at its input circuit. If each television broadcaster adheres to system standards concerning the relative levels of picture-chroma and burst information in its signals, the chroma signals will be maintained at the same color saturation level despite the viewer switching from one channel to another.

Too high a level of color saturation causes "oversaturation" -- at least on peaks of the color signals kinescope -- a condition in which the kinescope "blooms". "Blooming" is wheree the beam current in the kinescope increases so much that defocusing of the electron beam occurs and the color spot on the phosphor screen responsive to the electron beam is undesirably enlarged.

The ACC system desirably should provide for reducing chroma amplifier gain as the chroma signals become noisier, so that peaks of the combined chroma and noise signals will not cause oversaturation to occur. In an ACC system in which the burst information is detected by a noise-immune detector, unresponsive to noise accompanying the chroma signal, the chroma amplifier gain will not be reduced as the chroma signals become noisier. So undesirable oversaturation on peaks of noise is probable.

The detection of burst information for developing ACC signal may be done using a synchronous detector timed in response to the local color subcarrier source. The local color subcarrier source is itself synchronized with the incoming burst information which system may comprise, for example, automatic phase and frequency control (AFPC) or injection-locking of a crystal oscillator. Detection of the burst information by a synchronous detector provides an ACC substantially immune to noise signals accompanying the chroma signals to be controlled and will give rise to the problem of oversaturation during noisy signals.

However, in other ways a noise-immune ACC detector is desirable. It can provide, in combination with a simple threshold detector responsive to signals below a certain threshold level, for a noise-immune color killer circuit to disable the chroma demodulation processes during black and white television signal transmissions.

A noise-immune ACC detector is also advantageous when the local color subcarrier source is synchronized with burst information separated from chroma signal taken from the output circuit of the ACC'd chroma amplifier. This is because the burst information is not reduced in response to noisy signals, so synchronization of the local color subcarrier source is not consequently imparied. Further, a synchronous detector for developing ACC signal produces no gain reducing output until the local color source is brought into substantial synchronization with the burst information. This speeds the synchronization process.

Because of the advantages of using a synchronous detector for developing ACC signal, ways have been sought to augment its action with other circuitry to overcome its shortcomings. The picture-interval chroma information in the output signal of the ACC'd chroma amplifier may be detected to provide an ACC component signal to be added to the ACC component developed by the synchronous detector. This is disadvantageous to do when the local color subcarrier source is to be synchronized from the output signal of the ACC'd chroma amplifier, because the reduction of the signal during the reception of noisy signals impairs synchronization of the local color subcarrier source to the burst information contained therein.

Further, the detection of picture-interval chroma information to develop ACC information tends to produce control signals which are responsive to the chroma peaks of the broadcast scene, causing low-chroma scenes to have too high color saturation or high-chroma scenes to have too low color saturation. This is undesirable when strong, noise-free signals are being received.

An automatic chroma control system embodying the present invention includes a first chroma amplifier followed in cascade connection by a second chroma amplifier. The input circuit of the first amplifier is adapted to receive input chroma signals, having a burst information component and having a picture-interval information component, and provides in response thereto intermediate chroma signals at its output circuit. The second amplifier provides output chroma signals at its output circuit in response to intermediate chroma signals applied to its input circuit. A noise-immune detector means develops a control signal responsive to the amplitude of the burst information component of the input chroma signals. The control signal provided by the noise-immune detector means is applied to the first chroma amplifier to control its gain for chroma signals. A peak detector means develops a control signal responsive to peaks of the picture-interval component of the input chroma signals and accompanying noise. The control signal provided by the peak detector means is applied to the second chroma amplifier to control its gain for chroma signals.

When the noise level in the picture-interval component of the input chroma signals grows large enough to tend to cause peaks of the noisy output chroma signals to exceed the excursion permitted peaks of noise-free signals, the gain of the second amplifier is reduced by action of the peak detector means to maintain peaks of the noisy output chroma signals within the limits of excursion permitted peaks of noise-free signals. Accordingly, oversaturation during the reception of noisy signals is avoided. At the same time the noise immunity of the ACC of the first chroma amplifier is desirably unaffected.

The noise-immune detector is an amplitude detector in which the detector response for peaks of noise as compared to the response for the average level of burst information is less than that of a peak detector. Average detection, where the detector is responsive to the average energy of the signal peak detected rather than its peak energy, will provide for noise immunity since noise accompanying the burst information has a larger ratio of peak energy to average energy than the burst information itself does. Narrowing the bandwidth of the signals being admitted to an amplitude detector is an alternative or supplemental way to provide for noise immunity. Synchronous detection will afford additional noise immunity.

In a preferred embodiment of the present invention the peak detector means is provided with an offset threshold so that detection of peaks occurs only on peaks of the output chroma signals which exceed a certain threshold level, whereby the peak detection means is inoperative to reduce the gain of the second amplifier under conditions of reception by the television receiver of strong, noise-free television signals broadcast to proper standards.

In a further preferred embodiment of the present invention means are provided for manually controlling the gain of the second amplifier for chroma signals. The peak detector means operates to prevent oversaturation caused by setting the manual chroma gain control for too high gain.

The advantages of the present invention will be better understood from the detailed description of the drawings in which:

FIG. 1 is a block schematic of the present invention shown in a representative type of color TV receiver, and

FIG. 2 is a schematic of the cascaded first and second gain-controlled amplifiers and their associated circuitry as fabricated in integrated circuit form in a preferred embodiment.

Referring now to FIG. 1, television broadcast signals intercepted by an antenna 101 are applied to a "front end" 103 of the color television receiver comprising a tuner, mixer, intermediate-frequency amplifiers and video detector. Composite video signals from the video detector portion of "front end" 103 are applied as input signals to luminance circuitry 105 typically comprising trapping filters, contrast and brightness controls, and video amplifier stages. Output video signals from the luminance circuitry 105 and output color-difference signals from chroma demodulators 107 are combined and amplified in a color matrix and kinescope-driver amplifiers section 109. The output signals from the kinescope-driver amplifiers of the section 109 are red, green and blue drive signals which are applied to electrodes of a color kinescope 111. The color kinescope 111 is shown to have vertical magnetic deflection coils 113 and horizontal magnetic deflection coils 115.

Composite video signals from the video detector portion of the "front end" 103 are applied as input signals to a sync separator 117, which provides separated sync signals to a vertical sweep generator 119 and a horizontal sweep generator 121. The vertical sweep generator provides sweep signals to the vertical deflection coils 113; the horizontal sweep generator 121 provides sweep signals to the horizontal deflection coils 115.

Composite video signals from the video detector portion of the "front end" 103 are also applied to a chroma sidebands filter 123. Components of the composite video signals which are in the frequency range of the chroma sidebands, including those chroma sidebands, are selected by the filter 123 and applied as input signals to the gain-controlled amplifier 125. Output signals from the amplifier 125 are applied as input signals to another gain-controlled amplifier 127. Output signals from the amplifier 127 are applied to the chroma demodulators 107 as input signals to be detected.

Output signals from the amplifier 125 are applied as input signals to a burst gate 129. The burst gate 129 provides an output signal responsive to these signals during time intervals determined by gating pulses. These gating pulses are supplied to the burst gate 129 from the horizontal sweep generator 121. When the generator 121 is synchronized with the broadcast television signal, these gating pulses occur at intervals corresponding to the intervals in which burst information is present in the output chroma signals of amplifier 125, as applied to the input of burst gate 129.

The burst gate 129 provides separated burst signals, accordingly, during normal receiver operation. Separated burst signals from the burst gate 129 are applied to a local color subcarrier source with synchronizing circuitry 131 which provides a regenerated color subcarrier output signal timed in response to the separated burst signals. The source 131 may comprise a crystal oscillator synchronized by means of automatic phase and frequency control (AFPC) or by injection lock means, for example. The regenerated color subcarrier output signal from the source 131 is supplied to a phase-shift network 133, which provides appropriately phased color subcarrier signal outputs to time the chroma demodulators 107.

The burst gate 129 also provides separated burst signals to a noise-immune detector 135, which develops ACC signals therefrom for application to the amplifier 125 to control its gain for chroma signals. As the level of burst information at the output of the first amplifier tends to increase, the gain of the amplifier is reduced by the ACC signals. The noise-immune detector 135 may, for example, be a synchronous detector provided color subcarrier signals to time its detection processes either from the phase-shift network 133 as shown by solid connection or, alternatively, directly from the source 131 as shown by dotted connection. The ACC signals from the detector 135 may be applied to a color killer threshold detector 137, which provides a color kill instruction signal when the ACC signals at its input are smaller than a threshold level. This color kill instruction signal may be coupled to the amplifier 127 to reduce its gain substantially to zero, as shown by solid connection, or alternatively coupled to the chroma demodulators 107 to disable their operation, as shown by dotted connection.

The ACC of the first gain-controlled amplifier 125 will not sufficiently reduce its gain for chroma signals in response to accompanying noise, because of the noise-immunity of the detector 135. During the reception of weak, noisy signals the gain of the amplifier 125 will be increased to maintain the level of burst information in its output signal the same as when strong, noise-free signals are being received. The signal excursions of noise accompanying the chroma signals at the output of amplifier 125, a part of which noise is generated in the "front end" 103 of the receiver and a part of which is intercepted by the antenna 101, will exceed normal chroma signal excursions. Were the gain of the amplifier 127 fixed in value, oversaturation conditions would therefore obtain in the television receiver.

The output chroma signals from the amplifier 127 are supplied to a peak detector 139. The amplifier 127 may have gating pulses applied to it from the horizontal sweep generator, as shown, to reduce its gain to zero during horizontal blanking intervals or, alternatively, it may not. In either case, chroma is available during picture intervals at the output circuit of amplifier 127. The peak detector 139 is sensitive to peaks in the picture-interval chroma signals provided to it by amplifier 127 and develops a gain control signal in response thereto which is applied to control the gain of the amplifier 127. As the peaks in the picture-interval chroma tend to increase their excursion, the control signal provided by the peak detector 139 reduces the gain of the amplifier 127. The excessive signal excursions of noise accompanying the chroma signals, mentioned in the previous paragraph, cause the peak detector 139 to reduce the gain of amplifier 127. Therefore, the signal excursions of noise accompanying the output chroma signals of amplifier 127 will not exceed the normal signal excursions of strong, noise-free output chroma signals. Consequently, "blooming" of the color kinescope on noise peaks will be forestalled.

Cascading the gain controlled chroma amplifiers 125 and 127 and operating their gain control loops separately obviates the problem of the peak detector 139 being sensitive to the average chroma level of the scene during the reception of strong, noise-free signals if: 1. the maximum gain of the chroma amplifier 127 is correctly set, and 2. the peak detector 139 is made insensitive to signal excursions which do not exceed a certain threshold value. When these criteria are met, the normal level of strong, noise-free chroma signals maintained at the input of the amplifier 127 by the ACC of the amplifier 125 will never cause large enough signal excursions of the output chroma signals provided by the amplifier 127 to the peak detector 139 to develop a control signal to reduce the gain of amplifier 127. When receiving strong, noise-free television signals, the amplifier 127 will then operate at its predetermined gain. No control signals dependent upon picture interval chroma will be introduced into the ACC system.

A manual chroma gain control 141 may be connected to control the gain of the gain-controlled amplifier 127. Manual chroma gain controls are often mis-set by the viewer and when set for too high chroma gain will tend to cause blooming of the kinescope on objects having high color saturation. Placement of the manual chroma gain control prior to the output circuit of the amplifier 127 and the input circuit of the peak detector 139 permits the gain of the amplifier 127 to be reduced by a control signal from the peak detector 139 responsive to the overly large chroma signals. This automatic adjustment of the gain of the amplifier 127 will prevent kinescope "blooming" caused by mis-setting of the manual chroma gain control 141.

The peak detector 139 also acts to reduce the gain of the amplifier 127 if signals are received which depart from good broadcasting practice by reason of having insufficient burst information or excessive chroma modulation. This gain reduction prevents oversaturation during the reception of such signals.

The elements 125-139 shown enclosed by the dotted-line 143 are suitable for fabrication in primarily monolithic silicon integrated circuitry form. The burst gate 129 and noise immune detector 135 may be of the type referred to in the concurrently filed U.S. Pat. application Ser. No. 242,322, entitled "Detector Circuit With Self-Referenced Bias", filed in the name of the present inventor and assigned to RCA Corporation. The color subcarrier source with synchronizing circuitry may comprise a voltage-controlled crystal oscillator and an AFPC detector providing voltage to control the frequency of oscillations from the oscillator in response to separated burst signals provided by the burst gate 129. The AFPC detector may be of the type described in the concurrently filed U.S. Pat. Application Ser. No. 242,321 entitled "Electronic Signal Processing Circuit", filed in the name of the present inventor and assigned to RCA Corporation. It is desirable to provide to an integrated circuit AFPC detector input chroma signals taken from the output of an amplifier 125 provided with ACC from a noise-immune detector 135. This is because the constraint on operating supply voltages in an integrated circuit and the infrequency of burst information tend to make the AFPC detector output small with respect to direct-current biasing errors. This undesirable condition can be better tolerated if the detector is supplied as much input signal as possible without overloading the detector when the oscillator is synchronized to incoming burst information. The chroma signals at the output of the ACC'd amplifier 125 are suitably regulated in amplitude to provide these input signals.

FIG. 2 is a schematic diagram of integrated circuitry for performing the functions indicated by blocks 125, 127, 137, 139 and 141 of FIG. 1 in connection with the functions indicated by blocks 129, 131, 133, 135 as provided by the circuitry described in the previous paragraph. All elements shown except 230, 237-239 and 261 are considered within the confines of an integrated circuit, which is provided with terminals 229, 236 and 262 for connection to those external elements.

The first gain-controlled amplifier 200 is a differential amplifier employing emitter coupled NPN transistors 201 and 202. The second gain-controlled amplifier 220 is a differential amplifier employing emitter coupled NPN transistors 221 and 222. Resistive voltage dividers 203, 205 and 204,206 provide collector loads for the transistors 201 and 202, respectively, The base electrodes of transistor 221 and 222 are connected to respective ones of the resistive voltage dividers 203, 205 and 204, 206 to receive reduced output chroma voltages from the collector electrodes of transistors 201 and 202. This establishes a cascade connection for chroma signals of the amplifier 220 after the amplifier 200. The output chroma voltages at the collector electrodes of transistors 201 and 202, provided in response to composite chroma input signals applied to terminal 209 at the base electrode of transistor 201, are for application to the burst gate 129 and subsequently the noise-immune detector 135.

The resistive voltage divider formed by the series connection of resistors 219, 216, 217 and diode 218 between +5 volt operating supply and ground reference potential provides direct-current voltages intermediate therebetween for the biasing of the base electrodes of NPN transistors 212, 214. Direct-current bias for the base electrodes of transistors 201, 202 is provided via resistors 210, 211, respectively, from the emitter electrode of the common-collector transistor 212.

Emitter current is provided to the joined emitter electrodes of transistors 201, 202 from the collector electrode of an NPN transistor 214, which has its emitter electrode coupled to ground reference potential by a resistor 215. The collector current of the transistor 214 is varied, as explained below, to vary the gain of the gain-controlled amplifier 200.

Similarly, the gain of the second gain-controlled amplifier 220 is varied in response to the collector current variations of an NPN transistor 223, having its collector electrode connected to the joined emitter electrodes of NPN transistors 221, 222, 224 and 225. Transistors 224 and 225 have their base electrodes joined at a terminal 226 to which blanking signal input is applied. This blanking signal comprises positive-going pulses occurring during the horizontal blanking interval and swinging up from +2.5 volt to +5 volts. These positive-going pulses may be provided from the horizontal sweep generator 121 (shown in FIG. 1). When the base electrodes of the transistors 224 and 225 are at +2.5 volts, the more positive voltages at the base electrodes of transistors 221 and 222 cause the quiescent collector current of transistor 223 to flow in equal portions through themselves and they function as a gain-controlled differential amplifier. When the base electrodes of transistors 224 and 225 are at +5 volts, exceeding the voltages at the base electrodes of transistors 221 and 222, the quiescent collector current of transistor 223 is caused to flow in equal portions through the transistors 224 and 225. The diversion of current completely away from the transistors 221, 222 reduces their transconductance to zero and the gain of the emitter-coupled differential amplifier 220 they form to zero. This switching is unaccompanied by an appreciable direct potential shift at the base electrode of an NPN transistor 227 since the collector resistor connecting this point to the +11.2 volt operating supply conducts half the quiescent collector current flow of transistor 223, whether via the collector-to-emitter path of transistor 225 during the horizontal blanking interval or that of transistor 222 during the picture interval. The signal at the base electrode of transistor 227, is not composite chroma containing burst information then, but is picture-interval chroma signal, chroma signal from which the burst information has been removed. The transistor 227 is connected as an emitter-follower amplifier and so the picture-interval chroma signal appears at its emitter electrode which is connected to the output terminal 229.

The gain of the amplifier 220 for picture-interval chroma signals may be manually controlled by a potentiometer 238 labelled "MANUAL CHROMA GAIN CONTROL". The end terminals of potentiometer 238 are connected to the +11.2 volt operating supply and ground reference potential, respectively, and its slider arm terminal provides an adjustable potential therebetween coupled via resistor 237 to terminal 236 at the base electrode of NPN transistor 235. The potential at the emitter electrode of the emitter-follower transistor 235 is offset 0.7 volt approximately from the potential at its base electrode and is coupled via a resistive voltage divider comprising resistors 234, 232 and temperature-compensating diode 233 to the base electrode of the transistor 223. The emitter electrode of the transistor 223 is connected to ground reference potential by resistor 231, and the collector current of the transistor 223 is increased or decreased in response respectively to increase or decrease of the potential applied to its base electrode. Increasing the potential at the slider arm terminal of the potentiometer 238 increases the collector current of transistor 223 and consequently the gain of the amplifier 220 for picture-interval chroma signal. Decreasing the potential at the slider arm terminal decreases the gain of amplifier 220. The manual gain control system described in this paragraph provides the basic direct current biasing network to control the gain of the amplifier 220, the effects of which network are augmented to provide for color killer function and for gain control to avoid oversaturation. The capacitor 239 connected between terminal 236 and ground reference potential decouples any noise generated by slider arm movement of the potentiometer 238 from appearing at the base electrode of transistor 235 and affecting the gain of amplifier 220.

The gains of the gain-controlled amplifiers 200, 220 are affected by ACC and color killer delay circuitry 240 embodied in elements 241-251. The balanced chroma output provided from the terminals 207 and 208 is coupled to the input for signals to be detected of an in-phase keyed synchronous burst detector 131 (shown in FIG. 1), which provides ACC control signals which are coupled to the terminal 241 at the base electrode of a PNP transistor 242.

As the ACC signal supplied from the noise-immune synchronous detector 131 to terminal 241 approaches within approximately 600 millivolts of +11.2 volts, as will be the case when there is no detectable burst information the PNP transistor 242 is no longer biased into forward conduction. The collector electrode of the transistor 242 therefore no longer supplies base current to NPN transistor 245 through the resistor 244. Without base current, transistor 245 supplies no collector current to maintain a voltage drop across the resistor 246, which couples its collector electrode to a +1.6 volt potential. The base electrode of NPN transistor 247 connected to the collector electrode of transistor 245 seeks to rise to the +1.6 volt potential, which causes base current flow in transistor 247. Collector current flows in transistor 247 in response to its base current flow and causes a substantial voltage drop in resistor 237 and whatever resistance is offered by the potentiometer 238. The reduction of base voltage on transistor 235 is so substantial that it no longer supplies current to maintain transistor 223 in forward conduction. With no collector current from transistor 223, the gain of the differential amplifier 220 for chroma signals is reduced substantially to zero.

As detected ACC signal brings the potential at terminal 241 downward from +11.2 volts by more than the approximately 700 millivolts required to forward bias its base-emitter junction transistor 242 is biased into conduction. The consequent conduction of transistor 245 clamps the base electrode of grounded-emitter transistor 247 to ground reference potential. This prevents collector current flow in the transistor 247 and its modification of the potential at terminal 236, so the color killer circuitry exerts no influence on the gain of the amplifier 220.

The common-emitter amplifier transistors 242, 245, 247 thus function as a threshold detector providing output current from the collector electrode of transistor 247 only when the ACC signal developed by the noise-immune detector 131 exceeds a threshold amplitude of 700 millivolts, approximately, between terminal 241 and the +11.2 volt operating supply. Since the transistors 245 and 247 are grounded-emitter amplifiers with substantial forward gain, the switching into and out of color kill occurs over a small range of ACC signal potential. The capacitor 239 provides some additional noise immunity for the color killer, since sustained conduction of transistor 247 is required to discharge the capacitor 239 to kill the gain of the chroma amplifier 220. Sustained nonconduction of transistor 247 is required for capacitor 239 to charge when color is no longer killed.

The collector electrode of transistor 242 also is connected to the input terminal of a resistive voltage divider comprising resistors 249 and 250, the output terminal of which is connected to the base electrode of an NPN transistor 251. The emitter electrode of transistor 251 is coupled to ground reference potential by a resistor 252, and its collector electrode is connected to the base electrode of transistor 214.

The resistive divider 249, 250 prevents the application of sufficient voltage to the base electrode of transistor 251 to bias it into forward conduction to provide ACC control to the first gain-controlled amplifier 200 until the ACC signal applied to the input terminal 241 is more than large enough to bias transistor 245 into conduction, removing color kill from the amplifier 220. In the circuit shown in FIG. 2 color killer action is initiated when the output of the first gain-controlled amplifier 200 has fallen 6dB from the level maintained during operation of its ACC loop. The reason for doing this is best explained referring back to FIG. 1. If the amplitude of the burst information is in the composite chroma input signals to the amplifier 125 (corresponding to amplifier 200 in FIG. 2) is detected at a level below which ACC is exerted on the amplifier 125, the sensitivity of the burst detection process as carried out in the detectors 135, 137 is not reduced by that ACC action. This provides for better definition of the level of composite chroma input which will cause the threshold level of the threshold detector 137 (corresponding to transistor 245, resistor 246, transistor 247 in FIG. 2) to be exceeded by excursions of detected burst signal and which will subsequently cause color killer action to be inactivated. Accordingly, the need for a color killer threshold control is obviated.

The gain of the amplifier 220 is also controlled in response to peaks of the picture-interval chroma signal provided at the output terminal 229 which peaks are large enough to cause oversaturation in the television receiver and to result in blooming of the kinescope. A capacitor 261 having substantial capacitative reactance at color subcarrier frequency couples the picture-interval chroma from output terminal 229 to an input terminal 262 of the peak detector 260. The peak detector 260 comprises resistors 263 and 264 transistor 265, capacitor 239, and the resistance of resistor 237 and potentiometers 238.

The resistors 263, 264 provide the peak detector 260 with a threshold of response to input signals at terminal 262, so that it functions as a threshold peak detector. The terminal 262 is coupled by a resistor 263 to a 1 V_BE (approximately 0.7 volt) supply, as conventionally may be provided across a forward-biased semi-conductor junction and is coupled by a resistor 264 to ground reference potential. The connection of resistors 263, 264 places a quiescent bias potential of approximately 450 or 500 millivolts on terminal 262 and the base electrode of the grounded-emitter NPN transistor 265 connected thereto. This quiescent bias potential is insufficient of itself by approximately 200 millivolts to bias transistor 265 into conduction. This provides a threshold of some 200 millivolts which peaks of signal at terminal 262 must overcome in order that transistor 265 be biased into substantial conduction. Peaks of the picture-interval chroma signal superimposed upon this quiescent bias potential, which correspond to peaks of picture-interval chroma signal at terminal 229 sufficiently large to cause oversaturation, will overcome this threshold and provide sufficient forward bias to the base-emitter junction of transistor 265 to bias it into conduction during the duration of these peaks.

The setting of the potentiometer 238 determines the quiescent control voltage on the capacitor 239 and therefore the quiescent charge upon the capacitor, which quiescent conditions obtain when the base electrode of transistor 265 is not supplied signals with peaks large enough to bias the transistor 265 into conduction. The conduction of the transistor 265 during larger peaks will remove charge from the capacitor 239, which charge is only slowly replenished through the bleeder resistance afforded by resistor 237 and potentiometer 238. Accordingly, the potential across the capacitor 239 is reduced. This reduces the potential at the base electrode of transistor 235, which as previously explained reduces the gain of the amplifier 220.

The peak detector 260 can be constructed so that rapid fluctuations of the gain of the amplifier 220 due to its control are avoided, which most viewers of a television receiver incorporating the circuitry shown in FIG. 2 prefer. The capacitance of the capacitor 239 is chosen to be large enough so that appreciably sustained conduction of transistor 265 over the period of a field of television signal (1/30 to 1/25 of a second) or a few fields on peaks of the picture-interval chroma signals is required to reduce the gain of the amplifier 220 substantially. That is, a period of time longer than the duration of a single short noise or chroma signal peak is required for the peak detector to charge to the level of recurring such peaks. As shown, the discharge of the capacitor 239 may be accomplished during a sustained interval of recurring overly large noise or chroma signal peaks more rapidly than the quiescent charge of the capacitor 239 will be replenished through the bleeder resistance afforded by resistor 237 and potentiometer 238. The replenishment of this charge will be over a period of several fields, as provided by choosing the bleeder resistance to be suitably large.

NORDMENDE SPECTRA COLOR L2UT CHASSIS FFS 772.535.A 772.536.A (4.570.D) PHILIPS PAL CHROMA DELAY LINE:An improved ultrasonic delay line comprising a solid glass body having one or more slits in the side walls extending inwardly from the outer edge faces of the body. The slits are arranged in the path of the propagating ultrasonic energy so as to effectively increase the number of energy transmission paths in the body by acting as additional energy reflecting surfaces. The slits extend the effective length of the delay line. The slits also operate to reduce undesired cross-coupling between the input and output transducers.

1. An ultrasonic delay line comprising a solid body having a plurality of energy reflecting edge walls and composed of ultrasonic wave energy transmitting material, said edge walls being arranged to provide a first point for introducing ultrasonic energy and a second point for extracting said energy from the body and further providing a plurality of multiply reflected internal transmission paths for delaying said ultrasonic energy, and an energy reflecting surface positioned in the desired energy transmission path and formed by a wall slit arranged to block the passage therethrough of impinging ultrasonic energy and extending inwards from an outer edge wall of the solid body and positioned so as to provide substantially complete reflection of the desired energy from opposite faces thereof to the edge walls thereby to redirect said energy through 2. A delay line as claimed in claim 1 wherein the wall slit is arranged relative to one or more reflecting edge walls of the body so as to produce an odd number of said paths between a reflection from one face of the slit and a subsequent reflection from the same or opposite face of said slit. 3. A delay line as claimed in claim 2 further comprising first and second electromechanical transducers coupled to said body at said first and 4. A delay line as claimed in claim 1 further comprising first and second electromechanical transducers coupled to said body at said first and 5. A delay line as claimed in claim 1 wherein the wall slit is located in the plane of symmetry of the body, said delay line further comprising first and second electromechanical transducers coupled to said body at 6. A delay line as claimed in claim 1 wherein said body includes a second wall slit extending inwards from an outer edge wall and positioned so that the ultrasonic energy is reflected off of opposite faces of the second 7. A delay line as claimed in claim 1 wherein the wall slit is arranged in the body relative to one or more reflecting edge walls thereof so as to multiply reflect the desired energy from said wall slit to produce an odd number of said paths between a first reflection from one face of the slit 8. An ultrasonic delay line comprising a solid body having a plurality of energy reflecting edge walls and composed of ultrasonic wave energy transmitting material, said edge walls being arranged to provide a first point for introducing ultrasonic energy and a second point for extracting said energy from the body and further providing a plurality of multiply reflected internal transmission paths for delaying said ultrasonic energy, and an energy reflecting surface positioned in the desired energy transmission path and formed by a wall slit arranged to block the passage therethrough of impinging ultrasonic energy and extending inwards from an outer edge wall of the solid body to reflect the desired energy to the edge walls thereby to redirect said energy through the body, said wall slit being arranged in the body so as to intercept ultrasonic energy propagating along given undesired transmission paths between said first and second energy points of the body thereby to reduce any direct coupling of scattered secondary ultrasonic energy between said first and second 9. An ultrasonic delay line comprising a solid body having at least five energy reflecting edge walls and composed of ultrasonic wave energy transmitting material, two of said edge walls being parallel to each other and orthogonal to a third edge wall, the fourth and fifth edge walls each being at an angle of approximately 135° to a respective one of said parallel edge walls, said edge walls being arranged to provide a first point for introducing ultrasonic energy and a second point for extracting said energy from the body and further providing a plurality of multiply reflected internal transmission paths for delaying said ultrasonic energy, and an energy reflecting surface positioned in the desired energy transmission path and formed by a wall slit extending centrally inwards into the body orthogonal to said third edge wall and arranged to block the passage therethrough of impinging ultrasonic energy thereby to redirect 10. A delay line as claimed in cl

aim 9 wherein said wall slit extends 11. A delay line as claimed in claim 9 wherein said fourth and fifth edge walls intersect one another and said wall slit extends inwardly from the 12. A delay line is claimed in claim 1 wherein said body has a generally rectangular shape and a second wall slit extending inwards from an outer edge wall of the body so as to reflect the ultrasonic energy, the first and second wall slits extending inwards from opposite parallel edge walls 13. A delay line as claimed in claim 9 further comprising first and second electromechanical transducers coupled to said body at one or more edge 14. A delay line as claimed in claim 11 further comprising first and second electromechanical transducers coupled to said body at said fourth and fifth edge walls, respectively, thereby to reduce direct coupling of 15. An ultrasonic delay line comprising a solid body having a plurality of energy reflecting edge walls and composed of ultrasonic wave energy transmitting material, said edge walls being arranged to provide a first point for introducing ultrasonic energy and a second point for extracting said energy from the body and further providing a plurality of multiply reflected internal transmission paths for delaying said ultrasonic energy, an energy reflecting surface positioned in the desired energy transmission path and formed by a wall slit arranged to block the passage therethrough of impinging ultrasonic energy and extending inwards from an outer edge wall of the solid body thereby to redirect said energy through the body, and a second wall slit extending inwards from an outer edge wall of the body so as to reflect the ultrasonic energy, the first and second wall slits extending inwards from opposite parallel edge walls of the body, and wherein said body has a parallelogram cross-section and said first and second wall slits extend orthogonally inwards from the longer pair of 16. A delay line as claimed in claim 12 further comprising first and second electromechanical transducers coupled to said body at said first and second points which are located on edge walls other than the edge walls 17. An ultrasonic delay line comprising a solid body having at least five energy reflecting edge walls and composed of ultrasonic wave energy transmitting material, two of said edge walls being parallel to each other and two other edge walls being at right angles thereto, a fifth edge wall being at an angle of approximately 135° to each of two adjacent edge walls, said edge walls being arranged to provide a first point for introducing ultrasonic energy and a second point for extracting said energy from the body and further providing a plurality of multiply reflected internal transmission paths for delaying said ultrasonic energy, and an energy reflecting surface positioned in the desired energy transmission path and formed by a wall slit extending into the body centrally of and orthogonal to one of the edge walls located opposite to the fifth edge wall and arranged to block the passage therethrough of impinging ultrasonic energy thereby to redirect said energy through the 18. A delay line as claimed in claim 17 further comprising first and second electromechanical transducers coupled to said body at said first and 19. A delay line as claimed in claim 18 wherein said first and second energy points are located on the fifth edge wall.

Description:

This invention relates to ultrasonic delay lines of the type using a solid medium such as quartz or glass through which an acoustic signal wave is made to travel to provide a time delay between the application of the wave and its extraction. In such delay lines it is known to shape the solid medium so as to provide internal peripheral reflective surfaces for the ultrasonic wave in order to fold the wave over a plurality of legs to increase the length of the transmission path through the medium and thus increase the wave delay with a minimum mass of solid medium.

It is also known to increase the length of the transmission path of an ultrasonic wave by including specially shaped openings in the solid medium to provide additional reflective surfaces. In this case such openings have to be very accurately positioned and dimensioned to ensure proper operation.

In connection with such delay lines there arises a number of problems. Some of these concern the solid medium itself and its thermal properties. Delay lines using wavelengths equivalent to several Megahertz require very accurate dimensioning to reduce internal energy scatter and give an accurate source of extraction. This requires a solid medium having a very low temperature coefficient. A special glass having such properties is available but it is relatively costly for use in mass production so that any design steps that will allow an overall reduction in the mass of the delay medium will not only in itself reduce thermal problems but will also reduce overall costs.

In certain color television receiver systems a prescribed signal delay is required so that the delay line has to provide stable operation and yet lend itself to mass production at a very low cost.

Another problem which confronts the designer of such delay lines is the prevention of direct signal coupling between the application and extraction points of the signal which can result in the desired delayed signal being masked by a strong undelayed signal arriving at the extraction point. A further problem is the suppression of alternative signal paths which contribute a train of secondary spurious signals each having a different delay and which make extraction of the wanted delayed signal difficult.

The purpose of this invention is to provide a simple delay line construction in which the overall mass of the delay line medium is reduced in a manner which will also allow greater freedom from expensive manufacturing processes as well as providing enhanced electro-acoustical performance.

According to this invention there is provided an ultrasonic delay line using a solid medium through which an ultrasonic signal wave is made to travel and which is reflected over a plurality of paths to increase the time delay between the application point of the ultrasonic signal and its point of extraction, wherein the path followed by the ultrasonic waves includes at least one reflecting surface constituted by the side wall or face of a slit extending inwards from an edge face of the solid medium.

In order to make maximum utilization of a given delay line mass, the delay line may include several slits arranged so that both side walls of the slits can be used as reflective surfaces. Furthermore, if the geometrical pattern of the reflected signal legs or path is so arranged that an odd number of legs exists between reflections on the same or associated slit wall, this gives the advantage that the angular orientation of the slit is non-critical and it displays self-cancelling properties for minor errors.

Furthermore, the use of slits to provide reflective walls also has the advantage of reducing spurious secondary signals in that a greater control can be exercised over the required signal path by the very high damping barrier provided by the absence of any delay line medium forming the slit. This reduces any signal transference across the slit to a value far below the minimum requirements.

It should be noted that the use of notches introduced in the edge surfaces of a solid medium for a delay line to reduce secondary waves from reaching the output transducer is known per se. However, these notches do not constitute reflecting walls for the desired signal.

Examples of this invention will now be described with reference to the accompanying drawings in which FIG. 1 is a plan view of a substantially rectangularly shaped delay line showing a simplified embodiment of applicant's invention.

FIG. 2 is a plan view of a substantially rectangularly shaped delay line showing two slits for further increasing the length of the delay line of FIG. 1.

FIG. 3 is a plan view of a delay line having five reflecting faces for further increasing the length of the delay line of FIG. 1.

FIG. 4 is a plan view of a delay line shaped as a parallelogram having four slits.

FIG. 5 is a plan view of a delay line having five edges and a central slit.

FIGS. 1 to 5 show five different embodiments of delay lines according to this invention. Each Figure has certain design features which will be discussed below.

FIG. 1 shows a solid body 1 made, for example, of glass and having a substantially rectangular cross-section. Two corners of the body 1 are beveled and transducers A and B are arranged on the surfaces 14 and 15, respectively. The surfaces 14 and 15 are at respective angles of 135° to the surfaces 17, 18 and 18, 19 of the body 1. The input transducer A has an electric signal applied to it which is converted by the transducer into an acoustic ultrasonic signal. This acoustic signal propagates in the form of a wave through the body 1 and after a number of reflections it reaches the transducer B which reconverts it into an electric signal. The time required for the acoustic ultrasonic wave to cover the entire path (shown in dotted lines) from the transducer A to the transducer B determines the delay time between the application of the electric input signal at the transducer A and the electric output signal recovered at the transducer B. Use is preferably made of piezo-electric transducers which are so polarized that shear mode vibrations are produced so that the overall reflection at each of the reflective surfaces occurs without energy conversion of the shear vibrations into longitudinal vibrations.

According to this invention, a slit 2, in the form of a saw-cut having plane parallel walls, is provided at the plane of symmetry in the body 1 so that the waves originating from the transducer A first reflect at the left-hand wall of the slit 2 and then at the rectangular walls 16, 17, 18, 19, and 20 of the body 1, whereupon they are reflected from the right-hand wall of the slit 2 and finally strike the transducer B. The energy path from transducer A to transducer B is made up of eight reflected signal legs shown by dashed lines with arrowheads. It will be apparent from FIG. 1 that an increased path length for the ultrasonic wave is thus obtained in a simple manner. Moreover, secondary waves are suppressed by the slit 2. The angle at which the ultrasonic wave strikes the various reflective surfaces is always 45°. However, in this embodiment the angle 3 of 90° between the slit 2 and the surfaces 16 and 20 must be very accurately defined in order that the waves may follow the path indicated.

In the delay line of FIG. 2, the signal paths (shown in dotted lines) are obtained by providing two slits 2 and 4 at suitably chosen areas at right-angles to the long surfaces 21 and 22 of the delay line medium 1. In this embodiment the ultrasonic waves also strike the reflective surfaces at angles of 45°. However, after reflection at one wall of the slit 2, an odd number of signal legs (five) occurs before reflection at the other wall of the slit 2. As a result, the orientation of the angles 5 and 6 of 90° is not critical and the angular errors introduced into the reflected signals are cancelled automatically. In this construction, the slits 2 and 4 also cause a reduction of secondary (spurious) signals, and moreover the formation of any direct or secondary transmission path between the input transducer A and the output transducer B is prevented.

The delay line construction of FIG. 3 provides an increased length of the transmission path while retaining the advantages of the delay line constructions shown in FIGS. 1 and 2. In this case, the body 1 has a square cross-section (a corner of the square being denoted by x--x) and the opposite corner of the square is removed so that an additional wall 31 is formed on the body 1 which is at an angle of 135° to the walls 32 and 33. The transducers A and B are arranged side by side on the wall 31, while a slit 8 is provided at right angles to and approximately centrally of a wall 34 of the body 1 and extends approximately as far as half the length x into the body 1. The ultrasonic waves again follow the path indicated by dotted lines.

Either the transducer A or the transducer B may be used as input or output. Since the number of signal legs between the reflections at one wall and those at the other wall of the slit 8 is odd (five), the orientation of the angle 7 of 90° between the slit 8 and the surface 34 is not critical because the angular error introduced into the signal wave is automatically canceled. This self-canceling effect is illustrated in FIG. 3, in which the slit 8 is purposely slightly tilted. A practical embodiment of a glass delay line of this construction for use in a PAL color television receiver system has the following approximate dimensions:

x = 33 mm, y = 15 mm, and z = 6 mm.

The width of the slit 8 is approximately 1 mm and this slit extends over approximately 15 mm into the delay line 1. The electric characteristics give a delay of one line period, i.e., approximately 64 μ sec, at a band center frequency of 4.4 Mc/s.

FIG. 4 shows a body 1 in the form of a rectangular prism having a cross-section in the form of a parallelogram whose sides 41, 42 and 43, 44 respectively are at angles of 45° to each other. Slits 8, 10 and 9, 11, respectively, are provided at right angles to the side faces 42 and 44. In this delay line, only one side wall of each of the slits 8, 9, 10, and 11 is used at a time. An input transducer A is arranged for injecting an ultrasonic signal which follows the path shown in dotted lines and which is extracted by the output transducer B. In this construction, any angular displacements of the slits are not automatically canceled and the angles are therefore critical, but the remote positioning and interspersion of the slits between the input transducer A and the output transducer B provides a high degree of decoupling for spurious (secondary) signals when compared with known delay lines.

The surface of the delay line of FIG. 5 has a cross-section in the form of a pentagon having two parallel sides 51 and 52 and a third side 53 at right angles to the sides 51 and 52, while the fourth and fifth sides 54 and 55 are at angles of 135° to the sides 51 and 52, respectively. The latter sides 54 and 55 support the transducers A and B, respectively. According to the invention, a slit 56 is positioned at the intersection of the sides 54 and 55 and extends into the body 1 parallel to the sides 51 and 52 over a distance approximately equal to half the length of the sides 51 and 52. The path followed by the ultrasonic waves is shown in dotted lines. Small angular displacements of the surfaces 51 and 52 again substantially do not influence the overall delay time and the direction in which the waves strike the output transducer B. Also, the slit 56 prevents the direct coupling of scattered radiation from the input transducer A to the output transducer B.

It will be evident from the foregoing that delay lines constructed in accordance with this invention can be easily and economically mass produced. A comparatively long rod of delay line medium may be profiled, for example, in the desired shape, while the slits may be accurately arranged throughout its length. The method of manufacturing separate delay lines then merely resides in parting off portions of the rod to the desired thickness. This results in a high reproducibility of components of individual delay lines.

The invention is not limited to the delay line described consisting of a single layer, but the advantages of this invention may also be obtained in delay lines consisting of several layers, the path followed by the signal in one layer then being reflected at a suitable point to a further layer so that it can pass on through this further layer before it is extracted.

NORDMENDE SPECTRA COLOR L2UT CHASSIS FFS 772.535.A 772.536.A (4.570.D) PHILIPS PAL CHROMA DELAY LINE / ACOUSTIC DELAY LINE:
A glass for an acoustic delay line which consists of SiO₂, Al₂ O₃, B₂ O₃ and an oxide of a bivalent metal and satisfies the requirement that -5×10^-6 <Σ_i α_i x₁ < +5×10^-6 where α_i is the temperature coefficient of the rate of propagation in the range of 20°-70° C for the oxide component i and x_i is the molar fraction of that component.

1. In an acoustic delay line of the type having signal converting elements on the surface of a glass body for converting an input electric signal into an acoustic signal and an output acoustic signal into an electrical signal, the improvement comprising that said body of glass consist of the following compositions in wt. percent: 2. In an acoustic delay line of the type having signal converting elements on the surface of a glass body for converting an input electric signal into an acoustic signal and an output acoustic signal into an electric signal, the improvement comprising that said body of glass consist of the following composition in wt. percent:

Description:

The invention relates to an acoustic delay line in which the delay medium is glass.

Such delay lines are known per se for electronic uses in which delays of electric signals in the order 0.01-1 millisecond are to be obtained with bandwidths of a few tens of mc/s. The delay is produced in that an electric signal is converted, by means of a piezo-electric element, into an ultrasonic mechanical vibration, preferably a shear vibration, and after said acoustic signal has traversed the delay medium this is likewise converted again into an electric signal by a piezo-electric element, said signal having experienced the desired delay with respect to the original signal. The rate of propagation of the acoustic shear waves in a solid is approximately 10 ⁵ times smaller than that of electro-magnetic waves so that a comparatively large delay can be obtained over a comparatively small distance.

Delay lines are used inter alia in electronic computers, in radar technology and in television technology. In two color television systems delay lines are used for combining the color information of adjacent lines of a frame. The delay time required for this purpose is approximately 64 μsec. with 625 lines and a frequency of 50 c/s. At the frequency to be considered of 4.43 mc/s and the required bandwidth of approximately 2 mc/s, glass is a suitable delay medium.

A known glass which is excellency suitable for this purpose has the following composition in mol. percent:

SiO ₂ 70-78 PbO 15-30, of which maximally 5 mol. percent may be replaced by one or more of the oxides MgO, BaO, CaO and SrO, Na ₂ O + K ₂ O 0-7 Na ₂ O ≤0.5 SB ₂ O ₃ + As ₂ O ₃ ≤ 0.5

this glass is distinguished by the quality of various properties which are of importance for the end in view. Taking into account the temperature variations of ±30° C occurring in practice, the delay times does not vary more than 0.02 μsec. This means that the temperature coefficient of the delay time dτ/(τdτ) of these glasses is smaller than 10 × 10 ^-6 per ° C and in some cases even smaller than 1 × 10 ^-6 per ° C.

The damping of the acoustic vibrations in delay lines of this class is not too large. The mechanical attenuation of said glass is not more than 9 × 10 ^-3 dB/μs. Mc/s which is amply sufficient for delay lines in television receivers.

A further advantage of this glass consists in that it is very slightly sensitive to the previous thermal history of the glass which means that it has substantially no influence on the temperature coefficient of the delay time, whether the glass has been cooled relatively rapidly or slowly from temperatures in the proximity of the annealing te

mperature. Large variations in the treatment which consists of a heating for approximately 10 minutes at a temperature which lies approximately 50° C above the annealing temperature succeeded by cooling at a rate of approximately 1.5° C per minute, do substantially not influence the reproducibility.

Finally, a hysteresis effect is not present in this glass to any inconvenient extent, in contrast with some other known glasses. This hysteresis effect manifests itself in the delay time when the glass is heated from room temperature to a temperature between 60° and 80° C, is kept at said temperature for more than 1 hour, and is then cooled to room temperature again. The delay time at room temperature may be increased 1 to 10 ⁴ , said increase disappearing again gradually in the course of a few days. In the above-mentioned glasses said variation is at most 3 to 10 ⁵ at the temperature cycle described.

The rate of propagation for shear waves in these glasses is comparatively low and varies only slightly with the composition (2,400-2,600 m/sec.).

A difficulty in manufacturing the glass compositions required for delay lines is associated with the fact that small variations in the composition of a chosen glass may cause variations in the acoustic properties, notably in the temperature coefficient of the delay time. This is most undesirable, particularly when used in delay lines for color television. So this involves the necessity of keeping the content of the components of the glass constant between narrow limits. The known glasses have a high content of lead monoxide. However, lead monoxide has the property of partly evaporating at the surface of the glass melt so that there the PbO-content is considerably reduced. If such a glass, originating from the surface layer of the melt, forms part of the delay body, the good operation as a delay medium may be disturbed.

Possibilities are known, it is true, to restrict said evaporation of PbO. However, these requires special precautionary measures.

The invention provides a class of glasses of which the drawback of evaporation of one or more of the components with the resulting adverse influence on the acoustic properties of the glass is considerably smaller while the above-mentioned advantageous properties of the known glass are maintained therein.

According to the invention the acoustic delay line, the delay body of which consists of glass which contains the components SiO ₂ , K ₂ O and oxide of bivalent metal, is characterized in that the glass has the following composition in percent by weight:

SiO ₂ 50-75 K ₂ O + Na ₂ O 0-8 Na ₂ O ≤0.5 Sb ₂ O ₃ + As ₂ O ₃ ≤ 1.5 B ₂ O ₃ < 5 Al ₂ O ₃ < 15 PbO 0-10 CaO 0-20 BaO 0-40 MgO 0-10 ZnO 0-25 totally 20-50 CdO 0-35 SrO 0-30 Bi ₂ O ₃ 0-30

on the understanding, however, that the requirement is also satisfied, that -5 × 10 ^-6 <Σ _i α _i x _i < +5 × 10 ^-6 , where α _i is the factor for the temperature coefficient of the rate of propagation in the range of 20° to 70° C for the oxidic component i and x _i is the molar fraction in which said component is present in the glass.

During the experiments which led to the invention it was found that the temperature coefficient of the rate of propagation of acoustic shear waves is an additive quantity with respect to said quantity for the free oxidic components. In order that the temperature coefficient of the delay line be substantially zero, the above condition should be fulfilled. Within the above-mentioned range of compositions, only those glasses may be used as a delay medium in ultrasonic delay lines for the above-mentioned purposes in which the said condition is fulfilled without having to use additive ancillary means which have for their object to improve a delay line the temperature coefficient of which is not equal to zero, for example, by the combination with an electric transit time line the temperature coefficient of which is equal to but opposite to that of the glass delay line.

In the following Table I the values of the factors α _i are listed for the oxides to be considered.

TABLE I

Oxide i α _i + 10 ⁶ SiO ₂ - 100 B ₂ O ₃ - 90 Al ₂ O ₃ + 180 ZnO +165 PbO +285 CaO +340 BaO +350 MgO +325 CdO +210 Bi ₂ O ₃ + 350 SrO + 350 K ₂ O +300

as ₂ O ₃ and Sb ₂ O ₃ may be neglected in the calculation. The accuracy of the value of the temperature coefficient calculated by means of the formula is such that for glasses which have been cooled at a rate of approximately 1° C per minute from the highest annealing temperature or 50° C above said temperature said value does not differ from the experimentally determined value of the temperature coefficient more than ±5 × 10 ^-6 /° C over the temperature range of 20° - 70° C. With a desired greater accuracy a quantity of one or more components, starting from a previously chosen composition, may be varied until the desired value of the temperature coefficient has been reached. As a rule the desired value for glasses which are used as an acoustic medium will be equal to or substantially equal to zero but in some cases a value differing slightly from zero is desirable in order to obtain an optimum action of the delay line in a temperature range other than the said range of 20° to 70° C or to compensate for the temperature coefficient of the transducers and/or other components of the associated electric circuit. Alternatively, a different manner of cooling may result in a slightly differing value of the temperature coefficient.

The glasses according to the invention for the present use and a good stability, that is to say that the above-mentioned hysteresis effects do not occur to any inconvenient extent also after prolonged use.

Whereas for most of the known glasses the delay time τ in accordance with temperature has an approximately parabolic variation:

(Δτ)/τ = c ^. (T - T _o ) ²

in the temperature range in which │T-T _o │ ≤50° C and in which c is approximately +0.04 × 10 ^-6 /(° C) ² , the value of c for a large number of glasses according to the invention is only +0.02 × 10 ^-6 /(° C) ² , so that the constancy of the delay time as a function of the temperature for these glasses is still larger than for the known types of glass.

The rate of propagation of acoustic shear waves varies for the glasses with compositions within the range according to the invention from 2,800 to 3,500 m/sec. These values are somewhat higher than the above-mentioned known glasses (2,400-2,600 m/sec.) which means that for the same delay time a proportionally larger length of the acoustic beam is necessary. For delay lines having a small delay time of, for example, 64 μsec., however, that is no objection.

A preferred range of compositions is determined by the following limits (also in percent by weight).

SiO ₂ 60-70 K ₂ O+Na ₂ O 2-6 Na ₂ O ≤0.5 Sb ₂ O ₃ +As ₂ O ₃ ≤ 1.5 B ₂ O ₃ < 5 Al ₂ O ₃ < 15 PbO 0-5 CaO 0-10 BaO 0-25 MgO 0-5 together 25-38 i.e., the remainder not less than 25 ZnO 0-15 CdO 0-20 SrO 0-15 Bi ₂ O ₃ 0-20

a few examples of glass types which are used according to the invention as a delay medium in an acoustic delay line are the following which are stated in mol. percent and in wt. percent. Stated are the following properties: the average temperature coefficient TC = (Δτ)/(TΔT) in the temperature range of 20° - 70° C in 10 ^-6 per ° C, the variation (ΔTC) at 20° C of the temperature coefficient in 10 ^-6 per ° C after a cooling treatment in which the glass is heated from room temperature to 50° C above the annealing temperature of the glass and is then cooled to room temperature at a rate of 1 1/2° C per minute compared with that of the glass in which it is cooled at a rate of approximately 100° C per minute and the value of the constant c from the above formula in 10 ^-8 per (° C) ² . ------------------------------------------------------------ --------------- TABLE II

1 2 3 4 Mol Wt. Mol Wt. Mol Wt. Mol Wt. % % % % % % % % ____________________________________________________________ ______________ SiO ₂ 63.7 69.2 54.3 67.0 62.0 72.9 60.7 B ₂ O ₃ 3.0 2.7 3.0 3.2 Al ₂ O ₃ 5.0 6.7 5.0 7.9 K ₂ O2.5 3.4 2.5 3.1 2.5 3.6 2.5 3.3 PbO CaO 7.9 6.4 5.0 4.3 5.0 3.9 BaO 7.7 16.9 12.1 24.2 6.5 13.8 ZnO 7.7 9.0 8.0 8.5 12.3 15.3 7.9 8.9 MgO 5.0 3.1 CdO 5.0 8.9 As ₂ O ₃ 0.2 0.6 0.2 0.5 0.2 0.6 0.2 0.5 ____________________________________________________________ ______________ TC 0 ± 1 0 ± 1 0 ± 1 0 ± 1 ΔTC 4 3 6 6 c. 3 3 4 3 ____________________________________________________________ ______________ ____________________________________________________________ ______________ 5 6 7 8 Mol Wt. Mol Wt. Mol Wt. Mol Wt. % % % % % % % % ____________________________________________________________ ______________ SiO ₂ 53.9 73.3 58.5 70.1 60.7 72.6 62.5 B ₂ O ₅ 5.0 5.1 Al ₂ O ₃ 5.0 7.4 K ₂ O2.5 2.8 2.5 3.1 2.5 3.4 2.5 3.4 PbO CaO 5.0 3.3 5.0 4.0 7.0 5.6 BaO 5.5 9.9 11.0 22.4 7.2 15.9 7.0 15.4 ZnO 8.0 8.6 10.7 12.5 MgO 5.0 2.3 5.0 2.9 SrO 5.0 6.9 Bi ₂ O ₃ 5.0 27.3 As ₂ O ₃ 0.2 0.5 0.2 0.5 0.2 0.6 0.2 0.6 ____________________________________________________________ ______________ TC 0 ± 1 0 ± 1 0 ± 1 0 ± 1 ΔTC 6 3 3 5 c. 2 3 2 3 ____________________________________________________________ ______________

NORDMENDE SPECTRA COLOR L2UT CHASSIS FFS 772.535.A 772.536.A (4.570.D) Voltage multiplier:

A voltage multiplier formed by a pair of end contacted layer capacitors. The layer capacitors may be arranged side by side or may be stacked one upon the other. In each case a series of slits are cut in the capacitor to divide the unit into a plurality of individual capacitors, each being integrally connected by an unslit web. The diodes which are part of the voltage multiplier are then arranged in substantially parallel relation to extend across the capacitor body to make electrical contact between the end faces thereof.

1. A voltage multiplier having two rows of serially connected capacitors and terminals between capacitors on each row, a diode connecting each capacitor terminal on one row with a capacitor terminal on another row, all of said diodes being connected in series to pass pulses of one polarity only, said diodes being arranged to charge each capacitor, said two rows of serially connected capacitors consisting of end contacted layer capacitors which are integrally joined to one another, said end contacted layer capacitors being split into individual capacitors by slots which extend from one end face of the capacitor through the capacitive zone thereof to a point short of the opposite end face, the individual capacitors having unslotted portions of the metal coatings which serially connect each other, all of the terminals of a row of serially connected capacitors being formed in an end face, whereby said diodes may be readily connected from the terminals on the end face of one row of capacitors to the terminals on the end face of the other row of capacitors. 2. A voltage multiplier in accordance with claim 1 wherein said two rows of capacitors are arranged in side by side relation and spaced from one another, the majority of said diodes being arranged substantially parallel to one another. 3. A voltage multiplier in accordance with claim 1 wherein the terminals of each of the individual capacitors of each capacitor row lie substantially in the same plane. 4. A voltage multiplier in accordance with claim 1 wherein the two rows of capacitors are congruent in physical design, arranged on top of each other as a single integral unit separated by an intermediate insulating layer, the diodes arranged on a single side of the integral capacitor unit and extend perpendicularly to the end contact layers and are welded to the narrow edges of said end contact layer. 5. A voltage multiplier in accordance with claim 4 wherein said end contact layers are formed by the Schoop process. 6. A voltage multiplier in accordance with claim 1 wherein the two capacitor rows are arranged in spaced side by side relation, the end contact surfaces being arranged at the top of each row and lying generally in a single plane, the contact surfaces being arranged in a direction generally perpendicular to the longitudinal direction of the foils forming the capacitors, the majority of said diodes being arranged substantially parallel to one another and having their connection wires welded to the contact surfaces of said capacitor rows. 7. A voltage multiplier in accordance with claim 1 wherein one of said capacitor rows is formed on top of the other, said rows being separated by an insulating intermediate layer which does not project beyond the end contact layers, both rows of capacitors being subject in common to the Schoop process, the diodes spanning the end contact layers and being welded to the narrow edges thereof, the diodes extending in a direction generally parallel to the direction of the foils forming the capacitors. 8. A voltage multiplier in accordance with claim 1 wherein said diodes are silicon diodes. 9. A voltage multiplier comprising first and second end contacted layer capacitors, said capacitors being stacked upon each other, a plurality of substantially parallel slits extending alternately from opposite end faces of the stacked capacitor assembly to points intermediate of but not through the assembly, thereby electrically dividing the capacitor assembly into a number of individual capacitors which are serially connected by the remaining unslit portion of the metal coatings and which form the conductive layers thereof, an intermediate insulating layer separating the first and second end contacted capacitors, and a plurality of diodes arranged substantially parallel to each other and generally in a single plane which defines one side of the capacitor assembly, the diodes extending between the end contacts of the capacitor assembly and making electrical contact with the narrow edges of those end contacts. 10. A voltage multiplier in accordance with claim 9 wherein said first and second capacitors, being stacked to form an assembly, have opposite end faces contacted in common by the Schoop process, and the diodes are caused to span the capacitor assembly to make electrical contact between said opposite end faces. 11. A voltage multiplier in accordance with claim 1 wherein the two rows of serially connected capacitors are stacked one above the other and are separated by an intermediate insulating layer, said capacitors being end contacted by the schoop process, the intermediate insulating layer extending outwardly of the schoop layer, only one row of capacitors on each side thereof having free edge zones, diodes being arranged substantially parallel to said slots and making electrical contact with opposite end contact surfaces of said rows of capacitors. 12. A voltage multiplier in accordance with claim 11 wherein the portion of the intermediate insulating layer which extends outwardly of the schoop layer is free of schoop metal.

Description:

BACKGROUND OF THE INVENTION

1. Description of the Prior Art

Voltage multipliers of the type involved in the present invention are normally used in color television receivers to produce the high voltage for the picture tube anode. The known electrical voltage multipliers of this type are normally manually assembled from individual components and are soldered together. Such an arrangement is time consuming and it is not always possible to automate the soldering process in such a method of construction.

2. Field of the Invention

The field of art to which this invention pertains is solid state voltage multipliers and in particular to such voltage multipliers which are formed of an integral arrangement of layer type capacitors end contacted by the Schoop process.

SUMMARY OF THE INVENTION

It is an important feature of the present invention to provide an improved structure for a voltage multiplier.

It is another feature of the present invention to provide a voltage multiplier using a layer type capacitor arrangement.

It is a principle object of the present invention to provide an improved solid state voltage multiplier which is subject to easy production techniques.

It is another object of the present invention to provide a voltage multiplier formed of a layer type capacitor which is manufactured in such a way as to permit a series of diodes used in the multiplier to be readily easily soldered to the capacitor terminals in a single plane.

It is an additional object of the present invention to provide a voltage multiplier as described above wherein the capacitors of the multiplier consist of two layer type capacitors, each being slit in such a way as to form two sets of series connected individual capacitors.

It is also an object of this invention to provide a voltage multiplier as described above wherein the two sets of individual capacitors are arranged in side by side spaced relation.

It is a further object of the present invention to provide a voltage multiplier as described above wherein the two sets of serially connected individual capacitors are stacked one upon the other, and the diodes used in the multiplier are caused to span the body of the capacitor assembly and make electrical contact between the end faces thereof.

These and other features, advantages and objects of the present invention will be understood in greater detail from the following description and the associated drawings wherein reference numerals are utilized as to designate the preferred embodiments.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 shows a circuit diagram of a voltage multiplier in accordance with the present invention.

FIG. 2 shows a voltage multiplier in accordance with the present invention wherein two separate capacitor networks are used and arranged in side by side spaced relation.

FIG. 3 shows an arrangement having electrical characteristics similar to FIGS. 1 and 2 but wherein the capacitor networks or sets are stacked upon one another in a spaced saving arrangement.

FIG. 4 is a diagrammatic cross sectional view of the arrangement shown in FIG. 3.

FIG. 5 shows two capacitor networks which are separated by an insulating layer which extends beyond the Schoop layers.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

The present invention relates to an electric voltage multiplier of the type used in television receivers to produce the high anode voltage for the picture tube. Such multipliers usually consist of two rows of serially connected capacitors with a plurality of diodes connecting the capacitor terminals on one row with the like terminals on the other row. The diodes are connected in series to pass a unidirectional pulse and are arranged in such a way as to charge each of the capacitors.

By virtue of the present invention, there is provided a smaller physical design for a voltage multiplier and a physical construction which is easily suited to automation processes.

This is accomplished in the present invention by providing the two rows of serially connected capacitors to be formed from end contacted layer capacitors which are integrally joined together. The metal coatings of each layer of the capacitors have free edges at opposite sides, and the capacitor layers are split by a series of slits which form individual capacitors connected electrically in series. The slits pass from one end face to a point just beyond the capacitor zone of the capacitor. In each case the individual capacitors are connected in series with the other capacitors by an inner coating which is not severed in making the slits.

Capacitor networks of the type described are known from the German text laid open to public inspection No. 1,764,861. It is not possible however to achieve a simple production technique for voltage multipliers by the use of the teachings of this patent. Only by means of the combination of features of the present invention such as using an inner series connection has it been possible to develop an arrangement where all contact surfaces lie on the same end face of the capacitor network. In this way it has been possible to achieve a physical construction which enables the taking advantage of the most automated production techniques. Furthermore the use of capacitor networks which are based upon the principle of layer capacitors provides the added advantage that it is possible to use silicon diodes which are considerably smaller than the selenium rectifiers heretofore used.

In one arrangement of the invention, the two capacitor networks are arranged next to one another and the diodes are arranged substantially parallel to one another except for one of the diodes at the end of the circuit. This enhances the possibility of using automating processes. For the same reason, it is advisable for the contact surfaces to lie in a single plane which is possible according to the present invention.

A particularly space saving embodiment of the invention is provided in which the two capacitor networks are arranged one above another in a stacked manner and are separated from one another by an insulating intermediate layer which may extend beyond the Schoop layer as shown in FIG. 5.

Also, the projecting of the insulating layer and the removal of Schoop metal from the layers can be avoided if the two capacitor networks are stacked in such a way and separated by an insulating layer which does not project beyond the Schoop layers, if in the region of one end face, all the coatings of one capacitor network has free edge zones exposed. In such an arrangement each of the coatings is displaced in relation to the adjacent coatings. In such a manner the Schoop layer only covers the coatings of one of the two capacitor networks and the coatings of the other network is safely insulated from the Schoop layer by the free edge zones. At the same time, a mechanically stable structure is formed, since the Schoop layer also secures the parts of the end faces which they do not electrically contact.

A construction which is particularly advantageous is one in which the two capacitor networks are congruent and the diodes are arranged on one side of the capacitor network and welded to the Schoop layers. In this case the diodes are arranged perpendicularly to the end contact layers.

A simple process for the production of voltage multipliers according to the present invention is such that the capacitor networks are arranged in side by side spaced relation and the contact surfaces are generally in a single plane. In this arrangement the diodes are arranged in a direction perpendicular to the longitudinal direction of the foils in the capacitor, and the diodes span the end faces of the structure. A diode arranged at the beginning or end of the unit can be positioned at an angle, while the other diodes are arranged in a generally parallel orientation.

The preferred embodiment of the invention which is particularly small designed voltage multipliers consist of stacking the two layer capacitors upon each other and separating them by an insulating layer which does not project over the ends. The slits which are then formed on the capacitor layers, divide the capacitors into individual elements, and the diodes are placed in position on the top of the arrangement in such a way that the terminals of the diodes connect the end faces of the capacitor layers.

Referring to the drawings for greater detail FIG. 1 shows a schematic of a voltage multiplier according to the present invention in which capacitors 1 form one serially connected set and capacitors 2 form a second serially connected set. The capacitor network 4 of FIG. 2 is formed from the capacitors 1 of FIG. 1. The capacitor network 5 of FIG. 2 is formed in accordance with the invention from the capacitors 2 of FIG. 1. In FIG. 1 the diodes 3 are connected from the terminals of the capacitors 1 to the terminals of capacitors 2 as shown to produce a series diode arrangement which charges each of the capacitors as is well understood in the art of voltage multipliers.

The contact surface 6 (FIG. 2) is grounded, while the contact surface 7 is connected to the pulse input. Contact surface 8 is the tap for the high voltage output and the contact surface 9 serves to contact the diodes to two of the capacitors of one capacitor network and the contact surface 22 serves to contact the last capacitor of the capacitor network 5 to two of the diodes.

As in FIG. 2, the diodes 3 are placed on the contact surfaces 7, 8, 22, and 9 and are electrically connected thereto by spot welding. Slots 10 are provided and filled with synthetic material in the course of encasing the arrangement so that they possess the requisite dielectric strength. During operation these slots are connected at least temporarily with the full voltage of a capacitor in the case of television, for example, the voltage across one of these slots may be 8.5 K.V.

In FIG. 3, an arrangement is shown where the capacitor networks 4 and 5 are congruent and are stacked one above the other. In this case, the outer flanks 11 of the capacitor networks 5 are not used. The two end faces of the capacitor networks 4 and 5 are entirely covered with Schoop layers. The diodes 3 are arranged generally in a parallel layout with respect to the slots 10. The diodes connect the opposite contact surfaces 13 and 14 of the Schoop layers.

As illustrated in FIG. 4, the contact surfaces 13 and 14 contact only the corresponding metal coatings 15 and 17 of one of the capacitor networks 4 and 5. The coating 16 of the capacitor network 4 contacts neither of the two Schoop layers 13 and 14. However, these coatings extend beyond the slot depth 19, so that after the completion of the capacitor network, electrically conductive arms remain outside the slots and are integrally joined to the blind coatings. Accordingly, the blind coatings 18 of the capacitor 5 project beyond the slot depth 20. Thus, it is only possible to contact such a capacitor network on one end side.

The diodes 3 can thus only be connected by welding their two terminals to in each case one contact surface 13 and 14 in accordance with FIG. 1. This construction also must be sealed in order to achieve the required dielectric strength.

The intermediate layer 21 does not project beyond the end faces of the capacitor network 4 and 5 and is covered by the Schoop layer. It prevents sparkovers in the region of the cut edges and breakdowns through the dielectric, since, particularly in the use of a multiple inner series connection, the dielectric does not possess a dielectric strength sufficient to support the voltage of the overall capacitor.

In FIG. 5 the two capacitor networks 4 and 5 are separated from each other by an insulating intermediate layer 21 which extends beyond the Schoop layers 13 and 14. The Schoop layers 13 and 14 are interrupted by this intermediate layer and the upper and lower part of the Schoop layers 13 and 14 can always be contacted and wired separately.

NORDMENDE SPECTRA COLOR L2UT CHASSIS FFS 772.535.A 772.536.A (4.570.D)  ELECTRICAL COMPONENT PROTECTED AGAINST HIGH TENSION, PARTICULARLY FOR COLOR TELEVISION RECEIVERS AND METHOD OF ITS PRODUCTION:Voltage multipliers of the type involved normally used in color television receivers to produce the high voltage for the picture tube anode:

A grid-shaped electrical component is formed by molding a plurality of electrically interconnected capacitors and diodes, physically forming a grid structure, within a synthetic casting resin. The electrical components form the cores of the struts of the grid structure and each strut is provided with a plurality of indentations in the synthetic resin in the area of the walls of the components.

1. A grid-shaped electrical component protected against high tension comprising a plurality of molded struts extending in substantially two grid directions and in at least one grid direction oblique thereto, each of said struts formed of synthetic resin and having a large wall strength, a plurality of electrically interconnected electrical components individually cast within and forming the respective cores or said struts, the components extending in one of said grid directions disposed to lie in one plane, the components extending in the other grid direction disposed to lie in another plane parallel to said one plane, the components extending in the oblique direction disposed to lie in one of said planes, said components comprising a plurality of capacitors disposed in the struts which extend in one grid direction and a plurality of rectifiers which are disposed in said struts which extend in the other grid direction and in the oblique grid direction, said capacitors forming serially connected rows of capacitors and said rectifiers connected to the ends of said capacitors, a first group of conductors disposed at one end of said grid structure and having the individual conductors thereof connected to the junctions of said electrical components and each including a portion extending substantially parallel to the respective capacitors and a portion extending through said grid structure generally perpendicular to the first-mentioned portions, and a second group of electrical conductors at the other end of said grid structure connected to the junctions of said electrical components thereat, said grid structure further including a base having mounting shoulders formed thereon, said grid structure together with said base and said conductors forming a self-supporting structure for permitting the passage of cooling air between said struts.

Description:

BACKGROUND OF THE INVENTION

1. Field of the Invention

This invention relates to a grid-shaped electrical component having grid struts extending substantially in two main grid directions, and more particularly to grid-shaped components which are protected against high tension, particularly for color television receivers.

2. Description of the Prior Art

Heretofore, the prior art recognized a conventional expedient to provide a color television receiver with a line transformer to generate the high voltage required for the picture tube. This line transformer was advantageously utilized to provide high voltage pulses which were then rectified prior to being employed at the picture tube. Construction of the line transformer for protection against high voltages, as well as the associated rectifier arrangement, has proven difficult in situations wherein the transformer was required to deliver the full value of the required high voltage. This is particularly true in the case of color television receivers since a direct current voltage of approximately 25kv. is ordinarily required for proper operation of the picture tube.

In view of the foregoing it is therefore advisable to design the line transformer for a lower voltage and to generate a direct current voltage of the required magnitude by utilization of a multiplier cascaded with the line transformer. For example, it is significant that the line transformer delivers recoil pulses with an amplitude of 8.5kv., from which a DC voltage of 25kv. can be obtained in a multiplier cascade having 5 silenium rectifiers and four or five capacitors.

In electrical equipment technology, however, there is always the problem, as here in a multiplier cascade, to design components free from brush discharge and protected against high tension, so that neither adjacent components nor the operating personnel can be harmed. To this end, it is generally known in the art to combine such units which are exposed to high tension into a single component which can be built in or exchanged as a one unit component both during original manufacture of the apparatus and in case of maintenance or repair.

Arrangements of the type initially mentioned avoid a majority of the inconvenience experienced heretofore by components manufactured to meet the above conditions. Thus, for example, the corresponding components are frequently cast with plastic into one compact block. However, in addition to the problem of an effective heat dissipation there is the further difficulty that the wall strength of the grouting mass between the components is not sufficiently constant, or that the metallic connecting elements of the components or the wiring may appear on the surface and cause glow defects or flash-overs.

To avoid the latter drawbacks it is also an old expedient in the art to fixedly arrange the components of the unit on a base plate and to cast the entire unit in a beaker consisting of a material which is combined with the grouting mass. However, these solutions have the drawback that casting in a beaker is relatively expensive and that as a result of these steps the problem of cooling the component parts is rendered more difficult.

Although the component parts of the type initially mentioned have great advantages from the electrical point of view as well as with regard to the manufacturing cost, there is also an additional difficulty as a result of the size of the components thus produced. In many cases, particularly in the interior of electrical apparatus such as, for example, color television sets, the measurements of which one seeks to reduce through the use of integrated switching circuits, it is frequently undesirable or even quite impossible to accommodate the component parts.

Hence, it is an object of the invention to produce an electrical component protected against high tension which has the advantages of a component of a type mentioned initially, but which has smaller dimensions than heretofore known in employing the same individual components.

SUMMARY OF THE INVENTION

According to the invention, the component parts are disposed in a molded grid structure wherein certain component parts lie in one grid direction, other component parts lie in another grid direction, and still other components are disposed obliquely to the two main grid directions. Each of the individual components are molded in a strut of the grid structure and the components parts which are disposed in one grid direction and those disposed obliquely to the grid directions are located in one plane, while the component parts disposed in the other grid direction lie in a plane parallel to the first-mentioned plane. Through this arrangement with the component parts lying in different planes, one can substantially reduce the space required for the apparatus in question without adversely affecting the advantages of prior known constructions.

As shown in relatively long experiments, it is very important for an effective operation of the apparatus that the wall strength of the synthetic resin enveloping the component parts is substantially equally large throughout the entire apparatus. It has namely been shown that the important point is that the thickness of the layer of synthetic resin varies only relatively little throughout the apparatus, and in all respects the variation is not excessive, since otherwise there may be the danger that the sealing layer will crack upon cooling. The danger of a crack formation may also be responsible for the fact that with the component parts known heretofore and case in the form of a block with or without a beaker, only adverse results were obtained since the plastics suitable for the electrical components with respect to high tension do not lend themselves to casting into blocks.

A new type of apparatus which is protected against high tension in accordance with the principles of the present invention is particularly suitable for a high tension multiplier cascade in color television sets of the type mentioned initially which contains a plurality of rectifier elements and capacitors. A further development of the invention provides that the capacitors lie parallel one behind the other in an upper plane, that in the second plane lying therebeneath a portion of the rectifier elements is spaced at a distance from the other plane and lie perpendicular to the capacitors, and that in a diagonal line of each rectangle defined by two capacitors and two rectifiers there is arranged an additional rectifier in the second lower plane. For the electrical connection of the component part with the elements of a corresponding circuit, it is provided that electrical conductors are guided outwardly from adjacent ends of the rows of capacitors, the conductors first extending a distance within the synthetic casting resin substantially parallel to the capacitors at one end of the apparatus, and at the opposite end of the apparatus the electrical conductors extend from the rows of capacitors whose ends are connected with a rectifier element and guided outwardly therefrom through the envelope of synthetic resin.

BRIEF DESCRIPTION OF THE DRAWINGS

Other objects, features and advantages of the invention will become apparent from the following detailed description taken in conjunction with the accompanying drawings, in which:

FIG. 1 is a plan view, partially in section, illustrating apparatus constructed according to the invention; and

FIG. 2 is a sectional view taken along the line II--II of FIG. 1.

DESCRIPTION OF THE PREFERRED EMBODIMENT

FIG. 1 illustrates a high tension cascade constructed in accordance with the principles of the present invention to produce the accelerating voltage required for color television tubes amounting to approximately 25 kv. consisting of five silenium rectifiers 21, 31 and four capacitors 22, fifth capacitor not cast within the component part must be separately connected thereto. Therefore, this capacitor is not shown in the drawing. The reference numerals 25, 26 and 27 designate the electrical conductors which are guided outwardly through the synthetic resin 23. The electrical conductors 25 and 26 on the base side of the component part are first pulled a distance upwardly in the synthetic resin casing so that when they exit the component part (see FIG. 2) the conductors are already sufficiently separated from the base side which is generally connected with the base frame. The indentations 24 in the synthetic resin compound are shown somewhat exaggerated schematically, since the indentations generally end, at least partly, on the wall portions of the component parts which are protected against high tension. The indentations 24 are produced during the manufacture of the cascade in accordance with the invention, wherein the interconnected electrical components which form a loose grid are placed in a mold consisting of a material, such as polyethylene or polypropylene, which does not combine with the casting resin, for example, epoxy resin. Within this mold there are disposed spacing blocks which are preferably formed directly on the mold and which are arranged and dimensioned such that they assure a sufficient space between live metal parts of the electrical components and the inner wall of the mold when the loose grid is installed. Owing to the tolerances required for a simple insertion of the grid of component parts, not all component parts will rest against all spacing blocks provided for the centering thereof so that not all of the indentations 24 in the synthetic resin compound 23 reach as far as the high tension resistant wall of the component part accommodated within the corresponding strut. The fact is illustrated schematically in the drawing where some of the indentations do not reach quite as far as the component parts.

As apparent from FIG. 2, a plastic strip 28 is provided on the narrow side (base side) of the component part remote from the high tension conductor 27, which interconnects the parallel rows of capacitors and wherein recesses 29 are provided by means of which the component part can be screwed onto the apparatus or attached in a different manner.

The invention is not limited to the embodiment shown. For example, is it also possible to form separate feet on the component for attachment purposes. It is likewise possible to slide the capacitors 22 still further inwardly by way of the rectifiers whereby a lateral contacting of the connection conductors of the capacitors is advantageous so as to make the lateral expansion of the component part still smaller. It is however, essential that the component forms a punctured grid structure since in this way there is, first, comparatively uniform wall strengths assured and, second, the cooling of the component elements can be separately effected without forcing the heat to first penetrate a larger layer of synthetic resin.

When employed in television sets, particularly color television sets, component parts in accordance with the invention may be built into advantage in horizontal position in so-called knapsacks, where as a result of their grid-shaped construction, they not only do not impede the air circulation caused by heating of the other component parts of the set, but are at the same time effectively cooled as a result of the air currents flowing therethrough.

Changes and modifications may be made of the invention within the scope and spirit of the appended claims which define what is believed to be new and desired to have protected by Letters Patent.

NORDMENDE SPECTRA COLOR L2UT CHASSIS FFS 772.535.A 772.536.A (4.570.D) Television Voltage multiplier arrangement with capacitor rolls surrounded by diodes:

A voltage multiplier includes a plurality of capacitors and diodes in an integral unit. The capacitors are combined in capacitor rolls surrounded by diodes on the outside of the diodes can be positioned between two capacitor rolls. AC and dc-voltage-operated capacitors can be combined in separate or the same rolls. The ac capacitors may be located inside within the roll and the dc capacitors on the outside of the same roll. The capacitor plates are made of aluminum foil with intermediate polystyrene and polyester layers. A common capacitor electrode plate may be used for adjacent capacitors.


1. A voltage multiplier comprising a plurality of series connected diodes and a plurality of capacitors, each capacitor being connected between opposite ends of a pair of said diodes and including an intermediate thermoplastic dielectric layer and a pair of metal foil electrode layers on opposite sides of said dielectric layer, said electrode and dielectric layers being rolled into a plurality of overlapping layers including said plurality of capacitors within a common roll, said diodes being connected to said electrode layers and being disposed about opposite outer sides of said roll. 2. The device of claim 1 including means applying a.c. and d.c. voltages to different respective groups of said capacitors. 3. The device of claim 1 wherein said plurality of diodes surround the sides and one end of said roll, the other end of said roll having external connections thereto, and a common thermoplastic cover encapsulating said diodes and capacitors, said external connections extending from one end of said cover. 4. The device of claim 1 including means applying a.c. and d.c. voltages to different respective groups of said capacitors within said common roll, said a.c. voltage capacitors being within the inner layers and said d.c. voltage capacitors being within the outer layers. 5. The device of claim 1 wherein said roll has external connections to the two ends thereof. 6. The device of claim 1 wherein said thermoplastic dielectric layer includes two outer layers of polystyrene and a layer of polyester therebetween. 7. The device of claim 1 wherein one electrode layer is common to two capacitors within said roll. 8. The device of claim 6 wherein said electrode layers are of aluminum.

Description:

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a voltage multiplier arrangement with diodes and capacitors wherein at least two capacitors are combined into a unit.

2. Description of the Prior Art

Voltage multiplier arrangements serve to produce high voltages and are particularly useful for the operation of television picture tubes. The diodes and capacitors are connected in separate series paths with a capacitor in parallel with two series diodes and are generally embedded in plastic for voltage protection.

In one known arrangement, the diodes and capacitors are disposed in a lattice configuration to keep the volume to a minimum. At each long side, two capacitors are arranged one behind another in the longitudinal direction; a diode is located at each short side and in the middle therebetween, and an additional diode is disposed in each diagonal direction. Thus, the length of this arrangement is essentially determined by the length of the capacitors, while the width is determined by the length of the diodes.

In another known arrangement, in order to meet the requirement for optimum utilization of the space available and for technical simplification, the ac-voltage-operated capacitors are connected in series and potted to form a unit, and the same is done with the dc-voltage-operated capacitors.

SUMMARY OF THE INVENTION

The primary object of the present invention is to provide a simplified voltage multiplier that occupies less space and insures the necessary voltage protection.

According to the invention a plurality of capacitors are combined in a unit which forms a capacitor roll. The space occupied by the potted elements is reduced by at least one-half that of known arrangements.

According to one feature of the invention, the capacitors are combined into two separate capacitor rolls and the diodes are disposed between said rolls.

In a variation of the invention, all capacitors are combined into one capacitor roll and surrounded by diodes on three sides.

The invention will now be explained in further detail with reference to the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a circuit diagram of a voltage tripler using the present novel arrangement;

FIG. 2 schematically shows the structure of the arrangement of FIG. 1;

FIG. 2a shows the actual physical arrangement of the elements of FIG. 2;

FIG. 3 is a circuit diagram of a voltage doubler as in the present invention;

FIG. 4 schematically shows the structure of the arrangement of FIG. 3;

FIG. 4a shows the physical arrangement of the elements of FIG. 4; and

FIG. 5 is a section through a portion of a laminated structure of a capacitor roll.

DESCRIPTION OF THE PREFERRED EMBODIMENT

In FIGS. 1 and 2, the diodes of the voltage tripler are designated by the reference numerals 1, 2, 3, 4, and 5, and the reference numerals 6 and 7 denote ac-voltage-operated capacitors which come into operation only during charging, while 8, 9, and 10 are dc-voltage-operated capacitors. The tripler is constructed in the form of a five-stage cascade circuit, the capacitor of the first stage, formed by the diode 1 and the capacitor 8, being grounded, and is used, for example, to generate the high voltage for operating color picture tubes. The capacitors 6 and 7 of the ac-voltage portion of the circuit are combined into a single capacitor roll 11, and the capacitors 8, 9, and 10 of the dc-voltage portion are combined in another capacitor roll 12. As shown in FIGS. 2 and 2a, the diodes 1, 2, 3, 4, and 5 are disposed centrally between capacitor rolls 11 and 12. The arrangement is potted in a case 13 filled with plastic. The ac-voltage input terminal 14, the dc-voltage output terminal 15, the ground terminal 16, and an additional terminal 17 are brought out on one side of the case 13. In addition, a terminal 18 is provided at a diode 19 connected to the ac-voltage input terminal 14.

The high voltages necessary for operating the black-and-white picture tube in a television receiver are generated with a voltage doubler. In FIGS. 3, 4, and 4a, the diodes are designated by the reference numerals 20, 21, and 22, while 23 denotes an ac-voltage-operated capacitor, and 24 and 25 the dc-voltage-operated capacitors of a three-stage cascade circuit. The alternating voltage to be doubled is applied to the terminal 28, and the dc voltage to be doubled appears at the terminal 29. The terminal 30 represents the ground terminal, to which is connected the capacitor of the first stage of the cascade, formed by the diode 20 and the capacitor 24. Both the ac-voltage-operated capacitor 23 and the dc-voltage-operated capacitors 24 and 25, are all combined into one capacitor roll 26. As shown in FIGS. 4 and 4a, the capacitor roll 26 is disposed centrally and surrounded by diodes 20, 21, and 22 on three sides. The arrangement is potted in a case 27 filled with plastic. The terminals 28, 29, 30 are brought out on the fourth, free side. The capacitor 25, shown with a broken line in FIG. 3, is dispensed with in the arrangement of FIG. 4. This is primarily replaced by the self-capacitance of the picture tube when the multiplier arrangement is connected into the television receiver.

The individual capacitors used in the present multiplier arrangements are designed as high-voltage capacitors having a laminated dielectric of polystyrene and polyester, and metal foil electrode plates. As shown in FIG. 5, the plates of a capacitor roll of this type are preferably aluminum foils 31, 32. Disposed between these foils are two polystyrene foils 33 with an intermediate polyester foil 34. The various terminals may be brought out at one or the other end of the roll or at both ends. At least two capacitor rolls of this laminated structure are wound over one another so that at least one multiple capacitor roll is obtained.

Since ac-voltage- and dc-voltage-operated capacitors may be combined into one capacitor roll, as shown in the embodiments of FIGS. 3 and 4, and as can also be done in the embodiments of FIGS. 1 and 2, the ac-voltage-operated capacitors are advantageously located inside within the roll. With such an arrangement, the corona discharge at the surface of the plastic-filled case 13 or 27 is greatly reduced.

In addition, the various possibilities of combining the capacitors, in conjunction with the diodes, make it possible to achieve particular input capacitances for voltage multipliers. For example, the input capacitance of the voltage multiplier arrangement is increased if at least one dc-voltage-operated and at least one ac-voltage-operated capacitor are combined into one roll.















NORDMENDE SPECTRA COLOR L2UT CHASSIS FFS 772.535.A 772.536.A (4.570.D)  Television channel / NIXIE DISPLAY Program indicator:
Of the devices that were designed in the mid-1950's to meet this requirement the most successful was a gas-disc

harge device called the Nixie tube. At the time the Nixie tube was introduced it was not at all certain that it would become the dominant digital device for electronic instruments. There were two major competitors: incandescent lamps and electroluminescent numbers. There were several ways in which incandescent lamps could be driven from the outputs of vacuum-tube counters, and the lamps could be used to illuminate masks or to edge-light plastic panels to produce a number display. The circuitry required to power these displays was more complicated and more costly than what was needed for the Nixie tube. Moreover, the incandescent indicators themselves were relatively expensive. The electroluminescent numbers were made from powdered phosphors that emit light when they are subjected to an electric field. Unfortunately the early electroluminescent lamps had a short and unpredictable lifetimes and they gradually faded as serious competitors to the Nixie tube.
The name Nixie came about accidentally. A draftsman making drawings of the tube labeled it NIX 1, for numeric indicator experimental No. 1. His colleagues began referring to it as "Nixie," and the name stuck. The tube contains 10 metal cathodes, each shaped to form a different number. The cathodes are insulated from one another and are stacked one behind the other. The anode is a metal mesh. The entire assembly is in a glass bulb that contains neon gas with a small amount of mercury. When an electric potential of about 180 volts is applied between the anode and any cathode, the gas near the cathode breaks down and emits light. With a proper choice of gas pressure and cathode dimensions almost all the light comes from the immediate vicinity of the energized cathode, and the result is a luminous orange-red number.
The Nixie tube was first marketed commercially in 1956. It is still sold by its originator, the Burroughs Corporation, and by Burroughs' licensees in many countries. It is available in a variety off sizes and is widely used in measuring instruments of all kinds and in office equipment such as calculators and copying machines. This tube has been successful because it is reliable and has a long lifetime. Because it is a familiar device to design engineers the Nixie tube continues to be sold in large numbers.
The voltages to operate Nixie tubes are provided by circuits called drivers. Originally Nixie tubes were designed to be driven by vacuum tubes, which themselves operate at high voltages. Modern integrated circuits, however, operate at very low voltages, and interface circuits are required to drive Nixie-tube displays. These driving circuits are readily available from a number of sources, but the need for interface circuits, which provide a high voltage, is one reason why the Nixie tube is being challenged.

This invention relates to signaling devices and more specifically to glow lamp indicators for selectively signaling numerals, letters or other characters or symbols.

One object of the present invention is to provide a signaling device which is capable of selectively displaying one of a plurality of characters in substantially the same space. Another object is to provide a signaling device

for selectively displaying one of a plurality of characters in which the character to be displayed is selected by means of a momentary selecting impulse whereupon the selected character is maintained on display as long as desired by the

inherent characteristics of the indicator without requiring holding circuits externally of the signaling device.

A further object of the invention is to provide a control circuit for the above-mentioned signaling device which requires but one individual control wire for each of a large number of signaling devices.

Other objects will appear in the following description taken in conjunction with the accompanying drawings in which:

Fig. 1 illustrates one embodiment of the glow lamp indicator;

Fig. 2 shows certain parts of the gaseous discharge glow indicator in exploded fashion;

Fig. 3 shows the internal circuit connections of the glow lamp indicator;

Fig. 4 shows the fundamental operating and control circuits for a plurality of glow lamp indicators;

Fig. 5 shows the internal circuit connections of an alternative embodiment of the glow lamp indicator;

Fig. 6 shows the fundamental operating and control circuits for the alternative embodiment;

FIG7 shows an application of the glow lamp indicator and control circuit to a stock quotation system, illustrating the selecting equipment required on a subscriber's premises; and F'g. 8 shows the equipment required for one stock in the stock quotation system.

In the well-known space discharge devices or glow lamps, a pair of metallic electrodes are sealed within a glass bulb filled with neon, mercury, sodium or other suitable gases at a definite very low pressure. When a unidirectional (direct current) potential is applied to the electrodes and gradually increased, the glow discharge will set in at a certain definite potential called an "irniting potential". The luminous glow discharge is produced by negative electrons and positive gas ions and takes place within a certain small distance from the exposed surface of the cathode or negative electrode, which appears to be surrounded or coated with a thin film of light. This film of light follows the contours of the 5 cathode surface in all details.

When the potential is further increased, the glow discharge becomes somewhat brighter. When the potential is gradually reduced, the glow discharge is maintained down to a potential 10 considerably below the igniting potential, until at a certain definite minimum potential the discharge ceases.

If an intermediate potential somewhere between the igniting and minimum potential is ap- 15 plied to the electrodes, there will be no glow discharge, but if the potential is momentarily raised to or above the igniting potential and thereafter reduced to the intermediate potential, the discharge will be started by the igniting potential 20 and thereafter be maintained by the intermediate potential until the potential is reduced to or below the minimum potential. This characteristic of the glow lamp makes it possible to control the starting and stopping of the glow dis- 25 charge by means of brief momentary impulses of high and low potentials, with the lamp normally connected to an intermediate potential.

Thus, the glow lamp may be lighted by the application of an igniting impulse and thereafter 30 remains lit, until the potential is reduced momentarily below the minimum potential. This feature offers a means to control glow lamps without external holding relays or other means for keeping the lamp circuit closed when it is desired 35 to have the lamp glow.

The fact that the exposed parts of the cathode of a glow lamp are entirely surrounded by a thin film of luminous discharge may be utilized to display any desired character by means of properly 40 shaped cathodes. A cathode consisting of a wire shaped in the form of the numeral 1 will, when ignited, produce a luminous outline of the numeral 1, and similarly any other desired character may be formed.

In the present invention these two characteristics of the glow lamps are utilized as follows: In Fig. 1 the glass bulb 101 is filled with a suitable gas, such as neon, at the required pressure. The glass foot 102 has fused into it a number of 50 supports 103, which hold the disk assembly 104 near the forward part of the bulb. The disk assembly 104 consists of eleven very thin disks of glass, stacked one behind the other with a small separation between adjacent disks. In the in- 55 terstices between the disks the electrodes are arranged in the shape of fine metal wires, the cathodes being shaped In the form of the ten numerals 1, 2, 3, 4, 5, 6, 7, 8, 9, and 0, while the, 149,104 anodes are short pieces of wire near the lower part of each cathode. The anodes do not glow, and those parts of the cathode wires which are not desired to glow are covered by a suitable Insulation, such as enamel.
The bulb is mounted in a base I OB provided with external terminals 188. The connections from the terminals to the electrodes are made by means of connecting wires 181 and 188, and are carried through the glass foot 181 In a well

known manner by means of short connectors made of metal having the same coefficient of expansion as glass.

Fig. 2 shows the disk assembly 184 In an exploded view to illustrate the ten cathodes 281 and ten anodes 205. Each of ten glass disks 201 has the wires 201 and 20! forming the electrodes cemented to its surface in a suitable manner. The lead out wires, such as 202, which are not desired to glow, are covered with suitable insulation.

These ten disks with an additional front cover disk 204 are then stacked one upon the other, the wire electrodes serving to separate the disks from each other so as to permit access of the gas filling to the electrodes. After the disks are assembled, the interstices between them may be sealed in a suitable manner around the periphery to prevent interference from one electrode to another. A small aperture may be left at one point of the periphery by leaving out the sealing operation at this point, to provide communication with the main gas chamber formed by the glass bulb 101.

When the bulb 101 is subsequently exhausted and then filled with gas at the proper pressure,

the exhausting and filling process extends through this communicating aperture to the ten gas chambers formed by the eleven glass disks 203 and 204. The communicating aperture may be filled with a suitable sealing material which permits the air and gas to permeate during the exhausting and filling operation. After these operations are completed and the bulb 181 is sealed off, the sealing material in the communicating aperture may be rendered impervious to the gas by suitable procedures, such as heating by means of electronic bombardment, for the purpose of completely sealing the ten gas chambers from each other and from the main gas chamber formed by the bulb 101.

The entire disk assembly is very thin. If, for example, each glass disk is 0.008 inch thick and the electrode wires have a diameter of 0.002 inch, the assembly 104 is altogether only 0.108 inch thick. As a result, the rearmost cathode 8, when glowing, will be easily discernible through the ten disks in front, and the other nine cathodes in the shape of the numerals 1 to 9 will not obscure the glow surrounding the cathode 0 to a noticeable degree, inasmuch as the cathodes are only 0.002 inch in diameter while the glow discharge appearing on both sides of the glowing cathode is approximately %g inch wide.
Viewed from the front of the bulb, therefore, any one of the ten cathodes, when glowing, will
appear in approximately the same place. In this manner, any one of the ten numerals may be displayed by causing the corresponding cathode to glow. Fig. 3 shows the connections Inside the bulb, 181 being the ten cathodes, connected to ten terminals 182, the ten anodes Ml being connected to terminal 184. A resistance 181 may also be mounted in the base III and connected to terminals 184 and 181.

It will be obvious from the foregoing descrip- 5 Won of the characteristics of the glow lamp that If a potential between the minimum and igniting potential is applied between the common anode and all ten cathodes, any one of the ten numerals may be displayed by t

he momentary application 10 of the Igniting potential to the corresponding cathode. This initiates the glow discharge at the selected cathode which Is then maintained by the intermediate potential after the igniting potential Is removed while all other cathodes will i .-> remain dark, since the discharge of these cathodes had not been Initiated by the application of the Igniting potential. To extinguish the glowing cathode, the potential of this cathode, or of all cathodes, is momentarily reduced to a value be- -20 low the minimum potential or to zero. Thereafter, any other cathode may be caused to glow by momentarily applying to It the Igniting potential.

Thus the described glow lamp may be used to 25 display any one of the ten numerals at will, and it will be obvious that, instead of ten numerals, letters or any other desired characters may be displayed by giving the cathodes the required shape, and that the construction Is not limited to ten :::) characters, but permits the use of a larger or smaller number of different characters.

In the arrangement described above, one control wire is required for each cathode or character to be displayed. Where a large number of 35 glow lamp indicators are required to display the desired information, the number of control wires becomes considerable, and to reduce the necessary number of control wires to one individual wire per glow lamp indicator and a number of common ±3 control wires corresponding to the number of characters in each lamp, the invention makes use of the control circuit shown in Fig. 4.

In this circuit all cathodes corresponding to the numeral 1 are connected to the common wire 4.-, 481 and similarly the cathodes 2 to 8 and 8 are connected to common wires 482 to 488 and 418, respectively. Each of these ten wires is connected over a break contact of the ten number keys 411 to 428 to the negative pole of the battery 421, 50 which supplies the intermediate potential. The anodes of each of the glow lamps are connected through resistances 411 to 414 to the positive pole of the battery 421. In this manner Intermediate potential is applied to all cathodes. 05

If it is desired to light, for example, numeral 1 of glow lamp 441, the key 451 associated with this lamp is operated, thereupon number key 411 and then the common sending key 424. When key 451 is operated, all ten pairs of electrodes of glow GO lamp 441 are short-circuited from the anodes of lamp 441 over make contact of key 451, break contact of key 424, break contacts of the ten keys 411 to 428, wires 481 to 418, to the ten cathodes of lamp 441. This has no result if all lamps are 05 dark and will not affect any of the other lamps, such as 442, 441, 444, etc., which all remain connected to battery 421. Upon operation of key 411, cathodes I of all lamps 441, 442, etc. are disconnected from the negative pole of battery 70 421 at the break contact of key 411 and connected over the make contact of this key and rectifier 425 to the negative pole of battery 421. This has no effect upon any of the lamps, as the cathodes remain connected to the negative pole of 7« battery 421 and the rectifier 425 inserted in the circuit does not change the potential.


When the key 424 is operated, auxiliary battery 423 is connected in parallel with rectifier 425, 6 thus in effect placing battery 423 in series with battery 421 and thereby raising the potential on cathodes I on wire 401 to a value higher than the intermediate potential but not quite high enough to ignite the cathodes. This circuit is traced
from cathodes I of glow lamps 441 to 444 over wire 401, make contact of operated key 411, thence in parallel through rectifier 425 and through upper make contact of key 424 and battery 423 to battery 421, through battery 421 and resistances 431 to 434 to the anodes of glow lamps 441 to 444. Rectifier 425 serves to prevent short circuiting battery 423. At the same time the short-circuit on lamp 441 is opened at the break contact of key 424 and auxiliary battery 422 is connected in series with battery 421 over key 451 to lamp 441 only. This circuit is traced from cathode I of glow lamp 441 over wire 401, make contact of operated key 411, thence in parallel through rectifier 425 and through upper make contact of key 424 and battery 423 to battery 421, through battery 421, and thence in parallel through resistance 431 and through battery 422, lower make contact of key 424 wire 461 and make contact of key 451 to the anodes of glow lamp 441.

Battery 422 is of such potential that its addition to the potential of battery 421 is not quite sufficient to reach the igniting potential. At cathode I of lamp 441, however, the potential applied is that of batteries 421, 422 and 423 added together and
this is higher than the igniting potential, so that cathode I of lamp 441 is ignited. Cathodes I of all other lamps have impressed upon them the potential of battery 421 plus that of battery 423, which remains below the igniting potential, so that none of these cathodes will begin to glow. Cathodes 2 to 9 and 0 of lamp 441 have impressed upon them the potential of battery 421 plus that of battery 422, which is below the igniting potential, so that no one of these cathodes will begin to glow. The only cathode where the igniting potential is reached is cathode I of lamp 441 where the additional potentials of both auxiliary batteries 422 and 423 are added to that of battery 421. Consequently cathode I of lamp 441 is the only one that will light.

After this cathode is lighted, first key 451 and then keys 411 and 424 are released. The release of key 451 removes the additional potential of battery 422 from lamp 441, but cathode I of this

lamp remains illuminated through batteries 421 and 423 in series. This circuit is the same as that described above for connecting battery 423 in series with battery 421. When keys 411 and 424 are released, auxiliary battery 423 is also removed from the circuit, but cathode I of lamp 441 remains lit, in as much as the potential of battery 421 is above the minimum potential and is sufficient to maintain the glow discharge. The circuit for cathode I of lamp 441 is traced from this cathode over wire 401, normally closed contact of key 411,

battery 424, resistance 431 to the anodes of lamp

441. The control circuit is now back to normal

and cathode I of lamp 441 is lit.

If it is desired to extinguish cathode I of lamp 441 and to light cathode 2 of this lamp in its stead, first key 451 is operated and then keys 412 and 424. The operation of key 451, as described above, short-circuits lamp 441, thereby extinguishing cathode I of this lamp. The subsequent operation of keys 412 and 424 thereupon initiates
the discharge of cathode 2 of lamp 441 in the above described manner. Thus it will be evident that any desired cathode of any of the lamps may be lighted at will by means of the operation of the proper keys. The operation of the common 6 keys has no effect upon any lamp whose individual key, such as 451, 452, etc., is not operated. In the case described above, it is to be noticed that the potential of battery 421 plus that of battery 423 is impressed upon control wire 401 when keys 411 10 and 424 are operated. This potential is still below the igniting potential, and cathodes I of all lamps where this cathode is dark, remain dark. In those lamps where this cathode happens to be lit, the additional potential will cause a slight bright- 15 ening of the glow, but has no other effect upon their operation. It will be noticed that keys 411 to 420 are provided with make-before-break contacts, so that the operation of these keys never interrupts the battery circuit.

It is possible to control several lamps at the same time by operating several of the keys 451, 452 etc. before the keys 411 to 420 and 424 are operated. In this case the same numeral will be displayed on all the lamps which are controlled 25 simultaneously. It is not possible to light erroneously more than one cathode in each lamp inasmuch as the value of the series resistances 431, 432 etc. is such that the combined voltage drop occasioned by two or more cathodes glowing at 30 the same time brings the potential across the electrodes to a value below the minimum potential. In such a case all the cathodes of the lamp in question are extinguished as soon as the sending keys are released.

It will be obvious that this method of control can be applied to an unlimited number of lamps. Besides the common control wires 401 to 410, the number keys 411 to 421, the sending key 424, the batteries 421, 422 and 423, and the rectifier 425, 40 each lamp requires one individual control key, such as 451, 452, etc., one resistance such as 431, 432, etc., and one individual control wire such as 461, 462, etc. It will be obvious to those skilled in the art that relay contacts may be substituted 45 for the keys without affecting the method of operation.

In the well-known grid glow lamp a third electrode, the so-called grid, is interposed between the cathode and anode. When a negative bias g0 potential is applied to this grid, the result is an increase of the potential required for igniting the discharge. When the grid bias is gradually reduced, the discharge sets in at a certain definite value. Thereafter the grid bias may be increased 5g again without affecting the discharge, since the negative grid attracts a space charge of positive ions from the glow discharge, which effectively neutralizes the grid. This principle may also be used for the present invention. Fig. 5 shows the CO internal circuit of a glow lamp indicator using this principle. The mechanical

construction is substantially the same as illustrated in Pigs. 1 and 2. Electrically, however, all cathodes 50 f are connected to a common terminal 502, while the 65 anodes 503 are connected to terminal 504. Ten gr'ds 505 are interposed between the cathodes and anodes and connected individually to ten terminals 507. A potential below the igniting value impressed upon terminals 502 and 504 will not 70 cause the discharge to start. The ten grids 505 are normally connected to a negative grid bias potential. To start the discharge at any one of the cathodes, its corresponding grid bias is lowered to a point where the discharge will set in. 76 Thereafter, the grid bias may be returned to its normal value without affecting the discharge that has set in. In the actual construction of the glow lamp indicator, the grids may take the form of a

2,149,106

5 short piece of wire interposed between the cathodes and anodes.

The control circuit shown in Fig. 6 for the grid glow lamp indicator is similar in principle to that shown in Fig. 4 for the ordinary glow lamp indicator, the only changes being those made necessary by the characteristics of the grid control principle. The cathodes of all lamps 641, 642, etc. are connected to the negative pole of battery 621 and the anodes through individual resistances 631, 63J etc. to the positive pole of the same battery.

Battery 621 supplies a potential sufficient to maintain the glow discharge after it has once set in, but insufficient to initiate the glow disCharge.

Grids I of all lamps 641, 642, etc. are connected to the common control lead 601, and the other grids 2 to 9 and 0 similarly to control wires 602 to 610. All ten wires 601 to 610 are connected through break contacts of the associated keys 611 to 620 to point 625 of the main battery 621, this point being near the negative pole and thus impressing a negative grid bias upon all grids. In order to light cathode I of lamp 641, for example, first the control key 651 associated with this lamp is operated and then the common control key 611 associated with grids I and the sending key 624. The operation of key 651 short-circuits the lamp 641 from the anodes over makeS contact of key 651, individual control lead 661, break contact of key 624 to the cathodes. This short-circuit extinguishes any cathode of lamp 631 that may be lit at this time without affecting any of the other lamps. When key 611 is operAtIoNated, the grid bias on grids I of all lamps 641,642, etc. is disconnected from point 625 near the negative pole of the main battery 621 and connected to point 623 which is nearer the positive pole of this battery.

Keys 611 to 620 are provided with make-beforebreak contacts to prevent interruptions of the battery circuit. Rectifier 626 serves to prevent short-circuits between points 623 and 625 during the time while the make and break contacts of keys 621 to 620 are both closed.

Although the operation of key 611 changes the bias on grids I of all lamps, this change does not affect any of the lamps as long as their individual control keys 651 etc. are in the normal position.

In some of these lamps cathode I may be dark and in others it may be glowing, depending upon preceding control operations. In the lamps whose cathode I is dark, this cathode will remain dark, because the voltage of the main battery 621 is insufficient to start a discharge even with reduced grid bias. On the other hand, in the lamps' where cathode I is glowing, the discharge is not affected by changes in grid bias, so that these cathodes will continue to glow.

When key 624 is operated, the short-circuit on lamp 641 is opened at the break contact of key 624 and the anodes of lamp 641 are connected to the auxiliary battery 622 which is in series with the main battery 621 and raises the potential on the ten pairs of electrodes in lamp 641 to a value which in itself is not sufficient to initiate the discharge on those electrodes whose grid has the normal negative grid bias from point 625 of the main battery. However, where the increased potential on the anodes and the reduced grid biasfrom point 121 of the main battery come together, that is, at anode I, the combined effect of the increased potential on the lamp and the lowered grid bias is to cause the discharge to set in. As a result, the discharge sets in at cathode I of lamp a 641.

When key 651 is released, the Increased potential on lamp 641 is removed and this lamp now receives its potential over resistance 631 from the main battery 621. This potential is sufficient jo to maintain the discharge irrespective of the value of the grid bias. The release of keys 611 and 625, whereby the grid bias is restored to its normal value, therefore has no further effect upon the discharge at cathode I of lump 641.
In a similar manner al< other numerals in any of the lamps may be displayed at will by proper operation of the control keys. If it is desired to extinguish a lamp without lighting a new number, it is only necessary to operate the associated indi- 20 vidual control key, such as 4SI, 452, etc. or 651, 652 etc., whereby the associated lamp is shortcircuited in Figs. 4 and 6.

Figs. 7 and 8 illustrate the application of the new glow lamp indicator to a stock quotation sys- 25 tern, although it will be understood that the principle of this invention is by no means limited to stock quotation systems, but may be used to advantage in any system where it is necessary to display information by numerals, letters or any 30 other characters or symbols. It will also be understood that the new glow lamp indicator may be constructed in any desired shape or size up to the largest dimensions. The circuit shown in Figs. 7 and 8 makes use of the method of control ,•>.'> shown in Fig. 4, but it will be understood that it may be modified to the method of control shown in Fig. 6 by any one skilled in the art.

The stock quotation system illustrated is arranged for a maximum of 1500 different stocks, 40 giving for each stock the hundreds, tens and units digits and fractions (in eighths) of the closing price of the preceding day, and the tens and units digits and fractions (in eighths) of theopening, highest, lowest and last price of the current 45 day. It is capable of transmitting two quotations per second or 120 quotations per minute with the customary speed of telegraphic transmission over the line. Contrary to well-known stock quotation systems in use at the present time, 50 where the speed of transmission is governed chiefly by the time required for sending the necessary number of impulses into the mechanical indicators, the stock quotation system disclosed herein is limited in speed only by the transmission over the line, the local control of the new glow lamp indicators being accomplished practically instantaneously without recourse to a varying number of impulses.

I

n the system shown, first the desired stock is 60 selected by transmitting the hundreds, tens and units digits identifying the stock, next a code is transmitted to select the range, i. e. the close, open, high, low or last price or any desired combination thereof, and finally the tens and units r.r> digits and the fractions of the price are transmitted. The transmission is performed on the startstop principle by means of a four unit code, that is, each digit is represented by four line impulse spaces and the selected number is identified by 70 the absence, called "marking current", or presence, called "spacing current", of line current during each of these four spaces. The codes used are shown in the following table, but it will be understood that any other combination of.