PHILIPS 26C854/60R EL GRECO CHASSIS K12 (20AX) INTERNAL VIEW.

The PHILIPS CHASSIS K12 WAS the first PHILIPS tv chassis monocarrier type. Previous K11 was a dual panel development type.

The PHILIPS chassis K12 doesn't share technology from previous CHASSIS K11 except for the 20AX SYSTEM CRT TUBE.

The chassis is a pretty unique type because it was introducing improvements and technology philosophy kindly singular and unique in it's fashion.

From signal processing to Video Matrixing to power supply technology and many further aspects this chassis was a reference to understand PHILIPS development flexibility.

This chassis has known many further versioning and enhancements even in more sophisticated and complex types with different CRT TUBE like the 30AX FAMILY.

Modularity is the main concept of construction but some Units like the E/W and FRAME UNIT gave problems in the insertion slot which often was burning the contacts producing several faults.

The solution was simple: A complete hardening with soldering and reworking of the enpoints of the Units contacts was a definitive solution for long time. But when the damage was more extended the only solution was to direct wire all contacts from Unit to chassis !



Line Deflection + EHT, Line synchronized Supply, EW Correction + Supply, Frame Deflection, Signal Section Parts.

PHILIPS 26C854/60R CHASSIS K12 (20AX) VARACTOR TUNER BAND SWITCHING AND SIGNAL INDICATING CIRCUITRY:

A signal receiver having VHF and UHF varactor tuners and a channel indicating group for the upper and lower VHF channels and for the UHF channel includes band switching circuitry for selectively coupling a plurality of power sources to the tuners in accordance with activation of a particular channel indicating group. Also, the band switching circuitry is utilized in conjunction with a channel indicating meter circuit for providing a visual indication of the channel selected in each one of the channel indicating groups.

1. In a signal receiver having first, second, and third potential sources; VHF and UHF varactor tuners; first, second, and third channel indicating groups formed for selective coupling to the first potential source; and band switching circuitry having first, second, and third switching means for coupling the potential sources to the VHF and UHF tuners, channel indicating meter circuitry comprising: 2. The channel indicating meter circuitry of claim 1 wherein said means for coupling one of said pair of terminals of said channel indicating meter to a potential reference level and to the junction of said second switching means and third potential source includes a first unidirectional conduction device coupled intermediate said terminal and said potential reference level and a second unidirectional conduction device coupling the junction of said first unidirectional conduction device and potential reference level to said junction of said second switching means and third 3. The channel indicating meter circuitry of claim 1 wherein said means for coupling the other one of said pair of terminals of said channel indicating meter includes a first impedance coupling said terminal to a potential reference level and a second impedance coupling said terminal to 4. The channel indicating meter circuitry of claim 1 wherein said channel indicating meter provides a visual indication of a selected signal channel in accordance with the potential applied thereto from said channel 5. The channel indicating meter circuitry of claim 1 wherein said channel indicating meter includes a plurality of meter scales for visually indicating channels in the upper and lower bands of the VHF varactor tuner 6. In a signal receiver having first, second, and third potential sources; VHF and UHF varactor tuners; first, second, and third channel indicating groups each formed for selective coupling to a first potential source, band switching and channel indicating meter circuitry comprising in combination: 7. The band switching and channel indicating meter circuitry combination of claim 6 including a first impedance coupling said first terminal to said potential reference level and a second impedance coupling said first 8. The band switching and channel indicating meter circuitry combination of claim 6 including a first unidirectional conduction device coupling said 9. The band switching and channel indicating meter circuitry combination of claim 6 including a first unidirectional conduction device coupling said second terminal to said potential reference level and a second unidirectional conduction device coupled to said first unidirectional conduction device and potential reference level and to said second switching means and third potential source.

Description:

CROSS-REFERENCE TO OTHER APPLICATIONS

An application entitled "Pushbutton Tuning System" filed concurrently herewith in the names of William Lee Arrington and Lee Irving Merz (Ser. No. 326,759) and assigned to the assignee of the present application relates to a pushbutton tuner for selecting a signal channel wherein each one of the channels provide an output potential suitable for use with the band switching and channel indicating meter circuitry set forth in the present application.

BACKGROUND OF THE INVENTION

Generally, present-day television receivers utilize both VHF and UHF tuners with the VHF tuner having the usual upper band portion covering channels 7-13 and a lower band portion covering channels 2-6. Also, present-day tuners are commonly of the varactor type wherein channel or frequency selection is determined by the potentials applied to the tuners.

In the prior art it has been a common practice to provide ordinary mechanical switches for band switching the high and low bands of the VHF tuner and the UHF tuner. Moreover, printed circuit type switches have been utilized in an effort to overcome the usual problems associated with mechanical switches. However, printed circuits have not provided the answers for such mechanical problems as wear, contact failure, and lack of reliability even though the cost is undesirably increased.

Additionally, total electronic tuning systems using logic and digital techniques have been proposed for use with varactor tuners. In one known example, channel numbers are coded into binary numbers and the selection of a preset voltage on varactor diodes is made by a corresponding binary coded signal.

However, electronic tuning systems using logic and digital systems appear to be relatively expensive, require a large number of components, and have a reliability and repeatability factor which is relatively unknown in the industry. Moreover, repair and service problems associated with such a departure from known and more frequently employed techniques remains an unknown factor.

Additionally, it is highly desirable to provide some form of visual indicating apparatus whereby an operator is aware of the approximate tuning of the receiver. Obviously, the provision of such visual indicating apparatus at a minimum cost of materials and labor is of utmost importance.

OBJECTS AND SUMMARY OF THE INVENTION

An object of the present invention is to provide enhanced band switching circuitry for a multi-band signal receiver. Another object of the invention is to provide improved electronic band switching circuitry for varactor tuners in a signal receiver. Still another object of the invention is to provide improved visual indicating apparatus utilized with band switching circuitry in a multi-band signal receiver.

These and other and further objects, advantages and capabilities are achieved in one aspect of the invention by a multi-band signal receiver having VHF and UHF varactor tuners with channel indicating groups for the upper and lower VHF bands and the UHF band and band switching circuitry for selectively coupling a plurality of potential sources to the tuners in accordance with the particular channel indicating group selected. Also, channel indicating meter circuitry acts in conjunction with the band switching circuitry to provide a visual indication of the signal channel selected.

BRIEF DESCRIPTION OF THE DRAWING

FIG. 1 is a diagrammatic illustration, in block and schematic form, of electronic band switching and channel indicating meter circuitry suitable for use in a multi-band signal receiver; and

FIG. 2 is an alternative embodiment of a channel indicating meter circuit.

PREFERRED EMBODIMENT OF THE INVENTION

For a better understanding of the present invention, together with other and further objects, advantages and capabilities thereof, reference is made to the following disclosure and appended claims in conjunction with the accompanying drawings.

Referring to the drawings, band switching and channel indicating meter circuitry suitable for use in a television receiver includes first, second, and third channel indicating groups 3, 5, and 7 respectively. Each one of the channel indicating groups, 3, 5, and 7 includes a plurality of parallel coupled signal channels and each signal channel has a series connected neon bulb 9 and selector switch 11 coupled to a first potential source 13. Also, each channel includes a signal selector switch 14, ganged to the selector switch 11, and connected to an adjustable resistor 12 intermediate the first potential source 13 and a potential reference level such as circuit ground. Thus, the first channel indicating group 3, preferably includes signal channels 7-13, the second channel indicating group 5, includes signal channels 2-6, and the third channel indicating group 7 includes a selection of channels 14-83.

In operation, selection of a signal channel in any one of the first, second, or third channel indicating groups 3, 5, and 7 is effected upon closure of a selector switch 11 whereupon a potential from the first potential source 13 is applied to the series connected neon bulb 9 by the selector switch 11. Also, a potential representative of a signal channel within the channel indicating group 3, 5, and 7 is available at the ganged signal selector switch 14. Thus, there is provided a potential representative of the channel indicating group and another potential representative of a specific channel within the channel indicating group.

Coupled to the channel indicating groups 3, 5, and 7 respectively, is band switching circuitry including first, second, and third switching means 15, 17, and 19 respectively. The first switching means 15, in the form of a transistor 16 in this instance, has a base electrode coupled to the first channel indicating group 3, representative of the upper band of signal channels 7-13. A collector electrode is coupled to a second potential source B+ while an emitter electrode is coupled to a varactor or varicap VHF type tuner 21. Obviously, the transistor may be replaced by other forms of electronic switching apparatus.

The second switching means 17 includes a pair of DC coupled PNP and NPN transistors, 23 and 25 respectively. The PNP transistor 23 has an emitter coupled to the second potential source B+, a collector DC coupled to the base of the NPN transistor 25, and a base electrode coupled via a biasing resistor 27 to circuit ground and by way of a first unidirectional conduction device 29 to the second channel indicating group 5 (channels 2-6). Also, a second unidirectional conduction device 31 couples the junction of the first unidirectional conduction device 29 and second channel indicating group 5 to the junction of the first channel indicating group 3 and first switching means 15. The NPN transistor 25 has a collector electrode coupled to the second potential source B+ and an emitter electrode coupled via a resistor 33 to a third potential source B- and to the VHF varactor or varicap tuner 21.

The third switching means 19 includes a transistor 35 having a base electrode connected to the third channel indicating group 7 and a collector electrode coupled to the second potential source B+. The emitter electrode of the transistor 35 is coupled to a UHF varicap or varactor tuner 37.

Additionally, channel indicating meter circuitry includes a meter 39 having a first terminal 41 coupled via resistors 42 and 44 to the signal selector switches 14 of the series connected first, second, and third channel indicating groups 3, 5, and 7 respectively and to the junction of the VHF and UHF tuners 21 and 37. This first terminal 41 of the meter 39 is also connected via a series connected first unidirectional conduction device 43 and resistor 45 to circuit ground. The junction of this first unidirectional conduction device 43 and resistor 45 is coupled via a second unidirectional conduction device 47 to the junction of the second switching means 17, the resistor 33 connected to the third potential source B-, and to the VHF tuner 21. Moreover, a second terminal 49 of the indicating meter 39 is coupled by a first resistor 51 to a potential reference level and by a second resistor 53 to the second potential source B+.

FIG. 2 illustrates an alternative form of channel indicating circuitry wherein an indicating meter 39 has a first terminal 55 coupled to the signal indicating or tuning means 57 of the first, second, and third channel indicating groups 3, 5, and 7 and via a series connected diode 59, first alterable resistor 61, and second alterable resistor 63 to a potential reference level. The terminal 55 is also coupled to the second alterable resistor 63.

A second terminal 65 of the meter 39 is coupled to the second potential source B+ and to the potential reference level. Also, the second terminal 65 is coupled via a resistor 67 and diode 69 to the junction of the second switching means 17, to the resistor 33 coupled to the third potential source B-, and to the VHF tuner 21. Moreover, a third diode 71 couples the second diode 69 and junction of the second switching means 17 to the alterable resistor 61.

As to operation, selection of any one of the parallel connected signal channels (channels 7-13) of the first channel indicating group 3 causes application of a potential from the first potential source 13 to the first switching means 15. The first switching means 15 including the transistor 16 is normally nonconductive or open-circuited. However, application of a potential from the potential source 13 renders the transistor 16 conductive whereupon a potential from the second potential source B+ is applied to the VHF tuner 21.

Also, it is to be noted that the second switching means 17 is normally conductive whereupon the potential from the third potential source B- coupled to the VHF tuner 21 is "swamped-out" by the potential from the second potential source B+. Thus, activation of a signal channel (channels 7-13) of the first channel indicating group 3 causes application of a potential from the second potential source B+ to the VHF tuner 21 and "swamping-out" of a potential from the third potential source B- which is, in turn, coupled to the band switching terminal of the VHF tuner 21.

Activation of a parallel connected signal channel of the second channel indicating group 5 which includes the low portion (channels 2-6) of the VHF signal band, causes application of a potential from the first potential source 13 to the first switching means 15 via the diode 31 and to the second switching means 17 via diode 29. Thereupon, the first switching means 15 is rendered conductive and a potential from the second potential source B+ is applied to the VHF tuner 21. The second switching means 17 is rendered non-conductive, due to the bias potential developed across the resistor 27 via diode 29, whereupon the third potential source B- is no longer "swamped-out" but rather, is applied to the band switching terminal of the VHF tuner 21.

Selection of one of the parallel connected signal channels (channels 14-83) in the UHF band of signals causes activation of the third channel indicating group 7. Thereupon, a potential from the first potential source 13 is applied to the third switching means 19. This applied potential renders the transistor 35 conductive whereupon a potential from the second potential source B+ is applied to the UHF tuner 3. Thus, activation of the UHF tuner 3 is provided upon selection of a high frequency signal channel (channels 14-83).

It may be noted that each one of the switches 11 in each channel of the first, second, and third channel indicating groups 3, 5, and 7 is mechanically connected to the others in a manner such that only one of the switches 11 is operable at a time. In other words, activation of a second one of the switches 11 causes de-activation of a first one of the switches 11. Such a switching system is clearly set forth in the cross-referenced application entitled "Pushbutton Tuning System" filed concurrently herewith.

Additionally, the channel indicating meter circuitry is responsive to selection of a channel in any one of the first, second, and third channel indicating groups 3, 5, and 7. Upon selection of a channel (channels 7-13) in the upper portion of the VHF signal range, a positive potential provided by second potential source B+ will appear at the junction of the second switching means 17, resistor 33 coupled to the third potential source B-, and the band switching terminal of the VHF tuner 21. This positive potential appears at the junction of the first and second unidirectional conduction devices 43 and 47 back biasing the first unidirectional conduction device 43 whereupon conduction therethrough ceases and the potential of the first terminal 41 of the indicating meter 39 is raised to provide an approximate half-scale reading of the channel indicating meter 39. The indicating meter 39 is further advanced in accordance with a potential applied to the first terminal 41 via the series connected resistors 42 and 44 coupled to the particular channel (channels 7-13) selected. Thus, this further advance of the indicating meter 39 is effected by current flow via the series connected resistors 44 and 42, the indicating meter 39 and the resistor 51 coupled to circuit ground.

Upon selection of a channel (channels 2-6) in the lower portion of the VHF signal band, the third potential source B- is no longer "swamped-out" because of the non-conductivity of the second band switching circuitry 17 but rather, is applied to the second unidirectional conduction device 47 to effect a back-bias thereon. Thus, current supplied by the tuning voltage, derived via the first potential source 13 and second channel indicating group 5, flows through the indicating meter 39 via the resistors 44 and 42 and to circuit ground by way of the resistor 51. Also, a predetermined portion of this current flows to circuit ground by way of the first unidirectional conduction device 43 and resistor 45 coupled thereto. Thus, compression to the lower portion of the scale of the indicating meter 39 provides a maximum utilization of scale length.

In the UHF signal band (channels 14-83), a positive potential appears at the first terminal 41 as previously explained with respect to the upper portion (channels 7-13) of the VHF signal band. Also, a substantially equal positive potential is applied to the second terminal 49 of the indicating meter 39 as derived from the second potential source B+ via the resistor 53 coupled to the second terminal 49. Thus, by application of substantially equal potentials to the first and second terminals 41 and 49 of the indicating meter 39, the total length of the indicating scale may be employed. Moreover, the tuning voltage, as derived from the first potential source 13 and third channel indicating group 7, is applied via the resistors 44 and 42 to the first terminal 41 of the indicating meter 39 to provide a visual indication of the channel selected.

Thus, there has been provided a unique band switching circuit as well as a unique channel indicating meter circuit for a multi-channel signal receiver employing both VHF and UHF varicap or varactor tuners. This improved circuitry is inexpensive of components, circuitry, and assembly time while reliability and repeatability of results are enhanced. Not only is the undesired wear associated with mechanical systems eliminated but, more importantly, the electronic switching technique increases the time switching capabilities without appreciable increase in cost.

While there has been shown and described what is at present considered the preferred embodiment of the invention, it will be obvious to those skilled in the art that various changes and modifications may be made therein without departing from the invention as defined by the appended claims.

PHILIPS 26C854/60R CHASSIS K12 (20AX) Fine tuning indicator for TV with synchronous detector:

A fine tuning indicator for a television receiver uses a synchronous detector for generating a control voltage indicative of the tuning accuracy. The control voltage is sensed by an indicating circuit which produces a visual indication of the tuned or mistuned condition of the receiver.

1. In combination in a television receiver;
a synchronous detector receiving a variable frequency input signal and generating a variable DC control voltage when locked to the frequency of the input signal, said control voltage having a predetermined value when the input signal is of a nominal frequency indicative of proper tuning of the receiver and having values which deviate from said predetermined value in accordance with deviations from the nominal frequency of the input signal; and
indicating means responsive to said control voltage for generating a visual indication of the deviation of said control voltage from said predetermined value,
whereby said indicating means provides a visual indication of the deviation from proper tuning of the television receiver.

2. The combination set forth in claim 1 wherein the input signal to said synchronous detector is the intermediate frequency television signal, including video components, and said synchronous detector has an output comprising demodulated video. 3. The combination set forth in claim 2 wherein said indicating means includes at least one light-emissive element adapted to be energized when said control signal deviates from said predetermined value by a predetermined amount. 4. The combination set forth in claim 3 wherein said indicating means includes means for generating first and second reference voltages whose values are, respectively, greater than and less than the predetermined value of said control voltage, and including means for actuating at least one visual indicator when said control voltage is greater than said first reference voltage and less than said second reference voltage. 5. The combination set forth in claim 4 wherein said actuating means includes a first comparator receiving said control voltage and said first reference voltage, and a second comparator receiving said control voltage and said second reference voltage, said first and second comparators generating outputs representative of said control voltage being greater than and less than said first and second reference voltages, respectively, and said comparator outputs being coupled to at least one visual indicator for actuation thereof. 6. The combination set forth in claim 5 wherein said means for generating said first and second reference voltages includes a resistive voltage divider coupled across a voltage potential, said voltage divider including first and second variable resistors each having a wiper arm adapted to receive from its variable resistor one of said first and second reference voltages. 7. In combination in a television receiver;
a synchronous detector receiving an intermediate frequency television signal and generating a variable DC control voltage whose value deviates about a predetermined value in accordance with deviations from a nominal frequency of the intemediate frequency signal;
means for generating first and second reference voltages whose values are, respectively, greater than and less than the predetermined value of said control voltage;
means for comparing said first and second reference voltages to said control voltage and for generating a first indicating signal respresentative of said control voltage having a value between said first and second reference voltages and for generating a second indicating signal representative of said control voltage having a value outside said first and second reference voltages; and
means responsive to said first and second indicating signals for generating a visual indication of the results of the comparison of said control voltage with said first and second reference voltages,
whereby a visual indication is provided of the accuracy to which the television receiver is tuned.
8. The combination set forth in claim 7 wherein said means responsive to said first and second indicating signals includes at least one light-emissive element responsive to at least said second indicating signal for providing a luminance output when the value of said control voltage is outside the values of said first and second reference voltages.

Description:

BACKGROUND OF THE INVENTION
This invention relates generally to television receivers and particularly to fine tuning indicators for use with television receivers.
Television receivers generally provide a fine tuning control to permit a viewer to accurately tune the receiver to a selected television channel. However, most viewers are not able to determine precisely when the receiver is accurately tuned. Consequently, the television receiver may be inaccurately tuned and, as a result, not produce an optimum picture.
In the highly competitive television market, performance, reliability and cost are all of major concern. In this respect, adequate performing fine tuning indicators are generally cumbersome and expensive. Even though most "full feature" receivers include a form of automatic frequency control (AFC) for maintaining the tuner locked to the received signal, it is desirable to optimize receiver tuning prior to AFC activation. Consequently there is a need for fine tuning indicators to help the average viewer tune his receiver, at least initially before releasing the tuning system to AFC.
In video detection systems, the simple envelope detector has long been the most popular and have proven quite satisfactory. Recently however there has been a trend toward synchronous detectors, which offer enhanced performance.
The present invention offers attractive advantages with a receiver using a synchronous detector in that it uses the control voltage generated by the synchronous detector locking loop to control a fine tuning indicator, thus providing a reliable tuning indicator at a low cost.
OBJECTS OF THE INVENTION
Accordingly, it is a general object of the invention to provide an improved television receiver tuning indicator.
It is a more specific object of the invention to provide an accurate fine tuning indicator for a synchronous detector type television receiver.
SUMMARY OF THE INVENTION
The fine tuning indicator described herein utilizes the existing television synchronous detector for generating a control voltage of predetermined value when the receiver is properly tuned and for varying the value of the control voltage in accordance with the degree of mistuning of the receiver. The control voltage is sensed by an indicating circuit which produces a visual indication of the tuned or mistuned condition of the receiver.
BRIEF DESCRIPTION OF THE DRAWINGS
The above and other objects of the invention are more particularly set forth in the following detailed description and in the accompanying drawing which illustrates a preferred embodiment of a fine tuning indicator constructed in accordance with the invention.
DESCRIPTION OF THE PREFERRED EMBODIMENT
It is conventional to convert the received high frequency television signal to a lower frequency called the intermediate frequency (IF) signal. The IF frequency is generally 45.75 megahertz. After suitable amplification, the IF signal is demodulated to obtain the video and synchronizing components.
Most television receivers employ peak detectors for demodulating the IF signal. More advanced receivers incorporate a synchronous detector because it produces a demodulated signal of lower distortion. Such a synchronous detector includes a conventional APC (automatic phase control) loop which generates an error signal or control voltage for locking the frequency of the detector oscillator to the frequency of the IF signal. The detector oscillator and the IF signal are then fed to a multiplier whose output is a low distortion demodulated video signal. There are also so-called quasi-synchronous detectors which do not incorporate an APC loop. It should be remembered that for this invention only conventional synchronous detectors are of interest, that is, those which have a phase lock loop circuit.
It has been found that the control voltage developed by the synchronous detector can be advantageously used as an indicator of receiver tuning accuracy and in television receivers employing such synchronous detectors they may be used for driving a fine tuning indicator. Hence, the synchronous detector may be made to do double duty, namely, demodulate the IF signal and provide a control voltage for use as an indication of the accuracy of receiver tuning.
Referring now to the FIGURE, there is shown an exemplary embodiment of a fine turning indicator which employs a single synchronous detector 10 both for video demodulation and for providing a control voltage indicative of the accuracy of tuning of the receiver. An indicating circuit 12 responds to the synchronous detector control voltage and produces a visual indication of the accuracy of receiver tuning.

Referring more specifically to synchronous detector 10, it includes a voltage controlled detector oscillator 14, a first multiplier 16, an amplifier 18 (optional), a resistor 20, a second multiplier 22 and a series connection of a resistor 34 and a capacitor 36. Resistor 20, resistor 34 and capacitor 36 form a low pass filter network. Lead 24 supplies a conventional IF signal from an IF amplifier 26 and a limiter 28 to the input of the synchronous detector. The receiver circuitry preceding IF amplifier 26 is a conventional tuner and is, therefore, not shown.

The operation of synchronous detectors is well known. Suffice it to say that the various elements of detector 10 cooperate to lock the frequency of detector oscillator 14 to the frequency of the IF signal on lead 24. To control the oscillator frequency a control voltage is generated on a lead 30 which is at the junction of resistors 20 and 34. This control voltage is DC when lock has been achieved and, in the illustrated embodiment, has a predetermined magnitude when the IF signal is at its nominal frequency of 45.75 megahertz. Only when the receiver is properly tuned will the frequency of the IF signal be equal to 45.75 megahertz. Detuning of the receiver results in the frequency of the IF signal being either above or below nominal, depending on the direction of mistuning.
When the frequency of the IF signal deviates from nominal, the value of the control voltage on lead 30 deviates in a substantially linear manner. For example, when the IF frequency is above 45.75 megahertz, the control voltage is lower than the predetermined value. Similarly, when the IF frequency is low, the control voltage is higher than the predetermined value. Thus the control voltage on lead 30 is representative of the accuracy of receiver tuning.
To demodulate the IF signal, an output of oscillator 14 and the signal from IF amplifier 26 are fed to multiplier 22. At the output 32 of multiplier 22, demodulated video appears and is treated by conventional television circuitry (not shown) for creating a television picture.
The filtered control voltage at lead 30 is fed, via a resistor 38, to a pair of comparators 40 and 42. In general, comparators 40 and 42 compare the value of the control voltage with a pair of reference voltages which are, respectively, above and below the nominal or predetermined value. if the control voltage sensed by comparators 40 and 42 is outside the range of the reference voltages, one of a pair of visual indicators, in the form of light-emitting diodes 44 and 46, is energized to present a visual indication of the mistuning of the receiver.
More specifically, the comparison of the control voltage to the reference voltages is effected by coupling the control voltage to the negative input 48 of comparator 40 and to the positive input 50 of comparator 42, and by coupling one reference voltage to the positive input 52 of comparator 40 and the other reference voltage to the negative input 54 of comparator 42. Comparators 40 and 42 may include respective feedback resistors 41 and 43.
The voltages appearing at inputs 52 and 54 are developed by a voltage divider comprising fixed resistors 56 and 58 and a pair of variable resistors 60 and 62. The series combination of resistors 56, 58, 60 and 62 is coupled, as shown, between ground and a source of positive potention +V. A wiper arm 64, associated with variable resistor 60, picks off the desired reference voltage from resistor 60 and applies it to positive input terminal 52 of comparator 50 via a resistor 66. Wiper arm 68, associated with resistor 62, picks off the other reference voltage and applies it to negative input terminal 54 of comparator 42 via a resistor 70. Preferably, wiper arms 64 and 68 are adjusted such that the reference voltages appearing at input leads 52 and 54 of the comparators are above and below, respectively, the nominal or predetermined value of the control voltage. Thus a voltage range is formed within which the control voltage may vary without developing an indication of mistuning. For example, if the predetermined or nominal value of the control voltage is 5.8 volts, the reference voltage appearing at input 52 of comparator 40 may be 6.8 volts and the reference voltage appearing at input 54 of comparator 42 may be 4.8 volts. Accordingly, a tuning range is provided within which the control voltage may vary without generating an indication of mistuning. However, should the receiver be detuned such that the control voltage exceeds the level of voltage on input 52 of comparator 40, comparator 40 will develop an output for actuating light-emitting diode 46, thereby giving a visual indication of both the fact and direction of receiver mistuning. Should the receiver be mistuned in the opposite direction, the value of the control voltage at input 50 of comparator 42 will be less than the value of the reference voltage on input 54 and comparators 42 will develop an output for activating light-emitting diode 44.
It will be appreciated that when the receiver is properly tuned (within a predetermined tuning range) the value of the control voltage will be within the range of the reference voltages at inputs 52 and 54 of the comparators and neither of light-emitting diodes 44 or 46 will be activated. Thus, comparators 40 and 42 generate a first or non-actuating signal when the control voltage has a value between the reference voltages and a second or actuating signal when the control voltage is outside the reference voltages.
In some instances, synchronous detector 10 may require calibration in order to insure that the control voltage appearing at lead 30 is of a known or predetermined value when the receiver is properly tuned. Such calibration may be effected by applying a signal of precisely 45.75 megahertz to the synchronous detector at lead 24. With the synchronous detector now receiving a signal whose frequency is equal to the frequency of the IF signal when the receiver is properly tuned, oscillator 14 may be adjusted to develop at lead 30 a control voltage whose value is precisely equal to its predetermined or nominal value. In the example used, oscillator 14 may be adjusted to develop a control voltage at lead 30 of 5.8 volts.
The method of adjustment of oscillator 14 will vary according to its specific construction. However, where oscillator 14 includes an adjustable coil, the coil itself may be adjusted to achieve the desired level of control voltage on lead 30.
Because the value of control voltage on lead 30 generally varies in a known manner according to the deviation in frequency of the IF signal, the extent of its variation may be measured or calculated such that the reference voltages appearing at inputs 52 and 54 of comparators 40 and 42 may be set to correspond to known frequency deviations. For example, if the IF tuning limit for the receiver is ±150 kilohertz, the reference voltages appearing at inputs 52 and 54 of comparators 40 and 42 may be set to the values which the control voltage assumes with these IF frequency deviations. Within the ±150 kilohertz limits, neither of the light-emitting diodes will be activated. Should the IF frequency deviate by more than 150 kilohertz from nominal, one of diodes 44 or 46 will be activated to present to a viewer a visual indication of the mistuned condition. The viewer need only tune the receiver until neither of the light-emitting diodes is activated, at which point the viewer will know that the receiver is properly tuned (within a predetermined acceptable tuning range).
Although the embodiment described above has been illustrated as presenting a visual indication of mistuning by the activation of either of a pair of light-emitting diodes, many other visual methods of presenting an indication of mistuning are possible. For example, the outputs of the comparators may be applied to a logic gate whose output is coupled to a single light emissive element for activating that element only when the receiver is properly tuned. When the receiver is mistuned, the light emissive element will be off.
Other variations may include a tuning meter which may be calibrated to indicate proper tuning when the value of control voltage at lead 30 is at nominal and to indicate mistuning when the control voltage falls outside a predetermined range.
The major advantage of the fine tuning indicator described is that the synchronous detector can supply the control voltage for a visual tuning indicator in addition to demodulating the IF signal. Accordingly, the cost of implementing the fine tuning indicator is reduced.
Although the fine tuning indicator described has been illustrated in terms of a preferred embodiment, it will be obvious to those skilled in the art that many alterations and modifications may be made without departing from the spirit and scope of the invention. For example, many alternate visual indicators may be used to present an indication of receiver tuning in response to the control voltage developed by the synchronous detector. Tuning meters, one or more light-emitting diodes or lamps may be used and the reference voltages, if any, may be generated by any of many well-known circuits. Accordingly, all such modifications and alterations are deemed to be within the scope of the invention as defined by the appended claims.

PHILIPS  26C854/60R  EL GRECO  CHASSIS K12 (20AX) CIRCUIT ARRANGEMENT IN A PICTURE DISPLAY DEVICE UTILIZING A STABILIZED SUPPLY VOLTAGE CIRCUIT:

PHILIPS  26C854/60R  EL GRECO  CHASSIS K12 (20AX) Line synch Switched Mode Power Supply with Line deflection output Transistor Drive Circuit:PHILIPS POWER PACK
 A stabilized supply voltage circuit for a picture display device comprising a chopper wherein the switching signal has the line frequency and is duration-modulated. The coil of the chopper constitutes the primary winding of a transformer a secondary winding of which drives the line output transistor so that the switching transistor of the chopper also functions as a driver for the line output stage. The oscillator generating the switching signal may be the line oscillator. In a special embodiment the driver and line output transistor conduct simultaneously and in order to limit the base current of the line output transistor a coil shunted by a diode is incorporated in the drive line of the line output transistor. Other secondary windings of the transformer drive diodes which conduct simultaneously with the efficiency diode of the chopper so as to generate further stabilized supply voltages.

1. An electrical circuit arrangement for a picture display device operating at a given line scanning frequency, comprising a source of unidirectional voltage, an inductor, first switching transistor means for periodically energizing said inductor at said scanning frequency with current from said source, an electrical load circuit coupled to said inductor and having applied thereto a voltage as determined by the ratio of the ON and OFF periods of said transistor, means for maintaining the voltage across said load circuit at a given value comprising means for comparing the voltage of said load circuit with a reference voltage, means responsive to departures of the value of the load circuit voltage from the value of said reference voltage for varying the conduction ratio of the ON and OFF periods of said transistor thereby to stabilize said load circuit voltage at the given value, a line deflection coil system for said picture display device, means for energizing said line deflection coil system from said load voltage circuit means, means for periodically interrupting the energization of said line deflection coil comprising second switching means and means coupled to said inductor for deriving therefrom a switching current in synchronism with the energization periods of said transistor and applying said switching current to said switching means thereby to actuate the same, and means coupled to said switching means and to said load voltage circuit for producing a voltage for energizing said 2. A circuit as claimed in claim 1 wherein the duty cycle of said switching 3. A circuit as claimed in claim 1 further comprising an efficiency first 4. A circuit as claimed in claim 3 further comprising at least a second diode coupled to said deriving means and to ground, and being poled to 5. A circuit as claimed in claim 1 wherein said second switching means comprises a second transistor coupled to said deriving means to conduct simultaneously with said first transistor, and further comprising a coil coupled between said driving means and said second transistor and a third diode shunt coupled to said coil and being poled to conduct when said 6. A circuit as claimed in claim 1 further comprising a horizontal oscillator coupled to said first transistor, said oscillator being the 7. A circuit as claimed in claim 1 further comprising means coupled to said inductor for deriving filament voltage for said display device.

Description:

The invention relates to a circuit arrangement in a picture display device wherein the input direct voltage between two input terminals, which is obtained be rectifying the mains alternating voltage, is converted into a stabilized output direct voltage by means of a switching transistor and a coil and wherein the transistor is connected to a first input terminal and an efficiency diode is connected to the junction of the transistor and the coil. The switching transistor is driven by a pulsatory voltage of line frequency which pulses are duration-modulated in order to saturate the switching transistor during part of the period dependent on the direct voltage to be stabilized and to cut off this transistor during the remaining part of the period. The pulse duration modulation is effected by means of a comparison circuit which compares the direct voltage to be stabilized with a substantially constant voltage, the coil constituting the primary winding of a transformer.

Such a circuit arrangement is known from German "Auslegeschrift" 1.293.304. wherein a circuit arrangement is described which has for its object to convert an input direct voltage which is generated between two terminals into a different direct voltage. The circuit employs a switch connected to the first terminal of the input voltage and periodically opens and closes so that the input voltage is converted into a pulsatory voltage. This pulsatory voltage is then applied to a coil. A diode is arranged between the junction of the switch and the coil and the second terminal of the input voltage whilst a load and a charge capacitor in parallel thereto are arranged between the other end of the coil and the second terminal of the input voltage. The assembly operates in accordance with the known efficiency principle i.e., the current supplied to the load flows alternately through the switch and through the diode. The function of the switch is performed by a switching transistor which is driven by a periodical pulsatory voltage which saturates this transistor for a given part of the period. Such a configuration is known under different names in the literature; it will be referred to herein as a "chopper." A known advantage thereof, is that the switching transistor must be able to stand a high voltage or provide a great current but it need not dissipate a great power.
The output voltage of the chopper is compared with a constant reference voltage. If the output voltage attempts to vary because the input voltage and/or the load varies, a voltage causing a duration modulation of the pulses is produced at the output of the comparison arrangement. As a result the quantity of the energy stored in the coil varies and the output voltage is maintained constant. In the German "Auslegeschrift" referred to it is therefore an object to provide a stabilized supply

voltage device.

In the circuit arrangement according to the mentioned German "Auslegeschrift" the frequency of the load variations or a harmonic thereof is chosen as the frequency for the switching voltage. Particularly when the load fed by the chopper is the line deflection circuit of a picture display device, wherein thus the impedance of the load varies in the rhythm of the line frequency, the frequency of the switching voltage is equal to or is a multiple of the line frequency.




It is to be noted that the chopper need not necessarily be formed as that in the mentioned German "Auslegeschrift." In fact, it is known from literature that the efficiency diode and the coil may be exchanged. It is alternatively possible for the coil to be provided at the first terminal of the input voltage whilst the switching transistor is arranged between the other end and the second terminal of the input voltage. The efficiency diode is then provided between the junction of said end and the switching transistor and the load. It may be recognized that for all these modifications a voltage is present across the connections of the coil which voltage has the same frequency and the same shape as the pulsatory switching voltage. The control voltage of a line deflection circuit is a pulsatory voltage which causes the line output transistor to be saturates and cut off alternately. The invention is based on the recognition that the voltage present across the connections of the coil is suitable to function as such a control voltage and that the coil constitutes the primary of a transformer. To this end the circuit arrangement according to the invention is characterized in that a secondary winding of the transformer drives the switching element which applies a line deflection current to line deflection coils and by which the voltage for the final anode of a picture display tube which forms part of the picture display device is generated, and that the ratio between the period during which the switching transistor is saturated and the entire period, i.e., the switching transistor duty cycle is between 0.3 and 0.7 during normal operation.

The invention is also based on the recognition that the duration modulation which is necessary to stabilize the supply voltage with the switching transistor does not exert influence on the driving of the line output transistor. This resides in the fact that in case of a longer or shorter cut-off period of the line output transistor the current flowing through the line deflection coils thereof is not influenced because of the efficiency diode current and transistor current are taken over or, in case of a special kind of transistor, the collector-emitter current is taken over by the base collector current and conversely. However, in that case the above-mentioned ratios of 0.3 : 0.7 should be taken into account since otherwise this take-over principle is jeopardized.


As will be further explained the use of the switching transistor as a driver for the line output transistor in an embodiment to be especially described hereinafter has the further advantage that the line output transistor automatically becomes non-conductive when this switching transistor is short circuited so that the deflection and the EHT for the display tube drop out and thus avoid damage thereof.

Due to the step according to the invention the switching transistor in the stabilized supply functions as a driver for the line deflection circuit. The circuit arrangement according to the invention may in addition be equipped with a very efficient safety circuit so that the reliability is considerably enhanced, which is described in the U.S. Pat. No. 3,629,686. The invention is furthermore based on the recognition of the fact that the pulsatory voltage present across the connections of the coil is furthermore used and to this end the circuit arrangement according to the invention is characterized in that secondary windings of the transformer drive diodes which conduct simultaneously with the efficiency diode so as to generate further stabilized direct voltages, one end of said diodes being connected to ground.

In order that the invention may be readily carried into effect, a few embodiments thereof will now be described in detail by way of example with reference to the accompanying diagrammatic drawings in which:


FIG. 1 shows a principle circuit diagram wherein the chopper and the line deflection circuit are further shown but other circuits are not further shown.

FIGS. 2a, 2b and 2c show the variation as a function of time of two currents and of a voltage occurring in the circuit arrangement according to FIG. 1.

FIGS. 3a 3b, 3c and 3d show other embodiments of the chopper.

FIGS. 4a and 4b show modifications of part of the circuit arrangement of FIG. 1.

In FIG. 1 the reference numeral 1 denotes a rectifier circuit which converts the mains voltage supplied thereto into a non-stabilized direct voltage. The collector of a switching transistor 2 is connected to one of the two terminals between which this direct voltage is obtained, said transistor being of the npn-type in this embodiment and the base of which receives a pulsatory voltage which originates through a control stage 4 from a modulator 5 and causes transistor 2 to be saturated and cut off alternately. The voltage waveform 3 is produced at the emitter of transistor 2. In order to maintain the output voltage of the circuit arrangement constant, the duration of the pulses provided is varied in modulator 5. A pulse oscillator 6 supplies the pulsatory voltage to modulator 5 and is synchronized by a signal of line frequency which originates from the line oscillator 6' present in the picture display device. This line oscillator 6' is in turn directly synchronized in known manner by pulses 7' of line frequency which are present in the device and originate for example from a received television signal if the picture display device is a television receiver. Pulse oscillator 6 thus generates a pulsatory voltage the repetition frequency of which is the line frequency.

The emitter of switching transistor 2 is connected at one end to the cathode of an efficiency diode 7 whose other end is connected to the second input voltage terminal and at the other end to primary winding 8 of a transformer 9. Pulsatory voltage 3 which is produced at the cathode of efficiency diode 7 is clamped against the potential of said second terminal during the intervals when this diode conducts. During the other intervals the pulsatory voltage 3 assumes the value V _i . A charge capacitor 10 and a load 11 are arranged between the other end of winding 8 and the second input voltage terminal. The elements 2,7,8,10 and 11 constitute a so-called chopper producing a direct voltage across charge capacitor 10, provided that capacitor 10 has a sufficiently great value for the line frequency and the current applied to load 11 flowing alternately through switching transistor 2 or through efficiency diode 7. The output voltage V _o which is the direct voltage produced across charge capacitor 10 is applied to a comparison circuit 12 which compares the voltage V _o with a reference voltage. Comparison circuit 12 generates a direct voltage which is applied to modulator 5 so that the duration of the effective period δ T of switching transistor 2 relative to the period T of pulses 3 varies as a function of the variations of output voltage V ₀ . In fact, it is readily evident that output voltage V _o is proportional to the ratio δ :

V _o = V _i ^. δ

Load 11 of the chopper consists in the consumption of parts of the picture display device which are fed by output voltage V ₀ . In a practical embodiment of the circuit arrangement according to FIG. 1 wherein the mains alternating voltage has a nominal effective value of 220 V and the rectified voltage V i is approximately 270 V, output voltage V _o for δ = 0.5 is approximately 135 V. This makes it also possible, for example, to feed a line deflection circuit as is shown in FIG. 1 wherein load 11 then represents different parts which are fed by the chopper. Since voltage V _o is maintained constant due to pulse duration modulation, the supply voltage of this line deflection circuit remains constant with the favorable result that the line amplitude(= the width of the picture displayed on the screen of the picture display tube) likewise remains constant as well as the EHT required for the final anode of the picture display tube in the same circuit arrangement independent of the variations in the mains voltage and the load on the EHT generator (= variations in brightness).

However, variations in the line amplitude and the EHT may occur as a result of an insufficiently small internal impedance of the EHT generator. Compensation means are known for this purpose. A possibility within the scope of the present invention is to use comparison circuit 12 for this purpose. In fact, if the beam current passes through an element having a substantially quadratic characteristic, for example, a voltage-dependent resistor, then a variation for voltage V _o may be obtained through comparison circuit 12 which variation is proportional to the root of the variation in the EHT which is a known condition for the line amplitude to remain constant.

In addition this facilitates smoothing of voltage V _o since the repetition frequency of pulsatory voltage 3 is many times higher than that of the mains and a comparatively small value may be sufficient for charge capacitor 10. If charge capacitor 10 has a sufficiently high value for the line frequency, voltage V _o is indeed a direct voltage so that a voltage having the same form as pulsatory voltage 3 is produced across the terminals of primary winding 8. Thus voltages which have the same shape as pulsatory voltage 3 but have a greater or smaller amplitude are produced across secondary windings 13, 14 of transformer 9 (FIG. 1 shows only 2 secondary windings but there may be more). The invention is based on the recognition that one end of each secondary winding is connected to earth while the other end thereof drives a diode, the winding sense of each winding and the direction of conductance of each diode being chosen to be such that these diodes conduct during the same period as does efficiency diode 7. After smoothing, stabilized supply voltages, for example, at terminal 15 are generated in this manner at the amplitudes and polarities required for the circuit arrangements present in the picture display device. In FIG. 1 the voltage generated at terminal 15 is, for example, positive relative to earth. It is to be noted that the load currents of the supply voltages obtained in this manner cause a reduction of the switching power which is economized by efficiency diode 7. The sum of all diode currents including that of diode 7 is in fact equal to the current which would flow through diode 7 if no secondary winding were wound on transformer 9 and if no simultaneous diode were used. This reduction may be considered an additional advantage of the circuit arrangement according to the invention, for a diode suitable for smaller powers may then be used. However, it will be evident that the overall secondary load must not exceed the primary load since otherwise there is the risk of efficiency diode 7 being blocked so that stabilization of the secondary supply voltages would be out of the question.

It is to be noted that a parabola voltage of line frequency as shown at 28 is produced across the charge capacitor 10 if this capacitor is given a smaller capacitance so that consequently the so-called S-correction is established.

In FIG. 1 charge capacitors are arranged between terminals 15 etc. and earth so as to ensure that the voltages on these points are stabilized direct voltages. If in addition the mean value of the voltage on one of these terminals has been made equal to the effective value of the alternating voltage which is required for heating the filament of the picture display tube present in the picture display device, this voltage is suitable for this heating. This is a further advantage of the invention since the cheap generation of a stabilized filament voltage for the picture display tube has always been a difficult problem in transistorized arrangements.

A further advantage of the picture display device according to the invention is that transformer 9 can function as a separation transformer so that the different secondary windings can be separated from the mains and their lower ends can be connected to ground of the picture display device. The latter step makes it possible to connect a different apparatus such as, for example, a magnetic recording and/or playback apparatus to the picture display device without earth connection problems occurring.

In FIG. 1 the reference numeral 14 denotes a secondary winding of transformer 9 which in accordance with the previously mentioned recognition of the invention can drive line output transistor 16 of the line deflection circuit 17. Line deflection circuit 17 which is shown in a simplified form in FIG. 1 includes inter alia line deflection coils 18 and an EHT transformer 19 a secondary winding 20 of which serves for generating the EHT required for the acceleration anode of the picture display tube. Line deflection circuit 17 is fed by the output voltage V _o of the chopper which voltage is stabilized due to the pulse duration modulation with all previously mentioned advantages. Line deflection circuit 17 corresponds, for example, to similar arrangements which have been described in U.S. Pat. No. 3,504,224 issued Mar. 31, 1970 to J.J. Reichgelt et al., U.S. patent application Ser. No. 737,009 filed June 14, 1968 by W. H. Hetterscheid and U.S. application Ser. No. 26,497 filed April 8, 1970 by W. Hetterscheid et al. It will be evident that differently formed lined deflection circuits are alternatively possible.

It will now be shown that secondary winding 14 can indeed drive a line deflection circuit so that switching transistor 2 can function as a driver for the line deflection. FIGS. 2a and b show the variation as a function of time of the current i _C which flows in the collector of transistor 16 and of the drive voltage v ₁₄ across the terminals of secondary winding 14. During the flyback period (0, t ₁ ) transistor 16 must be fully cut off because a high voltage peak is then produced at its collector; voltage v ₁₄ must then be absolutely negative. During the scan period (t ₁ , t ₄ ) a sawtooth current i _C flows through the collector electrode of transistor 16 which current is first negative and then changes its direction. As the circuit arrangement is not free from loss, the instant t ₃ when current i _C becomes zero lies, as is known, before the middle of the scan period. At the end t ₄ of the scan period transistor 16 must be switched off again. However, since transistor 16 is saturated during the scan period and since this transistor must be suitable for high voltages and great powers so that its collector layer is thick, this transistor has a very great excess of charge carriers in both its base and collector layers. The removal of these charge carriers takes a period t _s which is not negligible whereafter the transistor is indeed switched off. Thus the fraction δ T of the line period T at which v ₁₄ is positive must end at the latest at the instant (t ₄ - t _s ) located after the commencement (t = 0) of the previous flyback.

The time δ T may be initiated at any instant t ₂ which is located between the end t ₁ of the flyback period and the instant t ₃ when collector current i _C reverses its direction. It is true that emitter current flows through transistor 16 at the instant t ₂ , but collector current i _C is not influenced thereby, at least not when the supply voltage (= V _o ) for line deflection circuit 17 is high enough. All this has been described in the U.S. Pat. No. 3,504,224. The same applies to line deflection circuits wherein the collector base diode does not function as an efficiency diode as is the case in the described circuit 17, but wherein an efficiency diode is arranged between collector and emitter of the line output transistor. In such a case the negative part of the current i _C of FIG. 2a represents the current flowing through the said efficiency diode.

After the instant t ₃ voltage v ₁₄ must be positive. In other words, the minimum duration of the period T when voltage v ₁₄ must be positive is (t ₄ - t _s ) - t ₃ whilst the maximum duration thereof is (t ₄ - t _s ) - t ₁ . In a television system employing 625 lines per raster the line period t ₄ is approximately 64 μus and the flyback period is approximately 12 μus. Without losses in the circuit arrangement instant t ₃ would be located approximately 26 μus after the instant t ₁ , and with losses a reasonable value is 22 μus which is 34 μus after the commencement of the period. If for safety's sake it is assumed that t _s lasts approximately 10 μus, the extreme values of δ T are approximately 20 and 42 μus and consequently the values for δ are approximately 0.31 and 0.66 at a mean value which is equal to approximately 0.49. It was previously stated that a mean value of δ = 0.5 was suitable. Line deflection circuit 17 can therefore indeed be used in combination with the chopper in the manner described, and the relative variation of δ may be (0.66 - 0.31) : 0.49 = 71.5 percent. This is more than necessary to obviate the variations in the mains voltage or in the various loads and to establish the East-West modulation and ripple compensation to be described hereinafter. In fact, if it is assumed that the mains voltage varies between -15 and +10 percent of the nominal value of 220 V, while the 50 Hz ripple voltage which is superimposed on the input voltage V _i has a peak-to-peak value of 40 V and V _i is nominally 270 V, then the lowest occurring V _i is:

0.85 × 270 V - 20 V = 210 V and the highest occurring V _i is

1.1 × 270 V + 20 V = 320 V. For an output voltage V _o of 135 V the ratio must thus vary between

δ = 135 : 210 = 0.64 and δ = 135 : 320 = 0.42.

A considerable problem presenting itself is that of the simultaneous or non-simultaneous drive of line output transistor 16 with switching transistor 2, it being understood that in case of simultaneous drive both transistors are simultaneously bottomed, that is during the period δ T. This depends on the winding sense of secondary winding 14 relative to that of primary winding 8. In FIG. 1 it has been assumed that the drive takes place simultaneously so that the voltage present across winding 14 has the shape shown in FIG. 2b. This voltage assumes the value n(V _i - V _o ) in the period δ T and the value -nVo in the period (1 - δ )T, wherein n is the ratio of the number of turns on windings 14 and 8 and wherein V _o is maintained constant at nominal mains voltage V _o = δ V _inom . However, if as a result of an increase or a decrease of the mains voltage V _i increases or decreases proportionally therewith, i.e., V _i = V _i nom + Δ V, the positive portion of V ₁₄ becomes equal to n(V _i nom - V _o +Δ V) = n [(1 -δ)V _i nom +ΔV] = n(0.5 V _inom +ΔV) if δ = 0.5 for V _i = V _i nom. Relatively, this is a variation which is twice as great. For example, if V _i nom = 270 V and V _o = 135 V, a variation in the mains voltage of from -15 to +10 percent causes a variation of V _i of from -40.5 V to +27 V which ranges from -30 to +20 percent of 135 V which is present across winding 8 during the period δ T. The result is that transistor 16 can always be bottomed over a large range of variation. If the signal of FIG. 2b would be applied through a resistor to the base of transistor 16, the base current thereof would have to undergo the same variation while the transistor would already be saturated in case of too low a voltage. In this case it is assumed that transformer 9 is ideal (without loss) and that coil 21 has a small inductance as is explained in the U.S. patent application Ser. No. 737,009 above mentioned. It is therefore found to be desirable to limit the base current of transistor 16.

This may be effected by providing a coil 22 having a large value inductance, approximately 100 μH, between winding 14 and the small coil 21. The variation of said base current i _b is shown in FIG. 2c but not to the same scale as the collector current of FIG. 2a. During the conducting interval δ T current i _b varies as a linear function of time having a final value of wherein L represents the inductance of coil 22. This not only provides the advantage that this final value is not immediately reached, but it can be shown that variation of this final value as a function of the mains voltage has been reduced, for there applies at nominal mains voltage that: If the mains voltage V _i = V _i nom +Δ V, then ##SPC1## because V _i nom = 2 V _o . Thus this variation is equal to that of the mains voltage and is not twice as great.

During switching off, t ₂ , of transistor 16 coil 22 must exert no influence and coil 21 must exert influence which is achieved by arranging a diode 23 parallel to coil 22. Furthermore the control circuit of transistor 16 in this example comprises the two diodes 24 and 25 as described in U.S. application Ser. No. 26,497 above referred to, wherein one of these diodes, diode 25 in FIG. 1, must be shunted by a resistor.

The control circuit of transistor 16 may alternatively be formed as is shown in FIG. 4. In fact, it is known that coil 21 may be replaced by the parallel arrangement of a diode 21' and a resistor 21" by which the inverse current can be limited. To separate the path of the inverse current from that of the forward current the parallel arrangement of a the diode 29' and a resistor 29" must then be present. This leads to the circuit arrangement shown in the upper part of FIG. 4. This circuit arrangement may now be simplified if it is noted that diodes 25 and 21' on the one hand and diodes 23 and 29' on the other hand are series-arranged. The result is shown in the lower part of FIG. 4 which, as compared with the circuit arrangement of FIG. 1, employs one coil less and an additional resistor.

FIG. 3 shows possible modifications of the chopper. FIG. 3a shown in a simplified form the circuit arrangement according to FIG. 1 wherein the pulsatory voltage present across the connections of windings 8 has a peak-to-peak amplitude of V _i - V _o = 0.5 V _i for δ = 0.5, As has been stated, the provision of coil 22 gives a relative variation for the base current of transistor 16 which is equal to that of the mains voltage. In the cases according to FIG. 3b, 3c and 3d the peak-to-peak amplitude of the voltage across winding 8 is equal to V _i so that the provision of coil 22 results in a relative variation which is equal to half that of the mains voltage which is still more favorable than in the first case.

Transistors of the npn type are used in FIG. 3. If transistors of the pnp type are used, the relevant efficiency diodes must of course be reversed.

In this connection it is to be noted that it is possible to obtain an output voltage V _o with the aid of the modifications according to FIGS. 3b, c and d, which voltage is higher than input voltage V _i . These modifications may be used in countries such as, for example, the United of America or France where the nominal mains voltage is 117 or 110 V without having to modify the rest of the circuit arrangement.

The above-mentioned remark regarding the sum of the diode currents only applies, however, for the modifications shown in FIGS. 3a and d.

If line output transistor 16 is not simultaneously driven with switching transistor 2, efficiency diodes 7 conducts simultaneously with transistor 16 i.e., during the period which is denoted by δ T in FIGS. 1 and 2b. During that period the output voltage V _o of the chopper is stabilized so that the base current of transistor 16 is stabilized without further difficulty. However, a considerable drawback occurs. In FIG. 1 the reference numeral 26 denotes a safety circuit the purpose of which is to safeguard switching transistor 2 when the current supplied to load 11 and/or line deflection circuit 17 becomes to high, which happens because the chopper stops. After a given period output voltage V _o is built up again, but gradually which means that the ratio δ is initially small in the order of 0.1.
 All this is described in U.S. patent No. 3,629,686. The same phenomenon occurs when the display device is switched on. Since δ = 0.1 corresponds to approximately 6 μs when T = 64 μs, efficiency diode 7 conducts in that case for 64 - 6 = 58 μus so that transistor 16 is already switched on at the end of the scan or at a slightly greater ratio δ during the flyback.
This would cause an inadmissibly high dissipation. For this reason the simultaneous drive is therefore to be preferred.

The line deflection circuit itself is also safeguarded: in fact, if something goes wrong in the supply, the driver voltage of the line deflection circuit drops out because the switching voltage across the terminals of primary winding 8 is no longer present so that the deflection stops. This particularly happens when switching transistor 2 starts to constitute a short-circuit between emitter and collector with the result that the supply voltage V _o for the line deflection circuit in the case of FIG. 1 becomes higher, namely equal to V _i . However, the line output transformer is now cut off and is therefore also safe as well as the picture display tube and other parts of the display device which are fed by terminal 15 or the like. However, this only applies to the circuit arrangement according to FIG. 1 or 3a.

Pulse oscillator 6 applies pulses of line frequency to modulator 5. It may be advantageous to have two line frequency generators as already described, to wit pulse oscillator 6 and line oscillator 6' which is present in the picture display device and which is directly synchronized in known manner by line synchronizing pulses 7'. In fact, in this case line oscillator 6' applies a signal of great amplitude and free from interference to pulse oscillator 6. However, it is alternatively possible to combine pulse oscillator 6 and line oscillator 6' in one single oscillator 6" (see FIG. 1) which results in an economy of components. It will be evident that line oscillator 6' and oscillator 6" may alternatively be synchronized indirectly, for example, by means of a phase discriminator. It is to be noted neither pulse oscillator 6, line oscillator 6' and oscillator 6" nor modulator 5 can be fed by the supply described since output voltage V _o is still not present when the mains voltage is switched on. Said circuit arrangements must therefore be fed directly from the input terminals. If as described above these circuit arrangements are to be separated from the mains, a small separation transformer can be used whose primary winding is connected between the mains voltage terminals and whose secondary winding is connected to ground at one end and controls a rectifier at the other end.

Capacitor 27 is arranged parallel to efficiency diode 7 so as to reduce the dissipation in switching transistor 2. In fact, if transistor 2 is switched off by the pulsatory control voltage, its collector current decreases and its collector-emitter voltage increases simultaneously so that the dissipated power is not negligible before the collector current has becomes zero. If efficiency diode 7 is shunted by capacitor 27 the increase of the collector-emitter voltage is delayed i.e., this voltage does not assume high values until the collector current has already been reduced. It is true that in that case the dissipation in transistor 2 slightly increases when it is switched on by the pulsatory control voltage but on the other hand since the current flowing through diode 7 has decreased due to the presence of the secondary windings, its inverse current is also reduced when transistor 2 is switched on and hence its dissipation has become smaller. In addition it is advantageous to delay these switching-on and switching-off periods to a slight extent because the switching pulses then contain fewer Fourier components of high frequency which may cause interferences in the picture display device and which may give rise to visible interferences on the screen of the display tube. These interferences occupy a fixed position on the displayed image because the switching frequency is the line frequency which is less disturbing to the viewer. In a practical circuit wherein the line frequency is 15,625 Hz and wherein switching transistor 2 is an experimental type suitable for a maximum of 350 V collector-emitter voltage or 1 A collector current and wherein efficiency diode 7 is of the Philips type BA 148 the capacitance of capacitor 27 is approximately 680 pF whilst the load is 70 W on the primary and 20 W on the secondary side of transformer 9. The collector dissipation upon switching off is 0.3 W (2.5 times smaller than without capacitor 27) and 0.7 W upon switching on.

As is known the so-called pincushion distortion is produced in the picture display tubes having a substantially flat screen and large deflection angles which are currently used. This distortion is especially a problem in color television wherein a raster correction cannot be brought about by magnetic means. The correction of the so-called East-West pincushion distortion i.e., in the horizontal direction on the screen of the picture display tube can be established in an elegant manner with the aid of the circuit arrangement according to the invention. In fact, if the voltage generated by comparison circuit 12 and being applied to modulator 5 for duration-modulating pulsatory voltage 3 is modulated by a parabola voltage 28 of field frequency, pulsatory voltage 3 is also modulated thereby. If the power consumption of the line deflection circuit forms part of the load on the output voltage of the chopper, the signal applied to the line deflection coils is likewise modulated in the same manner. Conditions therefore are that the parabola voltage 28 of field frequency has a polarity such that the envelope of the sawtooth current of line frequency flowing through the line deflection coils has a maximum in the middle of the scan of the field period and that charge capacitor 10 has not too small an impedance for the field frequency. On the other hand the other supply voltages which are generated by the circuit arrangement according to the invention and which might be hampered by this component of field frequency must be smoothed satisfactorily.

A practical embodiment of the described example with the reference numerals given provides an output for the supply of approximately 85 percent at a total load of 90 W, the internal resistance for direct current loads being 1.5 ohms and for pulsatory currents being approximately 10 ohms. In case of a variation of ± 10 percent of the mains voltage, output voltage V _o is stable within 0.4 V. Under the nominal circumstances the collector dissipation of switching transistor 2 is approximately 2.5 W.

Since the internal resistance of the supply is so small, it can be used advantageously, for example, at terminal 15 for supplying a class-B audio amplifier which forms part of the display device. Such an amplifier has the known advantages that its dissipation is directly proportional to the amplitude of the sound to be reproduced and that its output is higher than that of a class-A amplifier. On the other hand a class-A amplifier consumes a substantially constant power so that the internal resistance of the supply voltage source is of little importance. However, if this source is highly resistive, the supply voltage is modulated in the case of a class-B amplifier by the audio information when the sound intensity is great which may detrimentally influence other parts of the display device. This drawback is prevented by means of the supply according to the invention.

The 50 Hz ripple voltage which is superimposed on the rectified input voltage V _i is compensated by comparison circuit 12 and modulator 5 since this ripple voltage may be considered to be a variation of input voltage V _i . A further compensation is obtained by applying a portion of this ripple voltage with suitable polarity to comparison circuit 12. It is then sufficient to have a lower value for the smoothing capacitor which forms part of rectifier circuit 1 (see FIG. 3). The parabola voltage 28 of field frequency originating from the field time base is applied to the same circuit 12 so as to correct the East-West pincushion distortion.

The RGB Amplifier in the chassis PHILIPS K12 is operating on the G1 instead on katode in conventional schemes.
Cathode-ray tubes (CRTs) have been used extensively in various electronic products, such as television receivers, oscilloscopes, and other display units because of their high brightness, quick response, and high resolution, among many other inherent advantages. In raster scan display devices, such as television receivers, the control grid (G₁) is driven by the output signal from a video amplifier. The output signal will control the cathode (K)-to-grid G₁ voltage thereby controlling the beam current over a certain range, the beam current modulating the phosphor luminescence while the electron beam is scanning across the phosphor screen.
The cutoff bias voltage of a CRT is affected by the heater temperature of the cathode, the cathode-to-grid G₁ spacing, cathode aging, and the second grid (G₂) voltage. Any change in the cutoff bias voltage will result in a different brightness of the displayed image on the CRT, and it will affect the low light color balance of the display if a color CRT is used.
Some circuits regulate the control grid-to-cathode voltage within close tolerances. However, warm-up and aging characteristics of the cathode still tend to modify the grid-to-cathode cutoff voltage, thereby affecting the beam current and, accordingly, the brightness of the displayed image.
In some prior art video amplifiers, the cathode current is monitored. A feedback loop connects the monitored cathode current to the input of an amplifier. The amplifier is connected to the control grid G₁ of the CRT. When the cathode current varies from a desired level, the input to the amplifier is changed in such a way as to modify the control grid voltage in order that the brightness of the displayed image, or the color balance of the display will be maintained at its desired level. However, with such a circuit configuration, the amplifier has a limited bandwidth because it is necessary to maintain the dynamic range of the output signal at a certain level. This  amplifier has a wider bandwidth than the similar amplifiers of the prior art. It therefore generates a signal having a lower dynamic range than the dynamic ranges of the output signals associated with the similar amplifiers of the prior art. To compensate for the lower dynamic range of the output signal generated by the second amplifier, the drive circuit of the present invention measures the current in the cathode of the cathode-ray tube, converts this measured current into a voltage, stores the voltage in a capacitance, and amplifies the stored voltage in a fourth amplifier.

These are realized with HYBRID IC'S technology and they're in ceramic substrate.


PHILIPS CHASSIS K12 (20AX) CRT TUBE RGB AMPLIFIER DETAIL RGB HYBRID AMPLIFIER 4822 212 20307 3122 128 58455

PHILIPS  26C854/60R  EL GRECO  CHASSIS K12 (20AX) Hybrid RGB Amplifiers on CRT Socket.
Driving directly: G1 blue G1 red G1 green.Hybrid IC Technology on Ceramic substrate.



On a ceramic substrate, spiral-type inductors of a single layer wiring of a metal thin film are provided and respectively connected to a wiring pattern formed on another face of the substrate via through holes. A semiconductor chip is flip-chip mounted on the substrate in a face-down manner. On the face of the semiconductor chip, capacitors composed of a highly dielectric material, resistors formed by an ion implantation method or a thin-film forming method, and FETs are provided, respectively. Interconnection between the substrate and an external circuit board is achieved employing terminals formed at end faces of the substrate. The terminals have a concave shape with respect to the end face of the substrate. Thus, there is no need to use a package, and miniaturization and reduction in cost of a high-performance hybrid IC is achieved.

1. A hybrid IC comprising:
a substrate including a front face, a back face opposite the front face, and side faces interposed between the front face and the back face which define an outer perimeter of the substrate;
at least one inductor formed on at least one of the front face and the back face of the substrate;
a semiconductor chip mounted on the front face of the substrate by flip-chip bonding;
at least one terminal formed in a predetermined portion of the side faces of the substrate,
wherein the semiconductor chip comprises a plurality of circuit elements provided therein, at least one of the plurality of circuit elements being an MIM capacitor having a metal-insulation film-metal (MIM) structure, the insulation film being composed of a highly dielectric material.


2. A hybrid IC according to claim 1 further comprising at least one matching circuit for matching an input signal to the circuit elements provided inside the semiconductor chip, the matching circuit comprising at least one inductor.

3. A hybrid IC according to claim 2, wherein a wiring pattern is formed of a single metal layer on both the front and back faces of the substrate, the wiring patterns on the respective front and back faces of the substrate being interconnected with each other via through holes, and the at least one inductor comprised in the matching circuit is formed in the wiring pattern on one of the respective front and back faces of the substrate.

4. A hybrid IC according to claim 2, wherein the matching circuit is constituted only by inductors and comprises at least one serial inductor and at least one parallel inductor.

5. A hybrid IC according to claim 4, wherein the parallel inductor comprised in the matching circuit is a spiral-type inductor, outermost wiring of the spiral-type inductor being grounded.

6. A hybrid IC according to claim 2, wherein the inductors comprised in the matching circuit are a spiral-type inductor or a meander-type inductor.

7. A hybrid IC according to claim 2, wherein the matching circuit comprises an inductor and a capacitor, the capacitor being formed inside the semiconductor chip.

8. A hybrid IC according to claim 7, wherein the inductor comprised in the matching circuit is a spiral-type inductor or a meander-type inductor.

9. A hybrid IC according to claim 1, wherein the at least one terminal includes at least an RF terminal functioning as an input terminal for an RF signal, an LO terminal functioning as an input terminal for an LO signal, an IF terminal functioning as an output terminal for an IF signal, a ground terminal, and a supply terminal.


---------------------------------------

HERE BELOW A DETAILED DESCRIPTION OF THE PHILIPS CHASSIS K12 CRT DRIVING TECHNOLOGY:

PHILIPS  26C854/60R  EL GRECO  CHASSIS K12 (20AX) PHILIPS CHASSIS K12 Automatic gray scale control circuit for a color television receiver

The present invention relates to a novel automatic gray scale control circuit for a color television receiver. The circuit senses the cut-off voltage of each gun during the blanking interval, and uses a voltage equal to the cut-off voltage to energize the driver and bias the gun during the video field. The effect is to standardize the emission of each of the three guns against variation in gun cut-off voltage and to produce improved gray scale accuracy at the lowest emission levels. Since the gray scale adjustment is optimized at the lowest emission levels, where the eye is most intolerant to error in hue, one may avoid the need for manual adjustment of the cut-off point, and in cases where the gain does not vary widely from gun to gun, avoid the need for separate gain adjustment. Thus, the circuit may be used either to simplify or eliminate the color set up process at the factory when the receiver is manufactured. It may also reduce or avoid the need for readjustment after periods of use.CHASSIS K12 CRT Beam current control apparatus:

Introducing beam current control:

In a television picture display device wherein a cathode of a picture display tube is driven by an emitter-follower and a control signal for a beam current reference level control circuit is obtained from the collector circuit of this emitter-follower, measures are taken to compensate for leakage currents from and to the cathode. To this end a blacker-than-black current compensation circuit is provided while furthermore it is ensured that the blacker-than-black curent can be processed by the beam current reference level control circuit.




1. A television picture display device for displaying pictures derived from video signals, said display device comprising a picture display tube having a cathode; an emitter-follower device coupled to said cathode for producing a beam current for driving said picture display tube; a beam current reference level control circuit coupled to said emitter-follower device for controlling the black level of the beam current; a blacker-than-black current compensation circuit incorporated in said beam current reference level control circuit for compensating for a leakage current in said cathode caused by a blacker-than-black level in the video signal; and a blacker-than-black current conduction circuit coupled to said cathode to enable compensation of said leakage current when said beam current is blanked.

2. A television picture display device as claimed in claim 1, wherein the blacker-than-black current conduction circuit comprises a direct current source coupled to the cathode of the picture display tube for keeping said emitter-follower device conductive at the occurrence of a blacker-than-black current in the cathode circuit of the picture display tube thereby allowing for the compensation of said blacker-than-black signal.



3. A television picture display device as claimed in claim 2, wherein the direct current source comprises a first and second resistor serially connected to a supply voltage said first resistor being a.c. coupled to the base of the emitter-follower device.

4. A television picture display device as claimed in claim 1, which further comprises a second emitter-follower device, of an opposite conduction type as said first-mentioned emitter-follower device, also coupled to said picture display tube cathode, and wherein said blacker-than-black current conduction circuit comprises a difference-forming circuit, coupled to the collector of the second emitter-follower device and coupled to an input of said beam current reference level control circuit.

5. A television picture display device as claimed in claim 4, wherein said difference-forming circuit comprises a current mirror circuit having an input and an output, the input of which is coupled to the collector of the second emitter-follower device and the output to the collector of the first-mentioned emitter-follower device.

6. A television picture display device as claimed in claims 2, 3, 4, 5 or 1, which further comprises a heater, a wehnelt electrode connection of the picture display tube and a leakage current conducting circuit, said heater and said wehnelt electrode connection being coupled to the collector of the first-mentioned emitter-follower device through a said leakage current conducting circuit.

Description:

BACKGROUND OF THE INVENTION

The invention relates to a television picture display device having a picture display tube, a cathode of which is driveable by an emitter-follower, the collector of this emitter-follower being coupled to an input of a beam current reference level control circuit.
Dutch Patent Application No. 7604463 discloses a television picture display device of the above-defined type. In this device a control of the black level, serving as the reference level, of the beam current, to a constant value takes inter alia place by means of a voltage produced across the collector resistor of the emitter-follower. Although in principle this control should furnish a very constant black level this appears not to be the case.
SUMMARY OF THE INVENTION
It is an object of the invention to improve the constancy of the controlled reference level of the beam current.
A television picture display device of the above-defined type according to the invention is therefore characterized in that the beam current reference level control circuit comprises a blacker-than-black current compensation circuit by which the influence of the blacker-than-black current on the beam current reference level control circuit is compensated for while a blacker-than-black current conductive circuit is coupled to the cathode of the picture display tube so that the beam current reference level control circuit can also process a blacker-than-black current of the picture display tube occurring in case of a blanked beam current.

It should be noted that the use of a blacker-than-black current compensation circuit in a beam current reference level control circuit is known per se from the Dutch Patent Application No. 6903362. In that case, however, the picture display tube is controlled via the wehnelt electrode. When picture display devices to which the invention relates are controlled via the cathode, a blacker-than-black current compensation appears to be impossible without further measures. Applicants found, namely, that the cathode current of the picture display tube, in the case of a blanked beam current, may have a direction which may cut off the emitter-follower so that no measuring data about the collector resistance of the emitter-follower become available, and a blacker-than-black current compensation is not possible. By coupling a blacker-than-black current conduction circuit to the cathode of the picture display tube in such a manner that also the blacker-than-black current, occurring with a blanked beam current, can be processed by the beam current reference level control circuit, a blacker-than-black current compensation is possible. The blacker-than-black current conduction circuit may be a circuit supplying a constant direct current to the cathode of the picture display tube which ensures that the emitter-follower cannot be cut off if the picture display tube, in the case of a suppressed beam current, carries a cathode current which might cut off the emitter-follower, or a circuit which can take over the blacker-than-black current from the emitter-follower and pass it on to the beam current reference level control circuit.
DESCRIPTION OF THE DRAWINGS
The invention will now be further explained with reference to the drawing.
In the drawing
FIG. 1 shows a circuit of a picture display device according to the invention in which the emitter-follower can be kept conductive by means of a direct current and

FIG. 2 shows a further circuit of a picture display device according to the invention in which the emitter-follower current can be taken over by another circuit and passed to a control circuit.
DESCRIPTION OF THE PREFERRED EMBODIMENT
In FIG. 1 a video signal is applied to an input of an adder circuit 3. A control signal originating from an output 7 of a beam current reference level control circuit 9 appears at a further input 5 of the adder circuit 3. This control signal is added to the video signal and the sum signal is applied to the base of an emitter-follower 13 via an amplifier 11.
The emitter of the emitter-follower 13 drives a cathode of a picture display tube 14. The collector circuit of the emitter-follower 13 comprises a resistor 15, an end of which is connected to an input 17 of the beam current reference level control circuit 9. The beam current supplied by the emitter-follower 13 flows through this resistor 15.
The input 17 is, at the same time, the input of an amplifier 19, an output 21 of which is connected to an input 25 of an amplifier 27 via a switch 23. A capacitor 29 is also connected to the input 25. The switch 23 closes periodically under the influence of a signal derived from an output 31 of a switching signal generator 33 during the occurrence of the reference level, for example the black level, in the video signal.
Consequently a voltage, which is a measure of the cathode current of the picture display tube 14 during the occurrence of the reference level, is produced across the capacitor 29. This cathode current is composed of a beam current and a leakage current. To compensate for the influence of this leakage current, a switch 35, which is also connected to the output 21 of the amplifier 19, is periodically closed during the occurrence of a blacker-than-black level in the video signal, under the influence of a signal originating from an output 37 of the switching signal generator 33.
The switching signal generator 33 is synchronized with the video signal by a synchronisation signal, applied to an input 39 thereof, obtained from the video signal.
A capacitor 41, connected to the output of the switch 35, now has a voltage there across which is a measure of the leakage current of the cathode of the picture display tube 14. This voltage is supplied to an input 43 of the amplifier 27. The amplifier 27 amplifies the difference of the voltages across the capacitors 29 and 41 so that a control signal is produced at the output 7 thereof which is a measure of the beam current of the picture display tube during the occurrence of the reference level in the video signal. This control signal counteracts changes in the beam current reference level which corresponds to the video signal reference level.

The leakage current of the picture display tube 14 may sometimes be directed towards the cathode of that tube. In that case the emitter-follower 13 would be cut off and the leakage current could not be measured in the above described manner. Therefore a direct current is supplied to the emitter-follower 13 via two resistors 45, 47, which are connected to a positive supply voltage which may be in the order of approximately 40 μA. This direct current keeps the emitter-follower 13 in the conducting state thereof.
To enable also a compensation of this direct current in the the blacker-than-black compensation circuit, constituted by the switch 35, the capacitor 41 and the difference formation in the amplifier 27, the value of this direct current must not be dependent on the video signal. Therefore this video signal is applied, via a capacitor 49, to the junction of the resistors 45, 47 so that the voltage across the resistor 45 becomes independent of the video signal.

A second emitter-follower 51, which is also driven by a video signal, is also connected to the cathode of the picture display tube 14. The function of this second emitter-follower 51 is to enable a sufficiently rapid change of the charge of the cathode capacitance in the case of positive-going voltage transients in the video signal. This second emitter-follower plays no part in the measurements of the beam current reference value because the measurements are performed in periods in which the level in the video signal is constant for some time.
The variation in the leakage current in the cathode circuit of the picture display tube 14 can be many times larger than that in the beam current, which condition occurs if the reference level in the video signal is present. The variation in the voltage difference across the capacitors 28 and 41 would then be many times smaller than the variation in the voltage across one of the capacitors 28, 41. This might adversely affect the accuracy of the control system. To prevent this, two resistors 53 and 55 are provided which pass the leakage current from the wehnelt circuit and the heater circuit to the resistor 15 and, consequently, compensate the cathode leakage current to a large extent.
It will be obvious that the direct current supplied to the cathode of the picture display tube may be obtained, if so desired, by means of a transistor connected as a current source. This transistor should then be suitable for a rather high voltage because the voltage at the cathode of the picture display tube may change very much.
In FIG. 2 elements corresponding to elements of the circuit of FIG. 1 have been given the same reference numerals as in FIG. 1.
The blacker-than-black current compensation circuit of FIG. 2 is arranged somewhat differently then in FIG. 1. The capacitor 41 is now arranged in series with the output 21 of the amplifier 19 and the two switches 23 and 35, and the input 43 of the amplifier 27 is connected to ground. The switch 35 now operates as a clamping switch which ensures that the blacker-than-black level is connected to ground and the voltage across the capacitor 29 becomes a measure of the beam current occurring at the reference level in the video signal.
A further difference relative to the circuit of FIG. 1 is that the collector circuit of the second emitter-follower 51 includes a circuit which acts as blacker-than-black current conduction circuit and supplies any current flowing to the cathode to the input 17 of the beam current reference level control circuit 9 so that also these currents can be measured. The direct current supply circuit (45, 47) at the cathode of the picture display tube can then be dispensed with.
The collector current of the second emitter-follower 51 is supplied to the resistor 15 via two current mirror circuits. A first current mirror circuit is constituted by a series arrangement of a resistor 57 and a diode 59 in parallel with the series arrangement of the base-emitter path of a transistor 61 and a resistor 63 to a positive supply voltage. The collector current of the transistor 61 is supplied to a second current mirror circuit having a transistor 65 and a diode 67, which is in parallel with the base-emitter path of the transistor 65. The collector of the transistor 61 is connected to the resistor 15 through which the difference in the collector currents of the two emitter-followers 13, 51 now flows. Independent of the direction of the cathode currents of the picture display tube 14, a voltage, which is a measure of that cathode current, is now produced across the resistor 15.
Instead of determining the difference current by means of current mirror circuits in the described manner, it is alternatively possible, if so desired, to use other difference-determining circuits.
If so desired also the measuring data for a beam current limiting control of the resistor 15 can be obtained in the described circuits.
The amplifier 19 may comprise a circuit which limits the amplitude of the video signal outside the instants in which measuring takes place. This may be a circuit operated by an auxiliary signal or a self-switching circuit, for example a diode limiter circuit.
If the picture display tube is a color display tube having several electron guns, the emitter-followers for each of the guns may have the resistor 15 in common and a sequential measurement may take place at a reference level sequentially occurring in the different video signals, so that only a portion of the control circuits is not common.
It will be obvious that the choice of the measuring instants and the associated occurrence of the reference levels and blacker-than-black levels are not important for the essence of the invention and may be chosen in a suitable manner.
The blacker-than-black current compensation circuits 9 of the above-described embodiments are interchangeable.
The amplifier 27 may comprise a level reference circuit so that the value of the beam current is determined which is associated with the corresponding reference level in the video signal.

PHILIPS CHASSIS K12 : System for stabilizing cathode ray tube operation:
-------------------------------------------------------

The invention pertains to a video display apparatus which includes a cathode ray tube having at least one electron gun and means for deriving a source of excitatory voltage signal representative of picture information. In accordance with the invention there is provided a system for stabilizing the display intensity attributable to the electron gun comprising means coupling the source of voltage signal to the gun for generating a stabilized current in the gun, the level of current in the gun being substantially independent of variations in the operating characteristics of the gun. In the preferred embodiment of the invention a test signal is periodically applied at a predetermined level to the gun. A sampler samples the beam current of the gun during the test signal and generates a correction signal in accordance with the sample value. The correction signal is applied to the electron gun so as to stabilize its reference current level.



1. In a television display apparatus which includes a cathode ray tube having at least one electron gun and means for applying a television video signal to said electron gun; a system for stabilizing the display intensity attributable to said electron gun, comprising:
means for generating first and second test signals during the vertical blanking intervals of said television video signal;
means for applying said first and second test signals to said electron gun during first and second portions, respectively, of said vertical blanking intervals;
means for sampling the beam current of said electron gun during said first and second portions and for generating first and second correction signals, respectively, in accordance with the sampled values; and
means for applying said first and second correction signals to said electron gun so as to stabilize the reference current level of said electron gun.


2. The system as defined by claim 1 wherein said first and second correction signals are applied to grids of said electron gun.

3. The system as defined by claim 2 wherein means are provided for maintaining the correction signals during the intervals between successive test signals.

4. The system as defined by claim 1 wherein means are provided for maintaining the correction signals during the intervals between the successive test signals.

5. The system as defined by claim 1 wherein said first and second test signals are at white and black level viedo, respectively.

6. The system as defined by claim 5 wherein said first correction signal is applied as a multiplying factor to said television video signal.

7. The system as defined by claim 6 wherein the corrected television video signal is applied to the control grid of said electron gun.

8. The system as defined by claim 6 wherein the corrected television video signal is applied to the cathode of said electron gun.

9. The system as defined by claim 6 wherein said second correction signal is applied as a DC reference level to the corrected television video signal.

10. The system as defined by claim 1 wherein said means for sampling the beam current comprises a resistor in series with the cathode of said gun.

11. In a video display apparatus which includes a cathode ray tube having a plurality of electron guns and means for applying a plurality of television video signals representative of color picture information to said electron guns; a system for balancing the color screen temperature of said cathode ray tube, comprising:
means for generating first and second test signals during the vertical blanking intervals of said television video signal;
means for applying said first and second test signals to said electron guns during first and second portions, respectively, of said vertical blanking intervals;
means for sampling the beam current of each electron gun during said first and second portions and for generating first and second correction signals, respectively, for each electron gun in accordance with the sampled values; and
means for applying said first and second correction signals to their respective electron guns so as to balance the screen color temperature of said cathode ray tube.


12. The system as defined by claim 11 wherein said first and second correction signals are applied to grids of said electron guns.

13. The system as defined by claim 12 wherein means are provided for maintaining the correction signals during the intervals between successive test signals.

14. The system as defined by claim 11 wherein means are provided for maintaining the correction signals during the intervals between successive test signals.

15. The system as defined by claim 11 wherein said first and second test signals are at white and black level video, respectively.

16. The system as defined by claim 15 wherein said first correction signal is applied as a multiplying factor to each of said plurality of television video signals.

17. The system as defined by claim 16 wherein the corrected plurality of television video signals are applied to the control grids of their respective electron guns.

18. The system as defined by claim 16 wherein the corrected plurality of television video signals are applied to the cathodes of their respective electron guns.

19. The system as defined by claim 16 wherein said second correction signals are applied as DC reference levels to the corrected television video signals.

20. The system as defined by claim 11 wherein said means for sampling the beam current comprises a plurality of resistors in series with the cathode of said electron guns.

Description:

BACKGROUND OF THE INVENTION
This invention relates to improvements in video display apparatus and, more particularly, to a system for stabilizing the display intensity or "color temperature" of a cathode ray tube. The subject matter of this invention is related to subject matter disclosed in copending U.S. application Ser. No. 572,128 of C. W. Smith and R. H. McMann, filed of even date herewith and assigned to the same assignee.
Conventional television display systems employing kinescope cathode ray tubes are subject to performance degradation resulting from instabilities in the operating characteristics of the kinescope or the circuits which drive or bias the kinescope. Prior techniques have been developed which serve to stabilize the signals driving a kinescope. For example, the drive voltages applied to the cathodes of a color kinescope can be stabilized using a feedback scheme; e.g., circuitry which periodically senses the drive voltage at input "black" and "white" levels of operation and corrects for deviations from standard reference voltages by gain adjustment. DC voltages applied to the kinescope can also be stabilized by using precise voltage regulation techniques.
There remains, however, the recognized problem of kinescope electron gun drift which manifests itself as a drift in screen color temperature in a three gun color kinescope. As the electron guns age, their generated beam current per unit of applied voltage (which can be considered a transconductance function) varies, the variations being generally non-uniform in the three different guns. This is a cause of noticeable and undesirable drifts in the display screen color.
The major sources of drift are: aging or long term variations caused by a gradual decrease in cathode activity, not necessarily constant or uniform for each cathode; and cathode operating temperature. The relatively long term variations in emission are caused by filament voltage changes and heat build-up in the gun area, generally a function of how many hours a display tube has been operating. Dynamic heating of each gun depends on the ratio of gun currents drawn to provide the colored picture being instantaneously presented. For example, a long persisting mostly red field causes red gun current almost exclusively, thereby causing an unbalanced heating of the red cathode, which changes its emission characteristics to a different degree than the other cathodes, this change remaining until relative cooling occurs.
Cathode thermal current, I _th , is represented by the Dushman equation: I _th = SA ₀ T ² e ^{- b s} /T amperes
where S and A ₀ are constants and b ₀ = Dushman constant ≉ 11,600° for an oxide coated cathode.
The derivative of the natural logarithm of this equation gives the change in emission with respect to temperature change: dI _th /I th = 2 + (b ₀ /t ) (dT/T)
the temperature of the CRT cathode is approximately 1,160° K, which yields (dI _th /I th) = 12dT/T
typical ambient temperature variations, such as in a display monitor, are about 40° C, so that the net change of gun current is of the order of 12 . 40/1,160 ≉ 40%
Therefore, a 1° C change in cathode temperature yields about a 1% change in gun current, if the gun is fixed bias and not near cut-off.
It is an object of the present invention to provide a stabilizing system which overcomes the problems set forth.
SUMMARY OF THE INVENTION
The invention pertains to a video display apparatus which includes a cathode ray tube having at least one electron gun and means for deriving a source of excitatory voltage signal representative of picture information. In accordance with the invention there is provided a system for stabilizing the display intensity attributable to the electron gun comprising means coupling the source of voltage signal to the gun for generating a stabilized current in the gun, the level of current in the gun being substantially independent of variations in the operating characteristics of the gun.
In the preferred embodiment of the invention a test signal is periodically applied at a predetermined level to the electron gun. A sampler samples the beam current of the gun during the test signal and generates a correction signal in accordance with the sample value. The correction signal is applied to the gun so as to stabilize its reference current level.
Further features and advantages of the invention will become more readily apparent from the following detailed description when taken in conjunction with the accompanying drawings.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a simplified diagram of a color television display kinescope;
FIGS. 2A, 2B and 2C are block diagrams of embodiments of the invention which utilize periodically applied test signals;
FIG. 3 is a schematic representation of an embodiment of the invention which employs a "constant current" technique;
FIG. 4 is a schematic representation of another embodiment of the invention employing a differential amplifier; and
FIG. 5 is a schematic representation of another embodiment of the invention employing direct cathode temperature sensing and heater control.
DESCRIPTION OF THE PREFERRED EMBODIMENTS

Referring to FIG. 1, there is shown a simplified diagram of a color television display cathode ray tube or kinescope 10 as driven by excitatory video voltage signals designated R, G and B, these signals having typical ranges of about 100 volts peak-to-peak. The kinescope 10 has three electron guns, each including a cathode and associated grids. For clarity of illustration, only one of the three guns, designated by reference numeral 11, is represented in some detail, but it will be appreciated that two other complete electron guns (indicated in the Figure by only the two dashed cathodes coupled to the G and B inputs) are normally provided and are substantially identical to the gun 11. Hereinafter, and in the description of the embodiments of the invention, the circuitry associated with only one electron gun in a given kinescope will be described for illustrative clarity, but it will be understood that if the kinescope has two or more guns, similar circuitry can be employed in conjunction with the remaining guns.
The electron gun 11 comprises a cathode 21 and first, second and third grids, 22, 23, and 24, which are sometimes designated as "grid 1," "grid 2," and "grid 3," or as the "control electrode," the "accelerating electrode," and the "focusing electrode," respectively. Generated electrons impinge on an anode 25 near display screen 26 which is coated with an electron-sensitive phosphor as is conventional in the art. Typical voltages applied to the cathode, grid 1, grid 2, grid 3 and the anode are about 235 volts, 150 volts, 700 volts, 5000 volts and 25,000 volts, respectively. In alternate modes of operation, the excitatory voltage input signal may be applied to a control grid with the remaining grid and cathode voltages being set at appropriate values.
Referring to FIG. 2A, there is shown an embodiment of the invention which comprises a system for stabilizing the display intensity or "color temperature" of the electron guns in a kinescope 30. An excitatory voltage signal at input terminal 31, which may be the R, B or G signal in a color system or the luminance signal in a black and white system, is coupled through an adder 41 and DC restorer circuit 42 to grid 1 of the kinescope 30. Cathode 43 is coupled through a resistor R ₁ to ground reference. A bias voltage is applied to grid 2 via a voltage amplifier 44 which receives a signal on a line 58A which determines the level of the bias voltage applied to grid 2. Suitable focus and anode voltages are applied to grid 3 and the anode from sources not shown.
The vertical and horizontal synchronizing signals of the composite television signal, available in the television receiver, are applied to a line counter 51 which is adapted to count horizontal scanlines of the television field and to be reset to zero at the end of each television field. The counter generates a first output on a line 51A during the scanlines 15-17 of each television field and a signal on line 51B during lines 18-20 of each television field, all of the lines 15-20 occuring during the vertical blanking period. The signal on line 51A enables a gate 55 and also enables a sample-and-hold circuit 56. The signal on line 51B enables a gate 57 and a sample-and-hold circuit 58. The gates 55 and 57 respectively receive voltages at reference "black level" and "white level." The outputs of gates 55 and 57 are coupled over lines 55A and 57A, respectively, to inputs of the adder 41.
Operation of the system of FIG. 2A is as follows: During lines 15-17 of the vertical blanking interval the gate 55 is enabled so that black level voltage is coupled through adder 41 and circuit 42 to grid 1. With this voltage applied to grid 1 the cathode current should ideally have a certain nominal value that does not vary with the tube life or cathode temperature but, as indicated above in the Background, this is not generally the case in actual practice. The actual cathode current is sampled across resistor R ₁ , and a voltage representative of this current is coupled to the sample-and-hold circuit 56 which is enabled to sample the voltage across resistor R ₁ during the lines 15-17. The circuit 56 holds the sampled voltage through the subsequent video field and couples the held voltage to circuit 42 via line 56A, this voltage serving to adjust the DC reference level of the output of circuit 42. In this manner, the voltage on line 56A controls the bias level at grid 1 so as to correct for any variations in the cathode current at nominal black level. Thus, for example, if at some point in operation the cathode current for a black level input voltage is lower than its nominal value, the voltage drop across sampling resistor R ₁ will also be low. This will decrease the output of sample-and-hold circuit 56 fed to circuit 42 which, in turn, will cause the bias level at grid 1 to decrease (typically, to a less negative value with respect to the cathode). A lesser negative bias level on the control grid 1 will, in turn, cause a proportionate increase in the electron current flowing from cathode 43; the desired result.
Similarly, during lines 18-20 of the vertical blanking period white level voltage is applied via adder 41 and amplifier 42 to grid 1, and during this time the cathode current is sampled by circuit 58 which is enabled to sample by the signal on line 51B. During the remainder of the television field, the bias voltage applied to grid 2, via voltage amplifier 33, is a function of the voltage which had been sampled by circuit 58. For example, in an instance where the cathode current sensed at a white level voltage input is lower than the nominal value, the resultant low voltage sampled by circuit 58 will cause the grid 2 accelerating voltage to decrease. This causes the sampled voltage at black level to appear too negative (when next sampled during the succeeding vertical blanking interval) which, in turn, results in a decrease in grid bias by the black level circuit causing the desired increase in beam current over prior conditions, as previously described.
The embodiment of FIG. 2B is similar to that of FIG. 2A except that the output of sample-and-hold circuit 58 (which is a measure of the sampled white level current) is coupled to an analog multiplier circuit 59, which is in series with DC restorer circuit 52. In this embodiment, corrections resulting from both the white level and black level measurements are achieved via grid 1, with operation otherwise being substantially as described above.


In the embodiment of FIG. 2C the electron gun is driven by application of the video signal to the cathode 43 via a complementary emitter-follower 120 which comprises NPN transistor 121 and PNP transistor 122. (The system to the left of blocks 42 and 56 is the same as in FIG. 2B). The transistor emitters are coupled to the cathode 43 of kinescope 30 and the transistor bases receive the video signal from DC restorer circuit 42. The collector of transistor 121 is coupled to a suitable bias voltage, e.g., 150 volts, and the collector of transistor 122 is coupled to ground reference potential through sampling resistor R ₁ .
In operation, during the lines 15-20 the test signals are applied via circuit 42 and cathode 43 is driven while the cathode current is sampled by resistor R ₁ , a typical value for which is 1K ohm. Transistor 122 is "on" during the white level test signal (output of circuit 42 about 25 volts) and the black level test signal (output of circuit 42 about 125 volts), and the gun current-representative voltages sampled across resistor R ₁ are coupled to the appropriate sample-and-hold circuits as previously described. During the active portion of the television field the analog multiplier 59 and DC restorer circuit 42 apply appropriate corrections, with transistor 122 normally "on." During rapid lighter-to-darker transitions of the video signal the transistor 121 turns momentarily "on" and the tube capacitance and stray capacitance (collectively represented by C in the Figure can be thought of as charging. Diode D ₁ prevents inordinate voltage drops across R ₁ during the active picture area when R ₁ is not used for sampling.
In the embodiment of FIG. 3 the video voltage signal at terminal 61 is coupled to cathode 71 of a kinescope 75 by the parallel combination of capacitor 62 and amplifier 63 in a series with resistor R ₂ . Amplifier 63 comprises transistors 64 and 65 and has a voltage gain of about 5 and an output capability of about 500 volts. The resistor R ₂ is selected to be substantially greater than the input impedance of the cathode 71 and preferably has a resistance at least five times higher than the cathode resistance. Since the effective cathode resistance is the inverse of the gun transconductance (about 8.6 micromhos), a suitable value for R ₂ is of the order of 600K ohms. Accordingly, the amplifier 63 in conjunction with resistor R ₂ operates as a so-called "constant current" source, which effectively transforms the voltage signal at terminal signal to a current source input to the cathode 71, this current source input being relatively insensitive to variations in the kinescope characteristics. Since normal wiring capacitance and electron gun interelectrode capacitance render high frequency response impractical in a high impedance amplifier drive, the higher frequency portions of the video signal are shunted across the amplifier by capacitor 62 which may have a typical value of about 0.05 microfarads. The higher frequency signals arrive at substantially the same relative level as the low frequencies, thereby preserving their relationship. This is because the lower frequency signals are amplified by a factor of 5 and then undergo a one-fifth loss by virtue of the voltage divider action of resistor R ₂ and the cathode impedance.
FIG. 4 shows a further embodiment wherein the video voltage signal at an input terminal 81 is applied to one input of a differential amplifier comprising transistors 82, 83 and 84. The output stage 84 drives the grid 1 electrode of kinescope 90 through series peaking inductor L ₁ and shunt peaking inductor L ₂ . The cathode 91 of kinescope 90 is coupled to ground reference potential through resistor R ₃ which is used to continuously monitor the cathode current, the line 89 coupling a voltage representative of the cathode current to the other input of the differential amplifier; viz., the base of transistor 83.
In operation, the voltage developed across resistor R ₃ is proportional to the cathode current. This voltage, for a stable transconductance, should be in a stable relationship with respect to input voltage at terminal 81, and R ₃ is selected empirically at a value, typically about 2K ohms, which generates a sample voltage nominally equal to the input voltage at terminal 81. When a deviation exists between the inputs to transistors 82 and 83, the output of the differential amplifier adjusts up or down to correct for the difference, thereby adjusting control of the drive to grid 1 and correcting for drifts in the kinescope transconductance.
A characteristic of the circuit of FIG. 4 is that it linearizes the electron gun transfer function which normally is non-linear, the non-linear function conventionally being known as "gamma" of the kinescope. Television video signals are conventionally precorrected for the gamma of the kinescope. In a color kinescope the gamma may be different for each gun, making it difficult to match the effective light output attributable to each gun over the grey scale; a problem known as "tracking" in the prior art. The present invention allows use of an inverse gamma circuit (which eliminates the precorrection in the conventional television signal) and the linearized gun transfer functions reduce tracking problems.

The invention has been described with reference to particular embodiments, but it will be understood that variations within the spirit and scope of the invention will occur to those skilled in the art. For example, the circuits of the "constant current" generator of FIG. 3 or the differential amplifier of FIG. 4 may be of other suitable forms. Also, in the embodiment of FIG. 2, sampling could be achieved during any suitable blanking or active period. The beam could be deflected off the tube face during sampling time to avoid displaying the trace during this time. Finally, stabilization of cathode temperatures could be achieved directly, such as by providing heater/thermistor stabilization circuits for each cathode. A suitable circuit is shown in FIG. 5 wherein a negative temperature coefficient thermistor 101 is attached to the cathode metal. V ₀ is a precision voltage source providing a voltage typically in the range 5-12 volts and R ₀ is selected as being substantially equal to the resistance of the thermistor at nominal cathode temperature. If the cathode becomes unduly hot, the resistance of thermistor 101 will decrease which, in turn, causes the voltage at terminal 103 to decrease. This results in a decreased output of operational amplifier 102, so that the cathode heater drive is reduced, as desired. Insufficient cathode temperature can be seen to cause the opposite effect.

PHILIPS CHASSIS K12 Television display apparatus including a beam current clamping control circuit:

A beam current clamping control circuit in which in case of a too large deviation of the desired value of the field frequency measured beam current an accelerated correction of this deviation is effected.



1. Television display apparatus including a beam current clamping control circuit having a beam current measuring circuit operable by a pulse generator and coupled to a television display tube for measuring, during at least part of a line scan time of a field blanking time a beam current reference level to be corrected, a level insertion circuit for inserting the reference level during the measuring time into a video signal to be applied to the television display tube and a level correction circuit coupled to an output of the measuring circuit and to a control electrode of the television display tube, the measuring circuit including a threshold circuit and a storage circuit, characterized in that the threshold circuit is a circuit which applies a signal to an output thereof when a too large beam current occurs, said output being coupled to an operation signal input of a circuit means for extending the charge correction time of the storage circuit per field period.

2. Television display apparatus as claimed in claim 1, characterized in that the circuit extending the charge correction time is a circuit means for increasing the measuring frequency, whereby a greater number of measurements per unit time is effected.

3. Television display apparatus as claimed in claim 1, characterized in that the circuit extending the charge correction time is a change-over switch a first input of which is coupled to an input of the measuring circuit, a second input is coupled to a beam current independent charging circuit coupled to a supply source and an output is coupled to the storage circuit.

4. Television display apparatus as claimed in claim 1, characterized in that the pulse generator includes a counting circuit having a feedback which can be switched on by the threshold circuit.

5. A beam current clamping control circuit for a television display tube comprising a beam current measuring circuit means for measuring during at least part of a line scan time of a field blanking time a beam current reference level to be corrected, a pulse generator coupled to said measuring circuit, a level insertion circuit means coupled to said pulse generator for inserting the reference level during the measuring time into a video signal to be applied to the television display tube, a level correction circuit coupled to an output of the measuring circuit, the measuring circuit including a threshold circuit means for coupling to said tube for applying a signal to an output thereof when a beam current above a selected value occurs, a storage circuit, and a circuit means for extending the charge correction time of the storage circuit per field period having an input coupled to said threshold circuit output and an output coupled to said storage circuit.

Description:

The invention relates to television display apparatus including a beam current clamping control circuit comprising a beam current measuring circuit coupled to a television display tube and operable by a pulse generator for measuring during at least part of a line scan time of a field blanking period a beam current reference level to be corrected, a level insertion circuit for inserting during the measuring time the reference level into a video signal to be applied to the television display tube, and a level correction circuit coupled to an output of the measuring circuit and to a control electrode of the television display tube, the measuring circuit including a threshold circuit and a storage circuit.

Television display apparatus of this type is known from U.S. Patent No. 3,562,409 in which the threshold circuit ensures that in case of a too large deviation of the beam current relative to the desired value the drive of the storage circuit is limited.

It is an object of the invention to provide improved television display apparatus of the kind described in the preamble which is characterized in that the threshold circuit is a circuit which in case of occurrence of a too large beam current applies a signal to an output thereof which output is coupled to an operation signal input of a circuit extending the charge corrections time of the storage circuit per field period.

Due to the limiting circuit in the circuit arrangement described in the abovementioned U.S. Patent Specification, in combination with the short measuring time available it could occur in the known circuit arrangement that, for example, after switching on the apparatus a very brightly lit picture occurred for several seconds. The storage circuit must include a capacitor having a fairly large capacitance the charge of which must be corrected in the known circuit with a current limited by the threshold circuit during the measuring time.

As a result of the step according to the invention the threshold circuit does not serve to limit the current intensity during the correction of the charge but to activate a circuit in case of a too large deviation of the nominal beam current, which circuit accelerates a feedback of the beam current to the nominal value by realising an extended charge correction time for the storage circuit. Consequently a feedback of the beam current to a level which is not troublesome can be obtained in a very quick manner without the current intensity in circuit elements correcting the charge of the storage circuit becoming too large.

The extended charge correction time of the storage circuit can be obtained by applying a correction charge from a current source which is independent of the measuring value under the influence of the threshold circuit during a time which is considerably longer than the measuring time, or by increasing the measuring frequency so that a current source which is dependent on the measuring value can recharge the storage circuit more frequently per unit of time.

The invention will now be described with reference to the drawing and some embodiments.

In the drawing:

FIG. 1 illustrates by way of a block-schematic diagram a television display apparatus according to the invention.

FIG. 2 illustrates by way of a block-schematic diagram a further elaboration of a television display apparatus according to the invention,

FIG. 3 likewise illustrates by way of a block-schematic diagram a further embodiment of a television display apparatus according to the invention,

FIG. 4 illustrates partly by way of a block-schematic diagram and partly by way of a principle circuit diagram a colour television display apparatus according to the invention including a sequential measuring circuit and

FIG. 5 shows a number of wave forms as may occur in the circuit of FIG. 4.

In FIG. 1 a video signal is applied to an input 1 of a level insertion circuit 3 which signal is provided with a reference level every time during some line times of a field some time after a field flyback with the aid of a signal obtained from an output 7 of a pulse generator 9 and applied to a further input 5. The video signal with the reference level is passed from an output 11 of the level insertion circuit 3 to an input 13 of a beam current level correction circuit 15. A level correction quantity obtained from an output 19 of a measuring circuit 21 is applied to a further input 17 of the beam current level correction circuit 15. A video signal having a reference level and an added level correction quantity is obtained from an output 23 and is applied to an input 25 of an amplifier 27 an output 29 of which is connected to the wehnelt electrode of a television display tube 31. A cathode of the television display tube 31 is connected to an input 33 of the measuring circuit 21.

The measuring circuit 21 includes a gating circuit 35 connected to the input 33 which applies a quantity dependent on the beam current to an output 37 during the occurrence of the reference level. The gating circuit 35 is arranged with the aid of a gating signal applied to a gating signal input 39 and originating from an output 41 of the pulse generator 9.

The quantity dependent on the beam current produced during the occurrence of the reference level is applied from the output 37 of the gating circuit 35 to an input 43 of a detection circuit 45 and to an input 47 of a threshold circuit 49.

An output 51 of the detection circuit 45 is connected in the normal operating condition to a storage capacitor 53 which constitutes a storage circuit with the circuit arrangement connected thereto and which has a time constant which is long relative to the field period. The beam current level correction quantity occurring at the output 19 of the measuring circuit 21 is then obtained from the storage capacitor 53 which quantity readjusts the beam current to a constant value.

The threshold circuit 49 has an output 55 which is connected to an operation signal input 57 of a circuit 59 extending the charge correction time of the storage circuit. The circuit 59 extending the charge correction time includes a second storage capacitor 60 which can retain a voltage optionally provided by the threshold circuit for some time and a changeover switch 61 an operation signal input 63 of which is connected to the operation signal input 57 of the circuit 59 extending the recharge time. The changeover switch 61 has an input 67 which is connected to the output 51 of the detection circuit 45 and an input 69 which is connected through a resistor 71 to a positive supply voltage. An output 73 of the change-over switch is connected to the storage capacitor 53 and the output 19 of the measuring circuit 21.

In the normal operating condition the change-over switch 61 occupies the position shown and the above-described beam current control occurs.

When the beam current is very large during the measuring time, which means that the voltage at the storage capacitor 53 is very low, threshold circuit 49 applies a voltage to its output 55 which sets the change-over switch 61 to the position not shown and, dependent on the discharge time of the second storage capacitor 60, retains it in that position for some time, for example, several hundred line times. The resistor 71 then constitutes a current source independent of the measuring value which increases the voltage across the storage capacitor 53. During the next measurement the threshold circuit 49 determines whether this increase has been sufficient or not so that the change-over switch 61 can remain in the position shown or must be reset once more for some time.

The pulse generator 9 has an input 75 to which field frequency pulses derived from field synchronising or field flyback pulses are applied and an input 77 to which pulses derived from line synchronizing or line flyback pulses are applied. The input 75 is connected to an input 79 of a shift register 81 which in this case has four sections but of which the number of sections can of course be chosen arbitrarily. A line frequency pulse originating from the input 77 serves as a clock pulse for the shift register 81 into which, for example, at the end of the field flyback time a one is shifted which is shifted one section further at every subsequently occurring clock pulse. The last two sections of the shift register have outputs 83 and 85 which are connected to inputs 87 and 89, respectively, of a gating circuit 91 which has a further input 93 connected to the input 77. The gating circuit 91 supplies signals to two outputs 95 and 97 which signals are applied to the outputs 7 and 41 of the pulse generator 9 and which have the functions described hereinbefore.

In FIG. 2 corresponding components have the same reference numerals as those in FIG. 1. For the description of this Figure reference is made to FIG. 1.

The circuit arrangement of FIG. 2 differs from that of FIG. 1 in that the second storage capacitor 60 of the circuit 59 extending the recharge time of the first storage capacitor 53 is replaced by a trigger circuit 99 a set input of which is the input 57 which is connected to the output 55 of the threshold circuit and a reset input 101 of which is connected to the input 75 of the pulse generator 9 to which the field frequency pulses are applied. An output 103 of the trigger circuit is connected to the input 63 of the change-over switch 61 and to an operation signal input 105 of a second change-over switch 107 in the pulse generator 9. An input 109 of the second change-over switch 107 is connected to an output 111 of the shift register 81 and an input 113 is connected to the input 75 for field frequency pulses from the pulse generator 9. An output 115 of the change-over switch 107 is connected to the input 79 of the shift register 81.

In the position shown of the change-over switches 61, 107 the operation of the circuit arrangement is the same as in the case of FIG. 1. When a too high beam current is detected by the threshold circuit 49, the trigger circuit 99 will be set and the change-over switches 61 and 107 will be brought to the position not shown. The storage capacitor 53 will then be charged through the resistor 71 likewise as in the case of FIG. 1. The change-over switch 107 connects the output 111 of the shift register 81 to its input 79 so that always a one circulates in the shift register 81 and a continuous measurement of the beam current is effected until the threshold circuit 49 no longer detects a too high beam current and due to the output voltage dropping out or due to the next field frequency pulse at the reset input 101 the trigger circuit 99 is reset and the normal state is restored.

In FIG. 3 corresponding components have the same reference numerals as those in FIGS. 1 and 2 and reference is made to the description associated with the relevent Figures.

The circuit arrangement differs from that of FIG. 2 by the absence of the first change-over switch 61. Consequently, when a too high beam current occurs the normal control loop is maintained and only due to the second change-over switch 107 the shift register 81 is enabled so that a strongly increased measuring frequency and an accelerated feedback occurs through the normal control loop to the normal state. The recharge time of the storage capacitor 53 is in this case extended because per field period a larger number of measurements and associated recharges of this capacitor is effected.

FIG. 4 shows a colour television receiver including a circuit arrangement according to the invention in which the principle as described with reference to FIG. 2 is used. The circuit arrangement has a colour television display apparatus having a sequential beam current measuring circuit whose principle is described in prior application Ser. No. 402,159, filed Oct. 1, 1973, now abandoned.

When a colour television signal is applied to an input 201 of a high frequency-intermediate frequency and detection section 203, a luminance signal Y is produced at an output 205, a chrominance signal Chr is produced at an output 207 and a synchronizing signal S is produced at an output 209.

The synchronizing signal S is applied to an input 211 of a time base generator 213 connected to the output 209. Deflection currents for a television display tube 221 are obtained at two outputs 215 and 217 and an EHT for the supply of the display tube 221 is obtained at an output 223.

An output 227 of the time base generator 213 applies line flyback pulses to an input 229 and an output 231 applies field flyback pulses to an input 233 of a pulse generator 235. These pulses are shown in FIG. 5 by the waveforms 529 and 533, respectively.

Furthermore FIG. 5 shows a number of waveforms 537, 509, 514, 544, 548, 520, 539, 541, 543, 547, 549, 551, 553, 656, 555, 565 and 563 and a number of instants t ₁ , t ₁ +T ₁ , t ₃ to t ₁₀ , t _10a , and t ₁₀ + T ₂ , t ₁₀ + 2T ₂ , T ₁₁ , t ₁₁ + T ₁ , t ₁₁ + 2 T ₁ , and t ₁₂ which are important for explaining the operation of the circuit arrangement. The waveforms are not shown to scale.

The pulse generator 235 has a number of outputs 237, 239, 241, 243, 244, 245, 247, 249, 251, 253, 255 and an input 256 at which the waveforms 537, 539, 541, 543, 545, 547, 549, 551, 553, 555 and 556, respectively, are present.

The output 255 of the pulse generator 235 is connected to an input 257 of a level insertion circuit 259 an input 261 of which is connected to the output 205 of the section 201 and receives the luminance signal Y therefrom. The level insertion circuit 259 has an output 263 at which the modified luminance signal denoted by the waveform 563 of FIG. 5 is produced which includes a reference level 521 during three line times t ₅ -t ₆ , t ₇ -t ₈ , t ₉ -t ₁₀ at the commencement of the field, which level is inserted with the aid of the wave-form 555 applied to the input 257.

The waveform 565 is then produced at a further output 265 of the level insertion circuit 259, which waveform is applied to an input 267 of a chrominance signal amplifier 269 for suppressing the chrominance signal applied to an input 271 connected to the output 207 of the section 201.

The output 263 of the level insertion circuit 259 is connected to inputs 285, 287 and 289 of suppression and level correction circuits 291, 293 and 295, respectively, inputs 297, 299 and 301 of which are connected to outputs 247, 249 and 251 of the pulse generator 235 and inputs 303, 305 and 307 of which are connected to outputs 309, 311 and 313, respectively, of a demodulator and matrix circuit 315 to whose input 317 a chrominance signal originating from an output 319 of the chrominance signal amplifier 219 is applied.

The demodulator and matrix circuit applies a (B-Y), (G-Y) and (R-Y) colour difference signal to its outputs 309, 311 and 313, respectively. These colour difference signals are combined in the suppression and level correction circuits 291, 293 and 295 with the modified luminance signal applied to the inputs 285, 287 and 289 to form colour signals R, G and B, while as a result of the signals with the waveforms 547, 549, 551 at the inputs 297, 299 and 301 the reference level 521 originating from the luminance signal is suppressed in a special sequence. In this example a sequence is maintained in which only a reference level is left in the R-signal from t ₅ to t ₆ , in the G signal from t ₇ to t ₈ and in the B signal from t ₉ to t ₁₀ .

The suppression and level correction circuits 291, 293 and 295 furthermore have outputs 339, 341, 343 connected to the wehnelt electrode of the red, blue and green guns of the display tube 221 and inputs 345, 347 and 349 connected to the output 237 of the pulse generator 235 for receiving a suppression signal of the waveform 537 which prevent a signal supply to the display tube 221 during the time base flyback times. Inputs 351, 353 and 355 are furthermore connected to outputs 357, 359 and 361, respectively, of a measuring circuit 362 which outputs are connected to storage capacitors 363, 365 and 367, respectively, whose other ends are connected to earth. The storage capacitors 363, 365 and 367 are connected to switches 377, 378 and 379 respectively, operation signal inputs 383, 384 and 385 of which are connected to outputs 239, 241 and 243, respectively, of the pulse generator 235.

The switch 377 conducts from t ₅ to t ₆ , the switch 378 conducts from t ₇ to t ₈ and the switch 379 conducts from t ₉ to t ₁₀ . Each of these switches introduces a level applied to an input 381 thereof into the relevant storage capacitors 363, 365, 367.

The cathodes of the television display tube 221 are interconnected and are connected to an input 389 of the measuring circuit 362. Sequentially by each cathode voltages is generated across a resistor 393 connected to the input 389 which resistor is shunted by a diode 394 and is connected at its other end to a voltage of + 130 Volt during the said periods. This generated voltage produces a corresponding voltage across the storage capacitors 363, 365 and 367 which is retained during the next field flyback time and is passed on for beam current control to the inputs 351, 353 and 355 of the suppression and level correction circuits 291, 293 and 295. The two cathodes which are not measured then do not convey any beam current.

An input 391 of the measuring circuit is connected to the output 245 and an input 392 is connected to the output 253 of the pulse generator 235.

The screen grids of the guns of the television display tube 221 are each connected to an adjusting point of potentiometers 395, 396 and 397 between + 130 Volt and a higher voltage ++.

A signal occurring at the cathodes of the television display tube 221 produces a voltage across the resistor 393 which voltage is limited by the diode 394 blocked during the measuring periods. This voltage is applied to the base of a pnp transistor 401 whose emitter is connected to the other end of the resistor 393. The collector of the transistor 401 is connected to earth through a resistor 403, connected to earth through a capacitor 405 for high frequencies and has a negative feedback to the base through a resistor 407.

A signal amplified by the transistor 401 is applied through a series arrangement of a capacitor 409, a resistor 410 and a resistor 411 to the base of an npn transistor 413. The collector of an npn transistor 415 serving as a clamping switch is connected between the resistors 410 and 411. The emitter is connected to a clamping voltage V _kl and the base is connected to the input 391 of the measuring circuit 362.

As a result of the waveform applied to the input 391, which waveform is the inverse form of waveform 544, the transistor 415 is cut off during the occurrence of the signal produced by the reference level 521. During the preceding line periods the capacitor 511 has reached a constant charge condition (clamped) through the then conducting transistor 415 because both the luminance signal and the chrominance signal are suppressed and the cathodes of the display tube 221 do not convey any current.

During the period t ₄ -t ₁₀ signals caused by the reference level 521 occur successively during the periods t ₅ -t ₆ , t ₇ - t ₈ and t ₉ - t ₁₀ at the successive cathodes of the display tube 221 which signals are applied in an amplified manner to the base of the transistor 413 and which furthermore appear at the emitter of an npn transistor 412 connected to the base of the transistor 413.

When the signals have an amplitude which is not too large they are passed on by the transmitter 413 through its emitter and a resistor 423 to the base of an npn transistor 425.

When the signals at the base of the transistor 413 have a too large amplitude the emitter of the transistor 412 will become more negative than its base connected to a threshold voltage V _dr and this transistor 412 will start conducting and hence operate a set input of a trigger circuit 414 connected to its collector which then provides a positive voltage for its output 416. This positive voltage (waveform 556) is applied to an output 418 which is connected to the input 256 of the pulse generator 235 and which causes a variation to a free-running state which will be described hereinafter, while this voltage also controls the base of an npn transistor 422 through a resistor 420 and the emitter of this transistor is connected to earth while the collector is connected to the base of the transistor 413. The base of this transistor 413 then is no longer controlled because the transistor 422 acts as a short circuit for its input signal. The trigger circuit 414 furthermore has a reset input 424 to which a field flyback pulse (waveform 533) originating from the output 231 of the time base generator 213 is applied so that the trigger circuit 414 is reset every time at the commencement of the field flyback.

In the state in which the beam current is not too large the transistors 412 and 422 do not conduct and the signal is passed on through the transistor 413 to the base of the transistor 425 whose emitter is connected to the collector of an npn transistor 427 whose emitter is connected to earth and whose base is connected through a resistor 428 to the input 392 of the measuring circuit. The collector of the transistor 425 is connected through a resistor 430 to the base of an npn transistor 433 whose emitter is connected through a resistor 435 to the input 381 of the sequence switches 377, 378 and 379.

The collector of the transistor 425 is connected through a resistor 432 to the emitter of an npn transistor 434 arranged as an emitter follower whose base is connected through a resistor 436 to an input 438 of the measuring circuit 362 which is connected to the output 244 of the pulse generator 235 at which the waveform 544 occurs. This transistor 434 conducts during the measuring period and furthermore every time during a number of line times in the field scan time when the pulse generator 235 has acquired a free-running state (waveform 544) due to a too high beam current with the aid of the trigger circuit 414.

The transistor 427 conducts due to the waveform 553 during the measuring periods so that in the periods t ₅ -t ₆ , t ₇ -t ₈ , t ₉ -t ₁₀ the emitter of the transistor 425 is connected to earth through the transistor 427 and the transistor 425 acts as an amplifier and passes on a signal to the base of the transistor 433 and recharges through its emitter the storage capacitors 363, 365 and 367 during the corresponding periods t ₅ -t ₆ , t ₇ -t ₈ and t ₉ -t ₁₀ , respectively.

In the state of the circuit arrangement where a too large beam current has been detected and where the trigger circuit 414 is set, the transistor 425 does not conduct due to the short circuit of the base of the transistor 413 so that then the storage capacitors 363, 365 and 367 are recharged by the emitter follower 434, 433 every time during the periods when the waveform 544 is positive. This is effected a large number of times during field scan periods following the set of the trigger circuit 414 so that the storage capacitors are charged quickly to a higher voltage to reduce the beam current through the control loops as quickly as possible again.

The operation of a control loop will now be described. This control loop not only includes the television dislay tube 221 and the measuring circuit 362 but also the suppression and level correction circuit 295.

A modified luminance signal Y _m according to the waveform 563 is applied to the input 289 in which signal the reference level 521 is present during the periods t ₅ -t ₆ , t ₇ -t ₈ and t ₉ -t ₁₀ and which level is suppressed from t ₁ to t ₅ . The input 289 is connected to the base of an npn transistor 601 whose emitter is connected to earth through a resistor 603 and whose collector is connected through a resistor 605 to the emitter of an npn transistor 607. The collector of the transistor 607 is connected to a positive voltage. The base is connected to the input 307 and receives a red colour difference signal -(R-Y) which is suppressed from t ₁ to t _10a .

Furthermore the emitter of the transistor 601 is connected to the emitter of an npn transistor 609 whose collector is connected to a positive voltage and whose base is connected to the input 301.

A voltage having the waveform 547 is applied to the input 301 so that the transistor 609 conducts during the periods t ₇ -t ₈ and t ₉ -t ₁₀ and thus cuts off the transistor 601 so that also the transistor 607 does not convey current and the signal at the collector of the transistor 601 is suppressed.

When the transistor 609 does not conduct a signal is generated at the collector of the transistor 601 which signal is a combination of the -(R-Y) signal supplied through the emitter follower 607 and the -Y _m signal amplified through the transistor 601. This combination is a -R signal in which a level corresponding to the level 521 is present during the period t ₅ -t ₆ and which is suppressed in the line periods t ₁ -t ₅ and t ₆ -t _10a . This signal is applied to the base of an npn transistor 611 whose collector is connected to a positive voltage and whose emitter is connected through a resistor 613 to the collector of an npn transistor 615 whose emitter is connected to earth through a resistor 617 and whose base is connected to a reference voltage of 1.4 V.

The emitter of the transistor 615 is furthermore connected to the emitters of two npn transistors 619 and 621 whose collectors are connected to a positive supply voltage.

A level correction voltage originating from the capacitor 367 and applied to the input 355 is transferred to the base of the transistor 619 through an npn transistor 623 arranged as an emitter follower while a suppression signal having the waveform 537 applied to the input 349 is passed on to the base of the transistor 621. The latter signal ensures the common suppression of the beam current during the line and field flyback periods. Due to the fact that the otherwise cut-off transistor 621 is blocked during these flyback periods, the transistor 615 is cut off and no signal is passed to the collector of this transistor because then the emitter circuit of the transistor 611 is interrupted.

The level correction voltage applied to the input 355 is then applied through the emitters of the transistors 623 and 619 to the emitter of the transistor 615 and influences the direct current through the resistor 613 provided by the transistor 615 and hence of the direct current level of the -R signal passed through the emitter of the transistor 611 to the collector of the transistor 615.

The collector of the transistor 615 is connected to the base of an npn transistor 625 whose collector is connected to a positive voltage and whose emitter is connected through an adjustable resistor 629 to the base of an npn transistor 631. The emitter of the transistor 631 is connected to earth and the collector is connected to the emitter of an npn transistor 633 whose collector is connected through a resistor 635 to a positive supply voltage of +130 V and whose base is adjusted to a bias voltage of +5 V. The base of the transistor 631 receives a negative feedback voltage from the collector of the transistor 633 through a potential divider 637, 639 to a negative voltage.

A red colour signal amplified by the transistors 631 and 633 is obtained from the collector of the transistor 633 and is applied through the output 343 to the wehnelt electrode of the red gun of the display tube 221. This signal includes the correction level originating from the capacitor 363 with which the beam current in the red gun is adjusted to a desired value.

The amplification of the circuit arrangement is adjustable with the resistor 629 so as to perform for example, a white point correction. Due to the control loop such an adjustment has substantially no influence on the beam current which is produced by the reference level 521 so that the black level and hence the colour of dark picture parts does not change due to the adjustment.

The modified luminance signal Y _m for the input 289 is obtained in the level insertion circuit 259.

A luminance signal is applied to the input 261 which signal is applied to an input 701 of an amplifier 703. Furthermore the amplifier 703 has an input 707 to which an adjustable direct voltage is applied with the aid of a potentiometer 709 which serves for luminance adjustment and an input 771 to which a suppression signal is applied as is shown by the waveform 555. The black level of the picture 520 in the waveform 563 is adjustable with the aid of the potentiometer 709 relative to the level occurring during the suppression periods. The waveform 563 is the luminance signal which occurs at the output 263 across an emitter resistor 713 of an npn transistor 715 arranged as an emitter follower whose base is connected to an output 705 of the amplifier 703.

A signal having the waveform 555 is applied to the input 257 of the level insertion circuit through a capacitor 717, which signal is passed at one end through a resistor 721 to the base of an npn transistor 723 and at the other end through a potential divider 725, 727 to the base of an npn transistor 729.

During the most positive parts of the waveform 555 a low voltage is produced across a collector resistor 731 of the transistor 723. The level of the signal during the periods t ₅ -t ₁₀ lies below the cut-off point of the transistor 727 so that this is not found back in the collector signal 565 of this transistor. As a result of the potential divider 725, 727 connected to the positive voltage the transistor 729 only reacts to the most negative parts of the signal 555 and the transistor 729 is cut off during the periods t ₅ -t ₆ , t ₇ -t ₈ and t ₉ -t ₁₀ so that then positive going square-wave voltages occur at the collector of this transistor.

The collector of the transistor 729 is connected through a resistor 733 to a potential divider including a resistor 735 and a series arrangement of a diode 736 and a resistor 737. The voltage at the wiper of this potential divider has a constant value of + 2.2 V which also occurs at the collector of the transistor 729 during the said periods. This voltage is passed on through a resistor 738 to the base of an npn transistor 739 whose collector is connected to a positive supply voltage and whose emitter is connected to the emitter of the transistor 715. During the periods t ₅ -t ₆ , t ₇ -t ₈ and t ₉ -t ₁₀ the emitter of the transistor 739 is brought to a voltage of +1.5 V due to its base voltage so that the transistor 715 is cut off and the reference level 521 is produced at the output 263. The rest of the period the transistor 739 is cut off and a signal is applied to the output 263 through the transistor 715.

In the waveform 563 arrows show that the black level 520 of the signal is adjustable relative to the constant reference level 521 which as a result of the said control corresponds to a constant beam current valve in the guns of the display tube 221.

The pulses for the various parts of the circuit arrangement are supplied by the pulse generator 235.

This generator has four trigger circuits 801, 803, 805 and 807. A signal which relative to the signal 533 at the input is delayed over a time T ₁ is applied to a set input 809 of the trigger 801 from the input 233 through a delay circuit including a resistor 811 and a capacitor 813. This signal is shown in the waveform 509. Before this pulse appears the triggers 801, 803, 805 and 807 are assumed to be in the reset condition. At the instant t ₁ + T ₁ the trigger 801 is brought to the set condition. A signal represented by the waveform 514 then appears at an output 814 thereof which signal is applied to four AND gate circuits 817, 819, 821 and 823.

Furthermore an inverted set signal from the trigger 801 is applied to the gate 817 as well as a line frequency pulse signal originating from the input 229.

In addition a signal represented by the waveform 544 originating from an output 816 of the trigger 803 and a signal originating from an AND gate 817 are applied to the gate 819.

The gate 821 furthermore receives a signal having a waveform 518 from an output 818 of the trigger 805 and a signal originating from an AND gate 829.

Furthermore the gate 823 receives a signal having a waveform 520 from an output 820 of the trigger 807 and an inverted line frequency pulse signal originating from the input 229. An output of this gate 823 is connected through a delay circuit including a resistor 831 and a capacitor 833 to an AND gate 835 which also receives a line frequency pulse signal from the input 229. The output of this gate 835 is connected to the reset inputs of all four triggers. The gates 827 and 829 receive an inverted signal from the input 229 and an output signal from the triggers 805 and 807, respectively.

Furthermore a switchable feedback is present between the output 814 of the trigger 801 and the set input thereof, which feedback is only switched on when the beam current in the display tube 221 is too high. This feedback starts from the output 814 through a resistor 837 which is connected to the base of an npn transistor 839 whose emitter is connected to earth and whose collector is connected through a resistor 841 to the input 256 and is connected to the output of the trigger circuit 414, the base of a transistor 843 connected to the collector of the transistor 839 having its collector connected to a positive supply voltage and being connected through a resistor 845 connected to the emitter of this transistor 843 to the set input 809 of the trigger 801. Together with the capacitor 813 the resistor 845 constitutes a delay circuit having a time delay T ₂ . The feedback is not present when the beam current is so low that the trigger circuit 414 is not set; the voltage at the collector of the transistor 839 is then low and the transistor 843 is cut off. This condition is assumed to be present at the commencement of the time axis in FIG. 5.

When the voltage at the input 256 becomes high due to the trigger 414 coming in the set condition the feedback will be switched on. The transistor 839 will then act as an inverter.

When at the instant t ₁ + T ₁ the trigger 801 is set, the trigger 803 is blocked at the same time by the inverted set signal from the trigger 801 applied to the gate 817. The triggers 803, 805 and 807 then remain in the reset condition. After the end of the set pulse from the trigger 801, i.e. after the instant t ₃ = t ₂ +T ₁ the gate circuit 817 becomes conducting at the instant t ₄ due to the next line frequency pulse and brings the trigger 803 in the set condition. The gate 819 then becomes conducting after the end of the line pulse under the influence of the gate 827 and applies a pulse through a delay network 845, 847 to an input of an AND gate 849 to which also the line frequency signals of the input 229 are applied. At the next line pulse at the instant t ₆ the trigger 805 is set so that the gate 821 provides a pulse which is delayed by a network 851, 853 and is applied to an input of an AND gate 855 to which furthermore the line frequency pulse signal from the input 229 is applied. This gate 855 becomes conducting at the next line pulse at the instant t ₈ under the influence of the gate 829 and brings the trigger 807 to the set condition.

A pulse is obtained from the output of the gate 823 which begins after termination of the line pulse and which is applied delayed through the network 831, 833 to the gate 835 which then passes the next line pulse at the instant t ₁₀ so that the triggers 801, 803, 805 and 807 are reset.

When meanwhile the trigger circuit 414 has not reached the set condition during the time between t ₅ and t ₁₀ in which the beam currents in the display tube are measured, the triggers 801, 803, 805 and 807 remain in the reset condition until the next field pulse.

When the trigger circuit 414 changes to the set condition due to a too high beam current for example between t ₅ and t ₆ , the described feedback from the output 814 of the trigger 801 is provided at its set input 809 and a positive pulse will appear again at the instant t ₁₀ + T ₂ at the input 809 which pulse will bring the trigger 809 to the set condition again so that at an instant t ₁₀ + 2T ₂ the trigger 803 is set whereafter the triggers repeat the above-described conditions every time.

At the next field pulse at the instant t ₁₁ , however, the trigger 414 is reset so that the feedback in the pulse generator 235 is interrupted and a set cycle of the triggers 801, 803, 805 and 807 already started is terminated at the instant t ₁₂ when the gate 835 applies a line pulse to the reset input of the trigger 801 so that voltage at the output 814 remains equally low and all triggers are reset. After the line pulse drops out at its reset input the trigger 801 is brought immediately to the set condition so that at the next line pulse which lies after the instant which is T ₁ later than the trailing edge of the field pulse 533 the trigger 803 is set again and three line times later everything is reset and remains reset when the beam current has become sufficiently low due to the repeated recharge of the capacitors 363, 365, 367 in the measuring circuit 362.

A number of gate circuits supplying the output signals from the pulse generator 235 are coupled to the outputs of the trigger circuits.

The output 239 is connected to an output of an AND gate 857 which has an input connected to the output 816 of the trigger 803, an inverted input which is connected to the output 818 of the trigger 805 and an inverted input which is connected to the input 229 of the pulse generator. The output of this gate then applies the waveform 539.

The output 241 is connected to an output of an AND gate 859 and input of which is connected to the output 818 of the trigger 805, an inverted input is connected to the output 820 of the trigger 807 and an inverted input is connected to the input 229 of the pulse generator 235. The AND gate 859 provides an output signal having the waveform 541.

The output 243 is connected to an output of an AND gate 861 an input of which is connected to the output 820 of the trigger 807 and an inverted input is connected to the input 229 of the pulse generator 235. The AND gate 861 provides an output signal having the waveform 543.

The output 244 is connected to the output 816 of the trigger 803 and the output 245 is connected to an inverted output of this trigger.

The output 247 is connected to an output of an OR gate 863, the output 249 is connected to an output of an OR gate 865 and the output 251 is connected to an output of an OR gate 867. The inputs of the gate 863 are connected to the outputs of the gates 861 and 859, the inputs of the gate 865 are connected to the outputs of the gates 861 and 857 and the inputs of the gate 867 are connected to the outputs of the gates 857 and 859. The gates 863, 865 and 867 supply the signals 547, 549 and 551, respectively.

The output 253 is connected to an output of an AND gate 869 an input of which is connected to the output 816 of the trigger 803 and an inverted input is connected to the input 219 of the pulse generator 235. The waveform 553 is supplied by this gate.

The output 237 is connected to an output of an OR gate 871 whose inputs are connected to the inputs 229 and 233 of the pulse generator. The waveform 537 is provided by the gate 871.

The output 255 is connected to an output of a gate and superimposition circuit 873 an input of which is connected to an inverted output of the trigger 801, an inverted input is connected to the output of the gate 871 and an inverted input is connected to the output of the gate 869. The gate and superimposition circuit 873 provides the waveform 555.

Since the latter waveform is applied through the capacitor 717 to the input 257 of the level insertion circuit 259, an unsatisfactory amplitude selection will take place when the trigger 801, 803, 805 and 807 run free for a long period so that a charge of the capacitors 363, 365, 367 in the measuring circuit 362 is effected independently of the beam current when it has exceeded a given maximum value as described in the foregoing. The level shift due to this capacitor 717 and the results thereof are not important for the understanding of the invention and are therefore not shown in the waveforms.

Although the control voltage in the described embodiments is applied to a wehnelt electrode of the display tube it may alternatively be applied to a different control electrode.

The insertion of the reference level may of course alternatively be effected in a different manner such as, for example, by clamping the video signal on the control signal.

For switching over the described functions in case of a too large beam current value a schmitt trigger or a monostable multivibrator may be used if desired instead of a bistable trigger circuit having a set and reset, input, or an output signal from the threshold circuit may be used.

The type of measuring circuit is not important for using the step according to the invention. Instead of a sequence measuring circuit it is alternatively possible to use a simultaneous measuring circuit in which all beam currents are simultaneously measured in the case of a multigun display tube. Also the input circuit of the measuring circuit may of course be adapted as desired.

If desired other counting circuits may be used instead of a shift register.

PHILIPS CHASSIS K12 Television receiver including a beam current limiting circuit :

1. A beam current limiting circuit for a cathode ray display tube having at least one cathode, said circuit comprising an isolation circuit having a high impedance input means for coupling to said cathode and an output, and time constant circuit means for generating a beam current limiting signal in accordance with the cathode beam current having an input coupled to said isolation circuit output and an output means for providing said beam current limiting signal.

2. A circuit as claimed in claim 1 wherein said isolation circuit comprises an amplifier.

3. A circuit as claimed in claim 1, wherein the display tube has a plurality of interconnected cathodes, said beam current limiting circuit further comprising a parallel combination of a first resistor and a series circuit including a diode and a second resistor, said parallel combination being adapted to be coupled to a supply source, the junction of the diode and the second resistor being coupled to the isolation circuit input means, and a control circuit means for maintaining a reference level in the cathode current constant having an input means for coupling to said cathodes.

4. A circuit as claimed in claim 1, wherein the display tube has a plurality of interconnected cathodes, and further comprising a resistor coupled between said cathodes and a supply source, a capacitor coupled said cathodes, and a circuit means for recovering a direct-current component coupled between said capacitor and the input means of said isolation circuit.

5. A circuit as claimed in claim 1, wherein the display tube has a plurality of cathodes, and further comprising a plurality of resistors respectively coupled between said cathodes and the input means of said isolation circuit, a threshold circuit, a plurality of diodes coupled between said cathodes respectively and an input of said threshold circuit, the isolation circuit including a circuit means for obtaining a voltage which is a measure of the mean value of the overall cathode current and the threshold circuit including a circuit means for obtaining a voltage which is a measure of the peak values in each of the cathode currents.

6. A circuit as claimed in claim 1, further comprising a resistor and a diode coupled between said cathode and a source of a supply voltage, a measuring device means for measuring the black level in the beam current having an input coupled across the diode, and the resistor and the diode being coupled to the input means of the isolation circuit.

7. A circuit as claimed in claim 1, wherein said tube has a plurality of control electrodes, and further comprising a radio frequency, intermediate frequency, and detector section having luminance and chrominance signal outputs; first and second amplifiers each having signal inputs coupled to said outputs respectively, gain control inputs coupled to said time constant circuit output means, and outputs; a demodulator coupled to said second amplifier output; and a matrix circuit having inputs coupled to said first amplifier output and said demodulator, and a plurality of output means for coupling to said control electrodes respectively.

Description:

The invention relates to a television receiver having a beam current limiting circuit which includes a converting circuit with a time constant network for converting the current through the cathode of a display tube to a beam current limiting signal, by means of which overdriving of the display tube is avoided, characterized in that the time constant network is connected to the cathode via a separating circuit so that the converting circuit substantially cannot affect the current through the display tube.
U.S. Pat. No. 3,735,029 describes a television receiver of the abovementioned type in which the cathode of the display tube is connected to the mains voltage via a time constant network which is a parallel circuit of a capacitor with a series circuit of a zener diode and a resistor.
When proportioning the high voltage supply the properties of the time constant network in the cathode circuit of the display tube must be taken into account, whilst it is not possible to apply, for example, a peak detection circuit to prevent defocussing due to the occurrence of short-duration peak signals at the display tube.
It is an object of the invention to mitigate these drawbacks.
Hence a television receiver of the aforementioned type according to the invention is characterized in that the time constant network is connected to the cathode via a separating circuit so that the converting circuit substantially cannot affect the current through the display tube.

For example, there may be derived from a small resistor in a cathode circuit of the display tube a voltage which is a measure of the current carried by the cathode circuit concerned. Via a separating circuit, for example an amplifier and matched time constant networks, both information about the mean value of this current and information about any peak values which may occur in it can be derived from this voltage in simple manner, without the cathode circuit being affected. This enables a suitable beam current limiting quantity to be composed.
Embodiments of the invention will be described, by way of example, with reference to the accompanying diagrammatic drawings, in which:
FIG. 1 illustrates a television receiver in which a circuit for generating a beam current limiting voltage is included in the cathode circuit of a display tube according to the invention.
FIG. 2 illustrates a television receiver as shown in FIG. 1, representing the manner in which a clamping circuit which controls a beam current reference level can also be connected to the cathodes of the display tube, and
FIG. 3 shows another embodiment of a television receiver provided with a circuit for generating a beam current limiting quantity according to the invention.

Referring now to FIG. 1, a television signal is supplied to an input 1 of a high-frequency, intermediate-frequency and demodulation section 3 of a television receiver. The section 3 at a first output 5 provides a luminance signal Y and at a second output 7 delivers a chrominance signal CHR. The luminance signal Y is supplied via an amplifier 9 to an input 11 of a matrix circuit 13. The chrominance signal CHR is applied via an amplifier 15 to an input 17 of a demodulating circuit 19 which supplies colour difference signals (R-Y) and (B-Y) to inputs 21 and 23 respectively of the matrix circuit 13. The matrix circuit 13 then delivers colour signals R, G and B which are applied to the control electrodes of a display tube 25.
The cathodes of the display tube 25 are interconnected and through a resistor 27 have a supply voltage V_k applied to them. The resistor 27 can be so small as to cause no appreciable negative feedback.
Across the resistor 27 a voltage is produced which is proportional to the beam current flowing through the display tube 25. This voltage is applied via a capacitor 29 to the junction of the cathode of a diode 31 and a resistor 33. To the anode of the diode 31 is applied a positive voltage such, for example 1.4 volts, that the lowest signal voltages which may appear at the cathode of the diode 31 are clamped at a voltage level of about 0.7 volt. Via a resistor 35 the signal at the cathode of the diode 31 is supplied to the base of a transistor 37 which as a result produces across its emitter resistor 39 a signal voltage the lowest value is about zero volts when the voltage at a supply terminal 41 is zero volts.
The voltage at the emitter of the transistor 37 is applied via a resistor 43 to a capacitor 45 across which a direct voltage is set up which is a measure of the mean value of the cathode current of the display tube 25. Together with the capacitor 45 the resistor 43 forms a time constant network which is isolated, by means of the transistor 37 from the cathode circuit of the display tube and consequently does not affect it. The voltage across the capacitor 45 is applied via a resistor 47 to the base of a transistor 49 which acts as a threshold circuit and passes collector current only if its base voltage is of the order of 0.7 volt. The collector of the transistor 49 is connected to a supply terminal 55 via two resistors 51 and 53. The junction of the resistors 51 and 53 is connected to a capacitor 57 and to the collector of a transistor 59 the base of which is connected to the emitter of the transistor 37.
The transistor 59 acts as a threshold circuit for signal voltages at the emitter of the transistor 37 and passes current only if this signal voltages assumes high values. As a result a voltage which depends upon the amplitude of the said peak signal voltages is set up across the capacitor 57. The transistor 49 produces a further voltage across the capacitor 57 which depends upon the mean value of the current through the display tube so that there is set up across the capacitor 57 a control voltage which depends upon different forms of undesirable overloading of the display tube 25. Also the peak detection circuit with the transistor 49 and the capacitor 57 is isolated from the cathode circuit and cannot affect it.

The voltage across the capacitor 57 serves as beam current limiting quantity and is supplied to control inputs 60 and 61 of the amplifiers 9 and 15 respectively, enabling the contrast and the saturation of the displayed picture to be corrected. Obviously a brightness correction may, if desired, be effected by means of the control voltage obtained across the capacitor 57.
Thus the circuit in the cathode circuit of the display tube acts as a converting circuit by which the cathode current of the display tube is converted into a beam current limiting quantity.
To the cathode resistor 27 of the display tube 25 a direct voltage V_k is applied. As an alternative, a driving voltage may be applied to this cathode resistor. In this case a difference voltage measuring circuit must be connected across the resistor 27 to enable the voltage across the resistor 27 to be measured.
When a diode is so connected between the cathode resistor 27 and the supply voltage V_k as to have its pass direction from the resistor 27 to the supply voltage K_k, the input of a measuring device for measuring the black value of the beam current can be connected across the diode. This measuring device may deliver a control signal by means of which the said black value is maintained constant. In the following embodiments other possibilities of measuring the black value of the beam current are shown.
FIG. 2 shows how not only a circuit for obtaining a beam current limiting quantity but also a control circuit for obtaining a constant black value of the beam currents of the display tube which can be included in the cathode circuit of this tube.
Corresponding component parts of the circuit are designated by the same reference numerals as in FIG. 1.
The interconnected cathodes of the display tube 25 here are connected to the resistor 27 via a diode 63, the series combination of the resistor 27 and the diode 63 being shunted by a resistor 65. The value of the resistor 65 is much higher than that of the resistor 27, being, for example, 100 kΩ and 330 Ω respectively. For very small cathode voltages the diode 63 is nonconductive so that substantially only the large resistor 65 is operative in the cathode circuit. The said small cathode voltages are important with respect to the measurement of the black level and are applied to the input of an amplifier 67.
Larger cathode currents cause, via the diode 63, the small resistor 27 to become operative so that the voltage at the cathodes of the display tube is limited and a beam current limiting quantity is derived from the resistor 27 via a converting circuit 69, which may be of the construction as shown in FIG. 1.
The amplifier 67 supplies a signal to a sequential measuring circuit 71 which during, for example, three line periods in each field blanking period sequantially measures the black values in the beam currents at the different cathodes of the display tube and converts them into control voltages which are applied to level control inputs 73, 75, 77 of colour signal amplifiers 79, 81, 83 respectively.
In this circuit all the information about the beam currents of the display tube both with respect to beam current limiting and to level control is derived from the cathode circuit of the display tube substantially without affecting this cathode circuit which permits simple proportioning and independent and reliable operation of these and other circuits.

FIG. 3, in which corresponding parts are designated by the same reference numerals as in FIG. 1 and 2, shows a method of preventing defocussing of each individual beam. Each of the cathodes of the display tube 25 is connected to a resistor 85, 87, 89, respectively and to the anode of a diode 91, 93, 95 respectively. The cathodes of the diodes 91, 93, 95 are connected to the input of a threshold circuit in the form of a transistor 97 the emitter of which is connected via a series combination of two diodes 99 and 101 to the voltage V_k. Consequently the transistor 97 passes current only if the voltage at one of the cathodes of the display tube 25 exceeds a given threshold value, in the present case about 2.8 volts. The collector current of the transistor 97 and hence the direct voltage across a capacitor 103 included in its collector circuit are a measure of a peak currents which occur at the three cathodes. By means of this circuit the peak value of each of the cathode currents is measured so that defocussing of each beam is avoided.
The overall cathode current is further supplied from the junction of the three resistors 85, 87, 89 via a diode 105 to the base of a transistor 107 which is connected as a current amplifier and the emitter of which is connected to the base of a transistor 109. The collector circuit of the transistor 107 includes a capacitor 111 across which a direct voltage is produced which is a measure of the mean value of the joint cathode currents.
The voltages across the capacitors 103 and 111 are combined in a combining circuit 113 which, in its simplest form may be a resistor network to form a desired beam current control voltage or current.
From the collector circuit of the transistor 109 a signal is obtained by means of which the black level in the beam current can be measured. Between the collector of the transistor 109 and the base of the transistor 107 a resistor 115 is connected which has a large value and in the case of small beam current values of the display tube 25 determines the gain of the transistor 109.
If desired, the resistors 85, 87, 89 may be variable to enable the influence of differences in the characteristics of the display tube 25 to be eliminated.
In the embodiments shown, amplifier circuits were used as separating circuit, it is also possible to connect the time constant network, for example via a large resistor, to the cathode and to use a d.c. voltage amplification of a signal which is formed at the time constant network



PHILIPS  26C854/60R  EL GRECO  CHASSIS K12 (20AX) ONVERGENCE CORRECTION CIRCUIT 20AX PHILIPS CHASSIS K12
PHILIPS  26C854/60R  EL GRECO  CHASSIS K12 (20AX) Colour television display apparatus incorporating a television display tube

Colour television display apparatus provided with a display tube in which the electron beams are generated in one plane. The deflection coils are designed in such known manner that the landing points of the beams on the display screen coincide. Owing to tolerance angular errors of the orientation of the plane relative to one direction of deflection convergence errors are produced in the other direction of deflection. They are corrected by means of a magnetic quadripolar field the polar axes of which substantially coincide with the directions of deflection and the field strength of which is a substantially quadratic function of either deflection current or the sum of both functions.

1. Colour television display apparatus incorporating a television display tube having a display screen and two deflection coils for the deflection in two directions of electron beams which are generated in the tube substantially in one plane, a first direction of deflection being substantially parallel to the said plane whilst the second direction of deflection is substantially at right angles to the first direction, the field generated by the deflection coil for deflection in the first direction having a distribution in which its meridional image plane substantially coincides with the screen whilst the field generated by the deflection coil for deflection in the second direction has a distribution in which its sagittal image plane substantially coincides with the screen, the deflection errors due to comma and anisotropic astigmatism being substantially equal to zero, whilst at least one deflection coil is divided into two substantially equal coil halves, characterized in that in order to correct for tolerance angular errors in the orientation of the plane in which the electron beams are generated relative to the first direction of deflection the split deflection coil generates a magnetic quadripolar field the polar axes of which substantially coincide with the directions of deflection and the field strength of which is a substantially quadratic function of the instantaneous strength of the deflection current flowing through at least one deflection coil, and means for clamping the peak of said quadratic field. 2. Apparatus as claimed in claim 1, characterized in that a substantially parabolic correction current which is adjustable in amplitude and in polarity flows in the same direction as the deflection current in one coil half and in the opposite direction in the other coil half and is zero at the middle of the trace interval of the deflection current. 3. Apparatus as claimed in claim 2, in which one direction of deflection is horizontal and the other is vertical, characterized in that a line-frequency correcting current flows through the coil halves of the deflection coil for horizontal deflection and a field-frequency correction current flows through the coil halves of the deflection coil for vertical deflection. 4. Apparatus as claimed in claim 2, characterized in that a sawtooth current supplied by the deflection current generator which produces the deflection current flows through a potentiometer the setting of the slider on which determines the adjustment of the polarity and of the amplitude of the correcting current. 5. Apparatus as claimed in claim 4, in which the deflection current is of field frequency, characterized in that the setting of the slider on the potentiometer also renders symmetrical the deflection fields generated by the coil halves. 6. A display apparatus as claimed in claim 1 wherein said split coil field strength is substantially the sum of quadratic functions of the current flowing through both coils. 7. A color television deflection system for a television display tube having a display screen, said system comprising two deflection coils for the deflection in two directions of electron beams which are generated in the tube substantially in one plane, said first direction of deflection being substantially parallel to the said plane, the second direction of deflection being substantially at right angles to the first direction, the field generated by the deflection coil for deflection in the first direction having a distribution in which its meridional image plane substantially coincides with the screen, the field generated by the deflection coil for deflection in the second direction having a distribution in which its sagittal image plane substantially coincides with the screen, the deflection errors due to comma and anisotropic astigmatism being substantially equal to zero, at least one deflection coil comprising two substantially equal coil halves, means for correcting for tolernace angular errors in the orientation of the plane in which the electron beams are generated relative to the first direction of deflection comprising means for providing that the split deflection coil generates a magnetic quadripolar field the polar axes of which substantially coincide with the directions of deflection and the field strength of which is a substantially quadratic function of the instantaneous strength of the deflection current flowing through at least one deflection coil, and means for clamping the peak of said quadratic field. 8. A deflection system as claimed in claim 7 wherein said split coil field strength is substantially the sum of quadratic functions of the current flowing through both coils.

Description:

The invention relates to a colour television display apparatus incorporating a television display tube having a display screen and two deflection coils for the deflection in two directions of electron beams which are generated in the tube substantially in one plane, a first direction of deflection being substantially parallel to the said plane whilst the second direction of deflection is substantially at right angles to the first direction, the field generated by the deflection coil for deflection in the first direction having a distribution in which its meridional image plane substantially coincides with the screen whilst the field generated by the deflection coil for deflection in the second direction has a distribution in which its sagittal image plane substantially coincides with the screen, the deflection errors due to comma and anisotropic astigmatism being substantially zero, whilst at least one deflection coil is split into two substantially equal coil halves.

Such an apparatus is described by J. Haantjes and G. J. Lubben in "Philips Research Reports", Volume 14, February 1959, pages 65-97 and in U.S. Pat. No. 2,886,125. In this apparatus the landing points of the electron beams on the display screen coincide everywhere, in other words the various beams, which generally are three in number, which intersect the deflection plane along a straight line are imaged as points on the screen. It is assumed that both the construction of the device or devices which generate the beams, for example three cathodes, and the distribution of the deflection fields exactly satisfy the requirements derived in the said paper. In practice, however errors are produced which are due to tolerances so that the images of the beams on the screen are not points but lines which are substantially parallel to the second direction, i.e. convergence errors, for when a point is referred to what is actually meant is that each electron beam strikes a phosphor dot or stripe on the screen to cause it to luminesce in a given colour, the landing points being associated so as to be perceived as a single point. This is no longer the case if the aforementioned straight line, which is the projection of the plane of the three cathodes in the deflection plane, does not exactly coincide with the first direction of deflection but is at an angle thereto. This error is a tolerance error, i.e. it is small, and may be due to a slight misplacement of the cathodes and/or to a slightly incorrect field distribution within the display tube and hence to tolerances in the construction of the deflection coils.

If the first direction of deflection is horizontal and the second one is vertical, the said error entails a convergence error in the vertical direction. The aforementioned straight line in the deflection plane can be made to coincide with the horizontal direction of deflection by rotation. Attempts have been made to cancel the convergence errors due to this rotation by means of a coil which is axially arranged on the neck of the display tube and through which an adjustable direct current flows. The effect of this coil is comparable to that of a focussing coil; it exerts a force on the travelling electrons which causes their paths to be helical, so that some compensation is obtained. It has been found, however, that this solution has the following disadvantages: the residual errors in the corners are increased; the effect on the horizontal and vertical directions are different, so that satisfactory adjustment in both directions is difficult to realise; depending upon the axial position an undesirable influence may occur at the centre of the screen which in turn can be eliminated by the means, for example permanent magnets, provided for static convergence, requiring iterative and hence time-consuming trimming. Furthermore the coil is an expensive component.

The present invention is based on the recognition that the aforementioned convergence errors due to tolerance errors in the construction of the display tube and/or of the deflection coils can be eliminated by means of simple circuits without the need for additional components to be mounted on the neck of the display tube whilst avoiding the aforementioned disadvantages. For this purpose the apparatus according to the invention is characterized in that to correct for tolerance angular errors in the orientation of the plane in which the electron beams are generated relative to the first direction of deflection the split deflection coil generates a magnetic quadripolar field the polar axes of which substantially coincide with the directions of deflection and the field strength of which is a substantially quadratic function of the instantaneous strength of the deflection current flowing through either deflection coil or the sum of both quadratic functions.

It should be mentioned that it is known to use a split deflection coil to generate a quadripolar field the polar axes of which substantially coincide with the directions of deflection. This is described in U. S. Pat. No. 3,440,483 in which, however, the field strength of the quadripolar field is a function of the product of the values of the two deflection currents so that deflection errors due to anisotropic astigmatism can be corrected. In contradistinction thereto the present application described an apparatus having substantially no anistropic astigmatism whilst the quadripolar field generated according to the invention has a field strength which depends upon the value of either deflection current or upon the sum of the squares of the two deflection currents. For the sake of clarity it should be mentioned that in the apparatus according to the said U.S. Patent, in the absence of the correction quadripolar field described, the image of a beam on the screen is a tilted ellipse, whereas in the present application the corresponding image when not corrected is a vertical line.

The known apparatus has some isotropic astigmatism so that the vertical focal lines, i.e. the Meridional focal lines of the horizontal deflection plane and the sagittal focal lines of the vertical deflection plane, coincide with the display screen. Since the imaginary ribbon-shaped beam produced by the three beams together has substantially no dimension in the vertical direction, its image on the screen is a point. In these circumstances the term "isotropic astigmatism" as used herein in actual fact is to be understood to mean that the coefficients which determine the isotropic astigmatism differ from the desired values. Consequently the cross-sectional area on the screen of the imaginary thick beam of circular cross-section in the deflection plane (see FIG. 2 of the said paper in which, however, the three beams are generated in a vertical plane) does not degenerate into a straight line but takes the form of an ellipse the axes of which are parallel to the directions of deflection. Means for correcting such undesirable isotropic astigmatism is described in U.S. patent application Ser. No. 447,564 filed March 4, 1974. In this means a correcting quadripolar field which varies with the square of the strength of either deflection current is generated in the deflection region. However, the axes of said quadripolar field lie substantially along the diagonal between the axes of the deflection directions and the field is generated by separate windings and not by the deflection coil or coils. It should be noted that the apparatus according to the invention also may be subject to this defect which in this case may be corrected in the manner described in the said U.S. patent application. For the sake of simplicity this will be disregarded hereinafter, that is to say the deflection coil will be assumed to have the correct degree of isotropic astigmatism, causing the landing points of the beams on the screen to coincide in one point everywhere but for the abovementioned tolerance error.

In order that the invention may be more readily understood, embodiments thereof will now be described by way of example with reference to the accompanying diagrammatic drawings, in which:

FIG. 1 is a sectional view of a colour television display tube subject to the defect to be corrected,

FIG. 2 shows schematically the ensuring convergence error on the display screen of the tube,

FIGS. 3, 4, 5 and 7 are circuit diagrams of embodiments of correction circuits, and

FIG. 6 is a wave form obtained in the circuit of FIG. 5.

FIG. 1 is a simplified elevation of a cross-section of a colour television display tube 1 taken on the deflection plane at right angles to the axis of the tube in a direction opposite to the direction of propagation of the electron beams, the deflection coils being omitted for simplicity. Three electron beams L, C and R are generated in one plane, the beam C substantially coinciding with the axis of the tube 1 and the beams L and R being located to the left and to the right respectively of the beam C. If the construction of the devices, for example cathodes, which generate the beams and the field distribution within the tube 1 were exactly as desired, the points of intersection of the beams L, C and R with the deflection plane would be a straight line coinciding for example with the X axis, which coincides with the direction of horizontal deflection, the Y axis coinciding with the direction of vertical deflection. However, owing to tolerances the points of intersection lie on a straight line D which is at an angle α to the X axis which it intersects in C.

The paper mentioned in the second paragraph of this application shows that an imaginary thick beam may be considered the cross-section S of which with the plane of deflection is a circle. The line section LCR of FIG. 1 is a diameter of this circle. If the horizontal deflection field has a distribution in which the meridional image plane substantially coincides with the display screen of the tube 1 whilst the vertical deflection field has a distribution in which the sagittal image plane substantially coincides with the screen, and if moreover the deflection errors due to comma and both anisotropic and isotropic astigmatism are substantially equal to zero, all the points on and within the circle S are imaged on vertical line everwhere on the screen. It is supposed that the correct degree of isotropic astigmatism is actually obtained. Otherwise the image of the circle S would be an ellipse the axes of which are parallel to the X and Y axes, i.e. there would be a horizontal convergence error.

In these circumstances the beams L, C and R of FIG. 3 are imaged on the screen 2 of the tube 1 along vertical lines, some of which are shown (in exaggerated form) in FIG. 2, with the exception of the image at the midpoint of the screen, i.e. without deflection, where they coincide. In the ideal case in which the beams L and R of FIG. 1 would lie on the X axis, i.e. with α = 0, in each triplet L', C', R' in FIG. 2 the points L' and R' would coincide with the point C'. Consequently the error angle α results in a vertical convergence error on the screen. In FIG. 1 the beam L lies above the X axis and the beam R beneath the X axis. Because the beams cross within the tube, the points L' and R' in FIG. 2 always lie beneath and above the point C' respectively.

According to the invention a magnetic correction quadripolar field is generated the polar axes of which substantially coincide with the X and Y axes and four lines of force of which are shown in FIG. 1. The quadripolar field does not influence the beam C which is located at the centre of the deflection plane. The beams L and R are subject to forces F _L and F _R respectively which are superposed on the forces exerted by the deflection fields. FIG. 1 shows that as a result the angle α is effectively reduced to substantially zero so that the convergence error of FIG. 2 is cancelled.

Such a quadripolar field is obtainable by causing an additional current, the difference current, to flow through a deflection coil divided in two coil halves in a manner such that the said current is added to the deflection current in one coil half and subtracted from it in the other coil half. FIG. 2 shows that the convergence errors on the left-hand and right-hand halves of the screen 2 have the same sign and that they have the same sign in the upper and lower halves. Hence it is desirable for the value of the difference current to vary substantially as the square of each deflection. Because initially the value and polarity of the angle α are unknown, the current must be adjustable both in amplitude and in polarity. At the middle of the line and field trace intervals the angle must be zero. For this purpose either one or both deflection coils may be used.

Because the images L', C', R' in FIG. 2 are vertical, i.e. are not tilted, the convergence error to be corrected is to be considered as an isotropic astigmatic deflection error. Hence the line-frequency component of the difference current must be a function of horizontal deflection only and its field-frequency component must be a function of vertical deflection only. Thus it is simpler, but not necessary, to cause the line-frequency component of the difference current to flow through the split deflection coil for horizontal deflection and its field-frequency component to flow through the split deflection coil for vertical deflection.

FIG. 3 shows a simple circuit for generating a line-frequency difference current which satisfies the said requirements. A line deflection current generator 3 at one terminal supplies a line-frequency sawtooth current i _H to line deflection coil halves 4' and 4", which in this embodiment are connected in parallel for the current i _H . Adjustable coils 5' and 5" of low inductance are connected in series with the coil halves 4' and 4" respectively. The coils 5' and 5" may be adjusted jointly and oppositely so as to eliminate in knwon manner any asymmetry of the deflection fields generated by coil halves 4' and 4". The ends of the coils 5' and 5" not connected to the coil halves 4' and 4" respectively are connected to one another via a potentiometer 6 the slider on which is connected to the other terminal of the generator 3. The resistance of, for example, 4.7 ohms of the potentiometer 6 is low compared with the impedance of the coil halves 4' and 4" for the line repetition frequency. Thus a sawtooth voltage the polarity and amplitude of which depend upon the position of the slider is produced across the potentiometer 6. As a first approximation this voltage may be considered as being produced by a voltage source of low internal impedance. The coil halves 4' and 4" pass a current which is proportioned to the integral of the voltage across the potentiometer 6 and consequently is the required parabolic correction difference current i _KH . In one coil half it flows in the same direction as the current i _H /2 and in the other coil half it flows in the opposite direction. For this purpose it is required that the position of the slider on the potentiometer 6 should differ from the electric midpoint thereof.

The potentiometer 6 is shunted by the series combination of two resistors 7' and 7" the junction point of which is connected to the anode of a diode 8 the cathode of which is connected to the slider on the potentiometer 6. The diode 8 and the resistors 7' and 7" ensure that the peak of the parabola will be at zero. In actual fact the diode 8 produces a direct current which compensates for the sagging of the parabola, provided that the resistances of the resistors 7' and 7" are equal and have the correct value, for example 8.2 ohms. This direct current also is a difference current and since the diode 8 is connected to the slider on the potentiometer 6 the reversal of its polarity is automatically effected together with that of the parabolic component.

A disadvantage of the circuit of FIG. 3 may be that the obtainable amplitude of the current i _KH is limited because the permissible value of the potentiometer 6 is limited, for a comparatively large value of this potentiometer will increase dissipation and give rise to a linearity error of the deflection current whilst the current i _KH will no longer be parabolic but will also include higher-order components. The amplitude i _KH may be increased without increasing the resistance of the potentiometer 6 by coupling the latter to the remainder of the circuit by means of a transformer. This may be achieved by an autotransformer, as is shown in FIG. 4. Two windings 19' and 19" which are bifilarly wound on the same core and have the polarities shown are connected in series between the ends of the coils 5' and 5" not connected to the coil halves 4' and 4" respectively. The potentiometer 6 is connected between two tappings on the windings 19' and 19" which are symmetrical with respect to the junction point thereof and the potentiometer slider is connected to said junction point via the series combination of the diode 8 and a resistor 7.

In the circuit shown in FIG. 4 the operation of the balancing coils 5' and 5" is not disturbed, provided that the overall inductance value of the windings 19' and 19" between the junction point of the winding 19' and the coil 5' and the junction point between the winding 19" and the coil 5" is small compared with the inductance value of the coil halves 4' and 4" and the coils 5' and 5" measured between the same points. In a practical embodiment of the circuit of FIG. 4 the latter value is 3.55 mH and the former value is 1.25 mH. This means that the effect of the balancing coils 5' and 5" is reduced by only about one third. The tappings on the windings 19' and 19" are provided at the midpoints thereof, the value of the resistor 6 is about 3.3 ohms and that of the resistor 7 about 0.5 ohm. It should be noted that the resistance of the windings 19' and 19" should not be too small, for otherwise the direct component of the difference current would be short-circuited.

FIG. 5 shows a simple circuit for producing a field-frequency difference current in field-frequency deflection coil halves 9' and 9". Since these coil halves are predominantly resistive for the field repetition frequency, the circuits shown in FIGS. 3 and 4 cannot be used. A field deflection current generator 10 supplies a fieldfrequency sawtooth current i _V to coil halves 9' and 9" which are connected in series in this embodiment. The series combination of a diode 11', a potentiometer 12' and a second diode 13' and the series combination of a third diode 11", a potentiometer 12" and a fourth diode 13" are connected in parallel with the series combination of the said coil halves, the said four diodes having the polarities shown in FIG. 5. An isolating resistor 14' is connected between the slider on the potentiometer 12' and the junction point of the coil halves 9' and 9", and an isolating resistor 14" is connected between the slider of the potentiometer 12" and the said junction point, the values of the isolating resistors being high relative to the impedance of the coil halves, for example about 100 ohms.

During one half of the field trace interval the current i _V flows in the direction shown. Diodes 11' and 13' are conducting whereas diodes 11" and 13" are cutoff. Across the potentiometer 12' a sawtooth voltage is produced so that, if the position of the slider of the potentiometer 12' is different from the electric midpoint of the potentiometer, a sawtooth correction difference current i' _KV flows through the coil half 9', for example in a direction opposite to that of the current i _V , whilst the coil half 9" passes a sawtooth correction difference current i" _KV in the same direction as the current i _V , the currents i' _KV and i" _KV being substantially equal. It should be noted that a difference current, in this embodiment i" _KV , flows through a diode, in this embodiment 13', from the cathode to the anode. However, because the elements 9', 9", 11', 12', 13' and 14' form a Wheatstone bridge comprising resistors, the diodes cannot be cut off.

During the other half of the trace interval current i _V flows in the other direction. The diodes 11" and 13" are conducting and the diodes 11' and 13' are cut off. Sawtooth difference currents are produced which are derived from the slider on potentiometer 12". In FIG. 6 the variation of the extreme value i _KVmax of the difference currents is shown as a function of time, T denoting the field trace interval. At the middle of the interval T these currents are zero, because the current i _V and hence the voltage across the potentiometer 12' or 12" respectively are zero. Owing to the voltage drop across the diode the difference currents are zero for a certain time before and after the middle of the interval T. The resulting curves may be regarded as approximate parabolas, for practice has shown that the residual convergence error is negligibly small. Because the difference currents produced are sawtooth currents, the potentiometers 12' and 12" ensure also that the deflection fields generated by the coil halves 9' and 9" are symmetrical. An advantage of the circuit of FIG. 5 is that the adjustments of the upper half and of the lower half are independent of one another, which conduces to clarity. In the embodiment described both potentiometers have a resistance of about 330 ohms.

It will be appreciated that the quadripolar field generated will only be capable of correcting for the vertical convergence error if the angle α is very small. The error introduced by the incorrect position of the line D is compensated for by the quadripolar field according to the invention, it is true, however, at large values of the angle α this field in turn introduces new errors, especially in the corners of the screen. Practice has shown that an angle of from 2° to 3° still can be corrected.

Hereinbefore no statement has been made about the construction of the deflection coils. If they are in the form of saddle coils, no special steps are required. If, however, they are wound toroidally, a step as described in U.S. patent application Ser. No. 390,701 filed August 23, 1973 must be used which consists in the introduction of the difference currents into the deflection coil halves via tappings. In this case the simple circuits of FIGS. 3, 4 and 5 are to be replaced by circuits in which the parabolic difference currents are generated in a different manner, for example by separate generators.

In the embodiments described the coil halves 4' and 4" for horizontal deflection are connected in parallel for the line deflection current i _H , whereas the coil halves 9' and 9" for vertical deflection are connected in series for the field deflection current i _V . Obviously this is not of importance for the invention and the coil halves may be connected in a different manner. FIG. 7 shows an embodiment in which the coil halves 4' and 4" are connected in series for the current i _H . In this embodiment two diodes 8' and 8" are required. It will further be appreciated that the invention may also be applied if the electron beams are generated in a plane of substantially vertical orientation, in which case the convergence error to be corrected is horizontal.

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