GRUNDIG SUPER COLOR 1514 SERIE 8S15 CHASSIS GSC100 (29304-178.90(04)) INTERNAL VIEW.

The GRUNDIG GSC100/GSC200 chassis consists of a large, vertical main panel with a number of modules that take care of various circuit functions. The tubes are of the 90° in -line gun type while the line output stage is of the thyristor type. Unlike earlier Grundig colour chassis that employed a transductor for width/e.h.t. regulation, in this chassis a thyristor driven by a rather unusual circuit (more on this later) is used for the purpose. There are no less than six thyristors dotted around the chassis - e.h.t. regulator (Ty503), line scan and line flyback (Ty508 and Ty501), line generator start-up (Ty607), excess current trip (Ty615) and overvoltage trip (Ty2517).





 

Power Supply:

Fig. 1 shows the basic power supply arrangement (in addition, 200V and 18.6V supplies, fused by Si629 and Si627 respectively, are derived from the line output stage).

 As soon as people see a thyristor they throw up their hands 
in horror!

In normal operation, a supply (+B13.5V) derived from the combi coil is used to power the line generator and e.h.t. control modules. So a start-up supply is required. This is provided by Ty607 whose anode is fed via the fusible resistor R607. The output is limited by the 10V zener diode Di607. Once the line timebase has come into operation, Di511 rectifies pulses developed across a secondary winding on the combi coil, feeding the Darlington series regulator transistor Tr635 via fuse Si511 and the fusible resistor R632. Assuming that there's no fault condition, the regulator circuit produces a 15V supply and Tr608 switches on, shorting the gate of Ty607 to chassis to disable the start- up system.
One of the most common conditions is no results due to R607 having sprung open. In this event, check the following: Ty607 short-circuit; Tr608 open -circuit or low gain; R608 high in value; R633 high or open -circuit; no 311V h.t. supply; no output from the line generator module; no drive to Ty503 from the e.h.t. control module; Si511 or Di511 open -circuit; Ty503 open -circuit; Di636 low voltage. This covers 90 per cent of faults causing R607 to ping.
The cause of R621 in the h.t. supply being open -circuit is usually excess current trip operation due to a line output stage fault !

Excess Current Trip:

 The excess current trip module gives relatively few problems. If Ty615 has gone open -circuit there'll be no h.t. supply of course; if it's gone short-circuit there'll be no protection until R621 pings. The operation of this circuit is as follows. If a fault condition causing an increase of 100 per cent in the h.t. current occurs, the voltage developed across R621 will increase from approximately 9V to 13V plus. As a result zener diode Di619 will conduct, turning on Tr618 to short Ty615's gate -cathode junction so that it switches off. The time -constant of C618/R618 is approxi- mately 120msec, so that the trip "oscillates" until R621 pings. To check the operation of this module, connect a 10kfl resistor from the junction of R619 and D619 to chassis: the module should now oscillate at the trip frequency. EHT Control Module The e.h.t. control module (see Fig. 2) may come as a surprise. TTL in a line timebase! Very useful actually. Here's how it works. IC2511 is a monostable multivibrator which is triggered at pin 5 by pulses from the line generator module. The multivibrator's on time is set by the time -constant network R2514/C2513. It's output at pin 1 is capacitively coupled to Tr2506 which provides a transformer coupled drive to the regulating thyristor Ty503. Pulses from the line output transformer enter the module at pin 9 and are rectified by Di2521/C2522. The resultant supply controls transistor Tr2516 which in turn controls the supply to the monostable's time -constant network, thus providing e.h.t./width regulation. The same line output transformer derived pulses enter the module at pin 8. Under excess voltage conditions Di2517 and in turn Ty2517 conduct, shutting the whole operation down. The usual fault conditions are as follows. C2507 changes value, reducing the drive to Tr2506 which gets hot and dies due to the slower turn-off time. Ty2517 goes short-circuit, with the result that the monostable doesn't trigger. Zener diode Di2502 goes low which upsets things because TTL devices like a supply of 5V or thereabouts. If you can't adjust the set e.h.t. control R2523, change the 9.1V zener diode Di2516. Then set the control midway, reinsert the module, monitor tag b on the line output transformer with an AVO 8 or 9 and adjust R2523 for 49V d.c. This will give correct e.h.t. and width. If you wind the control too far Di2516 will snuff it, so be careful. Line Output Stage Thyristor line output stages are not the easiest circuits for fault finding.
The problem is that it either works or it doesn't, no half ways. A very useful tool is the transistor/ thyristor tester , since this enables you to check the power devices in situ before substitution. If you don't have a tester, the following checks and observations are worth making. R621 and maybe R607 in the power supply will usually have pinged. When resoldered, the trip module will "plop" repeatedly,  proving that excess current is flowing. If disconnecting the anode of the flyback thyristor Ty501 stops the tripping, it's probably short-circuit. If the set continues to trip, replace the scan thyristor Ty508 as it may be open -circuit. Also check the efficiency diode Line out put stage) Di508. Try disconnecting the tripler. Check the continuity of the scan coils, and the scan -correction capacitor C526 (2.3µF).
It's also worth inspecting the solder around R502 (18012, 11W) in the scan thyristor's gate drive circuit - it gets a bit hot and tends to get dry -jointed. I always replaced  devices in this area with exact Grundig  replacements and not other types, though alternatives may be o.k.
 The line output transformer and combi coil don't readily fail (Never changed one !), though I suppose some engineers will have found duff ones.
A useful tool, essential when running up a GSC100 line output stage, is a 2A variac. It saves on fuses and nerves.

Line Generator:

 The only problems I've had with the line generator module concern the TDA2591 chip. If the module fails to oscillate at start up the chip may be faulty - some are a bit funny about the voltage when cold. The Field Timebase The field time base module employs a TDA1170 i.c. There've been some odd faults in this area. C441 (0.22µF) leaky causes poor field sync. Tr467 leaky causes funny field flyback blanking - sometimes almost anywhere dur- ing the field period. The field scan coupling capacitor C473 is on the main panel: when it's leaky or short-circuit the result is field collapse with the line shifting upwards to near the top of the screen.



Audio Module:

There's little to report about the audio module apart from the TBA800 i.c. occasionally dying. IF Module The smoothing capacitor C2321 (10µF) on the i.f. module is a tantalum type and can go short-circuit, R607 eventually pinging. Don't forget that there are separate chroma and luminance outputs, with the chroma signal inverted. If you suspect that the SAWF is out of specification, if have  access to a sweep generator especially designed for use with this module should be used.





The Tuner:
 The tuner used  is of the v.h.f./u.h.f. type, part no. 29500.
For many people, delving into tuners is taboo. If certain ground rules are observed however many common faults can be cleared. Tuning drift or failure to tune is caused by one of the varicap diodes going leaky. These diodes normally have a very high impedance and any leakage at all will cause drift. The best method of tracing this fault is to connect a sensitive d.c. meter, switched to 50µA, in series with the varicap control line, disconnecting each of the diodes in turn until the current returns to zero. Replace these diodes with the exact type - no substitutes.
The r.f. amplifier transistor Tr118 tends to go sick after a thunderstorm. To confirm this, inject a signal via a loop into the output tuning area - some sort of signal should then be evident.

To check that the mixer circuit is operating, use the diode sniffer shown in Fig. 3. Insert the probe near the tuning elements and check for r.f. from ch. 21 to ch. 68. If any component has to be replaced, observe exactly how the original was fitted before removing it. Fit the replacement in the same way, otherwise severe mistuning may occur.



RGB Module:

The RGB module can present difficulties due to the feedback paths.
 If a number of panels need repairing it's worth finding a good one and fitting a 24 -pin i.c. socket so that the TDA2800 i.c. can be proved before making further investigations. Most faults occur in the RGB output stages however.
Here are one or two odd faults: no luminance, C907 (22µF) or the delay line (on the main panel) open -circuit; no luminance and low brightness, C977 (2.2µF) leaky.
There are obviously many internal faults that could occur in the i.c., causing obscure symptoms. Some less common faults I've had on the module are as follows. R1919 open -circuit, no contrast control. Zener diode D1948 open -circuit or L1920 high -resistance, uncontrolla- ble brightness.
Other faults depend on which output stage is involved.
 For the red output stage, R1904 open -circuit causes a tint of that colour on the background and loss of h.f. response; C1912 or C1914 leaky causes no red; R1911 open -circuit results in full beam current, as does T1908 going leaky or short-circuit; T1901 going short- circuit causes no colour. The relevant components in the blue and green output stages give analogoui faults.

Chroma Module:


Many chroma module faults are due to the two i.c.s (TDA2510 and TDA2521). There seem to have been difficulties with the TDA2521 as at least three versions were made, the TDA2521/3 being the latest. If there's no colour, check the colour burst level at pin 7 of IC861. It should be 0.5V peak -to -peak. Under fault conditions it may rise to 2.5V p -p and not be controllable with R827. Check C833 which could be leaky, IC861, and C823/832 which could be open -circuit.
If there's still no colour, check the reference oscillator and its tuning.
As with the RGB module, if you've many panels to look after it's worth fitting i.c. sockets to a known good one for use as an i.c. test bed. If the R -Y or B -Y signals are missing, suspect IC861 and either L854 or L857 for being open - circuit. If R828 has burnt up, check whether C831 is short-circuit. Di881 (12V zener diode) leaky causes green flashing lines while C809 causes weak flashing colours.
As mentioned in a letter (June issue) C843 must be changed to 0.0047µF if you have a colour locking problem with the Sinclair Spectrum microcomputer. this post has shed light on the problems that can be encountered with the GSC100 chassis. Most of the comments also apply to the GSC200 chassis which differs in only minor respects from the GSC100 (vision i.f. module, tuning system and the inclusion of a relay board).

GRUNDIG SUPER COLOR 1514 SERIE 8S15 CHASSIS GSC100 (29304-176.90) Modules - Units:

Horizontal Bst. 29301-008.90 (Horizontal oscillator + Synch sep. with TDA 2593)
Vertikal Bst. 29301-009.90 (Frame deflection with oscillator with TDA1170)
Nf Baustein 29301-004.90 (Audio sound amplifier with IC with TBA 800)
Sicherung Bst 29301-410.01 (Safety Unit)
Ton-ZF-Baustein (IF SOUND) (TBA 120)
Regel Baustein 29301-035.90 (Horizontal thyristor circuit regulator Stabiliser with SN74121N)

MATRIX + RGB AMPLIFIERS (TDA2800)

GRUNDIG SUPER COLOR 1514 SERIE 8S15 CHASSIS GSC100 (29304-176.90) Tuning unit with bandswitch for high frequency receivers:

Abstract:

A tuning unit with a bandswitch for high frequency receivers having a potentiometer system for the control of capacity diodes is disclosed. The potentiometer system includes a plurality of parallelly disposed resistance paths on which wipers can be moved by means of screw tuning spindles mounted beside one another in a common housing made of an insulating material. The bandswitch is formed of metal wires and is associated with each tuning spindle. The tuning spindles are joined for rotation with sleeves simultaneously forming the operating knobs which are carried in apertures in the front plate and each have a flange engaging the back side of the front plate about the apertures. The flange is slightly larger than the cross section of the apertures and tapers conically away from the back side of the front plate.

1. Tuning unit with bandswitch for high frequency receivers having potentiometer means for the control of capacity diodes composed of a plurality of parallelly disposed resistance paths on which wipers are moved by means of screw tuning spindle means mounted beside one another in a common housing of insulating material, bandswitch means formed of metal wires associated with each tuning spindle means, said tuning spindle means being joined for rotation with sleeve means simultaneously forming operating knobs which are borne in apertures in the front plate and each sleeve means having an axial flange surface engaging the back side of the front plate about one aperture therein, said flange surface being slightly larger than the cross section of the apertures and tapering conically away from the back side of the front plate.

2. Tuning unit of claim 1 wherein the sleeve means are joined telescopically and coaxially with the tuning spindle means, and the flange surface engages the back side of the front plate when the sleeve means are in the state wherein they are pulled out of the front plate.

3. Tuning unit of claim 1 wherein the ends of the tuning spindle means which are opposite the front plate have each an annular groove into which a spring bracket engages whose bent end is supported against the housing and which has two diametrically disposed spring arms having opposite spring curvature, the said spring arms in each case contacting the opposite axial walls of the groove.

4. Tuning unit of claim 3 wherein the spring bracket rests with its bent end against the housing and the spring arms additionally engage a bracket formed on the housing or an intermediate bracket formed in one piece with the connection soldering lugs.

5. Tuning unit of claim 3 wherein the spring bracket is formed in one piece with the connection soldering lugs and has spring arms curved both in the same direction which engage an axial wall of the annular groove in the spindle and the opposite axial wall rests against a housing wall.

6. Tuning unit of claim 1 wherein the pointers associated with each potentiometer means lie on the one hand in windows associated with each tuning spindle means in the front plate, and on the other hand are rotatably mounted with their ends opposite the front plate in pivot pins on the housing, and the guiding pin of the spindle nuts carried in a longitudinally displaceable manner on each tuning spindle is provided with a slit disposed parallel to the longitudinal axis of the tuning spindle and slides with its peripheral surface resiliently within the slide tract of the pointer.

7. Tuning unit of claim 1 wherein the bandswitches are formed each of a displaceable metal rod which is in working engagement with stationary metal rods common to all bandswitches of a tuning unit, contacting each of them individually.

8. Tuning unit of claim 7 wherein the metal rods are metal wires.

9. Tuning unit of claim 7 wherein the metal rods are stamped metal parts.

10. Tuning unit of claim 7 wherein levers of insulating material are placed on the front ends of the displaceable metal rods and extend through windows which are provided with detents and which are associated with each tuning spindle in the housing front plate, while the opposite ends are held fixedly in the rearward end of the housing, and the displaceable metal rods individually make contact with contact cams on the stationary metal rods, these cams being in an offset array corresponding to the detents in the windows, the corresponding rods extending parallel to the front plate and parallel to one another behind the front plate.

11. Tuning unit of claim 7 wherein insulating material bridges or insulating material slide pieces are inserted between the contact cams of two adjacent, stationary metal rods and within the free space between two such parallel metal rods.

12. Tuning unit of claim 7 wherein the displaceable metal rods have, in the vicinity of their mountings on the housing, an articulation in the form of a vertically disposed flat portion.

Description:

BACKGROUND
The invention relates to a tuning unit with bandswitch for high frequency receivers, especially radio and television receivers, having a potentiometer system for the control of capacity diodes, the said potentiometer system consisting of a plurality of parallel resistance paths along which wiper contacts can be driven by means of screw spindles disposed adjacent one another in a common insulating material housing in which a bandswitch formed of metal rods is associated with each tuning spindle.

In these tuning units, the working voltages of the capacity diodes in the tuning circuits are recorded once a precise tuning to the desired frequency has been performed. A potentiometer tuning system has great advantages over the formerly used channel selectors operating with mechanically adjustable capacitors (tuning condensers) or mechanically adjustable inductances (variometers), mainly because it is not required to have such great precision in its tuning mechanism.
Tuning units with bandswitches formed of variable resistances and combined with interlocking pushbuttons controlling the supply of recorded working voltages to capacity diodes are known. Channel selection is accomplished by depressing the knobs, and the tuning or fine tuning are performed by turning the knobs. The resistances serving as voltage dividers in these tuning units are combined into a component unit such that they are in the form of a ladderlike pattern on a common insulating plate forming the cover of the housing in which the tuning spindles and wiper contacts corresponding to the variable resistances are housed. The number of resistances corresponds to the number of channels or frequencies which are to be recorded. The wiper contact picks up a voltage which, when applied to the capacity diodes determines their capacitance and hence the frequency of the corresponding oscillating circuit. The adjustment of the wipers is performed by turning the tuning spindle coupled to the tuning knob. By the depression of a button the electrical connection between a contact rod and a tuning spindle is brought about and thus the selected voltage is applied to the capacity diodes. Since the push buttons release one another, it is possible simply by depressing another button to tune to a different receiving frequency or a different channel, as the case may be.
To permit the switching of a number of channels in a certain tuning range, bandswitches for a plurality of tuning ranges, such as UHF and VHF for example, are often provided in the tuning units described above. In the pushbutton tuning unit of the above-named type, the bandswitch consists of a printed circuit board which is fastened on the housing of the tuning unit, and a switch lever which is preset by means of the pushbutton by turning, and is operated by depressing the pushbutton while at the same time selecting the channel.
Where this combination of knobs and pushbuttons is not possible, the selection of the range is accomplished by means of an additional lever which can be set over to select the range.
However, since such tuning units require too many riveting operations when they are assembled, tuning units were later created in which the individual parts in the voltage divider and pushbutton housing were loosely inserted and/or held in place by projections, lugs, hooks or tabs of resilient plastic. In spite of these initial improvements, the bandswitch, especially the one associated with the tuning units, was still technically intricate and very expensive.
THE INVENTION
It is the object of the invention, therefore, to create an additionally improved and simplified tuning unit containing a bandswitch of simple, space-saving and reliably operating design.
In accordance with the invention, this object is accomplished in a tuning unit with bandswitch of the kind described in the beginning by joining the tuning spindles for rotation with sleeves simultaneously forming the control knobs, which are mounted in apertures in the front plate of the housing and have each a flange engaging the back of the front plate around the aperture, the said flange being slightly larger than the aperture and tapering conically away from the back of the front plate.
In further development, the sleeves can be joined telescopically for rotation with the tuning spindles, and the flange is able to engage the back side of the front plate when the sleeve is in the position in which it is drawn out of the front plate. The sleeves constructed in this manner, whose portions projecting from the apertures in the front plate form the control knobs for the tuning spindles, permit easy assembly of the tuning unit and at the same time assure positive co-rotation of sleeves and spindles. The sleeves can be pushed from the front side of the front plate through the apertures onto the clutch surfaces of the spindles, this inward pushing being easily accomplished on account of the taper, and the dropping out of the sleeve being prevented by the flange engaging the back of the front plate. If the control knobs project only slightly out of the front plate, they can be operated from the outside by inserting a tool into them. With the telescoping type of coupling, however, it is possible to draw the sleeves or control knobs further outwardly so that they can be rotated by hand without the use of tools.
To provide constant assurance of the axial fixation of the tuning spindles, the tuning spindle ends farthest from the front plate can each be provided with an annular groove engaged by a spring bracket whose one leg is supported against the housing and whose other leg is forked to form two spring arms, each bent in the opposite direction and each engaging one of the two opposite walls of the annular groove. The tuning spindles are secured against axial displacement by this construction of the invention alone, without the need for further measures. This facilitates the joining of the sleeves or control knobs to the tuning spindle, because in this case there is no need for precise axial fixation and extreme dimensional accuracy.
Furthermore, the indicators associated with each potentiometer can be mounted in windows in the front plate which are associated with each tuning spindle or tuning knob for visual indication at the front, the other extremities farthest from the front plate being mounted for pivoting on pins set in the housing; the guiding pin on the spindle nut that is driven longitudinally on each tuning spindle can be provided with a slit disposed parallel to the long axis of the tuning spindles and can slide within the indicator slide lever slot, with its surface resiliently engaging the walls of said slot.
In an especially advantageous embodiment, the tuning unit can have bandswitches each formed of a displaceable metal rod which is in contacting engagement individually with stationary metal rods which are common to all of the bandswitches of a tuning unit. It contrast to the bandswitches known hitherto, which as a rule consist of a printed circuit board with switchable contacts thereon, this frequency bandswitch of the invention is of great simplicity, can be manufactured simply and inexpensively, and at the same time is very reliable in operation.
The displaceable and stationary metal rods of the bandswitches can be formed of metal wires or they can be of stamped sheet metal. Also, in further expansion of the concept of the invention, the stationary metal rods thus formed can be all entirely alike and merely offset from one another, thereby further simplifying the manufacture and stocking thereof.
To permit connection also to audiovisual apparatus, one or more of the stationary metal rods can be divided electrically into at least two parts each.
In a special development of this concept, lugs of insulating material can be mounted on the front ends of the displaceable metal wires, these lugs extending through windows in the front plate of the housing which are associated with each tuning spindle and are provided with detents, while the opposite ends can be held fixedly at the rear end of the housing, and the displaceable metal wires can make contact with contact humps on the stationary metal wires, the humps being offset from one another to correspond to the detents in the windows, and the stationary metal wires extending in back of the front plate, parallel to the latter and parallel to one another.
To increase switching reliability, bridges or sliding pieces made of insulating material can be inserted between the contact humps of adjacent stationary wires within the free space between two such parallel lying metal wires.
To achieve easy displacement of the displaceable metal wires despite the fixed end mounting on the housing, the displaceable metal wires, in further embodiment of the invention, can have each an articulation adjacent their end mountings, in the form of a vertically disposed flattened portion. This flat permits the metal wires to be deflected horizontally against a weak spring bias.
DESCRIPTION OF THE DRAWING
As an example of the embodiment of the invention, there is represented in the drawings a tuning unit with bandswitch for television receivers. In these drawings,
FIG. 1 is a front elevational view of a tuning unit with bandswitch,
FIG. 2 is a plan view showing the bandswitch of the tuning unit of FIG. 1,
FIG. 3 is a side elevational, cross-sectional view of the tuning unit of FIG. 1,
FIG. 4 is a rear elevational view of the tuning unit of FIG. 1,
FIG. 5 is a plan view showing the indicator means of the tuning unit of FIG. 1,
FIG. 6 shows the sleeve with the operating knob and tuning spindle,
FIG. 7 shows the telescoping manner in which the sleeve is joined to the tuning spindle,
FIG. 8 is a fragmentary view of the bandswitch,
FIG. 9 is another fragmentary view of the bandswitch, and
FIG. 10 shows how the tuning spindle is fixed in position.
DESCRIPTION
The method of representation used in the drawings is greatly simplified, for the purpose of better delineating the features of the invention. The tuning unit with bandswitch consists of an insulating material housing 1 with a front plate 2, which is closed by a cover plate 3 accommodating the resistance paths. The housing 1 is divided by parallel sidewalls 4 into chambers in which the tuning spindles 5 are disposed.
The embodiments is an 8-fold tuning unit having eight bandswitches assocated with each tuning spindle, and eight indicators.
Accordingly, there are eight apertures 6 in a central row, through which the operating knobs 7 of the sleeves 8 coupled with the tuning spindles 5 are passed. The operating knobs 7 have recessed surfaces 9 for turning with a turning tool. In a row extending parallel above the row of the apertures 6 there are eight windows 10, whose upper edge is provided with notches 11. Lugs 12 of insulating material extend through the windows 10 and engage the upper notches 11 and are joined behind the front plate to displaceable metal wires 13 of the bandswitch. In a row located beneath the row of apertures 6 another eight windows 14 are provided, through which the ends of the pointers of the indicators 15 protrude.

Now, the bandswitch consists in each case of a displaceable metal wire 13 which can be brought into working engagement with stationary metal wires 16, which are all of the same construction and are only disposed offset from one another. While the displaceable metal wire 13 extends substantially parallel to the longitudinal axis and thus at right angles to the front plate 2, the stationary, parallelly disposed metal wires 16 are parallel to the front plate 2 and are thus inserted at a right angle to the displaceable metal wire. A departure from parallelism or from the right angle, as the case may be, takes place substantially only when the displaceable metal wire 13 is deflected to the two outer notches. The rearward end 18 of the displaceable metal wire, which forms a vertical loop, is tightly inserted into a receiver 17. Just ahead of the loop 18, the metal wire 13 is provided with a vertically disposed portion 19 by a flattening on the metal wire 13. The movement, when the metal wire 13 is deflected into the desired notches or detents, takes place horizontally by the flexing of these portions 19. The stationary metal wires 16 are held tightly in their positions in projections 20 on the housing, or by lugs or the like. Since three switch actions are provided, that is, three ranges, for each tuning spindle, a bandswitch consists of one displaceable metal wire and three stationary metal wires 16, which are used for all switches.
To permit each bandswitch to have exactly three switching actions, each of the three stationary metal wires 16 has one contact hump 21 corresponding to one of the detents 11 in the windows 10 of the front plate 2. The contact humps 21 are thus located one next to the other as seen from the front plate 2. So that the displaceable metal wire 13 will always come into mechanical and electrical contact only with the desired contact hump, and prevent short circuits, insulating bridges 22 are installed between the adjacent metal wires 16, said insulating bridges being stationary.
If more or less than three switching actions are desired, all that need be done in the case of the bandswitch of the invention is to change the number of stationary metal rods or wires accordingly.
The sleeves 8 with the operating knob 7 have a flange 23 engaging the back of the front plate 2 and tapering back to the point where it joins the tuning spindle. This enables the sleeves to be pushed in, in the case of a housing that has already been manufactured with the tuning spindle installed, without creating the possibility that the sleeves 8 might escape after they have been inserted. The sleeves 8 are connected to the tuning spindles 5 usually by means of driving surfaces. If manual operation without tools is to be possible, rather than requiring a tool for the operation of the sleeves, the coupling of the sleeve 8 to the tuning spindle will be a telescoping coupling (see FIG. 7).
The actual firm axial fixation of the tuning spindle 5 is located on the rear end of the housing. Here the tuning spindle 5 has an annular groove 24 which is engaged by a spring by means of two diametrically disposed spring arms 25 and 26. The spring arms 25 and 26 have oppositely curved lugs and are supported on the housing at their terminal and marginal surfaces and their lugs engage opposite axial walls 27 and 28 of the annular groove 24.
Additional support is provided by the common, bent foot 29 of the spring arms 25 and 26 against the cover plate of the housing.
The indicator means of the tuning unit with bandswitch consists of a pointer 15 which is movable within the window 14, and a cam 30 which is a prolongation of the pointer 15. At its rearward end, the pointer is mounted rotatably in the housing on pin 31. Within the cam 30 slides a guiding pin 32 which is attached to the spindle nut or carriage 40. Upon the rotation of the tuning spindle, the spindle nut is longitudinally displaceable therewith. In order to achieve good guidance and hence precise indication, the guiding pin has a slit 33 extending parallel to the longitudinal axis of the tuning spindle 5, so that it will resiliently engage the cam 30 within the slot thereof.
The necessary soldering lugs are indicated at 34.
On the basis of the design of the tuning unit with bandswitch in accordance with the invention, a desired frequency range--UHF, for example--can be selected by deflecting a displaceable metal wire 13 into one of the detents 11 by means of the lug 12 mounted thereon. Within this range, a transmitter or channel can then be selected by turning the tuning spindle 5. The transmitter preselected in this manner can then be tuned in by means of a keyboard or by electronic recall from a keyboard which is not shown. The fine tuning of this tuned-in transmitter, as well as the selection of a different transmitter within the same frequency range, is accomplished by turning the tuning spindle 5.
All of the details explained in the above description and represented in the drawings are important to the invention.

GRUNDIG SUPER COLOR 1514 SERIE 8S15 CHASSIS GSC100 (29304-176.90) Automatic peak beam current limiter:In a video signal processing system, apparatus for limiting excessive peak and average beam currents demanded by a kinescope in response to image representative video signals. A control signal derived by a sensing circuit in response to excessive beam current is applied to the kinescope in a manner to limit excessive beam currents above a threshold level. The sensing circuit includes a capacitor arranged with a conductive diode to form an average responding filter such that the sensing circuit derives a control voltage representative of excessive average beam current in a first beam current limiting mode. In a second beam limiting mode, when beam current exceeds a predetermined level, the diode is rendered nonconductive and the capacitor is decoupled for average detection purposes. In this mode, the control signal is free to vary in accordance with rapid variations in peak beam current.

1. In a video signal processing system including a kinescope for reproducing an image in response to video signals, apparatus for limiting excessive kinescope current conduction comprising:
means for deriving a control signal indicative of variations in the magnitude of kinescope current above a selected threshold level; and
means for utilizing said control signal to limit kinescope current in accordance with the magnitude of said control signal; and wherein
said control signal deriving means includes filter means subject to alternative operation in a first state and a second state; said filter means, when in said first state, exhibiting a restricted pass band such that said control signal is indicative of only relatively long term variations in the magnitude of said kinescope current; said filter means, when in said second state, exhibiting a pass band significantly wider than said restricted pass band such that said control signal additionally follows relatively short term variations in said kinetic current; and said filter means being subject to switching from said first state to said second state when said kinescope current exceeds a predetermined level.
2. Apparatus according to claim 1, wherein said signal deriving means additionally comprises a controlled conduction device exhibiting first and second conductive states in the presence of excess kinescope current exhibiting said relatively long and short term variations, respectively. 3. Apparatus according to claim 2, wherein:
said control signal is developed across the series combination of a filter capacitor and said controlled conduction device, said controlled conduction device exhibiting a high impedance in said second conductive state.
4. Apparatus according to claim 2 and further comprising a source of supply current representative of the magnitude of current conducted by said kinescope, wherein
said signal deriving means is coupled to said current source for sensing the level of said supply current to thereby derive said control signal when said supply current exceeds the threshold level; and
a source of reference current is coupled to said controlled device for maintaining said device in said first conductive state, said device exhibiting said second conductive state when said supply current exhibits said relatively short term variations and exceeds a predetermined level.
5. Apparatus according to claim 4, wherein
the magnitude of said reference current exceeds said threshold level;
said controlled device exhibits said first conductive state when the magnitude of supply current exhibits said long term variations and exceeds the threshold level; and
said controlled device exhibits said second conductive state when the magnitude of supply current exhibits said short term variations and exceeds the magnitude of said reference current by a predetermined amount.
6. Apparatus according to claim 4 and further comprising high voltage supply means for providing an operating supply for said kinescope; and wherein
said current source is coupled to an input of said high voltage means and said supply current corresponds to the current drawn by said kinescope from said high voltage means in accordance with the level of kinescope current conduction.
7. Apparatus according to claim 6, wherein:
said current source provides a given current with a magnitude corresponding to said threshold level, said last-named current being supplied to said high voltage means as said supply current in accordance with the level of kinescope current conduction; and
said controlled device exhibits said second conductive state when said supply current exceeds the sum of said reference current and said given current.
8. Apparatus according to claim 7, wherein: the level of said reference current is substantially equal to or less than the level of said given current. 9. Apparatus according to claim 4, wherein:
said controlled device comprises a semiconductor PN junction device coupled between a first point and a point of reference potential, said semiconductor being rendered conductive in response to said reference current; and
said filter means comprises a capacitor coupled to said first point and to said current source at a second point remote from said first point, for sensing said supply current.
10. Apparatus according to claim 9, wherein:
said semiconductor device is rendered non-conductive and said capacitor is decoupled from said point of reference potential when supply current exhibiting said relatively short term variations exceeds the magnitude of the said reference current by a predetermined amount.
11. Apparatus according to claim 1 and further comprising:
a channel for processing said video signals;
means for coupling video signals processed by said channel to said kinescope; and wherein
said control signal is coupled to said channel for translating the video signal in a direction to limit excessive kinescope current conduction in accordance with the magnitude of said control signal.
12. Apparatus according to claim 11, wherein:
said control signal is utilized to vary the D.C. level of signals processed by said channel in a direction to limit excessive kinescope current conduction.
13. In a television signal processing system including a kinescope for reproducing an image in response to processed television signals, apparatus for limiting excessive kinescope beam currents exhibiting relatively long term and short term variations, comprising:
a source of supply current representative of the magnitude of beam current demanded by said kinescope;
means including a filter capacitor, coupled to said current source for sensing said supply current to thereby derive a control signal representative of the magnitude of excessive beam current above a given threshold level as manifested by the level of said supply current;
means for coupling said control signal to said kinescope for limiting beam current above the threshold level in accordance with the magnitude of said control signal; and wherein
said signal deriving means additionally includes:
a threshold conducting device coupled between a terminal of said capacitor remote from said current source and a point of reference potential; and
a source of reference current for biasing said device for conduction between said capacitor terminal and said point of reference potential, the level of said reference current being in predetermined relationship with said threshold current level.
14. Apparatus according to claim 13, wherein:
said control signal is utilized to vary a luminance signal component of said television signal in a direction to limit kinescope beam current conduction above the threshold level; and
said threshold device comprises a semiconductor PN junction device poled for forward current conduction from said capacitor terminal to said reference potential.

Description:

This invention relates to apparatus for automatically limiting excessive beam currents drawn by an image reproducing device in a video signal processing system such as a television receiver. In particular, the invention relates to such apparatus capable of limiting short duration peak beam currents as well as average beam currents of relatively longer duration.
Excessive peak or average beam currents can cause a television receiver to produce a degraded image. In this regard, excess beam currents can cause degradation of the performance of the receiver deflection system which is operatively associated with an image reproducing kinescope, electron beam spot defocussing, and picture blooming. High beam currents can also exceed the safe operating current capability of the kinescope, possibly damaging the kinescope and associated circuit components which may be sensitive to high beam current levels.
Various automatic beam current limiter circuits responsive to average beam current levels are known. These circuits typically respond to excessive beam current levels occurring at a rate not exceeding the vertical image scanning rate of the kinescope. The average responding circuits essentially ignore peak increases in beam current levels occurring only for a few horizontal image lines of a vertical scanning interval, for example. Because of the inherent insensitivity of average responding circuits to transient or peak increases in beam current of less than average duration (e.g., less than a vertical image scanning period), a special problem exists for video signal processing systems wherein it is desired to limit such peak currents for a variety of reasons. Excessive peak beam current levels can be attributable to the information content of a received image-representative video signal, such as a signal representative of black-to-white image transitions occurring in one or more succeeding vertical image scanning intervals. Excessive peak beam currents can also occur as a result of transients produced when switching from one channel of the receiver to another. In any case, excessive peak beam current levels can adversely affect receiver circuits (e.g., deflection circuits) which may be sensitive to high levels of peak beam current, even if the maximum allowable average beam current level has not been exceeded.
Systems which provide for limiting both average and peak beam currents are known. For example, U.S. Pat. No. 3,980,822 (Suzuki et al.) discloses an arrangement wherein excess average beam currents are sensed and limited by means including a first limiter circuit with a first time constant. Peak or transient beam currents are sensed and limited by a second limiter circuit, distinct from the first circuit, exhibiting a short time constant relative to the first time constant. U.S. Pat. No. 4,017,681 (Smeulers et al.) also discloses an arrangement including a circuit for detecting excess peak beam currents, and a separate circuit for detecting excess average beam currents.
An analogous arrangement is disclosed in U.S. Pat. No. 3,914,545 (Engel). This patent describes a system wherein a derived control signal representative of the average level of a luminance signal varies the gain of a luminance signal amplifier inversely with changes in the average level. The control signal is modified by a peak limiter circuit whenever the instantaneous luminance signal exceeds a threshold level. The control signal is further modified by a signal from a beam current limiter network responsive to high levels of average beam current.
Because of the nature of a peak or transient excess beam current condition, a peak beam current limiter should exhibit a rapid response in order to provide appropriate compensation. It is also desirable in many instances for an automatic beam current limiter to be capable of limiting both excessive peak and average beam currents. Particularly in this instance, complex, uneconomical circuit arrangements should be avoided whenever possible.
Apparatus according to the present invention is included in a video signal processing system having a kinescope for reproducing an image in response to video signals, for limiting excessive kinescope current conduction. The apparatus includes a sensing circuit for deriving a control signal indicative of variations in the magnitude of kinescope current above a selected threshold level. The control signal is utilized to limit kinescope current in accordance with the magnitude of the control signal. The sensing circuit includes a filter network subject to alternative operation in first and second states. When in the first operating state, the filter exhibits a restricted pass band such that the control signal is indicative of only relatively long term variations in the magnitude of the kinescope current. When in the second operating state, the filter exhibits a significantly wider pass band such that the control signal additionally follows relatively short term variations in kinescope current. The filter network is subject to switching from the first to the second state when the kinescope current exceeds a predetermined level.
The single FIGURE of the drawing shows, partially in block diagram form and partially in schematic circuit diagram form, a general arrangement of a color television receiver employing apparatus constructed in accordance with the present invention.
The drawing depicts a color television receiver including a video signal processing unit 12 for receiving radio frequency signals from an antenna 10 and for translating these signals through intermediate frequency amplifying and detecting stages (not shown) to provide a composite video signal. The composite video signal contains luminance, chrominance, sound and synchronizing components.
A sync separator 15 serves to separate the synchronizing (sync) component from the composite video signal to provide periodic line sync pulses. These pulses are further processed by sync processing and deflection circuits 16 to provide horizontal flyback signals, and horizontal and vertical blanking and deflection signals as known.
A frequency selection unit 21 (e.g., a bandpass filter) selectively couples the chrominance component of the composite video signal to chrominance signal processing unit 24 (e.g., including amplifier and demodulator stages) to derive R-Y, B-Y and G-Y color difference signals. These signals are applied as inputs to a kinescope driver stage 60.
The luminance component of the composite video signal is amplified and otherwise processed by a luminance signal processing unit 35 in a luminance channel of the receiver. Luminance processing unit 35 includes a luminance signal clamping circuit for providing a clamped luminance output signal Y, as disclosed in a copending U.S. patent application Ser. No. 819,935 of R. P. Parker, now U.S. Pat. No. 4,110,787 entitled "Combined Blanking Level And Kinescope Bias Clamp For A Television Signal Processing System", assigned to the same assignee as the present invention. The periodic operation of the clamping circuit is controlled in response to periodic blanking pulses supplied during each image retrace blanking interval by a source of blanking pulses 54. Periodic auxiliary blanking pulses of predetermined magnitude supplied by an auxiliary blanking unit 45 are added to the luminance signal prior to clamping during each blanking interval. This and other aspects of unit 35 are described in greater detail in the last mentioned U.S. patent application.
The clamped luminance signal Y is supplied to an input of kinescope driver 60, where the luminance signal is combined with the color difference signals from unit 24 to form R, B and G color signals. These signals are then coupled to signal inputs (e.g., cathode electrodes) of a kinescope 66 for reproducing a color image.
High operating voltages for focus (not shown) and ultor electrodes of kinescope 66 are provided by a high voltage supply 68 (e.g., voltage tripler) in response to positive, periodic horizontal flyback pulses occurring during horizontal retrace scanning intervals. A current supply including a source of positive direct voltage (+27 volts) and a current determining resistor 72 provides a current I _S and is coupled to a D.C. input of high voltage unit 68 via a resistor 73. Current flowing in resistor 72 includes a component I _R representative of the beam current (i.e., ultor current) demand of the kinescope in response to the luminance and chrominance signals. This current flows into the D.C. input of high voltage unit 68 and is sometimes referred to as a "resupply" current (i.e., a current via the high voltage unit to recharge or resupply the ultor electrode voltage of the kinescope when depleted as a result of beam current conduction). The described current supply is typically associated with the high voltage supply in a television receiver for purposes of providing the resupply current. The resupply current typically consists of current pulses recurring at the horizontal line scanning rate. Some A.C. filtering of horizontal rate voltages which these current pulses tend to produce at the D.C. input terminal of high voltage unit 68 is provided by a filter capacitor 74.
Excessive levels of peak and average beam currents are sensed by a circuit 70. Sensing network 70 is operatively associated with the supply current source including resistor 72, and comprises a large value, average responding filter capacitor 75 and a normally conductive clamp diode 78. Network 70 also includes a normally conductive diode 77 for conductively coupling the negative plate of filter capacitor 75 to a point of reference potential (ground) when beam current demand does not exceed a predetermined level under normal operating conditions of the receiver, and also when a condition of excessive average beam current demand exists, as will be discussed. Diode 77 is forward biased into conduction by a reference current I _B of predetermined value, as supplied by a current source including a resistor 76 and a source of positive direct voltage (+27 volts).
A voltage representative of the level of resupply current (i.e., ultor current) is developed on the positive terminal of capacitor 75 when the resupply current exceeds a predetermined threshold level indicating the presence of excessive peak or average beam current demand, as will be explained. This voltage is supplied to an input of a gated automatic beam limiter (ABL) control network 90, which can be of the type disclosed in a copending, concurrently filed U.S. patent application of R. P. Parker, entitled "Gated Automatic Beam Current Limiter In A Video Signal Processing System", and assigned to the present assignee. Unit 90 then develops an output control signal in accordance with the magnitude of the excessive peak or average beam currents. This control signal is applied to luminance processor 35 in such a manner as to translate the luminance signal in a direction to limit excess beam currents.
In accordance with this invention, the otherwise average responding operation of sensing circuit 70 is modified in the presence of high levels of peak or momentary beam current demand, as manifested by resupply current I _R . This is accomplished by the coaction of filter capacitor 75 with diode 77 and reference bias current I _B .
During normal operating conditions, diode 78 clamps the voltage at the positive terminal of capacitor 75 to +11 volts plus the voltage drop across diode 78 (approximately 0.6 volts), or +11.6 volts. A portion of current I _S flows through clamp diode 78 when conducting. The control signal output from ABL network 90 is inhibited during this time, whereby the luminance signal is processed in normal fashion by unit 35.
The value of resistor 72 and the voltage drop thereacross (15.4 volts) determine a normal level of current I _S (0.7 milliamperes), which corresponds to a first threshold current level of beam limiter operation. This current divides between diode 78 when conducting and the D.C. input of high voltage source 68 in accordance with the beam current demand of kinescope 66 as manifested by the level of resupply current I _R .
Sensing circuit 70 exhibits dual mode operation for sensing both excessive average and peak beam current demand. Diode 77 and reference current I _B are specifically employed for this purpose. The threshold level at which peak beam currents are sensed and limited is determined by the magnitude of current I _B (approximately 2.25 milliamperes).
When the average resupply current exceeds the first threshold level determined by current I _S , current drive for clamp diode 78 is depleted and diode 78 ceases conducting. Since the positive terminal of capacitor 75 is no longer clamped by diode 78, the voltage at this terminal decreases to a less positive level at a rate determined by the amount by which the first threshold current level is exceeded. The ABL control network 90 responds to this less positive voltage by providing a corresponding output control signal which serves, for example, to translate the D.C. level of the luminance signal in a direction to cause kinescope 68 to conduct proportionally less average beam current. Under this condition, diode 77 remains conductive and current I _B flows to ground through diode 77, thereby preserving the role of capacitor 75 as a low pass, average responding filter.
The role of capacitor 75 as an average responding filter is altered when kinescope 66 momentarily demands high peak levels of resupply current. Specifically, as in the case of excessive average current demand, a sudden large increase in resupply current I _R in response to beam current demand causes the positive terminal of capacitor 75 to become less positive when the first threshold current level is exceeded.
Diode 77 ceases conducting since the forward bias current for diode 77 otherwise provided by current I _B now flows through capacitor 75 instead of through diode 77 to ground, in accordance with the rate of change of capacitor 75 voltage and the magnitude of the peak resupply current demand. This effect is produced since the current through a capacitor (e.g., capacitor 75) is determined by the product of the value of the capacitor and the rate of change of voltage across the capacitor. Thus for a given value of capacitance, the capacitor current increases as the rate of change of capacitor voltage increases. In this instance, the rapid rate of change of the voltage across capacitor 75 is produced in response to the sudden increase in peak resupply current. This rapid rate of voltage change is essentially unaffected by the small value of horizontal rate filter capacitor 74.
With capacitor 75 being decoupled in the presence of a rapid increase in peak resupply current demand as described, the current I _B then flowing through capacitor 75 thereafter flows primarily in the resupply current path to the resupply current input of high voltage supply 68. The beam current representative control voltage appearing at the positive terminal of capacitor 75 decreases rapidly in accordance with any further rapid increase in the level of resupply current I _R , since capacitor 75 no longer acts as a low pass, average responding filter during this condition. The described action permits the beam limiter control circuitry to respond quickly (i.e., track closely) and limit rapid or momentary increases in beam current demand, greater than the vertical scanning rate, in accordance with the magnitude of the control signal appearing at the positive terminal of capacitor 75.
It is noted that the level at which peak beam current limiting commences can be adjusted to suit the requirements of a particular system by tailoring the value of reference bias current I _B . Specifically, peak beam current limiting can be activated sooner by reducing the value of reference current I _B . When this rapidly increasing resupply current exhibits a magnitude equal to the sum of reference current I _B and supply current I _S , the current in capacitor 75 equals current I _B . Therefore, diode 77 ceases conducting and capacitor 75 is decoupled from ground. The sum of currents flowing through resistor 76 and resistor 72 then increases in accordance with the amount by which current I _R increases above the sum of currents I _S and I _B .
While the invention has been described in terms of a preferred embodiment, it should be recognized that various modifications can be made by persons skilled in the art without departing from the scope of the invention. Component values and other examples of operating parameters have been mentioned as an aid to understanding the invention and are not intended to be limiting.

TBA120T (Siemens) SIF (Sound IF)

















TDA1170 vertical deflection FRAME DEFLECTION INTEGRATED CIRCUITVertikal Bst. 29301-009.05 (Frame deflection with oscillator with TDA1170)

GENERAL DESCRIPTION
The TDA1170 and TDA1270 are monolithic integrated
circuits designed for use in TV vertical deflection systems. They are manufactured using
the Fairchild Planar* process.
Both devices are supplied in the 12-pin plastic power package with the heat sink fins bent
for insertion into the printed circuit board.
The TDA1170 is designed primarily for large and small screen black and white TV
receivers and industrial TV monitors. The TDA1270 is designed primarily for driving
complementary vertical deflection output stages in color TV receivers and industrial
monitors.
APPLICATION INFORMATION (TDA1170)
The vertical oscillator is directly synchronized by the sync pulses (positive or negative); therefore its free
running frequency must be lower than the sync frequency. The use of current feedback causes the yoke
current to be independent of yoke resistance variations due to thermal effects, Therefore no thermistor is
required in series with the yoke. The flyback generator applies a voltage, about twice the supply voltage, to
the yoke. This produces a short flyback time together with a high useful power to dissipated power
ratio.



TDA2521 synchronous demodulator for PAL

GENERAL DESCRIPTION
The TDA2521 is a monolithic integrated circuit designed as a synchronous demodulator for PAL color television receivers. It includes an 8.8 MHz oscillator and divider, to generate two 4.4 MHz reference signals, and provides color difference output.
The TDA2521 is intended to interface directly with the TDA251O with a minimum of external components and is constructed on a single silicon chip using the Fairchild Planar
epitaxial process.

ABSOLUTE MAXIMUM RATINGS
Supply Voltage 14 V
Internal Power Dissipation 600 mW ORDER INFQRMATIQN
Operating Temperature Range —2O°C to +6O°C TYPE PART NO.
Storage Temperature Range —55°C to +125°C 2521 TDA2521
Pin Temperature iSo|dering 10 si 260°C

Planar is a patented Fairchild process














TDA2510 CHROMINANCE COMBINATION

GENERAL DESCRIPTION —
The TDA2510 is a monolithic integrated circuit designed for the function of a color television receiver. It Is designed to Interface directly with the TDA2521, using a minimum number of external components.
TDA251O is constructed on a single silicon chip using the Fairchild Planar‘ epitaxial process.

ABSOLUTE MAXIMUM RATINGS

supply Voltage 15 V
Collector voltage of chroma output transistor (pin 7) 20 V
(PD I 100 mW max)
Collector current of chroma output transistor (pin 7) 20 mA
Collector current of color killer output transistor (pin 11) 10 mA
Power dissipation 500 mW
Operating temperature range —25°C 10 +6O°
Storage temperature range *55°C to +12!-3°C
GRUNDIG SUPER COLOR 1514 SERIE 8S15 CHASSIS GSC100 (29304-176.90) REGEL BAUSTEIN 29301-035.03 (Horizontal thyristor circuit regulator Stabiliser with SN74121N) LINE DEFL. REGULATION UNIT WITH SN74LS221N

The ’221 and ’LS221 devices are dual
multivibrators with performance characteristics
virtually identical to those of the ’121 devices.
Each multivibrator features a negative-transitiontriggered
input and a positive-transition-triggered
input, either of which can be used as an inhibit
input.

Pulse triggering occurs at a particular voltage level and is not directly related to the transition time of the input
pulse. Schmitt-trigger input circuitry (TTL hysteresis) for B input allows jitter-free triggering from inputs with
transition at rates as slow as 1 V/s, providing the circuit with excellent noise immunity, typically of 1.2 V. A high
immunity to VCC noise, typically of 1.5 V, also is provided by internal latching circuitry.
Once fired, the outputs are independent of further transitions of the A and B inputs and are a function of the timing
components, or the output pulses can be terminated by the overriding clear. Input pulses can be of any duration
relative to the output pulse. Output pulse length can be varied from 35 ns to the maximum by choosing
appropriate timing components. With Rext = 2 kΩ and Cext = 0, an output pulse typically of 30 ns is achieved
that can be used as a dc-triggered reset signal. Output rise and fall times are TTL compatible and independent
of pulse length. Typical triggering and clearing sequences are shown as a part of the switching characteristics
waveforms.
Pulse-width stability is achieved through internal compensation and is virtually independent of VCC and
temperature. In most applications, pulse stability is limited only by the accuracy of external timing components.
Jitter-free operation is maintained over the full temperature and VCC ranges for more than six decades of timing
capacitance (10 pF to 10 µF) and more than one decade of timing resistance (2 kΩ to 30 kΩ for the SN54221,
2 kΩ to 40 kΩ for the SN74221, 2 kΩ to 70 kΩ for the SN54LS221, and 2 kΩ to 100 kΩ for the SN74LS221).
Throughout these ranges, pulse width is defined by the relationship: tw(out) = CextRext In2 ≈ 0.7 CextRext. In
circuits where pulse cutoff is not critical, timing capacitance up to 1000 µF and timing resistance as low as 1.4 kΩ
can be used. Also, the range of jitter-free output pulse widths is extended if VCC is held to 5 V and free-air
temperature is 25°C. Duty cycles as high as 90% are achieved when using maximum recommended RT. Higher
duty cycles are available if a certain amount of pulse-width jitter is allowed.
The variance in output pulse width from device to device typically is less than ±0.5% for given external timing
components. An example of this distribution for the ’221 is shown in Figure 3. Variations in output pulse width
versus supply voltage and temperature for the ’221 are shown in Figures 4 and 5, respectively.
Pin assignments for these devices are identical to those of the SN54123/SN74123 or SN54LS123/SN74LS123
so that the ’221 or ’LS221 devices can be substituted for those products in systems not using the retrigger by
merely changing the value of Rext and/or Cext; however, the polarity of the capacitor must be changed.


FUNCTION TABLE
(each monostable multivibrator)
INPUTS OUTPUTS
CLR A B Q Q
L X X L H
X H X L H
X X L L H
H L ↑ † †
H ↓ H † †
↑‡ L H † †
† Pulsed-output patterns are tested during
AC switching at 25°C with Rext = 2 kΩ, and
Cext = 80 pF.
‡ This condition is true only if the output of
the latch formed by the two NAND gates
has been conditioned to the logic 1 state
prior to CLR going high. This latch is
conditioned by taking either A high or
B low while CLR is inactive (high).

absolute maximum ratings over operating free-air temperature range (unless otherwise noted)†
Supply voltage range, VCC 7 V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Input voltage range, VI (see Note 1): ’LS221 7 V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
’221 5.5 V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Package thermal impedance, θJA (see Note 2): D package 73°C/W . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
DB package 82°C/W . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
N package 67°C/W . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
NS package 64°C/W . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Storage temperature range, Tstg −65°C to 150°C.


TDA2591 SYNCHRO AND HORIZONTAL DEFLECTION CONTROL FOR COLOR TV SET

DESCRIPTION
The TDA2591 is a circuit intended for the horizontal
deflection of color TVsets, supplied with transistors
or SCR’S.

The TDA2591 and TDA2593 are integrated line
oscillator ‘_circuits for colour television receivers using
thyristor or transistor line deflection output stages.
The _circuits incorporate a line oscillator ‘which is
based on the threshold switching principle, a line de-
flection output stage capable of direct drive of thyristor
deflection circuits, phase comparison between the
oscillator voltage and both the sync pulse and line
flyback pulse. Also included on the chip is a switch for
changing the filter characteristic and the gate circuit
when used for VCR.
The TDA2593 generates a sandcastle pulse (at pin
7) suitable for use with the TDA.2532.

.LINE OSCILLATOR(two levels switching)
.PHASE COMPARISON BETWEEN SYNCHRO-
PULSE AND OSCILLATOR VOLTAGE Ø 1, ENABLED BY AN INTERNAL PULSE,
(better parasitic immunity)
PHASE COMPARISON BETWEEN THE FLYBACK
PULSES AND THE OSCILLATOR VOLTAGE Ø2
.COINCIDENCE DETECTOR PROVIDING A LARGE HOLD-IN-RANGE.
.FILTER CHARACTERISTICS AND GATE SWITCHING FOR VIDEO RECORDER APPLICATION.
.NOISE GATED SYNCHRO SEPARATOR
.FRAME PULSE SEPARATOR .BLANKING AND SAND CASTLE OUTPUT PULSES
.HORIZONTAL POWER STAGE PHASE LAGGING CIRCUIT
.SWITCHING OF CONTROL OUTPUT PULSE WIDTH
.SEPARATED SUPPLY VOLTAGE OUTPUT STAGE ALLOWING DIRECT DRIVE OF SCR’S CIRCUIT

.SECURITY CIRCUIT MAKES THE OUTPUT PULSE SUPPRESSED WHEN LOW SUPPLY
VOLTAGE.

Chassis GSC200 , Horizontal Thyristor deflection regulator Unit, (Regel Baustein 29301.035.03)
MATRIX + RGB amplifiers (with TDA 2800 , Philips)
View of Horizontal Thyristor deflection section including
Combined Commutating Coil - Transductor , Line + Eht transformer switching capacitor, various coils and bobbin.
Heatsink for Thyristors (Trace - return - regulation) Hinlauf - Ruecklauf - Regel Thyristor
( return/Ruecklauf thyristor RCA 17053 or BST CC 0146 R)
(Trace/Hinlauf thyristor RCA 17052 or BST CC 0143 H 21)
(Regulation thyristor / Regel Thyristor RCA 17127

GRUNDIG SUPER COLOR 1514 SERIE 8S15 CHASSIS GSC100 (29304-176.90) Electron beam deflection circuit including thyristors Further Discussion and deepening of knowledge, Thyristor horizontal output circuits:

1. An electron beam deflection circuit for a cathode ray tube with electromagnetic deflection by means of a sawtooth current waveform having a trace portion and a retrace portion, said circuit comprising: a deflection winding; a first source of electrical energy formed by a first capacitor; first controllable switching means comprising a parallel combination of a first thyristor and a first diode connected together to conduct in opposite directions, for connecting said winding to said first source during said trace portion, while said first switching means is turned on; a second source of electrical energy including a first inductive energy storage means coupled to a voltage supply; reactive circuit means including a combination of inductive and capacitive reactances for storing the energy supplied by said second source; second controllable switching means, substantially similar to said first one, for completing a circuit including said reactive circuit means and said first switching means, when turned on before the end of said trace portion, so as to pass through said first switching means an oscillatory current in opposite direction to that which passes through said first thyristor from said first source and to turn said first thyristor off after these two currents cancel out, the oscillatory current flowing thereafter through said first diode for an interval termed the circuit turn-off time, which has to be greater than the turn-off time of said first thyristor; wherein the improvement comprises: means for drawing, during at least a part of said trace portion, a substantial amount of additional current through said first switching means, in the direction of conduction of said first diode, whereby to perceptibly shift the waveform of the current flowing through said first switching means towards the negative values by an amount equal to that of said substantial additional current and to lengthen, in proportion thereto, said circuit turn-off time, without altering the values of the reactances in the reactive circuit which intervene in the determination of both the circuit turn-off and retrace portion time intervals.

2. A deflection circuit as claimed in claim 1, wherein said amount of additional current is greater than or equal to 5 per cent of the peak-to-peak value of the current flowing through the deflection winding.

3. A deflection circuit as claimed in claim 1, wherein said means for drawing a substantial amount of additional current through said first switching means comprises a resistor connected in parallel to said first capacitor.

4. A deflection circuit as claimed in claim 1, wherein said means for drawing an additional current is formed by connecting said first and second energy sources in series so that the current charging said reactive circuit means forms the said additional current.

5. A deflection circuit as claimed in claim 1, further including a series combination of an autotransformer winding and a second high-value capacitor, said combination being connected in parallel to said first switching means, wherein said autotransformer comprises an intermediate tap located between its terminals respectively connected to said first switching means and to said second capacitor, said tap delivering, during said trace portion, a suitable DC supply voltage lower than the voltage across said second capacitor; and wherein said means for drawing a substantial amount of additional current comprises a load to be fed by said supply voltage and having one terminal connected to ground; and further controllable switching means controlled to conduct during at least part of said trace portion and to remain cut off during said retrace portion, said further switching means being connected between said tap and the other terminal of said load.

Description:

The present invention relates to electron beam deflection circuits including thyristors, such as silicon controlled rectifiers and relates, in particular, to horizontal deflection circuits for television receivers.

The present invention constitutes an improvement in the circuit described in U.S. Pat. No. 3,449,623 filed on Sept. 6, 1966, this circuit being described in greater detail below with reference to FIGS. 1 and 2 of the accompanying drawings. A deflection circuit of this type comprises a first thyristor switch which allows the conenction of the horizontal deflection winding to a constant voltage source during the time interval used for the transmisstion of the picture signal and for applying this signal to the grid of the cathode ray tube (this interval will be termed the "trace portion" of the scan), and a second thyristor switch which provides the forced commutation of the first one by applying to it a reverse current of equal amplitude to that which passes through it from the said voltage source and thus to initiate the retrace during the horizontal blanking interval.

A undirectional reverse blocking triode type thyristor or silicon controlled rectifier (SCR), such as that used in the aformentioned circuit, requires a certain turn-off time between the instant at which the anode current ceases and the instant at which a positive bias may be applied to it without turning it on, due to the fact that there is still a high concentration of free carriers in the vicinity of the middle junction, this concentration being reduced by a process of recombination independently from the reverse polarity applied to the thyristor. This turn-off time of the thyristor is a function of a number of parameters such as the junction temperature, the DC current level, the decay time of the direct current, the peak level of the reverse current applied, the amplitude of the reverse anode to cathode voltage, the external impedance of the gate electrode, and so on, certain of these varying considerably from one thyristor to another.

In horizontal deflection circuits for television receivers, the flyback or retrace time is limited to approximately 20 percent of the horizontal scan period, the retrace time being in the case of the CCIR standard of 625 lines, approximately 12 microseconds and, in the case of the French standard of 819 lines, approximately 9 microseconds. During this relatively short interval, the thyristor has to be rendered non-conducting and the electron beam has to be returned to the origin of the scan. The first thyristor is blocked by means of a series resonant LC circuit which is subject to a certain number of restrictions (limitations as to the component values employed) due to the fact that, inter alia, it simultaneously determines the turn-off time of the circuit which blocks the thyristor and it forms part of the series resonant circuit which is to carry out the retrace. To obtain proper operation of the deflection circuit of the aforementioned Patent, especially when used for the French standard of 819 lines per image, the values of the components used have to subject to very close tolerances (approximately 2%), which results in high costs.

The improved deflection circuit, object of the present invention, allows the lengthening of the turn-off time of the circuit for turning the scan thyristor off, without altering the values of the LC circuit, which are determined by other criteria, and without impairing the operation of the circuit.

According to the invention, there is provided an electron beam deflection circuit for a cathode ray tube with electromagentic deflection by means of a sawtooth current waveform having a trace portion and a retrace portion, said circuit comprising: a deflection winding; a first source of electrical energy formed by a first capacitor; first controllable switching means comprising a parallel combination of a first thyristor and a first diode, connected together to conduct in opposite directions, for connecting said winding to said first source during said trace portion when said first switching means is turned on; a second source of electrical energy including a first inductive energy storage means coupled to a voltage supply; reactive circuit means including a combination of inductive and capacitive reactances for storing the energy supplied by the said second source; a second controllable switching means, substantially identical with the first one, for completing a circuit including said reactive circuit means and said first switching means, when turned on, so as to pass through said first thyristor an oscillatory current in the opposite direction to that which passes through it from said first source and to turn it off after these two currents cancel out, the oscillatory current then flowing through said first diode for an interval termed the circuit turn-off time which has to be greater than the turn-off time of said first thyristor; and means for drawing duing at least a part of said trace portion a substantial amount of additional current from said first switching means in the direction of conduction of said first diode, whereby said circuit turn-off time is lengthened in proportion to the amount of said additional current, without altering the values of the reactances in the reactive circuit by shifting the waveform of the current flowing through said first switching means towards the negative by an amount equal to that of said additional current.

A further object of the invention consists in using the supplementary current in the recovery diode of the first switching means to produce a DC voltage which may be used as a power supply for the vertical deflection circuit of the television receiver, for example.

The invention will be better understood and other features and advantages thereof will become apparent from the following description and the accompanying drawings, given by way of example, and in which:

FIG. 1 is a schematic circuit diagram partially in bloc diagram form of a prior art deflection circuit according to the aforementioned Patent;

FIG. 2 shows waveforms of currents and voltages generated at various points in the circuit of FIG. 1;

FIG. 3 is a schematic diagram of a deflection circuit according to the invention which allows the principle of the improvement to be explained;

FIG. 4 is a diagram of the waveforms of the current through the first switching means 4, 5 of the circuit of FIG. 3;

FIG. 5 is a circuit diagram of another embodiment of the circuit according to the invention;

FIG. 6 is a schematic representation of the preferred embodiment of the circuit according to the invention; and

FIG. 7 shows voltage waveforms at various points of the high voltage autotransformer 21 of FIG. 6.

In all these Figures the same reference numerals refer to the same components.

FIG. 1 shows the horizontal deflection circuit described and claimed in the U.S. Pat. No. 3,449,623 mentioned above, which comprises a first source of electrical energy in the shape of a first capacitor 2 having a high capacitance C ₂ for supplying a substantially constant voltage Uc ₂ across its terminals. A first terminal of the first capacitor 2 is connected to ground, whilst its second terminal which supplies a positive voltage is connected to one of the terminals of a horizontal deflection winding shown as a first inductance 1. A first switching means 3, consisting of a first reverse blocking triode thyristor 4 (SCR) and a first recovery diode 5 in parallel, the two being interconnected to conduct current in opposite directions, is connected in parallel with the series combination formed by the deflection winding 1 and the first capacitor 2. The assembly of components 1, 2, 4 and 5 forms the final stage of the horizontal deflection circuit in a television receiver using electromagnetic delfection.

The deflection circuit also includes a drive stage for this final stage which here controls the turning off of the first thyristor 4 to produce the retrace or fly-back portion of the scan during the line-blanking intervals i.e. while the picture signal is not transmitted. This driver stage comprises a second voltage source in the shape of a DC power supply 6 which delivers a constant high voltage E. The negative terminal of the power supply 6 is connected to ground and its positive terminal to one of the terminals of a second inductance 7 of relatively high value, which draws a substantially lineraly varying current from the power supply 6 to avoid its overloading. The other terminal of the second inductance 7 is connected, on the one hand, to the junction of the deflection winding 1 and the first switching means 3 by means of a second inductance 8 and a second capacitor 9 in series and, on the other hand, to one of the terminals of a second controllable bi-directionally conducting switching means 10, similar to the first one 3, including a parallel combination of a second thyristor 11 and a second recovery diode 12 also arranged to conduct in opposite directions.

The respective values of the third inductance 8 (L ₈ ) and of the second capacitor 9 (C ₉ ) are principally selected so that, on the one hand, one half-cycle of oscillation of the first series resonant circuit L ₈ - C ₉ , (i.e. π √ L ₈ ^. C ₉ ) is longer than the turn-off time of the first thyristor 4, but still is as short as possible since this time interval determines the speed of the commutation of the thyristor 4, and, on the other hand, one half-cycle of oscillation of another series resonant circuit formed by L ₁ , L ₈ and C ₉ , i.e. π √ (L ₁ + L ₈ ) ^. C ₉ , is substantially equal to the required retrace time interval (i.e. shorter than the horizontal blanking interval).

The gate (control electrode) of the second thyristor 11 is coupled to the output of the horizontal oscillator 13 of the television receiver by means of a first pulse transformer 14 and a first pulse shaping circuit 15 so that it is fed short triggering pulses which are to turn it on.

The gate of the first thyristor 4 fed with signals of a substantially rectangular waveform which are negative during the horizontal blanking intervals, is coupled to a winding 16 by means of a second pulse shaping circuit 17, the winding 16 being magnetically coupled to the second inductance 7 to make up the secondary winding of a transformer of which the inductance 7 forms the primary winding. It will be noted here that it is also possible to couple the secondary winding 16 magnetically to a primary winding connected to a suitable output (not shown) of the horizontal oscillator 13.

The operation of a circuit of this type will be explained below with reference to FIG. 2 which shows the waveforms at various points in the circuit of FIG. 1 during approximately one line period.

FIG. 2 is not to scale since one line period (t ₇ - t ₀ ) is equal to 64 microseconds in the case of 625 lines and 49 microseconds in the case of 819 lines, while the durations of the respective horizontal blanking intervals are approximately 12 and 9.5 microseconds.

Waveform A shows the form of the current i _L1 passing through deflection winding 1, this current having a sawtooth waveform substantially linear from t ₀ to t ₃ and from t ₅ to t ₇ , and crossing zero at time instants t ₀ and t ₇ , and reaching values of + I _1m and - I _1m , at time instants t ₃ and t ₅ respectively, these being its maximum positive and negative amplitudes.

During the second half of the trace portion of the horizontal deflection cycle, that is to say from t ₀ to t ₃ , the thyristor 4 of the first switching means 3 is conductive and makes the high value capacitor 2 discharge through the deflector winding 1, which has a high inductance, so that current i _L1 increases linearly.

A few microseconds (5 to 8 μ s) before the end of the trace portion, i.e. at time instant t ₁ , the trigger of the second thyristor 11 receives a short voltage pulse V _G11 which causes it to turn on as its anode is at this instant at a positive potential with respect to ground, which is due to the charging of the second capacitor 9 through inductances 7 and 8 by the voltage E from the power supply 6.

When thyristor 11 is made conductive at time t ₁ , on the one hand, inductance 7 is connected between ground and the voltage source 6 and a linearly increasing current flows through it and, on the other hand, the reactive circuit 8, 9 forms a loop through the second and first switching means 10 and 3, thus forming a resonant circuit which draws an oscillatory current i ₈ ,9 of frequency ##EQU1##

This oscillatory current i ₈ ,9 will pass through the first switching means 3, i.e. thyristor 4 and diode 5, in the opposite direction to that of current i _L1 . Since the frequency f ₁ is high, current i ₈ ,9 will increase more rapidly than i _L1 and will reach the same level at time t ₂ , that is to say i ₈ ,9 (t ₂ ) = -i _L1 (t ₂ ) and these currents will cancel out in the thyristor 4 in accordance with the well known principle of forced commutation. After time instant t ₂ , current i ₈ ,9 continues to increase more rapidly than i _L1 , but the difference between them (i ₈ ,9 - i _L1 ) passes the diode 5 (see wave form B) until it becomes zero at time instant t ₃ which is the turn off time instant of the first switching means 3, at which the retrace begins.

The interval between the time instant t ₂ and t ₃ , i.e. (t ₃ -t ₂ ), during which diode 5 is conductive and the thyristor is reverse biased will be termed in what follows the circuit turn-off time and it should be greater than the turn-off time of the thyristor 4 itself since the latter will subsequently become foward biased (i.e. from t ₃ to t ₅ ) by the retrace or flyback pulse (see waveform E) which should not trigger it.

At time instant t ₃ , the switching means 3 is opened (i ₄ and i ₅ are both zero -- see waveforms B and C) and the reactive circuit 8, 9 forms a loop through capacitor 2 and the deflection coil 1 and thus a series resonant circuit including (L ₁ + L ₈ ) and C ₉ , C ₂ being of high value and representing a short circuit for the flyback frequency ##EQU2## thus obtained.

The retrace which stated at time t ₃ takes place during one half-cycle of the resonant circuit formed by reactances L ₁ , L ₈ and C ₉ , i.e. during the interval between t ₃ and t ₅ . In the middle of this interval i.e. at time instant t ₄ , both i _L1 (waveform A) and i ₈ ,9 (waveform D) pass through zero and change their sign, whereas the voltage at the terminals of the first switching means 3 (V ₃ , waveform E) passes through a maximum. Thus, from t ₄ onwards, thyristor 11 will be reverse biased and diode 12 will conduct the current from the resonant circuit 1, 8 and 9 in order to turn the second thyristor 11 off.

At time instant t ₅ , when current i _L1 has reached - I _1m and when voltage v ₃ falls to zero, diode 5 of the first switching means 3 becomes conductive and the trace portion of scan begins.

Current i ₈ ,9 nevertheless continues to flow in the resonant circuit 8, 9 through diodes 5 and 12, which causes a break to appear in waveform D at t ₅ , and a negative peak to appear in waveform D and a positive one in waveform B in the interval between t ₅ and t ₆ , these being principally due to the distributed capacities of coil 1 or to an eventual capacitor (not shown) connected in parallel to the first switching means 3.

At time instant t ₆ , diode 12 of the second switching means 10 ceases to conduct after having allowed thyristor 11 time to become turned off completely.

The level of current i ₈ ,9 at time instant t ₅ (i.e. I _c ) as well as the negative peak I _D12 in i ₈ ,9 and the positive peak I _D5 in i ₅ depend on the values of L ₈ and C ₉ in the same way as does the turn-off time of the circuit (t ₃ - t ₂ ). If, for example, L ₈ and C ₉ , are increased I _D5 increases towards zero and this could cause diode 5 to be cut off in an undesirable fashion. I _c also increases towards zero, which is liable to cause diode 12 to be blocked and thyristor 11 to trigger prematurely.

From the foregoing it can be clearly seen that the choice of values for L ₈ and C ₉ is subject to four limitations which prevent the values from being increased to lengthen the turn-off time of the driver circuit of first switching thyristor 4 so as to forestall its spurious triggering.

Waveform F shows the voltage v _G4 obtained at the gate of thyristor 4 from the secondary winding 16 coupled to the inductor 7. This voltage is positive from t ₀ to t ₁ and from t ₆ to t ₇ and is negative between t ₂ and t ₆ i.e. while the second switching means 10 is conducting.

The present invention makes the lengthening of the turn-off time of thyristor 4 possible without altering the parameters of the circuit such as inductance 8 and capacitor 9.

In the circuit shown in FIG. 3, which illustrates the principle of the present invention, means are added to the circuit in FIG. 1 which enable the turn-off time to be lengthened by connecting a load to diode 5 so as to increase the current which flows through it during the time that it is conductive. These means are here formed by a resistor 18 connected in parallel with a capacitor 20 (which replaces capacitor 2) which is of a higher capacitance so that, in practice, it holds its charge during at least one half of the line period. FIG. 4, which shows the waveform of the current in the first switching means 3 for a circuit as shown in FIG. 3, makes it possible to explain how this lenthening of the turn-off time is achieved.

In FIG. 4, the broken lines show the waveform of the current in the first switch device 3 in the circuit of FIG. 1, this waveform being produced by adding waveforms B and C of FIG. 2. The current i ₄ above the axis flows through thyristor 4 and current i ₅ below the axis flows through diode 5. When the capacitance C ₂₀ of the capacitor in series with the deflector coil is increased to some tens of microfarads (C ₂ having been of the order of 1 μ F) and when there is connected in parallel with capacitor 20 a resistor 18 the value of which is calculated to draw a strong current I _R18 from capacitor 20, that is to say a current at least equal to 0,1 I _m (I _m being of the order of some tens of amperes), current I _R18 is added to that i ₅ which flows through diode 5 without in any way altering the linearity of the trace portion nor the oscillatory commutation of thyristor 4 which is brought about by the resonant circuit L ₈ , C ₉ .

The fact of loading capacitor C ₂₀ by means of a resistor 18 thus has the effect of permanently displacing the waveform of the current in the negative direction by I _R18 . Thus, during the trace portion of the scan, the transfer of the current from the diode 5 to the thyristor 4 begins at time t ₁₀ instead of t ₀ , that is to say with a delay proportional to I _R18 . The effect of the triggering pulse delivered by the horizontal oscillator (13 FIG. 1) to the second thyristor 11 at time instant t ₁ , will be to start the commutation process of the first thyristor 4 when the current it draws is less by I _R18 than that i ₄ (t ₁ ) which it would have been drawing had there been no resistor 18. Because of this, the turn-off time of the thyristor 4 proper, which as has been mentioned increases with the maximum current level passing throught it, is slightly reduced. Moreover, because the oscillatory current i ₈ ,9 (FIG. 2) from circuit L ₈ , C ₉ which flows through thyristor 4 in the opposite direction is unchanged, it reaches a value equal to that of the current i _L1 (FIG. 1) flowing in the coil 1 in a shorter time, that is to say at time t ₁₂ . Diode 5 will thus take the oscillatory current i ₈ ,9 (FIG. 2) over in advance with respect ro time instant t ₂ and will conduct it until it reaches zero value at a time instant t ₁₃ later than t ₃ , the amounts of advance (t ₂ - t ₁₂ ) and delay (t ₁₃ - t ₃ ) being practically equal.

It can thus be seen in FIG. 4 that the circuit turn-off time T _R of a circuit according to the invention and illustrated by FIG. 3 is distinctly longer than that T _r of the circuit in FIG. 1. This increase in the turn-off time (T _R - T _r ) depends on the current I _R18 and increases therewith.

It should be noted at this point that the current I _R18 produces a voltage drop at the terminals of the resistor the only effect of which is to heat up the resistor since the level of this voltage (40 to 60 volts) does not necessarily have a suitable value to be used as a voltage supply for other circuits in an existing transistorised television receiver.

In accordance with one embodiment of the invention, illustrated in FIG. 5, an application is proposed for the additional current which is to be drawn through diode 5. In FIG. 5, the positive terminal of capacitor 20 is connected by a conductor 19 to the negative pole of the power supply 6 and the voltage at the terminals of capacitor 20 is thus added to that E from the source 6.

In the preferred embodiment of the present invention, which is shown in FIG. 6, it is possible to cause a supplementary current of a desired value to flow through the first diode 5 while obtaining a voltage which has a suitable value for use in another circuit in the television receiver.

If the voltage at the terminals of capacitor 20 in FIG. 3 is not a usable value, it is possible to connect in parallel with the series circuit comprising the deflector coil 1 and the capacitor 2 in FIG. 1, i.e. in parallel with the terminals of the first switching means 3, a series combination of an autotransformer 21 and a high value capacitor 22 (comparable with capacitor 20 in FIGS. 3 and 5). The autotransformer 21 has a tap 23 is suitably positioned between the terminal connected to capacitor 22 at the tap 24 connected to the first switching means 3. This autotransformer 21 may be formed by the one conventionally used for supplying a very high voltage to the cathode ray tube, as described for example in U.S. Pat. No. 3,452,244; such a transformer comprises a voltage step-up winding between taps 24 and 25, which latter is connected to a high voltage rectifier (not shown).

The waveform of the voltage at the various points in the autotransformer is shown in FIG. 7, in which waveform A shows the voltage at the terminals of capacitor 22, waveform B the voltage at tap 24 and waveform C the voltage at tap 23 of the autotransformer 21.

The voltage V _c22 at the terminals of capacitor 22 varies slightly about a mean value V _cm . It is increasing while diode 5 is conducting and decreasing during the conduction of the thyristor 4.

The voltage v ₂₄ at tap 24 follows substantially the same curve as waveform E in FIG. 2, that is to say that during the retrace time interval from t ₁₃ to t ₅ to a positive pulse called the flyback pulse is produced and, during the time interval while the first switching means 3 is conducting, the voltage is zero. The mean valve of the voltage v ₂₄ at tap 24 of the auto-transformer 21 is equal to the mean value V _cm of the voltage at the terminals of capacitors 2 and 22.

Thus, there is obtained at tap 23 a waveform which is made up, during the retrace portion, of a positive pulse whose maximum amplitude is less than that of v ₂₄ at tap 24 and, during the trace portion, of a substantially constant positive voltage, the level V of which is less than the mean value V _cm of the voltage v _c22 at the terminals of capacitor 22. By moving tap 23 towards terminals 24 the amplitude of the pulse during fly-back increases while voltage V falls and conversely by moving tap 23 towards capacitor 22 voltage V increases and the amplitude of the pulse drops.

In more exact terms, the voltage V at tap 23 is such that the means value of v ₂₃ is equal to V _cm . It has thus been shown that by choosing carefully the position of tape 23, a voltage V may be obtained during the trace portion of the scan, which may be of any value between V _cm and zero.

This voltage V is thus obtained by periodically controlled rectification during the trace portion of the scan. For this purpose an electronic switch is used to periodically connect the tap 23 of trnasformer winding 21 to a load. This switch is made up of a power transistor 26 whose collector is connected to tap 23 and the emitter to a parallel combination formed by a high value filtering capacitor 27 and the load which it is desired to supply, which is represented by a resistor 28. The base of the transistor 26 receives a control voltage to block it during retrace and to unblock it during the whole or part of the trace period. A control voltage of this type may be obtained from a second winding 29 magnetically coupled to the inductance 7 of the deflection circuit and it may be transmitted to the base of transistor 26 by means of a coupling capacitor 30 and a resistor 31 connected between the base and the emitter of transistor 26.

It may easily be seen that the DC collector/emitter current in transistor 26 flows through the first diode 5 of the first switching means 3 via a resistor 28 and the part of the winding of auto-transformer 21 located between taps 23 and 24.

Experience has shown that a circuit as shown in FIG. 6 can supply 24 volts with a current of 2 amperes to the vertical deflection circuit of the same television set, the voltage at the terminals of capacitor 22 being from 50 to 60 volts.

It should be mentioned that, when the circuit which forms the load of the controlled rectifier 26, 27 does not draw enough current to sufficiently lengthen the circuit turn-off time T _R , an additional resistor (not shown) may be connected between the emitter of transistor 26 and ground or in parallel to capacitor 22, which resistor will draw the additional current required.

GRUNDIG SUPER COLOR 1514 SERIE 8S15 CHASSIS GSC100 (29304-176.90) INTEGRAL THYRISTOR-RECTIFIER DEVICEA semiconductor switching device comprising a silicon controlled rectifier (SCR) and a diode rectifier integrally connected in parallel with the SCR in a single semiconductor body. The device is of the NPNP or PNPN type, having gate, cathode, and anode electrodes. A portion of each intermediate N and P region makes ohmic contact to the respective anode or cathode electrode of the SCR. In addition, each intermediate region includes a highly conductive edge portion. These portions are spaced from the adjacent external regions by relatively low conductive portions, and limit the conduction of the diode rectifier to the periphery of the device. A profile of gold recombination centers further electrically isolates the central SCR portion from the peripheral diode portion.
That class of thyristors known as controlled rectifiers are semiconductor switches having four semiconducting regions of alternate conductivity and which employ anode, cathode, and gate electrodes. These devices are usually fabricated from silicon. In its normal state, the silicon controlled rectifier (SCR) is non-conductive until an appropriate voltage or current pulse is applied to the gate electrode, at which point current flows from the anode to the cathode and delivers power to a load circuit. If the SCR is reverse biased, it is non-conductive, and cannot be turned on by a gating signal. Once conduction starts, the gate loses control and current flows from the anode to the cathode until it drops below a certain value (called the holding current), at which point the SCR turns off and the gate electrode regains control. The SCR is thus a solid state device capable of performing the circuit function of a thyratron tube in many electronic applications. In some of these applications, such as in automobile ignition systems and horizontal deflection circuits in television receivers, it is necessary to connect a separate rectifier diode in parallel with the SCR. See, for example, W. Dietz, U. S. Pat. Nos. 3,452,244 and 3,449,623. In these applications, the anode of the rectifier diode is connected to the cathode of the SCR, and the cathode of the rectifier is connected to the SCR anode. Thus, the rectifier diode will be forward biased and current will flow through it when the SCR is reverse biased; i.e., when the SCR cathode is positive with respect to its anode. For reasons of economy and ease of handling, it would be preferable if the circuit function of the SCR and the associated diode rectifier could be combined in a single device, so that instead of requiring two devices and five electrical connections, one device and three electrical connections are all that would be necessary. In fact, because of the semiconductor profile employed, many SCR's of the shorted emitter variety inherently function as a diode rectifier when reverse biased. However, the diode rectifier function of such devices is not isolated from the controlled rectifier portion, thus preventing a rapid transition from one function to the other. Therefore, it would be desirable to physically and electrically isolate the diode rectifier portion from that portion of the device which functions as an SCR.





GRUNDIG SUPER COLOR 1514 SERIE 8S15 CHASSIS GSC100 (29304-176.90) Gating circuit for television SCR deflection system AND REGULATION / stabilization of horizontal deflection NETWORK CIRCUIT with Transductor reactor / Reverse thyristor energy recovery circuit.
In a television deflection system employing a first SCR for coupling a deflection winding across a source of energy during a trace interval of each deflection cycle and a second SCR for replenishing energy to the source of energy during a commutation interval of each deflection cycle, a gating circuit for triggering the first SCR. The gating circuit employs a voltage divider coupled in parallel with the second SCR which develops gating signals proportional to the voltage across the second SCR.


1. In a television deflection system in which a first switching means couples a deflection winding across a source of energy during a trace interval of each deflection cycle and a second switching means replenishes energy to said source of energy during a commutation interval of each deflection cycle, a gating circuit for said first switching means, comprising:
capacitive voltage divider means coupled in parallel with said second switching means for developing gating signals proportional to the voltage across said second switching means; and
means for coupling said voltage divider means to said first switching means to provide for conduction of said first switching means in response to said gating signals.
2. A gating circuit according to claim 1 wherein said voltage divider includes first and second capacitors coupled in series and providing said gating signals at the common terminal of said capacitors. 3. A gating circuit according to claim 2 wherein said first and second capacitors are proportional in value to provide for the desired magnitude of gating signals. 4. A gating circuit according to claim 3 wherein said means for coupling said voltage divider means to said first switching means includes an inductor. 5. A gating circuit according to claim 4 wherein said inductor and said first and second capacitors comprise a resonant circuit having a resonant frequency chosen to shape said gating signal to improve switching of said first switching means.

Description:

BACKGROUND OF THE INVENTION
This invention relates to a gating circuit for controlling a switching device employed in a deflection circuit of a television receiver.

Various deflection system designs have been utilized in television receivers. One design employing two bidirectional conducting switches and utilizing SCR's (thyristors) as part of the switches is disclosed in U.S. Pat. No. 3,452,244. In this type deflection system, a first SCR is










employed for coupling a deflection winding across a source of energy during a trace interval of each deflection cycle, and a second SCR is employed for replenishing energy during a commutation interval of each deflection cycle. The first SCR is commonly provided with gating voltage by means of a separate winding or tap of an input reactor coupling a source of B+ to the second SCR.

Various regulator system designs have been utilized in conjunction with the afore described deflection system to provide for uniform high voltage production as well as uniform picture width with varying line voltage and kinescope beam current conditions.

One type regulator system design alters the amount of energy stored in a commutating capacitor coupled between the first and second SCR's during the commutating interval. A regulator design of this type may employ a regulating SCR and diode for coupling the input reactor to the source of B+. With this type regulator a notch, the width of which depends upon the regulation requirements, is created in the current supplied through the reactor and which notch shows up in the voltage waveform developed on the separate winding or tap of the input reactor which provides the gating voltage for the first SCR. The presence of the notch, even though de-emphasized by a waveshaping circuit coupling the gating voltage to the first SCR, causes erratic control of the first SCR.
SUMMARY OF THE INVENTION
In accordance with one embodiment of the invention, a gating circuit of a television deflection system employing a first switching means for coupling a deflection winding across a source of energy during a trace interval of each deflection cycle and a second switching means for replenishing energy to said source of energy during a commutation interval of each deflection cycle includes a voltage divider means coupled in parallel with the second switching means for developing gating signals proportional to the voltage across the second switching means. The voltage divider means are coupled to the first switching means to provide for conduction of the first switching means in response to the gating signals.
A more detailed description of a preferred embodiment of the invention is given in the following description and accompanying drawing of which:
FIG. 1 is a schematic diagram, partially in block form, of a prior art SCR deflection system;
FIG. 2 is a schematic diagram, partially in block form, of an SCR deflection system of the type shown in FIG. 1 including a gating circuit embodying the invention;
FIG. 3 is a schematic diagram, partially in block form, of one type of a regulator system which employs an SCR as a control device and which is suitable for use with the SCR deflection system of FIG.2;
FIG. 4 is a schematic diagram, partially in block form, of another type of a regulator system suitable for use with the deflection circuit of FIG. 2; and
FIG. 5 is a schematic diagram, partially in block form, of still another type of a regulator system suitable for use with the SCR deflection system of FIG. 2.
DESCRIPTION OF THE PREFERRED EMBODIMENT
FIG. 1 is a schematic diagram, partially in block form, of a prior art deflection system of the retrace driven type similar to that disclosed in U.S. Pat. No. 3,452,244. This system includes a commutating switch 12, comprising a silicon controlled rectifier (SCR) 14 and an oppositely poled damper diode 16. The commutating switch 12 is coupled between a winding 18a of an input choke 18 and ground. The other terminal of winding 18a is coupled to a source of direct current voltage (B+) by means of a regulator network 20 which controls the energy stored in the deflection circuit 10 when the commutating switch is off, during an interval T3 to T0' as shown in curve 21 which is a plot of the voltage level at the anode of SCR 14 during the deflection cycle. A damping network comprising a series combination of a resistor 22 and a capacitor 23 is coupled in parallel with commutating switch 12 and serves to reduce any ringing effects produced by the switching of commutating switch 12. Commutating switch 12 is coupled through a commutating coil 24, a commutating capacitor 25 and a trace switch 26 to ground. Trace switch 26 comprises an SCR 28 and an oppositely poled damper diode 30. An auxiliary capacitor 32 is coupled between the junction of coil 24 and capacitor 25 and ground. A series combination of a horizontal deflection winding 34 and an S-shaping capacitor 36 are coupled in parallel with trace switch 26. Also, a series combination of a primary winding 38a of a horizontal output transformer 38 and a DC blocking capacitor 40 are coupled in parallel with trace switch 26.
A secondary of high voltage winding 38b of transformer 38 produces relatively large amplitude flyback pulses during the retrace interval of each deflection cycle. This interval exists between T1 and T2 of curve 41 which is a plot of the current through windings 34 and 38a during the deflection cycle. These flyback pulses are applied to a high voltage multiplier (not shown) or other suitable means for producing direct current high voltage for use as the ultor voltage of a kinescope (not shown).
An auxiliary winding 38c of transformer 38 is coupled to a high voltage sensing and control circuit 42 which transforms the level of flyback pulses into a pulse width modulated signal. The control circuit 42 is coupled to the regulator network 20.
A horizontal oscillator 44 is coupled to the gate electrode of commutating SCR 14 and produces a pulse during each deflection cycle slightly before the end of the trace interval at T0 of curve 21 to turn on SCR 14 to initiate the commutating interval. The commutating interval occurs between T0 and T3 of curve 21. A resonant waveshaping network 46 comprising a series combination of a capacitor 48 and an inductor 50 coupled between a winding 18b of input choke 18 and the gate electrode of trace SCR 28 and a damping resistor 52 coupled between the junction of capacitor 48 and inductor 50 and ground shapes the signal developed at winding 18b (i.e. voltage waveform 53) to form a gating signal voltage waveform 55 to enable SCR 28 for conduction during the second half of the trace interval occurring between T2 and T1' of curve 41.
The regulator network 20, when of a type to be described in conjunction with FIG. 3, operates in such a manner that current through winding 18a of input choke 18 during an interval between T4 and T5 (region A) of curves 21, 53 and 55 is interrupted for a period of time the duration of which is determined by the signal produced by the high voltage sensing and control circuit 42. During the interruption of current through winding 18a a zero voltage level is developed by winding 18b as shown in interval T4 to T5 of curve 53. The resonant waveshaping circuit 46 produces the shaped waveform 55 which undesirably retains a slump in region A corresponding to the notch A of waveform 53. The slump in waveform 55 applied to SCR 28 occurs in a region where the anode of SCR 28 becomes positive and where SCR 28 must be switched on to maintain a uniform production of the current waveshape in the horizontal deflection winding 34 as shown in curve 41. The less positive amplitude current occurring at region A of waveform 55 may result in insufficient gating current for SCR 28 and may cause erratic performance resulting in an unsatisfactory raster.
FIG. 2 is a schematic diagram, partially in block form, of a deflection system 60 embodying the invention. Those elements which perform the same function in FIG. 2 as in FIG. 1 are labeled with the same reference numerals. FIG. 2 differs from FIG. 1 essentially in that the signal to enable SCR 28 derived from sampling a portion of the voltage across commutating switch 12 rather than a voltage developed by winding 18b which is a function of the voltage across winding 18a of input choke 18 as in FIG. 1. This change eliminates the slump in the enabling signal during the interval T4 to T5 as shown in curve 64 since the voltage across the commutating switch 12 is not adversely effected by the regulator network 20 operation.
A series combination of resistor 22, capacitor 23 and a capacitor 62 is coupled in parallel with commutating switch 12, one terminal of capacitor 62 being coupled to ground. The junction of capacitors 23 and 62 is coupled to the gate electrode of SCR 28 by means of the inductor 50. The resistor 52 is coupled in parallel with capacitor 62.
Capacitors 23 and 62 form a capacitance voltage divider which provides a suitable portion of the voltage across commutating switch 12 for gating SCR 28 via inductor 50. The magnitude of the voltage at the junction of capacitors 23 and 62 is typically 25 to 35 volts. It can, therefore, be seen that the ratio of values of capacitors 23 and 62 will vary depending on the B+ voltage utilized to energize the deflection system. Capacitors 23 and 62 and inductor 50 form a resonant circuit tuned in a manner which provides for peaking of the curve 64 between T4 and T5. This peaking effect further enhances gating of SCR 28 between T4 and T5.
Since the waveshape of the voltage across commutating switch 12 (curve 21) is relatively independent of the type of regulator system employed in conjunction with the deflection system, the curve 64 also is independent of the type of regulator system.
When commutating switch 12 switches off during the interval T3 to T0' curve 21, the voltage across capacitor 62 increases and the voltage at the gate electrode of SCR 28 increases as shown in curve 64. As will be noted, no slump of curve 64 occurs between T3 and T5 because there is no interruption of the voltage across commutating switch 12.



















FIG. 3 is a schematic diagram, partially in block form, of one type of a regulator system which may be used in conjunction with the invention. B+ is supplied through a regulator network 20 which comprises an SCR 66 and an oppositely poled diode 68. The diode is poled to provide for conduction of current from B+ to the horizontal deflection circuit 60 via winding 18a of input choke 18. Current flows through the diode during the period T3 to T4 of curve 21 FIG. 1 after which current tries to flow through the SCR 66 from the horizontal deflection circuit to B+ since the commutating capacitor 25 is charged to a voltage higher than B+.
The horizontal deflection circuit 60 produces a flyback pulse in winding 38a of the flyback transformer 38 which is coupled to winding 38c. The magnitude of the pulse on winding 38c determines how long the signal required to switch SCR 66 on is delayed after T4 curve 21 FIG. 1. If the flyback pulse is greater than desirable, the SCR 66 turns on sooner than if the flyback pulse is less than desirable and provides a discharge path for current in commutating capacitor 25 back to the B+ supply. In this manner a relatively constant amplitude flyback pulse is maintained.
FIG. 4 is a schematic diagram, partially in block form, of another well-known type of a regulator system which may be used in conjunction with the invention shown in FIG. 2. B+ is coupled through winding 18a of input choke 18 and through a series combination of windings 70a and 70b of a saturable reactor 70 and a parallel combination of a diode 72 and a resistor 74 to the horizontal deflection circuit 60. Diode 72 is poled to conduct current from the horizontal deflection circuit 60 to B+.
Flyback pulse variations are obtained from winding 38c of the horizontal output transformer 38 and applied to a voltage divider comprising resistors 76, 78 and 80 of the high voltage sensing and control circuit 42. A portion of the pulse produced by winding 38c is selected by the position of the wiper terminal on potentiometer 78 and coupled to the base electrode of a transistor 82 by means of a zener diode 84. The emitter electrode of transistor 82 is grounded and a DC stabilization resistor 85 is coupled in parallel with the base-emitter junction of transistor 82. When the pulse magnitude on winding 38c exceeds a level which results in forward biasing the base-emitter junction of transistor 82, current flows from B+ through a resistor 86, a winding 70c of saturable reactor 70 and transistor 82 to ground. Due to the exponential increase of current in winding 70c during the period of conduction of transistor 82, the duration of conduction of transistor 82 determines the magnitude of current flowing in winding 70c and thus the total inductance of windings 70a and 70b. The current in winding 70c is sustained during the remaining deflection period by means of a diode 88 coupled in parallel with winding 70c and poled not to conduct current from B+ to the collector electrode of transistor 82. A capacitor 90 coupled to the cathode of diode 88 provides a bypass for B+. Windings 70a and 70b are in parallel with input reactor 18a and thereby affect the total input inductance of the deflection circuit and thereby controls the transfer of energy to the deflection circuit. The dotted waveforms shown in conjunction with a curve 21' indicate variations from a nominal waveform provided at the input of horizontal deflection circuit 60 by the windings 70a and 70b.













FIG. 5 is a schematic diagram of yet another type of a regulator system which may be used in conjunction with the invention. B+ is coupled through a winding 92a and a winding 92b of a saturable reactor to the horizontal deflection circuit 60. Windings 92a and 92b are used to replace the input choke 18 shown in FIGS. 1 and 2 while also providing for a regulating function corresponding to that provided by regulating network 20.
Flyback pulse variations are obtained from winding 38c and applied to the high voltage sensing and control circuit 42 as in FIG. 4. Current flows from B+ through resistor 86, a winding 92c and transistor 82 to ground. As in FIG. 4 the duration of the conduction of transistor 82 determines the energy stored in winding 92c and thus the total inductance of windings 92a and 92b which control the amount of energy transferred to the deflection circuit during each horizontal deflection cycle. The variations in waveforms of curve 21', shown in conjunction with FIG. 4, are also provided at the input of horizontal deflection circuit 60 by windings 92a and 92b.
For various reasons including cost or performance, a manufacturer may wish to utilize a particular one of the regulators illustrated in FIGS. 3, 4 and 5. Regardless of the choice, the gating circuit according to the invention may be utilized therewith advantageously by providing improved performance and the possibility of cost savings by eliminating taps or extra windings on the wound components which heretofore normally provided a source of SCR gating waveforms.

COMMON FAULTS  /  TROUBLESHOOTING  OF  GRUNDIG CHASSIS GSC100:

Grundig GCS100 Dead - blowing fusible resistor. Insulating pad under TY501 leaky.
Grundig GSC100 Dead - blowing 3.15 fuse - intermittently. EHT cap arcing.
Grundig GSC100 Dead - blowing 3.15amp fuse. EHT cap jumping across to degaussing coil.
Grundig GSC100 Dead. R607 o/c , faulty thyristors , insulation in line o/p stage.
Grundig GSC100 Dead. Di511 or C2502 ( 100u f , 16v ) on EHT control regel baust PCB.
Grundig GSC100 Frame - bottom foldover - cramping at top. R2522 ( 2.2k ) goes high.
Grundig GSC100 IF EHT IS VERY HIGH CHECK DI508 IS OPEN
Grundig GSC100 Line break up. C2502 ( 100uf , 16v ) on EHT control regel baust PCB.
Grundig GSC100 No frame. Di447 ( 1N4007 ) and/or TDA1170.
Grundig GSC100 R607 - repeated resoldering of R607. Di2502 ( 4.7v ) zener.