Richtige Fernseher haben Röhren!

Richtige Fernseher haben Röhren!

In Brief: On this site you will find pictures and information about some of the electronic, electrical and electrotechnical technology relics that the Frank Sharp Private museum has accumulated over the years .

Premise: There are lots of vintage electrical and electronic items that have not survived well or even completely disappeared and forgotten.

Or are not being collected nowadays in proportion to their significance or prevalence in their heyday, this is bad and the main part of the death land. The heavy, ugly sarcophagus; models with few endearing qualities, devices that have some over-riding disadvantage to ownership such as heavy weight,toxicity or inflated value when dismantled, tend to be under-represented by all but the most comprehensive collections and museums. They get relegated to the bottom of the wants list, derided as 'more trouble than they are worth', or just forgotten entirely. As a result, I started to notice gaps in the current representation of the history of electronic and electrical technology to the interested member of the public.


Following this idea around a bit, convinced me that a collection of the peculiar alone could not hope to survive on its own merits, but a museum that gave equal display space to the popular and the unpopular, would bring things to the attention of the average person that he has previously passed by or been shielded from. It's a matter of culture. From this, the Obsolete Technology Tellye Web Museum concept developed and all my other things too. It's an open platform for all electrical Electronic TV technology to have its few, but NOT last, moments of fame in a working, hand-on environment. We'll never own Colossus or Faraday's first transformer, but I can show things that you can't see at the Science Museum, and let you play with things that the Smithsonian can't allow people to touch, because my remit is different.

There was a society once that was the polar opposite of our disposable, junk society. A whole nation was built on the idea of placing quality before quantity in all things. The goal was not “more and newer,” but “better and higher" .This attitude was reflected not only in the manufacturing of material goods, but also in the realms of art and architecture, as well as in the social fabric of everyday life. The goal was for each new cohort of children to stand on a higher level than the preceding cohort: they were to be healthier, stronger, more intelligent, and more vibrant in every way.

The society that prioritized human, social and material quality is a Winner. Truly, it is the high point of all Western civilization. Consequently, its defeat meant the defeat of civilization itself.

Today, the West is headed for the abyss. For the ultimate fate of our disposable society is for that society itself to be disposed of. And this will happen sooner, rather than later.

OLD, but ORIGINAL, Well made, Funny, Not remotely controlled............. and not Made in CHINA.

How to use the site:

- If you landed here via any Search Engine, you will get what you searched for and you can search more using the search this blog feature provided by Google. You can visit more posts scrolling the left blog archive of all posts of the month/year,
or you can click on the main photo-page to start from the main page. Doing so it starts from the most recent post to the older post simple clicking on the Older Post button on the bottom of each page after reading , post after post.

You can even visit all posts, time to time, when reaching the bottom end of each page and click on the Older Post button.

- If you arrived here at the main page via bookmark you can visit all the site scrolling the left blog archive of all posts of the month/year pointing were you want , or more simple You can even visit all blog posts, from newer to older, clicking at the end of each bottom page on the Older Post button.
So you can see all the blog/site content surfing all pages in it.

- The search this blog feature provided by Google is a real search engine. If you're pointing particular things it will search IT for you; or you can place a brand name in the search query at your choice and visit all results page by page. It's useful since the content of the site is very large.

Note that if you don't find what you searched for, try it after a period of time; the site is a never ending job !

Every CRT Television saved let revive knowledge, thoughts, moments of the past life which will never return again.........

Many contemporary "televisions" (more correctly named as displays) would not have this level of staying power, many would ware out or require major services within just five years or less and of course, there is that perennial bug bear of planned obsolescence where components are deliberately designed to fail and, or manufactured with limited edition specificities..... and without considering........picture......sound........quality........

..............The bitterness of poor quality is remembered long after the sweetness of todays funny gadgets low price has faded from memory........ . . . . . .....
Don't forget the past, the end of the world is upon us! Pretty soon it will all turn to dust!

Have big FUN ! !
-----------------------

©2010, 2011, 2012, 2013, 2014 Frank Sharp - You do not have permission to copy photos and words from this blog, and any content may be never used it for auctions or commercial purposes, however feel free to post anything you see here with a courtesy link back, btw a link to the original post here , is mandatory.
All sets and apparates appearing here are property of
Engineer Frank Sharp. NOTHING HERE IS FOR SALE !

Wednesday, June 6, 2012

SINUDYNE COLOR 2202 (NORDMENDE COLOR 2921) CHASSIS FFS FIV (F4) (FCI25) INTERNAL VIEW.







































This is last NORDMENDE CHASSIS F4 which was replaced by the NORDMENDE F5

- The chassis NORDMENDE F4 I realized with 2 main sections:
- HORIZONTAL CHASSIS with all deflections and power supply.
- VERTICAL PANEL with all video and sound stages.


NOTE: THERE IS A CAPACITOR WICH HAS LEAKED PCB BUT EVEN IN SUCH CONDITIONS IS STILL FUNCTIONAL.


- 877/C FRAME DEFLECTION UNIT

- 090/A SYNCH UNIT

- 088/D RGB OUT SINGLE UNIT.

- 091/C TB540 COLOR CARRIER UNIT.

- 197/A SOUND OUTPUT UNIT.


SINUDYNE COLOR 2202 (NORDMENDE COLOR 2921) CHASSIS FFS FIV (F4) (FCI25) Power Supply CONSTANT-VOLTAGE CONVERTER EMPLOYING THYRISTOR:


A constant voltage converter having a rectifier for rectifying AC power and with a thyristor connected between the rectifier and a filter for selectively passing therethrough a rectified output to an output terminal. There is a wave generator connected to the output of the rectifier for producing a first signal and an intergrator circuit connected to the output of the wave generator for producing an integral output in response to this first signal. In addition there is a detector circuit for detecting a fluctuation of the rectified output power and for producing second signal. A comparison circuit is connected between the intergrator circuit and the detector circuit for producing third signal in accordance with the comparison. A trigger circuit is connected between the comparison circuit and the control gate of the thyristor for supplying a phase control signal to the thyristor to thereby obtain a constant voltage output regardless of the fluctuation of the rectified output.



1. A constant voltage converter comprising an input of a power supply means, an output terminal, filter means, rectifier means connected to said input for rectifying a.c. power and for supplying output thereof to said output terminal, thyristor means connected between said rectifier means and said filter means for selectively passing therethrough a rectified output to the output terminal by way of said filter means, saw-tooth wave generator means connected between the output of said rectifier means and at least one integrator circuit means for producing an integral output in response to a saw-tooth wave produced, a first transistor in said saw-tooth wave generator, the input of said integrator circuit means being connected to a collector of said first transistor, detector circuit means connected to said output terminal for detecting a fluctuation of the rectified output power and for producing an output signal, said detector circuit means having a second transistor, pulse generator circuit means connected between said saw-tooth wave generator means and said detector circuit means for producing a trigger pulse to said thyristor through a trigger means, a third transistor in said pulse circuit generator means, the base of said third transistor being connected to the output of said integrator circuit means, the emitter thereof being connected to the emitter of said second transistor in said detector circuit means, and the collector thereof being connected to the gate of the thyristor means so as to supply a phase control signal thereto, thereby obtaining a constant voltage output regardless of the fluctuation of the rectified output.

Description:
This invention relates to constant-voltage converters and more particularly to a constant-voltage converter employing a thyristor.

Conventional constant-voltage converters of the type employing a thyristor are arranged to phase shift and full-wave-rectify an input a.c. power applied thereto and to maintain the output voltages constant by regulating the firing angle of the thyristor in comparison of the output voltages with the phase-shifted and rectified input a.c. power. When, however, these converters are connected to a common a.c. source having a relatively high internal impedance, the waveform of the phase-shifted and rectified a.c. input power is distorted thereby causing undesired operations of the converters.

It is therefore an object of the present invention to provide a constant-voltage converter which correctly operates notwithstanding the distortion of the input a.c. voltage.

Another object of the invention is to provide a constant-voltage converter which effectively suppress an undesired rush current.

Another object of the invention is to provide a constant-voltage converter having an improved feed-back circuit of a substantially constant loop gain .

In the drawings:

FIG. 1 is a schematic view of a converter according to the present invention;

FIG. 2 is a diagram showing a circuit arrangement of the converter of FIG. 1;

FIG. 3 is a diagram showing various waveforms of signals appearing in the circuit of FIG. 2;

FIG. 4 is a diagram showing various waveforms appearing in the circuit of FIG. 2 when an a.c. power is supplied to the circuit;

FIG. 5 is a diagram showing another circuit arrangement of the converter of FIG. 1;

FIG. 6 is a diagram showing waveforms of signals appearing in the circuit of FIG. 5; and

FIG. 7 is a diagram showing further another circuit arrangement of generator the of FIG. 1.

Referring now to FIG. 1, a constant-voltage converter 10 according to the present invention comprises a rectifier 11 having two input terminals 12 and 13 through which an a.c. power is supplied. The rectifier 11 is preferably a full-wave rectifier although a half-wave rectifier may be employed. An output 14 of the rectifier 11 is connected through a line 15 to an anode of a thyristor 16. The thyristor 16 passes therethrough the rectified a.c. power in only one direction from its anode to cathode when triggered by a trigger pulse through its gate. The cathode of the thyristor 16 is connected through a line 17 to an input of a smoothing filter 18. The smoothing filter 18 smoothes the power from the thyristor 16. An output of the smoothing filter 18 is connected through a line 19 to an output terminal 20. The output 14 of the rectifier 11 is also connected through a line 21 to a saw-tooth wave generator 22 which generates a saw-tooth wave signal having the same repetition period as the rectified input a.c. power. An output of the saw-tooth wave generator 22 is connected through a line 23 to one input of a trigger pulse generator 24. The other input of the trigger pulse generator 24 is connected through a line 25 to the line 19. An output of the trigger pulse generator 24 is connected through a line 26 to the gate of the thyristor 16. The trigger pulse generator 24 produces a trigger pulse on its output when the voltage of the saw-tooth wave signal reaches a level which is varied in response to the output voltage on the terminal 20. The trigger pulse generator 24 may be variously arranged and in this case arranged to comprise rectangular generator 27 having one input connected through the line 23 to the saw-tooth wave generator 22 and the other input connected through a line 28 to an output voltage detector 29. The detector 29 produces a reference signal representing the output voltage on the terminal 20. The pulse generator 27 is adapted to produces a rectangular pulse when the saw-tooth wave signal to the one input reaches a level which defined is in accordance with the reference signal. An output of the rectangular pulse generator 27 is connected through a line 30 to an input of a trigger circuit 31. The trigger circuit 31 is adapted to convert the rectangular pulse into a spike pulse. An output of the trigger circuit 31 is connected through the line 26 to the gate of the thyristor 16.

FIG. 2 illustrates a preferred circuit arrangement of the converter shown in FIG. 1 which comprises a rectifier 11 of a full-wave rectifier consisting of rectifiers 40, 41, 42 and 43. Inputs of the rectifier are connected to terminals 12 and 13 through which an a.c. power is applied. The output 14 of the rectifier 11 is connected through a line 15 to an anode of a thyristor 16. A cathode of the thyristor 16 is connected through a line 17 to a smoothing filter 18 which includes a capacitor C4 having one terminal connected to the line 17 and the other terminal grounded. The output of the smoothing filter 18 is connected through a line 19 to an output terminal 20.

The saw-tooth wave generator 22 includes a resistor R 1 having one terminal connected to the line 21 and the terminal connected through a junction J 1 to one terminal of a resistor R 2 . The other terminal of the resistor R 2 is grounded. The junction J 1 is connected through a coupling capacitor C 1 to a base of a transistor T 1 of PNP type. An emitter of the transistor T 1 is connected through a resistor R 3 to the line 21. A resistor R 4 is provided between the emitter and the base of the transistor T 1 so as to apply a bias potential to the base. A collector of the transistor T 1 is grounded through a parallel connection of a resistor R 5 and capacitor C 2 . To the emitter is connected a capacitor C 3 which is in turn grounded and passes therethrough only a.c. signals to the ground.

The rectangular pulse generator 27 comprises a transistor T 2 of PNP type having a base connected through a resistor R 6 to the collector of the transistor T 1 . An emitter of the transistor T 2 is connected through a resistor R 7 to the emitter of the transistor T 1 . A collector of the transistor T 2 is grounded through a resistor R 8 and connected through the line 30 to one terminal of a capacitor C 4 of the trigger circuit 31. The other terminal of the capacitor C 4 is connected through a line 26 to the gate of the thyristor 16.

The output voltage detector 29 includes a transistor T 3 of NPN type having an emitter grounded through a zener diode ZD. A collector of the transistor T 3 is connected through a line 28 to the emitter of the transistor T 2 and, on the other hand, connected through a capacitor C 5 to the grounded. A base of the transistor T 3 is connected to a tap of an adjustable resistor R 9 connected through a resistor R 10 and a line 25 to the line 19 and connected, in turn, to the ground through a resistor R 11 .

When, in operation, an a.c. electric power is applied through the input terminals 12 and 13 of the rectifier 11, a full-wave rectified power as shown in FIG. 3 (a) appears on the output 14. The rectified power is applied through the line 15 to the anode of the thyristor 16. The thyristor 16 passes therethrough the rectified power while its firing angle is regulated by the trigger signal applied to the gate. The rectified power passed through the thyristor 16 is applied through the line 17 to the smoothing filter 18. The smoothing filter smoothes the power by removing the ripple component in the power. The smoothed power appears on the line 19 which is to be supplied to a load through the output terminal 20. The smoothed power on the line 19 is, on the other hand, delivered through the line 25 to the resistor R 10 of the output voltage detector 29. The resistor R 10 constitutes a voltage divider in cooperation with the resistors R 9 and R 11 . The output of the voltage divider is applied through the tap of the resistor R 9 to the base of the transistor T 3 . When the potential of the base of the transistor T 3 exceeds the zener voltage of the zener diode ZD, a base current flows through the transistor T 3 so as to render the transistor T 3 conductive. The potential of the collector of the transistor T 3 then varies in accordance with the voltage of the smoothed output power on the line 19. The potential variation at the collector of the transistor T 3 is then applied through the line 28 to the trigger pulse generator 27 and utilized to regulate the triggering timing of the thyristor 16.

The full-wave rectified power is, on the other hand, applied through the line 21 to the saw-tooth wave generator 22. Since the resistors R 1 and R 2 consistute a voltage divider to reduce the voltage of the full-wave rectified power to a potential at the junction J 1 , a charging current to the capacitor C 1 flows from the emitter to the base of the transistor T 1 whereby the transistor T 1 repeats ON-OFF operation in accordance with the voltage of the rectified power. If the transistor T 1 is conductive when the voltage of the full-wave rectified power is lower than a threshold voltage v 1 as shown in FIG. 3(a), then the potential at the collector of the transistor T 1 is varied as shown in FIG. 3 (b) due to the charge and discharge of the capacitor C 2 . The variation of the potential at the collector of the transistor T 1 is supplied through the line 23 to the resistor R 6 of the trigger pulse generator 27.

As long as the voltage of the smoothed power on the line 19 equals to the rated output voltage, the transistor T 2 is adapted to become conductive when the voltage of the saw-tooth wave signal falls below a threshold value v 3 shown in FIG. 3(b). Therefore, a potential at the collector of the transistor T 2 varies as shown in FIG. 3(c). The potential variation, that is, a pulse signal at the collector of the transistor T 2 is supplied through the line 30 to the capacitor C 4 of the trigger circuit trigger 31. The trigger circuit 31 converts the pulse signal into a spike pulse or a trigger pulse shown in FIG. 3(d) which is then applied through the line 25 to the gate of the thyristor 16. Upon receiving the spike pulse, the thyristor 16 becomes conductive until the voltage of the rectified power on the line 15 falls below the cut-off voltage of the thyristor 16.

When the voltage of the smoothed power on the line 19 exceeds the rated output voltage, the collector current of the transistor T 3 increases with the result that the current flowing through the resistor R 7 increases. The threshold voltage of the transistor T 2 therefore reduces to a voltage v 2 as shown in FIG. 3(b). At this instant, leading edge of the pulse signal delays as shown by dot-and-dash lines in FIG. 3(c), so that each trigger pulse delays as shown by dot-and-dash line in FIG. 3(d). When on the contrary, the voltage of the smoothed signal on the line 19 lowers below the rated output voltage, the collector current of the transistor T 3 decreases whereby the threshold voltage rises to a voltage v 4 in FIG. 3(b). Each leading edge of the signal pulse now leads as shown by dotted line in FIG. 3(d). Being apparent from the above description, the appearance timing of each trigger pulse is regulated in accordance with the voltage of the smoothed power on the line 19 so that the voltage of the output voltage at the terminal 20 is held substantially constant.

Referring now to FIG. 4, start operation of the converter 10 is discussed hereinbelow in conjunction with FIG. 2. When an a.c. voltage is applied to the input terminals 12 and 13, the capacitor C 3 begins to be charged by the voltage on the line 15, and the capacitor C 5 also begins to be charged through the resistors R 3 and R 7 . It is important that the time constant of power supply circuit constituted by the resistor R 3 and the capacitor C 3 is selected to be much larger than that of the time constant of another power supply circuit constituted by the resistor R 7 and the capacitor C 5 . Thus, the emitter potential of the transistor T 1 is built up more quickly than that of the transistor T 2 . Upon completion of the charging of the capacitor C 3 , the saw-tooth wave generator 22 begins to generate saw-tooth wave signal as shown in FIG. 4(b). Since the capacitor C 5 is, on the other hand, slowly charged, the emitter voltage of the transistor T 2 slowly rises as shown in FIG. 4(c), so that, the threshold voltage of the transistor T 2 gradually rises as shown by a dotted line in FIG. 4 (b). Accordingly, the trigger pulses is produced on the gate of the thyristor 16 as shown in FIG. 4(d), whereby the firing angle of the thyristor 16 is gradually reduced as shown in FIG. 4(a) which illustrates the voltage at the output terminal 14 of the rectifier 11. The output voltage on the output terminal 20 therefore gradually rise up as shown in FIG. 4(e). It is to be understood that since the output voltage of the converter 10 starts to gradually rise up as shown in FIG. 4(e), an undesired rush current is effectively suppressed.

FIG. 5 illustrates another form of the converter 10 which is arranged identically to the circuit arrangement of FIG. 1 except that an integrator 50 is interposed between the output of the saw-tooth wave generator 22 and the input of the trigger pulse generator 27. The integrator 50 includes a resistor R 12 having one terminal connected to the output of the saw-tooth wave generator 22 and the other terminal connected to the input of the rectangular pulse generator 27, and a capacitor C 7 having one terminal connected to the other terminal of the resistor R 12 and the other terminal grounded.

In operation, the saw-tooth wave generator 22 produces on its ouput a saw-tooth wave signal having decreasing exponential wave form portion as shown in FIG. 6 (a), although the saw-tooth wave signal ideally is illustrated in FIG. 3. This saw-tooth wave signal is converted by the integrator 50 into another form of saw-tooth wave having a increasing exponential wave form portion as shown in FIG. 6(b).

It should be noted that the saw-tooth wave signal of FIG. 6(a) has a smaller inclination near 180°. Hence, when the integrator 50 is omitted and the saw-tooth wave signal as shown in FIG. 6(a) is applied to the trigger pulse generator 27, the rate of change of the output voltage of the converter 10 become larger at a firing angle near to 180°. On the other hand, it is apparent from FIG. 6(c) that the rate of change the output voltage of the thyristor 16 with respect to the firing angle become large at a firing angle near to 180°. Therefore, the loop gain of the trigger pulse generator 24 increases when the firing angle of the thyristor 16 is near to 180°. It is apparent through a similar discussion that the loop gain of the trigger pulse generator 24 decreases when the firing angle is near to 90°. Such non-uniformity of the loop gain of the trigger pulse generator invites a difficulty of the regulation of the output voltage of the converter. It is to be noted that the saw-tooth wave signal shown in FIG. 6(b) has a large inclination at an angle near 180°. Therefore, when the saw-tooth wave signal of FIG. 6(b) is applied to the trigger pulse generator 24, the loop gain of the trigger pulse generator 24 is held substantially constant, whereby the output voltage of the converter is effectively held constant.

It is to be understood that the integrator 50 may be substituted for by a miller integrator and a bootstrap integrator. Furthermore, a plurality of integrator may be employed, if desired.

FIG. 7 illustrates another circuit arrangement of the converter according to the present invention, which is arranged identically to the circuit of FIG. 2 except for the trigger circuit 31 and the smoothing circuit 18.

The trigger circuit 31 of FIG. 7 comprises a transformer TR with primary and secondary coils. One terminal of the primary coil is connected to the resistor R 7 of the pulse generator 27. The other terminal of the primary coil is connected to a collector of a transistor T 4 of NPN type. The secondary coil has terminals respectively connected to the gate and cathode of the thyristor 16. An emitter of the transistor T 4 is grounded through a resistor R 13 . A base of the transistor T 4 is grounded through a resistor R 14 and connected through a capacitor C 8 to the collector of the transistor T 2 of the pulse generator 27.

The smoothing filter 18 of FIG. 7 comprises a choke coil CH connected to the lines 17 and 19, and to capacitors C 9 and C 10 which are in turn grounded. The circuit of FIG. 7 operates in the same manner as the circuit of FIG. 2.

Obviously many modifications and variations of the present invention are possible in the light of the above teachings. It is therefore to be understood that within the scope of the appended claims the invention may be practiced otherwise than as specifically described.


TBA920 line oscillator combination

DESCRIPTION
The line oscillator combination TBA920 is a monolithic
integrated circuit intended for the horizontal deflection of the black and white
and colour TV sets
picture tube.

FEATURES:
SYNC-PULSE SEPARATION
OPTIONAL NOISE INVERSION
GENERATION OF A LINE FREQUENCY VOL-
TAGE BY MEANS OF AN OSCILLATOR
PHASE COMPARISON BETWEEN SYNC-
PULSE AND THE OSCILLATOR WAVEFORM
PHASE COMPARISON BETWEEN THE OS-
CILLATOR WAVEFORM AND THE MIDDLE OF
THE LINE FLY-BACK PULSE
AUTOMATIC SWITCHING OF THE VARIABLE
TRANSCONDUCTANCE AND THE VARIABLE
TIME CONSTANT TO ACHIEVE NOISE SUP-
PRESSION AND, BY SWITCHING OFF, POS-
SIBILITY OF TAPE-VIDEO-REGISTERED RE-
PRODUCTION
SHAPING AND AMPLIFICATION OF THE OS-
CILLATOR WAVEFORM TO OBTAIN PULSES
FOR THE CONTROL OF DRIVING STAGES IN
HORIZONTAL, DEFLECTION CIRCUITS
USING EITHER TRANSISTORS OR THYRISTORS,

BU208(A)

Silicon NPN
npn transistors,pnp transistors,transistors
Category: NPN Transistor, Transistor
MHz: <1 MHz
Amps: 5A
Volts: 1500V
HIGH VOLTAGE CAPABILITY
JEDEC TO-3 METAL CASE.

DESCRIPTION
The BU208A, BU508A and BU508AFI are
manufactured using Multiepitaxial Mesa
technology for cost-effective high performance
and use a Hollow Emitter structure to enhance
switching speeds.

APPLICATIONS:
* HORIZONTAL DEFLECTION FOR COLOUR TV With 110° or even 90° degree of deflection angle.

ABSOLUTE MAXIMUM RATINGS
Symbol Parameter Value Unit
VCES Collector-Emit ter Voltage (VBE = 0) 1500 V
VCEO Collector-Emit ter Voltage (IB = 0) 700 V
VEBO Emitter-Base Voltage (IC = 0) 10 V
IC Collector Current 8 A
ICM Collector Peak Current (tp < 5 ms) 15 A
TO - 3 TO - 218 ISOWATT218
Ptot Total Dissipation at Tc = 25 oC 150 125 50 W
Tstg Storage Temperature -65 to 175 -65 to 150 -65 to 150 oC
Tj Max. Operating Junction Temperature 175 150 150 °C



A vertical deflection circuit for use in a television receiver, comprising a control circuit for stabilizing the width of a pulse either in a vertical oscillator circuit or between a vertical oscillator circuit and vertical output circuit to thereby stabilize the width of a pulse component included in the vertical deflection output signal.



1. A transformerless output vertical deflection circuit, comprising a vertical oscillator circuit for generating a vertical pulse train in response to vertical synchronizing pulses applied thereto, a sawtooth signal generator for generating a series of sawtooth signals, each cycle of said sawtooth signal including a pulse component, a vertical output circuit coupled to said sawtooth generator for amplifying said sawtooth signal including said pulse component and loading a vertical deflection coil, and stabilizing means connected between said vertical oscillator and said sawtooth signal generator for varying the width of the pulse component which is to be fed to said vertical output circuit in response to the average level of DC output voltage fed from the vertical output circuit. 2. A transformerless output vertical deflection circuit claimed in claim 1, wherein said stabilizing means comprises a control circuit means for receiving a series of pulses from the vertical oscillator and a feedback signal from the vertical output circuit and for varying the width of the pulse which is to be fed to the vertical output circuit in response to a DC control signal proportional to the width of the pulse component included in the vertical output signal and smoothing circuit means connected between said vertical output circuit and said stabalizing means for smoothing said feedback signal. 3. A transformerless output vertical deflection circuit claimed in claim 2, wherein said control circuit comprises a charging capacitor which is parallel to a transistor, said transistor being switched on in response to pulses fed from the vertical oscillator wherein said capacitor is charged by the voltage fed from said smoothing circuit, and discharged in response to conduction of the transistor, a differential amplifier circuit which receives the voltage on said capacitor and a fixed voltage, and a gating circuit for producing a pulse which has a width equal to the difference between the width of the pulse fed from the vertical oscillator circuit and the width of pulse fed from the differential amplifier circuit. 4. A transformerless output vertical deflection circuit claimed in claim 2, wherein said control circuit comprises a capacitor which is charged by a fixed power source and is discharged by means of a switching transistor operated by the pulses fed from the vertical oscillator circuit and a differential amplifier circuit receiving the voltage on the capacitor and the output of said smoothing circuit. 5. A transformable output vertical deflection circuit comprising a vertical oscillator for generating a vertical pulse train in response to vertical synchronizing pulses applied thereto, a sawtooth signal generator for generating a series of sawtooth signals each cycle of said sawtooth signal including a pulse component, a vertical output circuit for amplifying said sawtooth signal including said pulse component and loading a vertical deflection coil, and pulse stabilizing means coupled between the vertical oscillator circuit a
nd the sawtooth signal generator, said stabilizing means comprising a capacitor which is charged by a fixed power source and discharged by means of a discharging means operated in response to the vertical pulse fed from the vertical oscillator, a circuit means for generating a train of output pulses each starting at the time when the voltage appearing on the capacitor exceeds a predetermined value and terminating in synchronism with termination of the pulse fed from the vertical oscillator, and gating means for generating pulses having a width equal to the difference between the width of the pulse fed from the vertical oscillator and the width of the output pulse of the circuit means. 6. A transformerless output vertical deflection circuit, comprising a vertical oscillator circuit for generating a vertical pulse train in response to vertical synchronizing pulses applied thereto, a sawtooth signal generator for generating a series of sawtooth signals, each cycle of said sawtooth signal including a pulse component, a vertical output circuit coupled to said sawtooth generator for amplifying said sawtooth signal including said pulse component and loading a vertical deflection coil, and stabilizing means, comprising a control circuit connected between said vertical output circuit and said vertical oscillator circuit for varying the width of each pulse produced by the vertical oscillator circuit in response to a DC control signal having a value corresponding to the width of the pulse component applied to the vertical deflection coil of the vertical output circuit for controlling the pulse width of the output of said vertical oscillator circuit and thereby the pulse width of said pulse component.
Description:
BACKGROUND OF THE INVENTION
The present invention relates to a vertical deflection circuit for use in a television receiver and, more particularly, to a vertical deflection circuit of a type wherein no vertical output transformer is employed. This type of vertical deflection circuit with no output transformer is generally referred to as an OTL (Output Transformerless) type vertical deflection circuit.
It is known that variation of the pulse width of the flyback pulse produced in a vertical output stage of the vertical deflection circuit is the cause in the raster on the television picture tube, of a white bar, flicker, jitter, line crowding and/or other raster disorders. In addition thereto, in the vertical deflection output circuit where the output stage is composed of a single-ended push-pull amplifier having a vertical output transistor, an excessive load is often imposed on the output transistor and, in an extreme case, the output transistor is destroyed.

SINUDYNE COLOR 2202 (NORDMENDE COLOR 2921) CHASSIS FFS FIV (F4) (FCI25) Amplifier suitable for use as a color kinescope driver:

A color kinescope matrix amplifier has a first input coupled through a capacitor to a source of color difference signals. Another input is coupled to a source of luminance signals. The matrix amplifier includes a cascode output stage direct current coupled to a cathode of a kinescope. A portion of a direct voltage developed at the cascode output amplifier is coupled to one input of a comparator circuit. The other input of the comparator circuit is coupled to a temperature compensated direct voltage reference source. The comparator is rendered operative during horizontal retrace intervals to provide a current to either charge or discharge the input capacitor in accordance with the difference between the voltage at the output of the cascode output amplifier and the reference voltage to compensate for voltage variations at the output of the cascode amplifier due to power supply variations and the like. To compensate for droop caused by the discharge of the input capacitor during the scanning interval, one input of a differential amplifier is included between the input capacitor and the input of the cascode output stage. Negative signal feedback is provided from the output stage to the other input of the differential amplifier via a capacitor arranged to be charged during the horizontal retrace interval. The two capacitors discharge at substantially the same rates during the scanning interval. By virtue of the common mode operation of the differential amplifier droop effects are minimized.


1. In a tel
evision receiver including an image reproducing device, a source of chrominance signals, a source of luminance signals and a source of horizontal blanking pulses, said horizontal blanking pulses occurring during the time interval during which said image reproducing device is horizontally retraced, the apparatus comprising:
amplifying means for combining said chrominance signals and said luminance signals, said amplifying means including first and second input terminals and an output terminal, said output terminal being direct current coupled to said image reproducing device, said second input terminal being direct current coupled to said source of said luminance signals;
first capacitive means for coupling said chrominance signals to said first input terminal;
comparator means having first and second input terminals for comparing voltages applied thereto, said comparator means being normally inoperative;
a relatively low level stabilized reference voltage source coupled to said first input terminal of said comparator means;
means coupled to said second input terminal of said comparator means for providing a direct voltage proportional to the direct voltage developed at said output terminal;
means for selectively rendering said comparator operative in response to said horizontal blanking pulses; and
current converting means coupled to said comparator and to said first capacitive means for charging and discharging said capacitive means to a direct voltage level in relation to the difference in voltage between said first and second input terminals of said comparator means so as to counteract the changes of the voltage developed at said output terminal.
2. The apparatus recited in claim 1 wherein said amplifying means includes:
a differential amplifier having first and second input terminals and an output terminal, said first input terminal being coupled to sai
d first input terminal of said amplifying means, said output terminal of said differential amplifier being coupled to said output terminal of said amplifying means;
second capacitive means coupled to said second input terminal of said differential amplifier; and
means for selectively charging said second capacitive means during said horizontal retrace interval, said first and second capacitive means being selected to have substantially equal discharging rates during the time intervals between said horizontal retrace intervals.
3. The apparatus recited in claim 2 wherein said second capacitive means is coupled between said output terminal of said amplifying means and said second input terminal of said differential amplifier. 4. The apparatus recited in claim 3 wherein said amplifying means includes a cascode amplifier coupled between the output of said differential amplifier and said output terminal of said amplifying means. 5. The apparatus recited in claim 3 wherein said amplifying means includes first and second transistors, the emitter of said first transistor being direct current coupled to the collector of said second transistor, the base of said first transistor being coupled to said first input terminal of said amplifying means, the base of said second transistor being coupled to said second input terminal of said amplifying means, the emitter of said first transist
or being coupled to said first input terminal of said differential amplifier. 6. The apparatus recited in claim 3 wherein said means for selectively charging said second capacitive means includes means for clamping the second input terminal of said differential amplifier to a predetermined voltage during said horizontal retrace interval. 7. The apparatus recited in claim 3 wherein means are provided for adjusting the portion of the voltage developed at said output terminal of said amplifying means which is coupled to said second capacitive means. 8. The apparatus recited in claim 1 wherein said means coupled to said second input terminal of said comparator means for providing a direct voltage proportional to the direct voltage developed at said output terminal of said amplifying means includes means for adjusting the voltage coupled to said second input terminal of said comparator means. 9. The apparatus recited in claim 1 wherein said comparator means includes:
a differential amplifier having two input terminals and two output terminals, one of said input terminals being coupled to said reference voltage source, the other of said input terminals being coupled to said output terminal of said amplifier means; and
a current mirror circuit having an input and an output, one of said output terminals of said differential amplifier being coupled to said input terminal of said current mirror circuit, the other of said output terminals of said differential amplifier being coupled to the output of said current mirror circuit and to said first capacitor means.
10. The apparatus recited in claim 1 wherein said voltage reference source is temperature compensated. 11. In a television receiver including a color kinescope leaving a plurality of electron beam forming apparatus, a source of luminance signals, a source of a plurality of color difference signals, and a source of horizontal blanking pulses, said horizontal blanking pulses corresponding to the time interval during which said electron beams are horizontally retraced, the apparatus comprising:
a plurality of amplifiers, each of said amplifiers including
amplifying means for com
bining one of said plurality of color difference signals with said luminance signals, said amplifying means including first and second input terminals and an output terminal, said output terminal being direct current coupled to a respective one of said plurality of electron beam forming apparatus, said second input terminal being direct current coupled to said source of said luminance signals, capacitive means for coupling said one of said plurality of color difference signals to said first input terminal,
comparator means having first and second input terminals for comparing voltages applied thereto, said comparator means being normally inoperative,
means coupled to said second input terminal of said comparator means for providing a direct voltage proportional to the direct voltage developed at said output terminal,
means for selectively rendering said comparator operative in response to said horizontal blanking pulses, and
current converting means coupled to said comparator and to said capacitive means for charging and discharging said capacitive means to a direct voltage level in relation to the difference in voltage between said first and second input terminals of said comparator means so as to counteract the changes of the voltage developed at said output terminal; and a relatively low level stabilized reference
voltage source coupled to said first input terminals of each of said plurality of comparator means.
Description:
The present invention is directed to the field of amplifiers and is particularly directed to the field of amplifier arrangements utilized to drive color image reproducing devices such as kinescopes.
The electron guns of a color kinescope are typically driven by separate amplifier stages. Variations of the operating conditions of an amplifier stage, such as variations of the stage's supply voltage, tend to produce variations in the brightness of a reproduced image. Furthermore, because each of the stages tends to operate at different power dissipation levels the operating conditions of the stages vary with respect to each other and hence color imbalances may occur.
Athou
gh supply voltage regulators and high level clamping circuits have been employed in conjunction with kinescope amplifier stages to inhibit the aformentioned problems, it is desirable to provide kinescope driver amplifier arrangements which maintain their operating point stability with variations in operating conditions such as power supply variations without the need of supply voltage regulators or high level clamping circuits.
Furthermore, it is desirable, because of the trend toward miniaturization in electronic art, that at least a portion of the kinescope amplifier driver should be able to be constructed in integrated circuit form.
It is also desirable to provide kinescope driver amplifier arrangements which include independent controls for adjusting the DC level and the AC amplitude of the signals coupled to the kinescope. This is particularly desirable where "precision-in-line" kinescopes or the like, in which the electron guns have common control electrodes, are employed since, in these types of kinescopes, it is difficult to independently adjust the operating conditions associated with the respective guns because of the commonality of control electrodes.
Furthermore, it is desirable that a kinescope driver amplifier which is to be utilized with a precision-in-line type of kinescope provide a relatively wide bandwidth without the requirement of high frequency peaking coils. Peaking coils tend to be bulky. In addition, undesirable voltages may be developed across a peaking coil due to the large magnetic fields which may be produced by the yokes associated with a precision-in-line kinescope. These undesirable voltages may produce disconcerting brightness and/or hue changes.
In accordance with the present invention, one input terminal of amplifying means is coupled to a source of chrominance signals through capacitive means. A second input of the amplifying means is direct current coupled to a source of luminance signals. The output terminal of the amplifying means is direct current coupled to a color image reproducing device such as a precision-in-line kinescope of the like. The amplifying means includes means for combining the luminance and chrominance signals to provide the image reproducing device with color signals. The amplifying means also includes comparator means for comparing the voltage developed at the output terminal to a reference voltage to generate a current to control the charging of the capacitive means in a manner so as to counter-act the changes of the voltage developed at the output due, for example, to changes in the power supply voltage. The comparator means is arranged to be normally inoperative and is selectively rendered operative during the horizontal retrace interval.
In accordance with another aspect of the present invention, the amplifying means includes a differential amplifier having first and second input terminals and an output terminal. The output terminal of the differential amplifier is coupled to the output terminal of the amplifying means. The first input terminal of the differential amplifier is coupled to the input terminal of the amplifying means. The second input terminal of the differential amplifying means is coupled to a second capacitive means. Means are provided for selectively charging the second capacitive means during the horizontal retrace interval. The first and second capacitive means are selected to have substantially equal discharging rates so as to compensate for any decrease in the DC content (i.e., droop) at the output terminal of the amplifying means during the scanning interval.
In accordance
with still another feature of the present invention, the second capacitive means is coupled to the output terminal of the amplifying means in a manner so as to allow adjustment of the AC gain of the amplifying means. The DC conditions of the output of the amplifying means may be controlled by controlling the portion of the voltage developed at the output terminal coupled to the comparator means.
The present invention may best be understood by reference to the following detailed description and accompanying drawing which shows, partially in block diagram form and partially in schematic form, the general arrangement of a color television receiver employing a kinescope driver amplifier arrangement constructed in accordance with the present invention .
The color television receiver includes a video signal processing unit 141 responsive to radio frequency (RF) signals, received by an antenna, for receiving in a known manner, a composite video signal comprising chrominance, luminance, sound and synchronizing signal components.
The output of video processing unit 141 is coupled to a chrominance channel 142 including a chrominance processing unit 143 and a color demodulator 144. Chrominance processing unit 143 separates chrominance signals from the composite video signal. Color demodulator 144 derives signals of the appropriate polarity representing, for example, R-Y, G-Y and B-Y color difference signal information from the chrominance signals. The TAA630 integrated circuit or similar circuit is suitable for use as color demodulator 144.
The output of video processing unit 141 is also coupled to a luminance channel 145 including a luminance processing unit 146 which amplifies and processes luminance components of the composite signal to form an output signal of the appropriate polarity representing luminance, Y, information. A brightness control unit 147 to control the DC content of luminance signal Y and a contrast control unit 148 to control the amplitude of luminance signal Y are coupled to processing unit 146.
The composite video signal is also coupled to a sync separator 149 which, in turn, is coupled to a horizontal deflection unit 151 and a vertical deflection unit 152. Horizontal deflection unit 151 is also coupled to a high voltage unit 154 which generates operating voltages for kinescope 153. Outputs from horizontal deflection unit 151 and vertical deflection unit 152 are coupled to luminance pr
ocessing unit 146 to inhibit or blank luminance signal Y during the horizontal and vertical retrace intervals. Similarly, an output from horizontal deflection unit 151 may be coupled to chroma processing unit 143 or color demodulator 144 to inhibit the color difference signals during the horizontal retrace interval. Furthermore, first and second signals including positive going pulses, the pulses of each signal being coincident with the horizontal retrace or blanking interval, are coupled to matrix unit 100 to control its operation, as will appear below, via conductors 159 and 167, respectively.
The R-Y output signal and luminance signal Y are coupled to a matrix unit 100 where they are combined to form a color signal representing red (R) information. Similarly, the B-Y and G-Y color difference signals are respectively coupled to matrix-driver units 150 and 157, similar to the combination of matrix unit 100 and kinescope driver 199, where they are matrixed with luminance signal Y to produce color signals representing blue (B) and green (G) information. Since the matrix units for the various color difference signals are similar, only matrix unit 100 will be described in detail.
Matrix unit 100, enclosed within dotted line 160, is suitable for construction as an integrated circuit. The R-Y color difference signal is coupled through a capacitor 110 to the base of an NPN transistor 101 which is a
rranged as a common collector amplifier for color difference signals. Transistor 101, NPN transistor 102, resistors 178 and 184 form a summing circuit 161 for the color difference signal and luminance signal Y, the latter being direct current coupled to the base of transistor 102. The combined output of circuit 161, taken at the collector of transistor 102, is coupled to the base of an NPN transistor 105. Transistor 105 and an NPN transistor 106 form a differential amplifier 162 to which bias current is supplied from a current source including a suitably biased transistor 182. The output of differential amplifier 162, taken at the collector of transistor 105, is coupled through a level shifter, shown as the series connection of a zener diode 163, and a diode 165 to a kinescope 199. Bias current is provided for zener diode 163 and diode 165 through a resistor 183, which serves as the load resistor of transistor 105, and resistors 176 and 177.
Kinescope driver 199 comprises a cascode amplifier 164 including NPN transistors 120 and 119. The output of matrix unit 100 is coupled to the base of transistor 119 while a positive supply voltage (e.g. +12 volts) is coupled to the base of transistor 120. The output of kinescope driver 199, taken at the collector of transistor 120 is direct current coupled through a resistor 179 to the red (R) cathode of kinescope 153. The collector of transistor 120 is coupled to a source of supply voltage B+ through a load resistor 165. Supply voltage B+ is a relatively high voltage, typically, in the order of 200 to 300 vdc.
The collector of transistor 120 is also coupled to a series combination of a resistor 166 and a black level setting potentiometer 167, the latter being returned to ground. A direct voltage proportional to that at the collector of transistor 120 is developed at the wiper arm of potentiometer 167 and is coupled to one input of a voltage comparator circuit 168. Comparator 168 comprises NPN transistors 103 and 104 coupled as a differential amplifier. A second input of comparator 168, at the base of transistor 103, is coupled to a temperature compensated voltage reference (TCVR) unit 169. Voltage reference unit 169, which may, for example, be similar to that employed in the CA3085 integrated circuit manufactured by RCA Corporation, supplies a regulated reference voltage of approximately 1.6 vdc.
Voltage reference unit 169 is also coupled to the matrix portions of units 150 and 157 via conductor 155 so that a common reference voltage is coupled to the respective comparators of units 100, 150 and 157. It is noted that matrix unit 100 and the matrix portions of units 150 and 153 may be constructed as a single integrated circuit.
A current source including an NPN transistor 170 is coupled to the jointly connected emitters of transistors 103 and 104. The first horizontal blanking pulse signal generated by horizontal deflection unit 151 is coupled to the base of transistor 170 via conductor 159.
The output of differential amplifier 168 provided at the collector of NPN transistor 103 is converted to a bidirectional current by means of a current mirror circuit 180 comprising a diode-connected PNP transistor 172 and a PNP transistor 173. The collector of transistor 173 is coupled to the collector of transistor 104 and to the base of transistor 101.
The junction of resistors 166 and 167 is coupled to a signal feedback circuit comprising a series connection of a potentiometer 174 and a resistor 175. Feedback voltage developed at the wiper arm of potentiometer 174 is coupled through a capacitor 120 to the base of transistor 106 (i.e., one input of differential amplifier 162). The base of transistor 106 is returned to ground through resistor 181 and the collector-emitter junction of a transistor 108. The base of transistor 108 is coupled to horizontal deflection unit 151 to receive the first horizontal blanking pulse signal via conductor 159. An NPN transistor 107, the emitter of which is coupled to the base of transistor 106, is arranged together with resistor 181 and the collector-emitter junction of transistor 108 as an emitter follower. The base of transistor 107 is coupled to horizontal deflection unit 151 to receive the second horizontal blanking pulse signal via conductor 167. It is noted that this signal may also be generated within the IC device.
Kinescope 153 may be a precision-in-line kinescope such as the RCA type 15VADTCO1. As is described in U.S. Pat. No. 3,817,397, issued May 21, 1974, there is no provision for separate adjustment of red, green and blue gun screen and grid potentials and only the cathodes of the three guns of such a kinescope are available for separate adjustment of the cut off point of the guns. As will become apparent in the following description, matrix unit 100 and kinescope driver 199 are particularly suited to a kinescope of the precision-in-line type but it should be appreciated that they may be utilized for other types of kinescopes such as delta-gun, shadow mask or other slotted mask types.
In operation, the signal supplied to the base of transistor 107 during the scanning interval by horizontal deflection unit 151 is of sufficiently low amplitude (e.g., less than +4vdc) in relationship to the voltage at its emitter (controlled by the charge on capacitor 120 as will be explained) that it is non-conductive. Because of relatively low voltage applied to the bases of transistors 108 and 170 during the scanning interval, transistors 108, 170, 103 and 104 are also non-conductive and do not affect the operation of matrix circuit 100 during the scanning interval.
The signal -(R-Y), representing red color difference information, and the signal Y, representing luminance information, are coupled to amplifier 161 where they are combined in the emitter circuit of transistor 101 to form a signal -R, representing red information. The signal -R is further amplified and inverted twice by differential amplifier 162 and cascode amplifier 164 for application to kinescope 153.
It is noted that resistors 183, 176 and 177 should be selected so that zener diode 163 is biased well into its reverse breakdown region to inhibit noise.
The portion of the output signal of cascode amplifier 164 developed at the wiper arm of potentiometer 174, is capacitively fed back to one input of differential amplifier 162. This negative feedback arrangement, in conjunction with the use of cascode amplifier 199, provides for a relatively wide bandwidth, thereby eliminating the need for peaking coils or the like to improve high frequency response. The AC gain (or drive) of the matrix unit-kinescope driver arrangement may be adjusted by adjustment of the wiper arm of potentiometer 174 (normally a service or factory adjustment).
During the horizontal retrace interval, a relatively high voltage (e.g., approximately +6 vdc plus the base to emitter voltage of transistor 107 when transistor 107 is rendered conductive) is applied to the base of transistor 107 from horizontal deflection unit 151. Horizontal deflection unit 151 also applies a relatively high voltage to the bases of transistors 108 and 170. As a result transistors 107, 108, 170, 103 and 104 are rendered conductive and the base of transistor 106 is clamped to a voltage substantially equal to the voltage at the base of transistor 107 less the base emitter voltage of transistor 107 (e.g., +6 vdc). The voltage to which the base of transistor 106 is clamped is sufficiently lower than that at the base of transistor 105 so that transistor 106 will be rendered non-conductive and transistor 105 will be rendered fully conductive. Under these conditions, the voltage developed at the collector of transistor 120 will rise toward B+ to a voltage determined by t
he conduction of transistors 119 and 120 and the voltage division action of resistors 165, 166 and the impedance of potentiometer 167 in parallel combination with the series combination of potentiometer 174 and resistor 175.
While the base of transistor 106 is clamped to the voltage applied to the base of transistor 107 less the voltage developed between the base and emitter of transistor 107, the AC feedback provided by capacitor 120 is effectively disconnected and capacitor 120 is provided with a charging path including resistor 166 and a portion of potentiometer 174 by which it is rapidly charged to a voltage determined by the voltage at the emitter of transistor 107 and DC voltage developed at the collector of transistor 120.
The voltage developed at the wiper arm of potentiometer 167 is coupled to the base of transistor 104 and, during each horizontal retrace interval, is compared to the voltage developed at the base of transistor 103 by TCVR 169. A difference in voltage is converted by virtue of the current mirror configuration of transistors 172 and 173 into an error current at the junction of the collectors of transistors 104 and 173. The error current acts, depending on the relative levels at the bases of transistors 103 and 104, to charge or discharge capacitor 110.
Potentiometer 167 initially is adjusted to provide a voltage at the collector of transistor 120 sufficient to cut off the red gun of kinescope 153 when a black image signal is present. Therefore, it is desirable to select the values of resistors 165 and 166 and potentiometer 167 to ensure that the full range of black level control at the red cathode of kinescope 153 is available.
Matrix circuit 100 is arranged so that capacitor 110 will be charged or discharged in a manner to compensate for any change in B+. For example, if B+ decreases, the voltage developed at the base of transistor 104 will decrease relative to the stable reference voltage developed at the base of transistor 103. Therefore, the collector current of transistor 103 and the substantially equal currents flowing through the emitter-collector circuits of transistors 172 and 173 will increase, causing capacitor 110 to be charged. As a result, the voltage at the base of transistor 101 will increase, the voltage at the bas
e of transistor 105 will increase, the voltage at the collector of transistor 105 will decrease and the voltage at the collector of transistor 120 will increase.
It is noted that transistor 173 and transistor 104 operate in what may be termed a push-pull fashion in that the change in current flowing between the emitter and collector of transistor 173 is inversely related to the change in current flowing between the collector and the emitter of transistor 104. Thus, if the current flowing through the emitter-collector of transistor 104 increases, the current through the collector-emitter of transistor 173 decreases, so that capacitor 110 is discharged by the excess of current flowing through transistor 104 rather than being charged by current from transistor 173.
Thus, the feedback arrangement including TCVR 169 of matrix unit 100 adjusts the charge on capacitor 110 to compensate for, and therefore substantially eliminate, the effect on the direct voltage applied to the kinescope cathodes of variations in B+. Furthermore, it is noted that variations in other portions of the matrix amplifier driver arrangement (such as variations caused by temperature or component tolerance changes) affecting the DC conditions at the collector of transistor 120 will be compensated for by the arrangement in a similar manner.
The charge stored on capacitor 110 during the horizontal retrace interval serves to control the bias on cascode amplifier 164 during the succeeding scanning interval. It is noted that the charge on capacitor 110 is not affected by the color difference signals or luminance signals during the horizontal retrace interval, since these signals are arranged to be constant during the horizontal retrace interval.
After the horizontal retrace interval, transistors 103, 104, 170, 172, 173, 107 and 108 are rendered nonconductive (as previously described) and capacitors 110 and 120 begin to discharge. While capacitor 110 controls the bias voltage at the base of transistor 105, capacitor 120 controls the bias voltage at the base of transistor 106. Capacitors 110 and 120 and their associated discharging circuitry preferably are selected so that capacitors 110 and 120 discharge at substantially equal rates. The similar changes in voltage are applied to opposite sides of differential amplifier 162. The common mode rejection characteristics of differential amplifier 162 will prevent the discharging of capacitor 110 to be reflected in the DC conditions at the collector of transistor 120. This "droop" compensation feature provided by capacitor 120 in junction with differential amplifier 162 is desirable, since in its absence, capacitor 110 would have to be a relatively large value to prevent droop. This is especially undesirable if it is desired to construct matrix unit 100 as an integrated circuit because large currents, not compatible with integrated circuit technology, would be required to charge and discharge capacitor 110.
Typical values for the arrangement are shown on the accompanying drawing.
It should be noted that although the present invention has been described in terms of a particular configuration shown in the diagram, modifications may be made which are contemplated to be within the scope of the invention. For instance, cascode driver 199 may be placed with other driver stages well known in the art. Furthermore, the current mirror configuration comprising transistors 172 and 173 may be modified in accordance with other known current mirror configurations.


SINUDYNE COLOR 2202 (NORDMENDE COLOR 2921) CHASSIS FFS FIV (F4) (FCI25) CONTACTLESS TOUCH SENSOR PROGRAM CHANGE KEYBOARD CIRCUIT ARRANGEMENT FOR ESTABLISHING A CONSTANT POTENTIAL OF THE CHASSIS OF AN ELECTRICAL DEVICE WITH RELATION TO GROUND :




Circuit arrangement for establishing a reference potential of a chassis of an electrical device such as a radio and/or TV receiver, such device being provided with at least one contactless touching switch operating under the AC voltage principle. The device is switched by touching a unipole touching field in a contactless manner so as to establish connection to a grounded network pole. The circuit arrangement includes in combination an electronic blocking switch and a unidirectional rectifier which separates such switch from the network during the blocking phase.


1. A circuit arrangement for establishing, at the chassis of an electrical device powered by a grounded AC supply network, a reference potential with relation to ground, said device having at least one contactless touching switch operating on the AC voltage principle, the switch being operated by touching a unipole touching field in a contactless manner, said arrangement comprising an electronic switch for selectively blocking the circuit of the device from the supply network, a half-wave rectifier including a pair of diodes individually connected in series-aiding relation between the terminals of the supply network and the terminals of the device for separating the electronic blocking switch from the supply network during a blocking phase defined by a prescribed half period of the AC cycle, and a pair of condensers individually connected in parallel with the respective diodes. 2. A circuit arrangement according to claim 1, wherein the capacitances of the two condensers are of equal magnitude.
Description:
This invention relates to a circuit arrangement for establishing a constant reference potential on the chassis of an electrical instrument such as a radio and/or a TV receiver. Such instrument includes at least one contactless touching switch operating under the AC voltage principle, whereby by touching a single pole touching field the contactless switch is operated.

In electronic devices, for example TV and radio receivers, there are used in ever increasing numbers electronic touching switches for switching and adjusting the functions of the device. In one known embodiment of this type of touching switch, which operates on a DC voltage principle, the function of the electronic device, is contactlessly switched by touching a unipole touching field, the switching being carried out by means of an alternating current voltage. When using such a unipole touching electrode, one takes advantage of the fact that the AC current circuit is generally unipolarly grounded. In order to close the circuit by touching the touching surface via the body of the operator to ground, it is necessary to provide an AC voltage on the touching field. In one special known embodiment there is employed a known bridge current rectifier for the current supply. This type of arrangement has the drawback that the chassis of the device changes its polarity relative to the grounded network pole with the network frequency. With such construction considerable difficulties appear when connecting measuring instruments to the device, such difficulties possibly eventually leading to the destruction of individual parts of the electronic device.

In order to avoid these drawbacks, the present invention provides a normal combination of a unidirectional rectifier with an electronic blocking switch that separates the chassis of the electronic device from the network during the blocking phase. In accordance with the present invention, the polarity of the chassis of the electronic device does not periodically change, because the electronic device is practically separated from the network during the blocking phase of the unidirectional rectifier by means of the electronic blocking switch.

In a further embodiment of the invention a further rectifier is connected in series with the unidirectional rectifier in the connection between the circuit and the negative pole of the chassis. Such further rectifier is preferably a diode which is switched in the transfer direction of the unidirectional rectifier. According to another feature of the invention there are provided condensers, a respective condenser being connected parallel with each of the rectifiers. Preferably the two condensers have equal capacitances. Because of the use of such condensers, which are required because of high frequency reasons, during the blocking phase there is conducted to the chassis of the electronic device an AC voltage proportional to the order of capacitances of the condensers. Thus there is placed upon the touching field in a desired manner an AC voltage, and there is thereby assured a secure functioning of the adjustment of the device when such touching occurs.

In the embodiment of the invention employing two rectifiers there is the further advantage that over a bridging over of the minus conduit of the rectifier that is connected between the network and the negative pole of the chassis connection, no injuries can be caused by a measuring instrument in the electronic device itself and in the circuit arrangement connected thereto.

In the accompanying drawing:

The sole FIGURE of the drawing is a circuit diagram of a preferred embodiment of the invention.

In the illustrated embodiment the current supply part of the device, shown at the left, is connected via connecting terminals A and B to an AC voltage source, the terminal B being grounded at 8. The current supply part consists of a unidirectional rectifier in the form of a diode 1 with its anode connected to the terminal I, the cathode of diode 1 being connected to one input terminal 9 of an electronic device 2. In the device 2 there is also arranged a sensor circuit 3, shown here mainly as a block, circuit 3 being shown as including a pnp input transistor the emitter of which is connected to an output terminal 11 of the device 2. The collector of such transistor is connected to the other output terminal 12 of the device 2. The base of the transistor is connected by a wire 13 to a unipolar touching field 4 which may be in the form of a simple metal plate instead of the pnp transistor shown, the sensor circuit itself may consist of a standard integrating circuit which controls, among other things, the periodic sequential switching during the touching time of the touching field 4. All of the circuits of the electronic device 2 are isolated in a known manner from the chassis potential. Between the network terminal B and the negative pole 10 of the chassis there is arranged in the direction opposite that of diode 1 a further diode 5, the anode of diode 5 being connected to the terminal 10, and the cathode of diode 5 being connected to the terminal B of the current supply. To provide for HF type bridging of the diodes 1 and 5 there are arranged condensers 6 and 7 respectively, which are connected in parallel with such diodes.

The invention functions by reason of the fact that in an AC network separate devices radiate electromagnetic waves which produce freely traveling fields in the body of the person who is operating and/or adjusting the device, thereby producing an alternating current through his body to ground, as indicated by the - line at the right of the circuit diagram. If now the person operating the device touches the switching field 4, then the pnp type input transistor of the sensor circuit 3, which is placed on a definite reference potential (for example 12 Volts) and is connected with the negative halfwave of the AC voltage potential, is made conductive. There is thereby released a control command in the sequential switching, for example, for switching the electronic device to the next receiving channel. It is understood that the most suitable connection is formed between ground and the touching field 4 by means of a wire. By the use of such wires it would be assured that in all cases the base of the transistor in circuit 3 is connected to ground. This would, however, not permit anyone to operate the switch without the use of an auxiliary means such as a wire. It will be assumed that the touching almost always results directly via the almost isolated human body. For this reason the AC current fields are necessary, because otherwise there cannot always be provided a ground contact. Thus this connection is established via the body resistance of the person carrying out the touching of the switch.

The positive half wave of the alternating current travels to the terminal 9 of the electronic device 2 after such current has been rectified and smoothed by the devices 1, 6. Such positive halfwave is also conducted to the sensor circuit 3. The thus formed current circuit is closed by way of the chassis of the electronic device 3, the diode 5, and the terminal B. When there is a negative halfwave of the alternating current delivered by the current supply, both diodes 1 and 5 remain closed so that the chassis of the device 2 remains separated from the network during the blocking phase. Nevertheless, by means of condensers 6 and 7 the chassis is placed in a definite network potential, which depends on the relationship of the order of magnitude of the two condensers 6 and 7. When the capacitances of such condensers are equal, there is placed upon the chassis of the device 2 the constant reference potential, and simultaneously there is present via the sensor circuit 3 the required AC voltage at the touching field 4 for adjusting the function or functions of the device 2 upon the touching of the touching field 4.

The reference character 15 indicates a terminal or point at which the potential of the chassis of the device 2 may be measured. As above explained, the diode 5 causes the potential of the chassis at 15 to be separated from the network ground when a negative AC halfwave arrives. It will be noted that the return conduit of the circuit is held at a fixed chassis potential. The input transistor of the sensor circuit 3 remains, however, locked because it is subjected to a DC current of about 12 volts. If now, by means of touching the touching field 4, the chassis potential is connected to ground, then the transistor switches through and releases a switching function.

If the connecting terminals AB of the current source are exchanged, as by changing the plug, then there is still secured the condition that the chassis of the device is separated from the network ground via the diode, in this case the diode 1. The reference potential of the chassis consequently remains constant and the changing AC fields which are superimposed on the condensers can produce in the touching human body an AC current voltage due to the fields which are radiated by the device.

A suitable sensor which may be employed for the circuit 3 herein may be a sensor known as the "SAS 560 Tastatur IS," manufactured and sold by Siemens AG.

It is to be understood that the present invention is not limited to the illustrated environment. They can also be used in electronic blocking switch including a Thyristor circuit, which in the same manner separates the electronic device during the blocking phase from the network rectifier. With such Thyristor circuit the drawbacks described in the introductory portion of the specification of known circuit arrangements are also avoided.

Although the invention is illustrated and described with reference to a plurality of preferred embodiments thereof, it is to be expressly understood that it is in no way limited to the disclosure of such a plurality of preferred embodiments, but is capable of numerous modifications within the scope of the appended claims.


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