It's a TDA4600
(SIEMENS) BASED SMPS Circuit which delivers +151V and St-by supply of +20V also used for sound amplifier.
Siemens Co., "Switched-Mode Power Supplies Using the TDA 4600", 2/81, pp. 6-15.
Funkschau 1975, "Ein Sperrwandler-Netzmodul mit Netztrennung", Dangschat et al., pp. 40 to 43.
"Schaltnetzteile", by Wuestehube et al., pp. 182 to 183.
"Schaltnetzteile mit der IS TDA 4600", Siemens publication, pp. 6 and 7.
What is a Flyback Transformer?
The
low cost, simplicity of design and intrinsic efficiency of flyback
transformers have made them a popular solution for power supply designs
of below 100W to 150W. Other advantages of the flyback transformer over
circuits with similar topology include isolation between primary and
secondary and the ability to provide multiple outputs and a choice of
positive or negative voltage for the output.
Flyback transformer, or,
line output transformers are a part of the power supplies in cathode
ray tubes. The flyback transformer generates a high voltage, as needed
by the CRT display or similar devices (e.g. plasma lamps). A flyback
transformer generates a voltage between a few kilovolts to 50 kilovolts
and uses high frequency switched currents between 17 kHz and 50 kHz.
The
chief difference between a flyback transformer and main/audio
transformer is that flybacks transfer as well as store energy, for a
just a fraction of an entire switching period. The secret behind that is
the coil winding on a ferrite core that has an air gap; it increases
the magnetic circuit reluctance for storing the energy.
The
reason it is called a flyback transformer is because the primary winding
uses a relatively low-voltage saw-tooth wave. The wave gets
strengthened first and then gets switched off abruptly; this causes the
beam to fly back from right to left on the display.
Applications
Cathode ray tube.
Televisions.
Plasma Lamps.
Any display requiring high voltage to operate and much more.
A
very compact CHASSIS THE GRUNDIG CUC70KT for that era and was used even
for models of 26 Inches (66Cm) sets with different wooden cabinet
colors types.
Highly and friendly serviceability we can see here, and it's not highly complex.
Siemens "Switched-Mode Power Supplies Using the TDA 4600":
A switching mode power supply (SMPS) may be used as an apparatus for
supplying power to electronic products. The SMPS converts input
alternating current (AC) voltages and outputs static voltages to operate
electronic products.The present invention relates to a switched-mode power supply. Such a
switched-mode power supply operates on the flyback converter principle,
in which a switching transistor is switched through during a switched-on
phase and magnetization is in consequence built up in a transformer,
and the switching transistor is switched off during a switched-off phase
and the magnetization is dissipated again via coupled windings of the
transformer.
In a typical switch mode power supply (SMPS) of a television receiver,
for example, the AC mains supply voltage is coupled directly to a bridge
rectifier for producing an unregulated direct current (DC) input supply
voltage that is, for example, ref
erenced to a common conductor, referred to as "hot" ground, and that is conductively isolated from the cold ground conductor. A pulse width modulator controls the duty cycle of a chopper transistor switch that applies the unregulated supply voltage across a primary winding of an isolating flyback transformer. In principle a switched-mode power supply comprises at least the following components: a switch, an inductor, a rectifier, capacitor and a load. The load may be considered as a resistance which is in parallel with the capacitor. During the part of the period in which the switch conducts a current originating from the input voltage source passes through the inductor so that energy which is derived from this source is stored in the inductor. During the other part of the period, in which the switch is not conducting, the energy stored in the inductor produces a current through the rectifier which current recharges the capacitor and, consequently, replenishes the energy losses caused by the load. By the adjustment or control of the conducting period of the switch relative to the cycle, the output D.C. voltage across the load can be independent of variations of the input D.C. voltage, for example, it can be kept constant. Such variations are caused by, for example, fluctuations in the electric AC supply where the input voltage is derived therefrom by rectification. A flyback voltage at a frequency that is determined by the modulator is developed at a secondary winding of the transformer and is rectified to produce a DC output supply voltage such as a voltage B+ that energizes a horizontal deflection circuit of the television receiver. The primary winding of the flyback transformer is, for example, conductively coupled to the hot ground conductor. The secondary winding of the flyback transformer and voltage B+ may be conductively isolated from the hot ground conductor by the hot-cold barrier formed by the transformer. Such a switched mode power supply is generally called SMPS. A SMPS as it is commonly used, for example, in consumer devices like television receivers, video recorders, audio equipments etc. generally includes a main switching transistor connected in series with the primary winding of a transformer, a base drive circuit for periodically switching said switching transistor between ON and OFF, and a control circuit for controlling the base drive current for said main switching transistor in such a way that output voltages derived from several secondary windings of said transformer are stabilized.On the other hand such a SMPS generally includes a protection circuit for case of overloading or a short circuit or any other failures within the operating voltages. Said protection circuit is needed since without protection means the collector-emitter current of the main switching transistor can reach excessively high values in case of a failure which might damage said switching transistor or cause any other damages of circuit components.
For some uses a D.C. isolation between the input voltage source and
the output voltage is absolutely required. This is the case, for
example, with power supplies of television receivers especially where it
is desirable to connect additional apparatus to the receiver, such as,
for example, video storage devices or television game circuits. A
switched-mode power supply is eminently suitable for this purpose as the
transformer which must effect that isolation passes signals which
usually have a much higher frequency, for example 15 to 20 kHz, than
those of the electric AC supply source so that said transformer may be
relatively small in size.
With a switched-mode power supply of
the flyback converter type the inductor of the converter can be
implemented in a simple manner as a transformer. A primary winding
thereof is connected in series with the switch between the terminals of
the input voltage source whereas a secondary winding is in series with
the rectifier. The publication "Philips Application Information" 472:
"properties of d.c.-to d.c. converters for switched-mode power supplies"
of Mar. 18, 1975 describes such a circuit. Of the three types the
flyback converter has the best control properties which is evidenced by
the formula which expresses the output voltage as a function of the
input voltage and of the ratio of the time of conduction of the switch
to the entire cycle. However, it should be noted that the entire energy
which is supplied to the load by a flyback converter must be passed on
by the transformer which imposes higher requirements both on the
transformer and, particularly, on the storage capacity thereof as well
as on the switch.
HERE PICTURED
It's a TDA4600 (SIEMENS) BASED SMPS Circuit which delivers +151V and 12 VOLT St-by supply.
SIEMENS TDA 4600-2 TDA 4600-2D
ControlIer for Switched-Mode Power Supplies / BipolarlC
In addition to their use with TV receivers and video recorders, the ICs TDA 4600-2 and TDA 4600-2 D can be applied in power supplied of hi-fi sets and active speakers due to their wide operational ranges and superior voltage stability during high load changes.
Features
• Direct driving of switching transistor
• Low start-up current
• Reversing linear overload characteristic
• Collector current - proportional to base-current input
TDA 4600-2 Circuit description
During start-up, normal and overload operations the TDA 4600-2; or -2D regulates, controls and protects the switching transistor installed in the flyback converter power supplies.
I) Start-up operation
The start-up operation is divided into three consecutive phases: 1. An internal reference voltage is built up which supplies the voltage regulator and effects the charging of the coupling electrolytic capaCitor and the switching transistor. During these procedures an 19 current less than 3.2 mA will be maintained, if the supply voltage Vg does not exceed", 12 V. 2. At Vg '" 12 V an internal reference voltage V1 = 4 V is suddenly released to provide all IC components with the exception of the control logic with a thermally stable and overload-resistant current. 3. In concurrence with the release of the reference voltage the control logic is activated by an additional stabilization circuit, and the IC is now ready for operation. Above sequential start-up phases ensure the charging of the switching transistor by the coupling electrolytic capacitor and subsequent precision switching.
II) Normal operation
Zero passages of the feedback coil are registered at pin 2 and forwarded to the control logic. At pin 3 (input control, overload, and standby recognition) the rectified amplitude variations of the feedback coil are applied. The regulating (control) amplifier operates with an input voltage of about 2 V and a current of about 1.4 mA. According to the internal reference voltage, ttie operating region of the regulating amplifier will be defined by the collector current simulation pin 4 and the overload recognition. The simulation of the collector current is generated by an external RC network at pin 4 and an internally set voltage level. By increasing the capacitance (10 nFl, the collector current of the switching transistor is increased as well and establishes the desired control range. The control range extends between a 2 V clamped dc voltage and an ac voltage rising as a sawtooth wave, which may vary up to a maximum amplitude of 4 V (reference voltage). By reducing the secondary load to 20 W, the switching frequency increases to about 50 kHz at an almost constant pulse duty factor (on-time to period approx. 1/3). During additional secondary load reduction to about 1 W, the switching frequency will change to approx. 70 kHz, while the pulse duty factor falls to approx. 1/11. At the same time, the collector peak current falls below 1 A. The output level of the regulating (control) amplifier, the overload recognition, and the collector current simulation are compared in the trigger and the control logic is instructed accordingly. Pin 5 will provide additional blocking alternatives, i.e. the output at pin 8 is blocked at a voltage of equal to or less than 2.2 Vat pin 5. Based on the start-up circuit, the zero crossing identification, and the trigger-activated release, the control logic flipflops are set which control both the base current amplification and shut-down. The base current amplifier forwards the sawtooth voltage V 4 to pin 8. Also, a current feedback with an external resistance of R "" 0.68 Q is inserted between pin 8 and pin 7. The resistance value determines the maximum amplitude of the base current for the switch ing transistor. III) Safety features The base current shut-down, released by the control logic, clamps the output of pin 7 at 1.6 V and thus blocks the driving of the switching transistor. This preventive method will go into effect, if the voltage at pin 9 falls below typo 7.4 V or if voltages of less than typo 22 V are present at pin 5. In case of short-circuited secondary windings in the SMPS, the fault condition will be continuously monitored by the IC. With the load completely removed from the secondary winding in the SMPS, the IC is set at a small pulse duty factor. The total power consumption of the SMPS is kept below n = 6 to 10 W during both operating conditions. After the output has been blocked at a supply voltage Vg of less than or equal to typo 7.4 V, an additional voltage reduction of .1V g = 0.6 V will switch off the reference voltage (4 V).
Thermal resistance (only applicable to TDA 4600-2 D)
Standardized, ambience-related thermal resistance Rth JA 1 versus lateral length 1of a square
copper-clad cooling area (35 IJ.m copper lamination).
Rth JA
(I = 0) = 60 K/W
Tamb:S;; 70°C
Pv= 1 W
PCB in Ifertical position circuit in vertical position static air.
Measurement diagram for overload operations
Pin configuration
(TDA 4600-2: Plastic Power Package - 9 pin SIP package)
(TDA 4600-2D: Plastic 18 pin DIP package)
Pin No,
Function
1 Vre! output
2 Zero passage identification
3 Input regulating amplifier, overload amplifier
4 Collector current simulation
5 Possible connection for additional protective circuit
6 Ground
7 DC voltage output for charging the coupling capacitor
8 Pulse output - driving the switching transistor
9 Current supply input
only applicable to TDA 4600-2 D
10
11
12
13
14
interconnected (ground)
15
16
17
18
siemens TDA 4601/TDA 4601 D
Controller for Switched-Mode Power Supplies
During start-up, normal and overload operations the TDA 4601 or TDA 46010 regulates, controls and protects the switching transistor installed in the flyback converter power supplies. It also protects the complete SMPS by preventing an increase in the secondary voltage in case of errors. In addition to their use with TV receivers and video recorders, these ICs can be applied in power supplies of hi-fi sets and active speakers due to their wide operatiGnal ranges and superior voltage stability during high load changes.
Features
• Direct driving of switching transistor
* Low start-up current
• Reversing linear overload characteristic
* Collector current - proportional to base-current input
• Protective circuit for the event of errors
siemens TDA 4601/TDA 4601 D Circuit description
During start-up, normal, overload, and disturbed operations the TDA 4601 10 regulates, controls and protects the switching transistor installed in the flyback converter power supplies. If an error occurs, the driving of the switching transistor is blocked and the voltage on the secondary side is prevented from increasing.
I) Start-up operation
The start-up operation is divided into three consecutive phases: 1. An internal reference voltage is built up which supplies the voltage regulator and effects the charging of the coupling electrolytic capacitor and the switching transistor. During these procedures an Ig current less than 3.2 mA will be maintained, if the supply voltage V g does not exceed '" 12 V. 2. At V g '" 12 V an internal reference voltage V1 = 4 V is suddenly released to provide all IC components with the exception of the control logic with a thermally stable and overload-resistant current. 3. In concurrence with the release of the reference voltage the control logic is activated by an additional stabilization circuit, and the IC is now ready for operation. Above sequential start-up phases ensure the charging of the switching transistor t)y the coupling electrolytic capacitor and subsequent precision switching.
II) Normal operation
Zero passages of the feedback coil are registered at pin 2 and forwarded to the control logic. At pin 3 (input control, overload, and standby recognition) the rectified amplitude variations of the feedback coil are applied. The regulating (control) amplifier operates with an input voltage of about 2 V and a current of about 1.4 mAo According to the internal reference voltage, the operating region of the regulating amplifier will be defined by the collector current simulation pin 4 and the overload recognition. The simulation of the collector current is gen~rated by an external RC network at pin 4 and an internally set voltage level. By increasing the capacitance (10 nFl, the collector current of the switching transistor is increased as well and establishes the desired control range. The control range extends between a 2 V clamped dc voltage and an ac voltage rising as a sawtooth wave, which may vary up to a maximum amplitude of 4 V (reference voltage). By reducing the secondary load to 20 W, the switching frequency increases to about 50 kHz at an almost constant pulse duty factor (on-time to period approx. 1/3). During additional secondary load reduction to about 1 W, the switching frequency will change to approx. 70 kHz, while the pulse duty factor falls to approx. 1/11. At the same time, the collector peak current falls below 1 A. The accordingly. collector blocked output at current a level vciltage Pin 5 simulation of will of the equal provide regulating are to or additional compared less (control) than blocking in Vref the amplifier, /2 trigger - 0.1 alternatives, V the and at pin overload the 5.
Based on the start-up circuit, the zero crossing identification, and the trigger-activated release, the control logic flipflops are set which control both the base current amplification and shut-down. The base current amplifier forwards the sawtooth voltage V 4 to pin 8. Also, a current feedback with an external resistance of R '" 0.68 Q is inserted between pin 8 and pin 7. The resistance value determines the maximum amplitude of the base current for the switching transistor.
III) Safety features
The base current shut-down, released by the control logic, clamps the output of pin 7 at 1.6 V and thus blocks the driving of the switching transistor. This preventive method will go into effect, if the voltage at pin 9 falls below typo 6.7 V or if voltages of equal to or less than Vref/2 - 0.1 V are present at pin 5. In case of short-circuited secondary windings in the SMPS, the fault condition will be continuously monitored by the IC. With the load completely removed from the secondary winding in the SMPS, the IC is set at a small pulse duty factor. The total power consumption of the SMPS is kept below n = 6 to 10 W during both operating conditions. After the output has been blocked at a supply voltage V g of less than or equal to typo 6.7 V, an additional voltage reduction of L1 Vg = 0.6 V will switch of the reference voltage (4 V). Protective operation for faults with pin 5 For protection against disturbances such as primary undervoltages and/or secondary overvoltages (e.g. as a result of alterations in the parameters of components of the SMPS), it is possible to implement applications of the following kind:
• Protective operation with periodic sampling In the event of the fault condition, falling below the protective threshold V5 of typically V1/2 causes the output pulses on pin 8 to be inhibited and pin 5 to be clamped internally to ground across typically 300 Q. The current consumption of the IC reduces (Ig :2: 14 rnA for V g = 10 V). . With a suitably high-impedance starting resistor') the supply voltage Vg then falls below the minimal turn-off threshold (5.7 V) for the reference voltage V1. As a result V1 is turned off and the blocking of pin 5 is cancelled. Because of the renewed reduction in the current consumption of the IC (Ig S 3.2 rnA for Vg s 10 V) the supply voltage can again climb to the turn-on threshold Vg :2: 12.3 V. The protective threshold on pin 5 is released and the switched-mode power supply attempts to turn on. If the same fault is still present or another (V5 s V 1I2 - 0.1 V), the turn-on will be interrupted by the above, periodic protective operation, i.e. pin 8 is disabled, pin 5 is blocked, Vg falls off, etc .
• Protective operation with capture circuit The starting resistor on pin 9 is chosen sufficiently low-impedance so that in the event of a fault Vg does not fall below the maximum turn-off threshold (7.5 V) for V1• The blocking of pin 5 is preserved because V 1 will not have been turned off. A one-time fault is thus captured and turning the SMPS on again is not possible, for example, until the supply voltage has been manually turned off (power switch). In the designing of the starting resistor it should be considered that in protective operation the current consumption reduces to 19 :s; 28 mA for Vg = 10 V. IV) Turn-on in wide-range power supply (90 to 270 Vac) (application circuit 2) Free-running flyback converters used as wide-range power supplies call for a power supply to the TDA 4601 that is independent of the rectified line voltage, thus the sense of the winding 11/13 corresponds to the secondary side of the flyback-converter transformer. Turning on is hampered by the fact that the TDA 4601 must be supplied by the start-up circuit until the entire load secondary side is charged. This leads to long turn-on times, especially with a low line voltage. If the special start-up circuit is used (marked by dashed lines) this time can be shortened. The unregulated phase of the feedback control winding 15/9 is used as a turn-on aid. The transistor T1 blocks after turn-on, when the winding 11/13 has taken over the power supply to the TDA 4601, thus eliminating any effects on the control circuit during operation.
(TDA 4601: Plastic Power Package - 9 pin SIP package)
Pin configuration
(TDA 4601D: Plastic 18 pin DIP package)
Pin No. Function
1 Vref output
2 Zero-passage identification
3 Input regulating amplifier, overload amplifier
4 Collector-current simulation
5 Possible connection for additional protective circuit
6 Ground (rigidly connected to substrate mounting plate)
7 DC voltage output for charging the coupling capacitor
8 Pulse output, driving the switching transistor
9 Power supply
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| tda4601 block diagram |
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| Test and measurement circuit 2 Application circuit 1 |
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| SIEMENS TDA4601 Application circuit 2 |
Thermal resistance (only applies to TDA 4601 D)
Standardized, ambience-related thermal resistance Rth JA 1 versus lateral length I of a square
copper-clad cooling area (35 IJom copper lamination).
Rth JA (I = 0) = 60 K/W
Tamb ~ 70°C
Pv=1 W
PCB in vertical position
circuit in vertical position
static air
Power supply Description based on TDA4601d (SIEMENS)
TDA4601 Operation. * The TDA4601 device is a single in line, 9 pin chip. Its predecessor was the TDA4600 device, the TDA4601 however has improved switching, better protection and cooler running. The (SIEMENS) TDA4601 power supply is a fairly standard parallel chopper switch mode type, which operates on the same basic principle as a line output stage. It is turned on and off by a square wave drive pulse, when switched on energy is stored in the chopper transformer primary winding in the form of a magnetic flux; when the chopper is turned off the magnetic flux collapses, causing a large back emf to be produced. At the secondary side of the chopper transformer this is rectified and smoothed for H.T. supply purposes. The advantage of this type of supply is that the high chopping frequency (20 to 70 KHz according to load) allows the use of relatively small H.T. smoothing capacitors making smoothing easier. Also should the chopper device go short circuit there is no H.T. output. In order to start up the TDA4601 I.C. an initial supply of 9v is required at pin 9, this voltage is sourced via R818 and D805 from the AC side of the bridge rectifier D801, also pin 5 requires a +Ve bias for the internal logic block. (On some sets pin 5 is used for standby switching). Once the power supply is up and running, the voltage on pin 9 is increased to 16v and maintained at this level by D807 and C820 acting as a half wave rectifier and smoothing circuit. PIN DESCRIPTIONS Pin 1 This is a 4v reference produced within the I.C. Pin 2 This pin detects the exact point at which energy stored in the chopper transformer collapses to zero via R824 and R825, and allows Q1 to deliver drive volts to the chopper transistor. It also opens the switch at pin 4 allowing the external capacitor C813 to charge from its external feed resistor R810. Pin 3 H.T. control/feedback via photo coupler D830. The voltage at this pin controls the on time of the chopper transistor and hence the output voltage. Normally it runs at Approximately 2v and regulates H.T. by sensing a proportion of the +4v reference at pin 1, offset by conduction of the photo coupler D830 which acts like a variable resistor. An increase in the conduction of transistor D830 and therefor a reduction of its resistance will cause a corresponding reduction of the positive voltage at Pin 3. A decrease in this voltage will result in a shorter on time for the chopper transistor and therefor a lowering of the output voltage and vice versa, oscillation frequency also varies according to load, the higher the load the lower the frequency etc. should the voltage at pin 3 exceed 2.3v an internal flip flop is triggered causing the chopper drive mark space ratio to extend to 244 (off time) to 1 (on time), the chip is now in over volts trip condition. Pin 4 At this pin a sawtooth waveform is generated which simulates chopper current, it is produced by a time constant network R810 and C813. C813 charges when the chopper is on and is discharged when the chopper is off, by an internal switch strapping pin 4 to the internal +2v reference, see Fig 2. The amplitude of the ramp is proportional to chopper drive. In an overload condition it reaches 4v amplitude at which point chopper drive is reduced to a mark-space ratio of 13 to 1, the chip is then in over current trip. The I.C. can easily withstand a short circuit on the H.T. rail and in such a case the power supply simply squegs quietly. Pin 4 is protected by internal protection components which limit the maximum voltage at this pin to 6.5v. Should a fault occur in either of the time constant components, then the cho
pper transistor will probably be destroyed. Pin 5 This pin can be used for remote control on/off switching of the power supply, it is normally held at about +7v and will cause the chip to enter standby mode if it falls below 2v. Pin 6 Ground. Pin 7 Chopper switch off pin. This pin clamps the chopper drive voltage to 1.6v in order to switch off the chopper. Pin 8 Chopper base current output drive pin. Pin 9 L.T. pin, approximately 9v under start-up conditions and 16v during normal running, Current consumption of the I.C. is typically 135mA. The voltage at this pin must reach 6.7v in order for the chip to start-up.
The integrated circuit TDA 4601/D is designed for driving, controlling and protecting the switching transistor in self-osci"ating flyback converter power supplies as we" as for protecting the overall power supply unit. In case of disturbance, the rise of the secondary voltage is prevented. In addition to the Ie's application range including TV receivers, video tape recorders, hifi devices and active loudspeakers, it can also be used in power supply units for professional applications due to its wide control range and high voltage stability during increased load changes.
Features:
• Direct control of the switching transistor
• Low start-up current
• Reversing linear overload characteristic
• Base current drive proportional to collector current
• Protective circuit for case of disturbance
SIEMENS TDA4601 Circuit description
The TDA 4601 is designed for driving, controlling and protecting the switching transistor in flyback converter power supplies during start-up, normal and overload operations as well as during disturbed operation. In case of disturbance the drive of the switching transistor is inhibited and a secondary voltage rise is prevented.
I. Start-up
The start-up procedures (on-mode) include three consecutive operating phases as follows: 1. Build-up of internal reference voltage The internal reference voltage supplies the voltage regulator and effects charging of the coupling electrolytic capacitor connected to the switching transistor. Current consumption will remain at Ig < 3.2 rnA with a supply voltage up to Vg approx. 12 V. 2. Enabling of .internal voltage - reference voltage V; = 4 V
Simultaneously with V g reaching approx. 12 V, an internal voltage becomes available, providing all component elements, with the exception of the control logic, with a thermally stable and overload-resistant current supply.
3. Enabling of control logic
In conjunction with the generation of the reference voltage, the current supply for the control logic is activated by means of an additional stabilization circuit. The integrated circuit is then ready for operation. The above described start-up phases are necessary for ensuring the charging of the coupling eieGiroiytic capacitor, wmcn In turn supplies the switching transistor. Only then is it possible to ensure that the transistor switches accurately.
II. Normal operating mode/control operating mode
At the input of pin 2 the zero passages of the frequency provided by the feedback coil are registered and forwarded to the control logic. Pin 3 (control input, overload and standby identification) receives the rectified amplitude fluctuations of the feedback coil. The con- trol amplifier operates with an input voltage of approx. 2 V and a current of approx. 1.4 mAo Depending on the internal voltage reference, the overload identification limits in conjunction with collector current simulator pin 4 the operating range of the control amplifier. The collec- tor current is simulated by an external RC combination present at pin 4 and internally set threshold voltages. The largest possible collector current applicable with the switching transistor (point of return) increases in proportion to the increased capacitance (10 nF). Thus the required operating range of the control amplifier is established. The range of control lies between a dc voltage clamped at 2 V and a sawtooth-shaped rising ac voltage, which can vary up to a max. amplitude of 4 V (reference voltage). During secondary load reduction to approx. 20 W, the switching frequency is increased (approx. 50 kHz) at an almost constant pulse duty factor (1 :3). During additional secondary load decreases to approx. 1 W, the switching frequency increases to approx. 70 kHz and pulse duty factor to approx. 1 :11. At the same time collector peak current is reduced to < 1 A.
The output levels of the control amplifier as well as those of the overload identification and collector current simulator are compared in the trigger and forwarded to the control logic. Via pin 5 it Is possible to externally inhibit the operations of the IC. The output at pin 8 will be inhibited when voltages of::,; V 2EF -0.1 V are present at pin 5. Flipflops for controlling the base current amplifier and the base current shut-down are set in the control logic depending on the start-up circuit, the zero passage identification as well as on the enabling by the trigger. The base current amplifier forwards the sawtooth-spahed V 4 voltage to the output of pin 8. A current feedback with an external resistor (R = 0.68 Q) is present between pin 8 and pin 7. The applied value of the resistor determines the max. amplitude of the base driving current for the switching transistor. III. Protective operating mode The base current shut-down activated by the control logic clamps the output of pin 7 to 1.6 V. As a result, the drive of the switching transistor is inhibited. This protective measure is enabled if the supply voltage at pin 9 reaches a value::,; 6.7 V or if voltages of V~F -0.1 V are present at pin 5.
In case of short-circuits occurring in the secondary windings of the switched-mode power supply, the illtegrated circuit continuously monitors the fault conditions. During secondary, completely load-free operation only a small pulse duty factor is set. As a result the total power consumption of the power supply is held at N = 6 ... 10 W during both operating modes. After the output has been inhibited for a voltage supply of ::,; 6.7 V, the reference voltage (4 V) is switched off if the voltage supply is further reduced by .:1V g = 0.6 V.
Protective operating mode at pin 5 in case of disturbance The protection against disturbances such as primary undervoltages and/or secondary over- voltages (e.g. by changes in the component parameters for the switched-mode power supply) is realized as follows:
• Protective operating mode with continuous fault condition monitoring In case of disturbance the output pulses at pin 8 are inhibited by falling below the protective threshold V 5 , with a typical value of V/2. As a result ·current consumption is reduced (Ig ~ 14 mA at V g = 10 V). With a corresponding high-impedance start-up resistor"), supply voltage V g will fall below the minimum shut-down threshold (5.7 V) for reference voltage V1• V 1 will be switched off and current consumption is further reduced to Ig $; 3.2 mA at Vg $; 10 V. Because of these reductions in current consumption, the supply voltage can rise again to reach the switch-on threshold of V g ~ 12.3 V. The protective threshold at pin 5 (is released and the power supply is again ready for operation. In case of continuing problems of disturbance (V5 $; V 1/2 -0.1 V) the switch-on mode is interrupted by the periodic protective operating mode described above, i.e. pin 8 is inhibited and V g is falling, etc.
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| SIEMENS TDA4601 DLOCK DIAGRAM |
IV. Switch-on in the wide range power supply (90 Vac to 270 Vac) (application circuit 2)
Self-oscillating flyback-converters designed as wide range power supplies require a power source independent of the rectified line voltage for TDA 4601. Therefore the winding polarity of winding 11/13 corresponds to the secondary side of the flyback converter transformer. Start-up is not as smooth as with an immediately available supply voltage, because TDA 4601 has to be supplied by the start-up circuit until the entire secondary load has been charged. This leads to long switch-on times, especially if low line voltages are applied. However, the switch-on time can be shortened by applying the special start-up circuit (dotted line). The uncontrolled phase of feedback control winding 15/9 is used for activating purposes. Subsequent to activation, the transistor T1 begins to block when winding 11/13 generates the current supply for TDA 4601. Therefore, the control circuit cannot be influenced during operation.
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| SIEMENS TDA4601 APPLICATION CIRCUIT 1 |
Notes on application circuit 1
Protective circuit against secondary voltage rise even in case of disturbance During standby this circuit type is necessary only under certain conditions. If switch 81 is open and the secondary side is loaded with no more than 1 to 5 W, a secondary voltage overshoot of approx. 20% will occur. In case of disturbance (e.g. if the potentiometer is loosely contacted resulting in 10 kQ (2), if the capacitor exhibits a 1 J.1F loss in capacitance, or if the 2 kQ resistor increases to a high-impedance value of 32 kQ), the protective effect of the standard turn-off is not active before the point of return has been reached. The result is that during disturbance energy is pumped into the secondary side, which will not ease off before reaching the point of return and, in the worst case, entails an instantaneous doubling of the voltage to 300 V (endangering the secondary electrolytic capacitors). This additional protective circuit, which identifies the energy surge as voltage overshoot, is directly active at control winding 9/15. Through the 56 Q resistor and the 1N4001 rectifier the negative portion is deducted and stored in the 10 J.1F capacitor. If the amplitude exceeds the voltage of Z-diode BZX 83/39, pin 5 is drawn below the turn-off threshold, inhibiting further control pulses at pin 8. During disturbance conditions the voltage overshoot on the secondary side will assume maximum values of approx. 30%.
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| SIEMENS TDA4601 APPLICATION CIRCUIT FOR WIDE RANGE INPUT SUPPLY |
Notes on application circuit 2 Wide range SMPS:
Filtering of the rectified ac voltage has been increased up to 470 [IF to ensure a constant and hum-free supply at lIIine = 80 Vac. The stabilized phase is tapped for supplying the IC. In order to ensure good start-up conditions for the SMPS in the low voltage range, the non-stabilized phase of winding 13/15 is used as a starting aid (BD 139). which is turned off after start-up by means of Z diode C12. In comparison to the 220 Vac standard circuit, however, the collector-emitter circuit had to be altered to improve the switching behavior of BU 208 for the entire voltage range (80 to 270 Vac.) Diode BY 231 is necessary to prevent inverse operation of BU 208 and may be integrated for switching times with a secondary power < 75 W (BU 208 D). Compared to the IC TDA 4600-2, the TDA 4601 has been improved in turn-off during under- voltage at pin 5. The TDA 4601 is additionally provided with a differential amplifier input at pin 5 enabling precise turn-off at the output of pin 8 accompanied by hysteresis. For wide range SMPS, TDA 4601 is recommendable instead of TDA 4600-2. If a constant quality standard like that of the standard circuit is to be maintained, wide range SMPS (80 to 270 Vac) with secondary power of 120 W can only be implemented at the expense of time.
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| SIEMENS TDA4601 2ND APPLICATION CIRCUIT FOR WIDE RANGE INPUT SUPPLY |
Notes on application circuit 3
Fully insulated, clamp-contacted PTC thermistor suitable for SMPS applications at increased start-up currents The newly developed PTe thermistor Q63100-P2462-J29 is designed for applications in SMPS as well as in various other electronic circuits, which, for example, receive the supply voltage directly from the rectified line voltage and require an increased current during turn-on. Used in the flyback converter power supply of TV sets, an application proved millions of times over, the new PTC thermistor in the auxiliary circuit branch has resulted in a power saving of no less than 2 W. This increase in efficiency has a highly favorable effect on the standby operation of TV sets. The required turn-on current needs only 6 to 8 s until the operating temperature of the PTC thermistor is reached. Low thermal capacitance of the PTC thermistor allows the circuit to be operated again after no more than 2 s. Another positive feature is the improved short- circuit strength. The clamp contacts permit more or less unlimited switching operations and thus guarantee high reliability. A flame-retardant plastic package and smarr dimensions are additional advantages of this newly developed PTC thermistor.
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| SIEMENS TDA4601 3TH APPLICATION CIRCUIT FOR WIDE RANGE INPUT SUPPLY |
Notes on application circuit 4
Improved load control and short-circuit characteristics Turn-on is the same as for circuit 3. To make the price more attractive, switching transistor au 50BA was selected. To ensure optimum standby conditions, the capacitance between pins 2 and 3 was increased to 100 pF. Z diode C6.2 transfers control voltageLlV cont directly to pin 3 resulting in improved load control. Design and coupling conditions of various flyback transformers were sometimes a reason for overshoot spectra, which, despite the RC attenuating element 33 Q x 22 nF and the 10 kQ resistor, even penetrated across the feedback winding 9/15 to the zero passage indicator input (pin 2) and activated double and multiple pulses in the IC. Double and multiple pulses, however, lead to magnetic saturation in the flyback transformer and thus increase the risk of damaging the switched-mode power supply. The larger the quantities of power to be passed, the more easily overshoots are generated. This can be observed around the point of return. The switched-mode power supply, however, reduces its own power to a minimum for all cases of overload or short-circuit A series resonant circuit, whose' resonance corresponds to the transformer's self-oscillation, was created through combination of the 4.7 IJ.H inductance and the 22 nF capacitance. This resonant circuit short-circuits overshoots via a 33 OHM resistor.
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| SIEMENS TDA4601D APPLICATION CIRCUIT FOR WIDE RANGE INPUT SUPPLY |
Notes on application circuit 5 Highly stable secondary side:
Power supplies for commercial purposes require highly constant low voltages and high currents which, on the basis of the flyback converter principle, can be realized only under certain conditions, but, on the other hand, are implemented for economical reasons. An electrically isolated flyback converter with a highly stable secondary side must receive the control information from this secondary side. There are only two possibilities of meeting this requirement: either through a transformer which is magnetically isolated from the flyback converter or by means of an optocoupler. The development of CNY 17 has enabled the manufacture of a component suitable for electrical isolation and characterized by high reliability and long-term stability. The IC TDA 4601D is the sucessor of the TDA 4600 D. It is compatible with its predecessor in all operational functions and in the control of a self-oscillating flyback converter. Pin 3 is the input for the control information, where the latter is compared with the reference voltage prevailing at pin 1 and the control information from the optocoupler and subsequently transformed into a frequency/pulse width control. The previous feedback and control information winding is not necessary. The feedback information (zero passage) is obtained from winding 3/4 - supply winding. The time constant chain 330 0/3.3 nF and 330 0/2.2 nF was implemented in series with 150 J.LH to prevent interference at pin 2. The LC element forms a series resonant circuit for overshoots of the flyback converter and short-circuits them.
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| SIEMENS TDA4601D APPLICATION CIRCUIT FOR WIDE RANGE INPUT SUPPLY N°2 |
Notes on application circuit 6 Wide range plug SMPS up to 30 W
Due to their volume and weight, plug SMPS have so far been limited to a restricted primary voltage and a secondary power of no more than 6 W. The line-isolated wide range flyback converter presented here has a variable frequency and is capable of producing a secondary power of 30 W. It is characterized by a compact design with an approx. weight of 400· g. The entire line voltage range of 90 to 260 Vac is stabilized to ± 1.5% on the secondary side. Load fluctuations between 0.1 and 2 A are regulated to within 5%. The output (secondary side) is overload, short-circuit, and openloop proof.
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| SIEMENS TDA4601 APPLICATIONS CIRCUIT FOR WIDE RANGE INPUT SUPPLY |
Notes on application circuit 7 Wide range SMPS with reducing peak collector current Ie BU 208 for rising line voltage (variable point of return)
Wide range SMPS have to be dimensioned at line voltages of 90 to 260 Vac. The difference between the maximum collector current Ie BU 208 max and the largest possible limit current Ie BU 208 limit which causes magnetic saturation of the flyback transformer and flows through the primary inductance winding 5/7 is to be determined atVaCmin (Ie BU 3081imit~ 1.2 X IeBU208max). Then, the transmissible power of the flyback transformer and its value at Vacmax is to be determined. In the standard circuit the collector current Ie BU 208 max is almost constant at the point of return independently of the line voltage. The transmissible power on the secondary side, however, increases at the point of return in proportion to the rising rectified line voltage applied (figures 1 and 2). In the wide range SMPS a line voltage ratio of 270/90 = 3/1 is obtained causing doubling of the transmissible power on the secondary side, i.e. in the wide range SMPS a flyback transformer had to be implemented that was much too large. The point of return protecting the SMPS against overloads or short circuits, is derived from the time constant at pin 4 r4 = 270 kQ x 4.7 nF. Thus, the largest possible pulse width is determined. With the introduction of the 33 kQ resistor this time constant is reduced as a function of the control voltage applied to winding 13/15, rectified by diode BY 360 and filtered by the 1 ~F capacitance, which means that the pulse time becomes shorter. By means of the Z diode C18 the line voltage level can be defined at which the influence of the time constant correction becomes noticeable. The change in the rectified voltage of winding 13/15 is proportional to the change in the rectified line voltage. At the point of return Ie BU 208 the peak collector current has been reduced with the aid of the given values from 5.2 A at 90 Vac to 3.3 A at 270 Vac. The transmissible power at the point of return remains stable between 125 and 270 Vac due to the set activation point of the point of return correction (unbroken curve in fig. 2).
Control lC for Switched-Mode Power Supplies THE SIEMENS TDA4600-3
Preliminary data SIP9 TYPE IC
The integrated circuit TDA 4600-3 is designed for driving, controlling, and protecting the switching transistor in self-oscillating flyback converter power supplies. In addition to its application in TV receivers and video tape recorders, this IC can also be used in hifi devices and active loud speakers due to its wide control range and high voltage stability.
•Direct control of the switching transistor
•Low start-up current
•Reversing linear overload characteristic
•Base current drive proportional to collector current
SIEMENS TDA4600-3 Description of functions and application:
This IC is designed for driving a bipolar power transistor and for performing all necessary control and protective functions in self-oscillating flyback converter power supplies. Owing to the IC's outstanding voltage stability, which is maintained even at major load fluctuations, the IC is suited for consumer as well as for industrial applications. The rectified line voltage is applied to the series connection of the power transistor and the primary winding of the flyback transformer. During the on-phase of the transistor, energy is stored in the primary winding and released to the consumer via the secondary winding. The IC controls the power transistor in such a way that the secondary voltage is kept at a constant value independently of changes in the line voltage or load. The control information required is derived from the rectified line voltage during the on-phase as well as from a secondary winding during the off-phase. Load differences are compensated by altering the frequency, line voltage fluctuations are additionally counteracted by changing the pulse duty factor. This results in the following load-dependent modes of the SMPS:
- Open-loop or small load: Secondary voltage slightly above the desired value
- Control: Load-independent secondary voltage
- Overload: In case of a secondary overload or short circuit, the secondary voltage is decreased at the point of return as a function of the load current, following a reversing characteristic.
Description of use
A flyback converter designed for color TV sets, applicable between 30 Wand 120 Wand for line voltages ranging from 160 V to 270 V, is described on one of the following pages. On the subsequent pages the major pulses and diagrams can be found. The line voltage is rectified by bridge rectifier Gr1 and smoothed by C3. During start-up the IC current is supplied via the combination Gr2+Rl1 while, in the post-transient condition, it is additionally supplied via winding 13/11 and rectifier Gr3. The size of filter capacitor Cg determines the turn-on behavior. Switching transistor T1 is a BU 208. Parallel capacitance Cll and primary winding 1/7 form a resonant circuit, thus limiting the frequency and amplitude of collector-emitter voltage overshoots upon turn-off of n. R12 , Gr4, C 10, R 15 and Dr2 are elements to improve the switching behavior of T1. The inductance of the primary winding determines the current increase in T1. This sawtooth- shaped current rise is simulated at network R5CS and applied to pin 4 of the IC. Depending of the dimensions of the primary inductance, timing element R5CS is to be adapted to the current rise angle in T1. Thus, during the on-phase, the IC receives control information at pin 4 in the form of the simulated energy content of the primary winding as a function of the line voltage versus time.
Fluctuations at pin 3 are recognized by control winding 9/15. This measure requires fixed coupling to secondary winding 2/16. The control winding is also used for feedback and permits self-oscillating conditions in parallel circuit C11 /primary inductance if power transistor T1 is blocked. In this way the maximum open-loop frequency is determined. The control voltage required at pin 3 is rectified by diode Gr5 and smoothed by capacitor C6• Furthermore, resistor Rs and C6 form a timing element. Due to these circumstances, fast changes in the control voltage are filtered out, i.e. the controlling element does not respond until several periods have occured. The secondary voltage can be set by means of the voltage divider formed of resistors Ry, R 6, R3 and R 2
• Reason: in the IC the control voltage at pin 3 is compared with a stable, internal reference voltage. According to the result of this comparison, frequency and pulse duty factor are corrected until the secondary voltage selected by Ry has established itself. In the case of overload or short circuit on the secondary side, only a small voltage portion is passed to control winding 9/15; the reference voltage at pin 1 becomes directly active at control input pin 3 and activates an overload amplifier (point of return). which drives power transistor T1 down to a smaller pulse duty factor. The line power output is reduced to 6 VA. For all operating ranges of the SMPS, the zero passages of the voltage at the control winding contain information on pulse duty factor and switching frequency of switching transistor T1, or on the open-loop frequency. Conditioning of the corresponding signal at pin 2 is performed by series resistor R 4 , and by integrated limiter diodes. Timing network RS C 4 suppresses HF spikes at pin 2.
Before the line voltage drops below its minimum value, the SMPS must be switched off in order to obtain defined on/off conditions. Winding 11/13 is configured in such a way that the voltage at pin 9 changes linearly with the rectified line voltage. The IC goes into on-state if Vg ~12.3 V, and into off-state if Vg:S; 5.7 V. The drive of the power transistor will be blocked as soon as V g :S; 6.7 V. Pin 5 is connected to pin 9 via resistor R g, since the IC's output is not enabled until voltages V5 ~2.7 V prevail. On the secondary side start-up voltages from V'sec to V4sec are available. If switch S1 is put into open position, standby is set automatically, with a secondary effective power of approx. 3 W being tapped from winding 12/16. Resistors R'3 and R'4 form a basic load of voltages V'sec and V2sec. They contribute to maintaining standby conditions, i.e. Vsec rise :S; 20%. Capacitors C'2 through C'5 prevent spikes caused by reversing rectifiers Gr6 and Gr9. The secondary voltages are smoothed by the charging electrolytic capacitors C'6 through C,g.
After the line voltage has been applied at time to, the following voltages start to increase:
- V g according to the half-cycle charge via R".
- V 4 to V 4 max (typ. 6.2 V)
- V5 to the value determined by Rg
In this case the current consumption of the IC is smaller than 3.2 mA. If Vg reaches the threshold 12.3 V, the IC will switch on the reference voltage of pin 1. The current consumption rises to typically 80 mA. The primary current voltage transformer adjusts V4 down to VREF/2 and the start pulse generator produces the start pulse. Feedback to pin 2 starts a subsequent pulse and so forth. The width of all pulses, including the start pulse, is controlled by the control voltage at pin 3. During turn-on the control voltage corresponds to standby conditions, i.e. V3 = VREF/2 + 50 mV. The IC begins with narrow pulses, which become wider depending on the feedback control voltage. Instantly, the IC operates in the control mode. The control loop is in a post-transient state. If, during start-up, voltage V g drops below the turn-off threshold V g :S; 7.8 V, the start- up phase will be terminated (pin 8 is switched to Low). Since the IC remains in the on-state, Vg drops further to Vg:S; 5.7 V. The IC switches to the off-state, Vg is now able to rise again and a new start-up phase may begin. After the IC has been started, it will operate in the control mode. The voltage at pin 3 is typically VREF/2 + 0.2 V. If the output is loaded, the control amplifier allows wider charge pulses to occur (Va = H). The peak value of the voltage at pin 4 rises to V 4 = V REF• Upon an increase in the secondary load the overload amplifier begins adjusting the pulse width down. Since altering of the pulse width is reversed, this is referred to as the reverse point of the SMPS or point of return. In case of a short circuit on the secondary side, the overload amplifier will adjust the pulse width to typically 1.6 ~s and reduces the pulse duty factor to < 1 :100. The SMPS decreases the line power consumption to typically 6 VA. A small pulse duty factor entails a drop in supply voltage V g below the threshold V g :S; 6.7 V causing a drive interrupt of the switching transistor and a continued drop of supply voltage V g. If supply voltage Vg:S; 5.7 V, the IC is turned off and enters into a new start-up phase.
This intermittent periodic duty operation is continued until the short circuit on the secondary side has been eliminated. If the secondary side is unloaded (standby), the control pulse width becomes narrower. The frequency rises. During open-loop operation the approximate natural frequency of the system (75 kHz) is obtained; pulse duty factor 1 :11. The rise of the secondary voltages is approx. 20%. If resistors R131R14 were absent, the IC would have to perform adjustment beyond the natural frequency of the system, with the zero passage identification only recognizing every 2nd, 3rd or 4th zero passage as a pulse start, i.e. the frequency would divide down to the 2nd, 3rd or 4th subharmonic. The pulse duty factor is thus diminished to 1 : 22, 1 : 33, or 1 : 44, respectively. The pulse width remains constant at approx. 1.2 j.l.sec. A certain small pulse duty factor causes supply voltage V9 to drop below the threshold voltage V 9 ::;: 6.7 V. Then, the interrogation intermittent periodic duty operation begins as already described for the short circuit case. Constant open-loop operation will not continue until resistors R131R14 have been loaded.
SIEMENS TDA4600-3 pin Circuit description:
Pin 1 :Reference voltage output, overload-protected.
I 1max
= 5 mA. All modules, excluding the IC's output stage, are supplied by the
internal reference voltage.
Pin 2:The zero passage identification driving the control logic identifies the discharged
status of the transformer at the zero passage of voltage V 2 from negative to positive
values and enables the logic for pulse start, which is driven by trigger start.
Pin 3:The control voltage supplied to this pin is compared with two stable reference
potentials in the control amplifier, in overload identification and during standby.
The outputs of these stages operate onto the trigger hold, thus terminating the
pulse.
Pin 4:A voltage proportional to the collector current of the switching transistor is generated
on the basis of the external RC combination in conjunction with the collector
current simulation block. This voltage introduces the beginning of a pulse at a stable
voltage via trigger start and determines at a second stable voltage (reverse point)
the absolute maximum pulse (with respect to time length) in trigger hold. At the
same time the rise angle of the voltage proportional to the collector current of the
switching transistor is impressed onto the base current amplifier, and, in accordance
with the smallest current amplification B of the switching transistor to be expected,
the base of the switching transistor is driven via pin 8.
Pin 5:If a voltage :?2.7 V is applied, the control logic is enabled via the trigger. Pins 7/8
are driven by the coupling capacitor charge circuit and the base current. In case
a voltage ~ 1.8 V prevails, base current switch-off pin 7 is clamped at a voltage
V 7 ~1.3 V; driving of the switching transistor is impossible. The IC will not be
enabled again until the voltage at pin 9 has dropped below 5.7 V, the IC has
been turned off and the SMPS has entered a new start-up phase.
Pin 6:GND
Pin 7/8:Via the voltage controller and the coupling capacitor charge circuit, the output
stage of the IC is dc-adjusted to the switching transistor. The switching transistor
is driven via a base current amplifier and pin 8, while it is blocked via the basic
current switch-off and pin 7.
Pin 9:Current supply of the IC.
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| TDA4600-3 BLOCK DIAGRAM |
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| SIEMENS TDA4600-3 MEASUREMENT APPLICATION CIRCUIT |
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| SIEMENS TDA4600-3 APPLICATION CIRCUIT |
THE SIEMENS TDA 4600 Semiconductor circuit description for supplying power to electrical equipment, comprising a transformer having a primary winding connected, via a parallel connection of a collector-emitter path of a transistor with a first capacitor, to both outputs of a rectifier circuit supplied, in turn, by a line a-c voltage; said transistor having a base controlled via a second capacitor by an output of a control circuit acted upon, in turn by the rectified a-c line voltage as actual value and by a reference voltage; said transformer having a first secondary winding to which the electrical equipment to be supplied is connected; said transformer having a second secondary winding with one terminal thereof connected to the emitter of said transistor and the other terminal thereof connected to an anode of a first diode leading to said control circuit; said transformer having a third secondary winding with one terminal thereof connected, on the one hand, via a series connection of a third capacitor with a first resistance, to the other terminal of said third secondary winding and connected, on the other hand, to the emitter of said transistor, the collector of which is connected to said primary winding; a point between said third capacitor and said first resistance being connected to the cathode of a second diode; said control circuit having nine terminals including a first terminal delivering a reference voltage and connected, via a voltage divider formed of a t
hird and fourth series-connected resistances, to the anode
of said second diode; a second terminal of said control circuit serving
for zero-crossing identification being connected via a fifth resistance
to said cathode of said second diode; a third terminal of said
control-circuit serving as actual value input being directly connected
to a divider point of said voltage divider forming said connection of
said first terminal of said control circuit to said anode of said second
diode; a fourth terminal of said control circuit delivering a sawtooth
voltage being connected via a sixth resistance to a terminal of said
primary winding of said transformer facing away from said transistor; a
fifth terminal of said control circuit serving as a protective input
being connected, via a seventh resistance to the cathode of said first
diode and, through the intermediary of said seventh resistance and an
eighth resistance, to the cathode of a third diode having an anode
connected to an input of said rectifier circuit; a sixth terminal of
said control circuit carrying said reference potential and being
connected via a fourth capacitor to said fourth terminal of said control
circuit and via a fifth capacitor to the anode of said second diode; a
seventh terminal of said control circuit establishing a potential for
pulses controlling said transistor being connected directly and an
eighth terminal of said control circuit effecting pulse control of the
base of said transistor being connected through the intermediary of a
ninth resistance to said first capacitor leading to the base of said
transistor; and a ninth terminal of said control circuit serving as a
power supply input of said control circuit being connected both to the
cathode of said first diode as well as via the intermediary of a sixth
capacitor to a terminal of said second secondary winding as well as to a
terminal of said third secondary winding.
The
invention relates to a blocking oscillator type switching power supply
for supplying power to electrical equipment, wherein the primary winding
of a transformer, in series with the emitter-collector path of a first
bipolar transistor, is connected to a d-c voltage obtained by
rectification of a line a-c voltage fed-in via two external supply
terminals, and a secondary winding of the transformer is provided for
supplying power to the electrical equipment, wherein, furthermore, the
first bipolar transistor has a base controlled by the output of a
control circuit which is acted upon in turn by the rectified a-c line
voltage as actual value and by a set-point transmitter, and wherein a
starting circuit for further control of the base of the first bipolar
transistor is provided. Such a blocking oscillator switching
power supply is described in the German periodical, "Funkschau" (1975)
No. 5, pages 40 to 44. It is well known that the purpose of such a
circuit is to supply electronic equipment, for example, a television
set, with stabilized and controlled supply voltages. Essential for such
switching power supply is a power switching transistor i.e. a bipolar
transistor with high switching speed and high reverse voltage. This
transistor therefore constitutes an important component of the control
element of the control circuit. Furthermore, a high operating frequency
and a transformer intended for a high operating frequency are provided,
because generally, a thorough separation of the equipment to be supplied
from the supply naturally is desired. Such switching power supplies may
be constructed either for synchronized or externally controlled
operation or for non-synchronized or free-running operation. A blocking
converter is understood to be a switching power supply in which power is
delivered to the equipment to be supplied only if the switching
transistor establishing the connection between the primary coil of the
transformer and the rectified a-c voltage is cut off. The power
delivered by the line rectifier to the primary coil of the transformer
while the switching transistor is open, is interim-stored in the
transformer and then delivered to the consumer on the secondary side of
the transformer with the switching transistor cut off. In the blocking converter described in the aforementioned reference in the literature, "Funkschau" (1975), No. 5, Pages 40 to 44, the power switching transistor is connected in the manner defined in the introduction to this applic
ation. In addition, a so-called starting
circuit is provided. Because several diodes are generally provided in
the overall circuit of a blocking oscillator according to the definition
provided in the introduction hereto, it is necessary, in order not to
damage these diodes, that due to the collector peak current in the case
of a short circuit, no excessive stress of these diodes and possibly
existing further sensitive circuit parts can occur. Considering the operation of a blocking oscillator, this means that, in the event of a short circuit, the number of collector current pulses per unit time must be reduced. For this purpose, a control and regulating circuit is provided. Simultaneously, a starting circuit must bring the blocking converter back to normal operation when the equipment is switched on, and after disturbances, for example, in the event of a short circuit. The starting circuit shown in the literature reference "Funkschau" on Page 42 thereof, differs to some extent already from the conventional d-c starting circuits. It is commonly known for all heretofore known blocking oscillator circuits, however, that a thyristor or an equivalent circuit replacing the thyristor is essential for the operation of the control circuit.
It is accordingly an object of the invention to provide another starting circuit. It is a further object of the invention to provide a possible circuit for the control circuit which is particularly well suited for this purpose. It is yet another object of the invention to provide such a power supply which is assured of operation over the entire range of line voltages from 90 to 270 V a-c, while the secondary voltages and secondary load variations between no-load and short circuit are largely constant.
In accordance with another feature of the invention, there is provided a second bipolar transistor having the same conduction type as that of the first bipolar transistor and connected in the starting circuit with the base thereof connected to a cathode of a semiconductor diode, the semiconductor diode having an anode connected to the emitter of the first bipolar transistor, the second bipolar transistor having a collector connected via a resistor to a cathode of the first diode in the starting circuit, and having an emitter connected to the input serving to supply power to the control circuit and also connected to the cathode of the second diode which is connected to the other secondary winding of the transformer.
In accordance with a further feature of the invention, the base of the second bipolar transistor is connected to a resistor and via the latter to one pole of a first capacitor, the anode of the first diode being connected to the other pole of the first capacitor.
In accordance with an added feature of the invention, the input serving to supply power to the control circuit is connected via a second capacitor to an output of a line rectifier, the output of the line rectifier being directly connected to the emitter of the first bipolar transistor.
In accordance with yet another feature of the invention, the control circuit has a control output connected via a fifth capacitor to the base of the first bipolar transistor for conducting to the latter control pulses generated in the control circuit.
In accordance with a concomitant feature of the invention, there is provided a sixth capacitor shunting the emitter-collector path of the first transistor.
Other features which are considered as characteristic for the invention are set forth in the appended claim.
Although the invention is illustrated and described herein as embodied in a blocking oscillator type switching power supply, it is nevertheless not intended to be limited to the details shown, since various modifications and structural changes may be made therein without departing from the spirit of the invention and within the scope and range of equivalents of the claims.
The construction and method of operation of the invention, however, together with additional objects and advantages thereof will be best understood from the following description of specific embodiments when read in connection with the accompanying drawings, in which:
FIGS. 1 and 2 are circuit diagrams of the blocking oscillator type switching power supply according to the invention; and
FIG. 3 is a circuit diagram of the control unit RS of FIGS. 1 and 2.
The cathode of the diode D1 and, therewith, the power supply terminal 9 of the control circuits RS are furthermore connected to one pole of a capacitor C3, the other pole of which is connected to the output 3' of the rectifier G. The capacitance of this capacitor C3 thereby smoothes the positive half-wave pulses and serves simultaneously as an energy storage device during the starting period. Another secondary windi
ng
III of the transformer Tr is connected by one of the leads thereof
likewise to the emitter of the first transistor T1, and by the other
lead thereof via a resistor R2, to one of the poles of a further
capacitor C4, the other pole of which is connected to the
first-mentioned lead of the other secondary winding III. This second
pole of the capacitor C4 is simultaneously connected to the output 3' of
the rectifier circuit G and, thereby, via the capacitor C3, to the
cathode of the diode D1 driven by the secondary winding II of the
transformer Tr as well as to the power supply input 9 of the control
circuit RS and, via a resistor R9, to the cathode of a second diode D4.
The second pole of the capacitor C4 is simultaneously connected directly
to the terminal 6 of the control circuit RS and, via a further
capacitor C 6, to the terminal 4 of the control circuit RS as well as,
additionally, via the resistor R6, to the other output 4' of the
rectifier circuit G. The other of the poles of the capacitor C4 acted
upon by the secondary winding II is connected via a further capacitor C5
to a node, which is connected on one side thereof, via a variable
resistor R4, to the terminals 1 and 3 of the control circuit RS, with
the intermediary of a fixed resistor R5 in the case of the terminal 1.
On the other side of the node, the latter and, therefore, the capacitor
C5 are connected to the anode of a third diode D2, the cathode of which
is connected on the one hand, to the resistor R2 mentioned hereinbefore
and leads to the secondary winding III of the transformer Tr and, on the
other hand, via a resistor R3 to the terminal 2 of the control circuit
RS.
The nine terminals of the control circuit RS have the following purposes or functions:
Terminal 1 supplies the internally generated reference voltage to ground i.e. the nominal or reference value required for the control or regulating process;
Terminal 2 serves as input for the oscillations provided by the secondary winding III, at the zero point of which, the pulse start of the driving pulse takes place;
Terminal 3 is the control input, at which the existing actual value is communicated to the control circuit RS, that actual value being generated by the rectified oscillations at the secondary winding III;
Terminal 4 is responsive to the occurrence of a maximum excursion i.e. when the largest current flows through the first transistor T1 ;
Terminal 5 is a protective input which responds if the rectified line voltage drops too sharply; Terminal 6 serves for the power supply of the control process and, indeed, as ground terminal;
Terminal 7 supplies the d-c component required for charging the coupling capacitor C2 leading to the base of the first transistor T1 ;
Terminal 8 supplies the control pulse required for the base of the first transistor T1 ; and
Terminal 9 serves as the first terminal of the power supply of the control circuit RS.
Further details of the control circuit RS are described hereinbelow.
The diode D4, together with the resistor R9 in cooperation with the diode D1 and the secondary winding II, forms the starting circuit provided, in accordance with the invention. The operation thereof is as follows:
In FIG. 2, there is shown a further embodiment of the circuit for a blocking oscillator type switching power supply, according to the invention, as shown in FIG. 1. Essential for this circuit of FIG. 2 is the presence of a second bipolar transistor T2 of the type of the first bipolar transistor T1 (i.e. in the embodiments of the invention, an npn-transistor), which forms a further component of the starting circuit and is connected with the collector-emitter path thereof between the resistor R9 of the starting circuit and the current supply input 9 of the control circuit RS. The base of this second transistor T2 is connected to a node which leads, on the one hand, via a resistor R10 to one electrode of a capacitor C7, the other electrode of which is connected to the anode of the diode D4 of the starting circuit and, accordingl
y,
to the terminal 1' of the supply input of the switching power supply G.
On the other hand, the last-mentioned node and, therefore, the base of
the second transistor T2 are connected to the cathode of a Zener diode
D3, the anode of which is connected to the output 3' of the rectifier G
and, whereby, to one pole of the capacitor C3, the second pole of which
is connected to the power supply input 9 of the control circuit RS as
well as to the cathode of the diode D1 and to the emitter of the second
transistor T2. In other respects, the circuit according to FIG. 2
corresponds to the circuit according to FIG. 1 except for the resistor
R11 which is not necessary in the embodiment of FIG. 2, and the missing
connection between the resistor R9 and the cathode of the diode D1,
respectively, and the protective input 5 of the control circuit RS.
Because of the time constant R9.C3, several positive half-waves are necessary in order to increase the voltage U9 at the supply terminal 9 of the control circuit RS to such an extent that the control circuit RS is energized. During the negative sine half-wave, a partial energy chargeback takes place from the capacitor C3 via the emitter-base path of the transistor T2 of the starting circuit and via the resistor R10 and the capacitor C7, respectively, into the supply network. At approximately 2/3 of the voltage U9, which is limited by the diode D3, the control circuit RS is switched on. At the terminal 1 thereof, the reference voltage Uref then appears. In addition, the voltage divider R5 /R4 becomes effective. At the terminal 3, the control amplifier receives the voltage forming the actual value, while the first bipolar transistor T1 of the blocking-oscillator type switching power supply is addressed pulsewise via the terminal 8.
Because the capacitor C6 is charged via the resistor R6, a higher voltage than Uref is present at the terminal 4 if the control circuit RS is activated. The control voltage then discharges the capacitor C6 via the terminal 4 to half the value of the reference voltage Uref, and immediately cuts off the addressing input 8 of the control circuit RS. The first driving pulse of the switching transistor T1 is thereby limited to a minimum of time. The power for switching-on the control circuit RS and for driving the transistor T1 is supplied by the capacitor C3. The voltage U9 at the capacitor C3 then drops. If the voltage U9 drops below the switching-off voltage value of the control circuit RS, the latter is then inactivated. The next positive sine half-wave would initiate the starting process again.
By switching the transistor T1, a voltage is transformed in the secondary winding II of the transformer Tr. The positive component is rectified by the diode D1, recharing of the capacitor C3 being thereby provided. The voltage U9 at the output 9 does not, therefore, drop below the minimum value required for the operation of the control circuit RS, so that the control circuit RS remains activated. The power supply continues to operate in the rhythm of the existing conditions. In operation, the voltage U9 at the supply terminal 9 of the control circuit RS has a value which meets the condition U9 >[UDs -UBE (T2)], so that the transistor T2 of the starting circuit remains cut off.
For the internal layout of the control circuit RS, the construction shown, in particular, from FIG. 3 is advisable. This construction is realized, for example, in the commercially available type TDA 4600 (Siemens AG).
The block diagram of the control circuit according to FIG. 3
shows
the power supply thereof via the terminal 9, the output stage being
supplied directly whereas all other stages are supplied via Uref. In the
starting circuit, the individual subassemblies are supplied with power
sequentially. The d-c output voltage potential of the base current gain
i.e. the voltage for the terminal 8 of the control circuit RS, and the
charging of the capacitor C2 via the terminal 7 are formed even before
the reference voltage Uref appears. Variations of the supply voltage U9
at terminal 9 and the power fluctuations at the terminal 8/terminal 7
and at the terminal 1 of the control circuit RS are leveled or smoothed
out by the voltage control. The temperature sensitivity of the control
circuit RS and, in particular, the uneven heating of the output and
input stages and input stages on the semiconductor chip containing the
control circuit in monolithically integrated form are intercepted by the
temperature compensation provided. The output values are constant in a
specific temperature range. The message for blocking the output stage,
if the supply voltage at the terminal 9 is too low, is given also by
this subassembly to a provided control logic.Within a provided trigger circuit, the driving pulse length is determined as a function of the sawtooth rise at the terminal 4, and is transmitted to the control logic. In the control logic, the commands of the trigger circuit are processed. Through the zero-crossing identification at input 2 in the control circuit RS, the control logic is enabled to start the control input only at the zero point of the frequency oscillation. If the voltages at the terminal 5 and at the terminal 9 are too low, the control logic blocks the output amplifier at the terminal 8. The output amplifier at the terminal 7 which is responsible for the base charge in the capacitor C2, is not touched thereby.
The base current gain for the transistor T1 i.e. for the first transistor in accordance with the definition of the invention, is formed by two amplifiers which mutually operate on the capacitor C2. The roof inclination of the base driving current for the transistor T1 is impressed by the collector current simulation at the terminal 4 to the amplifier at the terminal 8. The control pulse for the transistor T1 at the terminal 8 is always built up to the potential present at the terminal 7. The amplifier working into the terminal 7 ensures that each new switching pulse at the terminal 8 finds the required base level at terminal 7.
Supplementing the comments regarding FIG. 1, it should also be mentioned that the cathode of the diode D1 connected by the anode thereof to the one end of the secondary winding II of the transformer Tr is connected via a resistor R11 to the protective input 5 of the control circuit RS whereas, in the circuit acco
rding to FIG. 2, the protective input 5 of
the control circuit RS is supplied via a voltage divider R8, R7 directly
from the output 3', 4' of the rectifier G delivering the rectified line
a-c voltage, and which obtains the voltage required for executing its
function. It is evident that the first possible manner of driving the
protective input 5 can be used also in the circuit according to FIG. 2,
and the second possibility also in a circuit in accordance with FIG. 1.The control circuit RS which is shown in FIG. 3 and is realized in detail by the building block TDA 4600 and which is particularly well suited in conjunction with the blocking oscillator type switching power supply according to the invention has 9 terminals 1-9, which have the following characteristics, as has been explained in essence hereinabove:
Terminal 1 delivers a reference voltage Uref which serves as the constant-current source of a voltage divider R5.R4 which supplies the required d-c voltages for the differential amplifiers provided for the functions control, overload detection, short-circuit detection and "standby"-no load detection. The dividing point of the voltage divider R5 -R4 is connected to the terminal 3 of the control circuit RS. The terminal 3 provided as the control input of RS is controlled in the manner described hereinabove as input for the actual value of the voltage to be controlled or regulated by the secondary winding III of the transformer Tr. With this input, the lengths of the control pulses for the switching transistor T1 are determined.
A sawtooth-shaped voltage, the rise of which corresponds to the collector current of the switching transistor T1, is present at the terminal 4 and is minimally and maximally limited by two reference voltages. The sawtooth voltage serves, on the one hand as a comparator for the pulse length while, on the other hand, the slope or rise thereof is used to obtain in the base current amplification for the switching transistor T1, via the terminal 8, a base drive of this switching transistor T1 which is proportional to the collector current.
The terminal 7 of the control circuit RS as explained hereinbefore, determines the voltage potential for the addressing pulses of the transistor T2. The base of the switching transistor T1 is pulse-controlled via the terminal 8, as described hereinbefore. Terminal 9 is connected as the power supply input of the control circuit RS. If a voltage level falls below a given value, the terminal 8 is blocked. If a given positive value of the voltage level is exceeded, the control circuit is activated. The terminal 5 releases the terminal 8 only if a given voltage potential is present.
Foreign References:
DE2417628A1 1975-10-23 363/37
DE2638225A1 1978-03-02 363/49
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61. Clarke, J. C., “The Design of Small Current Transformers,” Electrical Review, January 1985.
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67. Jansson, L. E., “Power-handling Capability of Ferrite Transformers and Chokes for Switched-Mode
Power Supplies,” Mullard Technical Note 31, 1976.
68. Hirschmann, W., Macek, O., and Soylemez, A. I., “Switching Power Supplies 1 (General, Basic Circuits),”
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69. Ackermann, W., and Hirschmann, W., “Switching Power Supplies 2, (Components and Their Selection
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70. Schaller, R., “Switching Power Supplies 3, (Radio Interference Suppression),” Siemens Application Note.
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72. Bulletin SFB, Buss Small Dimension Fuses, Bussmann Division, McGraw-Edison Co., Missouri.
73. Catalog #20, Littlefuse Circuit Protection Components, Littlefuse Tracor, Des Plaines, III.
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Further References:
1. G. Aboud, Cathode Ray Tubes, 1997, 2nd ed., San Jose, CA, Stanford Resources, 1997.
2. G. Aboud, Cathode Ray Tubes, 1997, Internet excerpts, available http://www.stanfordresources.com/
sr/crt/crt.html, Stanford Resources, February 1998.
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4. N. H. Lehrer, The challenge of the cathode-ray tube, in L. E. Tannas, Jr., Ed., Flat Panel Displays
and CRTs, New York: Van Nostrand Reinhold, 1985.
5. P. Keller, The Cathode-Ray Tube, Technology, History, and Applications, New York: Palisades Press,
1991.
6. D. C. Ketchum, CRT’s: the continuing evolution, Society for Information Display International
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7. L. R. Falce, CRT dispenser cathodes using molybdenum rhenium emitter surfaces, Society for
Information Display International Symposium Digest of Technical Papers, 23: 331–333, 1992.
8. J. H. Lee, J. I. Jang, B. D. Ko, G. Y. Jung, W. H. Kim, K. Takechi, and H. Nakanishi, Dispenser
cathodes for HDTV, Society for Information Display International Symposium Digest of Technical
Papers, 27: 445–448, 1996.
9. T. Nakadaira, T. Kodama, Y. Hara, and M. Santoku, Temperature and cutoff stabilization of
impregnated cathodes, Society for Information Display International Symposium Digest of Technical
Papers, 27: 811–814, 1996.
10. W. Kohl, Materials Technology for Electron Tubes, New York, Reinhold Publishing, 1951.
11. S. Sugawara, J. Kimiya, E. Kamohara, and K. Fukuda, A new dynamic-focus electron gun for color
CRTs with tri-quadrupole electron lens, Society for Information Display International Symposium
Digest of Technical Papers, 26: 103–106, 1995.
12. J. Kimiya, S. Sugawara, T. Hasegawa, and H. Mori, A 22.5 mm neck color CRT electron gun with
simplified dynamically activated quadrupole lens, Society for Information Display International
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13. D. Imabayashi, M. Santoku, and J. Karasawa, New pre-focus system structure for the trinitron gun,
Society for Information Display International Symposium Digest of Technical Papers, 27: 807–810,
1996.
14. K. Kato, T. Sase, K. Sasaki, and M. Chiba, A high-resolution CRT monitor using built-in ultrasonic
motors for focus adjustment, Society for Information Display International Symposium Digest of
Technical Papers, 27: 63–66, 1996.
15. S. Sherr, Electronic Displays, 2nd ed., New York: John Wiley, 1993.
16. N. Azzi and O. Masson, Design of an NIS pin/coma-free 108° self-converging yoke for CRTs with
super-flat faceplates, Society for Information Display International Symposium Digest of Technical
Papers, 26: 183–186, 1995.
17. J. F. Fisher and R. G. Clapp, Waveforms and spectra of composite video signals, in K. Benson and
J. Whitaker, Television Engineering Handbook, Featuring HDTV Systems, New York: McGraw-Hill
Reinhold, 1992.
18. D. Pritchard, Standards and recommended practices, in K. Benson and J. Whitaker, Television
Engineering Handbook, Featuring HDTV Systems, New York: McGraw-Hill Reinhold, 1992.
19. A. Vecht, Phosphors for color emissive displays, Society for Information Display International Sym-
posium Conference Seminar Notes F-2, 1995.
20. Optical Characteristics of Cathode Ray Tube Screens, EIA publication TEP116-C, Feb., 1993.
21. G. Wyszecki and W. S. Stiles, Color Science: Concepts and Methods, Quantitative Data and Formulae,
2nd ed., New York: John Wiley & Sons, 1982.
© 1999 by CRC Press LLC
22. A. Robertson and J. Fisher, Color vision, representation, and reproduction, in K. Benson and J.
Whitaker, Television Engineering Handbook, Featuring HDTV Systems, New York: McGraw-Hill
Reinhold, 1992.
23. M. Maeda, Trinitron technology: current status and future trends, Society for Information Display
International Symposium Digest of Technical Papers, 27: 867–870, 1996.
24. C. Sherman, Field sequential color takes another step, Inf. Display, 11 (3): 12–15, March, 1995.
25. L. Ozawa, Helmet mounted 0.5 in. crt for SVGA images, Society for Information Display Interna-
tional Symposium Digest of Technical Papers, 26: 95–98, 1995.
26. C. Infante, CRT display measurements and quality, Society for Information Display International
Symposium Conference Seminar Notes M-3, 1995.
27. J. Whitaker, Electronic Displays, Technology, Design, and Applications, New York: McGraw-Hill, 1994.
28. P. Keller, Electronic Display Measurement, Concepts, Techniques, and Instrumentation, New York:
John Wiley & Sons, 1997.
Further Information
L. Ozawa, Cathodoluminescence: Theory and Applications, New York: Kodansha, 1990.
V. K. Zworykin and G. A. Morton, Television: The Electronics of Image Transmission in Color and Mono-
chrome, New York: John Wiley & Sons, 1954.
B. Wandell, The foundations of color measurement and color perception, Society for Information Display
International Symposium, Conference Seminar M-1, 1993. A nice brief introduction to color science
(31 pages).
Electronic Industries Association (EIA), 2500 Wilson Blvd., Arlington, VA 22201 (Internet: www.eia.org).
The Electronic Industries Association maintains a collection of over 1000 current engineering publi-
cations and standards. The EIA is an excellent source for information on CRT engineering, standards,
phosphors, safety, market information, and electronics in general.
The Society for Information Display (SID), 1526 Brookhollow Dr., Suite 82, Santa Ana, CA 92705-5421
(Internet: www.display.org). The Society for Information Display is a good source of engineering
research and development information on CRTs and information display technology in general.
Internet Resources:
The following is a brief list of places to begin looking on the World Wide Web for information on CRTs
and displays, standards, metrics, and current research. Also many of the manufacturers listed in Table
91.3 maintain Web sites with useful information.
The Society for Information Display
The Society of Motion Picture and Television Engineers
The Institute of Electrical and Electronics Engineers
The Electronic Industries Association
National Information Display Laboratory
The International Society for Optical Engineering
The Optical Society of America
Electronics & Electrical Engineering Laboratory
National Institute of Standards and Technology (NIST)
The Federal Communications Commission
www.display.org
www.smpte.org
www.ieee.org
www.eia.org
www.nta.org
www.spie.org
www.osa.org
www.eeel.nist.gov
www.nist.gov
www.fcc.gov
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The determination of whether any comment is in compliance is at the sole discretion of this blog’s owner.
Comments on this blog may be blocked or deleted at any time.
Fair people are getting fair reply. Spam and useless crap and filthy comments / scrapers / observations goes all directly to My Private HELL without even appearing in public !!!
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This means that there may be a delay between the submission and the eventual appearance of your comment.
Requiring blog comments to obey well-defined rules does not infringe on the free speech of commenters.
Resisting the tide of post-modernity may be difficult, but I will attempt it anyway.
Your choice.........Live or DIE.
That indeed is where your liberty lies.
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