- Deflection Board on the right called large signal board. Line deflection output (BU508A) + EHT, E/W
Correction, FRAME Deflection Output with IC TDA3650 (PHILIPS)
- Signal processing board + Tuning control drive TRD (Tuning Remote Digital)
Chrominance + Luminance with TDA3561A, Synchronization With TDA3576B.
CHASSIS K40 Chrominance + Luminance with TDA3561A,
The TDA3561A is a decoder for the PAL colour television standard. It combines all functions required for the identification
and demodulation of PAL signals. Furthermore it contains a luminance amplifier, an RGB-matrix and amplifier. These
amplifiers supply output signals up to 5 V peak-to-peak (picture information) enabling direct drive of the discrete output
stages. The circuit also contains separate inputs for data insertion, analogue as well as digital, which can be used for
text display systems (e.g. (Teletext/broadcast antiope), channel number display, etc. Additional to the TDA3560, the
circuit includes the following features:
· The peak white limiter is only active during the time that the 9,3 V level at the output is exceeded. The start of the limiting function is delayed by one line period. This avoids peak white limiting by test patterns which have abrupt
transitions from colour to white signals.
· The brightness control is obtained by inserting a variable pulse in the luminance channel. Therefore the ratio of
brightness variation and signal amplitude at the three outputs will be identical and independent of the difference in gain
of the three channels. Thus discolouring due to adjustment of contrast and brightness is avoided.
· Improved suppression of the internal RGB signals when the device is switched to external signals, and vice versa.
· Non-synchronized external RGB signals do not disturb the black level of the internal signals.
· Improved suppression of the residual 4,4 MHz signal in the RGB output stages.
· Cascoded stages in the demodulators and burst phase detector minimize the radiation of the colour demodulator
· High current capability of the RGB outputs and the chrominance output.
The function is described against the corresponding pin
1. + 12 V power supply
The circuit gives good operation in a supply voltage range
between 8 and 13,2 V provided that the supply voltage for
the controls is equal to the supply voltage for the
TDA3561A. All signal and control levels have a linear
dependency on the supply voltage. The current taken by
the device at 12 V is typically 85 mA. It is linearly
dependent on the supply voltage.
2. Control voltage for identification
correct operation. The voltages available under various
signal conditions are given in the specification.
3. Chrominance input
The chroma signal must be a.c.-coupled to the input.
Its amplitude must be between 55 mV and 1100 mV
peak-to-peak (25 mV to 500 mV peak-to-peak burst
signal). All figures for the chroma signals are based on a
colour bar signal with 75% saturation, that is the
burst-to-chroma ratio of the input signal is 1 : 2,25.
4. Reference voltage A.C.C. detector
This pin must be decoupled by a capacitor of about 330
nF. The voltage at this pin is 4,9 V.
5. Control voltage A.C.C.
The A.C.C. is obtained by synchronous detection of the
burst signal followed by a peak detector. A good noise
immunity is obtained in this way and an increase of the
colour for weak input signals is prevented. The
recommended capacitor value at this pin is 2,2 mF.
6. Saturation control
The saturation control range is in excess of 50 dB.
The control voltage range is 2 to 4 V. Saturation control is
a linear function of the control voltage.
When the colour killer is active, the saturation control
voltage is reduced to a low level if the resistance of the
external saturation control network is sufficiently high.
Then the chroma amplifier supplies no signal to the
demodulator. Colour switch-on can be delayed by proper
choice of the time constant for the saturation control
When the saturation control pin is connected to the power
supply the colour killer circuit is overruled so that the colour
signal is visible on the screen. In this way it is possible to
adjust the oscillator frequency without using a frequency
counter (see also pins 25 and 26).
7. Contrast control
The contrast control range is 20 dB for a control voltage
change from + 2 to + 4 V. Contrast control is a linear
function of the control voltage. The output signal is
suppressed when the control voltage is 1 V or less. If one
or more output signals surpasses the level of 9 V the peak
white limiter circuit becomes active and reduces the output
signals via the contrast control by discharging C2 via an
internal current sink.
8. Sandcastle and field blanking input
The output signals are blanked if the amplitude of the input
pulse is between 2 and 6,5 V. The burst gate and clamping
circuits are activated if the input pulse exceeds a level of
The higher part of the sandcastle pulse should start just
after the sync pulse to prevent clamping of video signal on
the sync pulse. The width should be about 4 ms for proper
9. Video-data switching
an input pulse between 1 V and 2 V. In that condition, the
internal RGB signals are switched off and the inserted
signals are supplied to the output amplifiers. If only normal
operation is wanted this pin should be connected to the
negative supply. The switching times are very short
(< 20 ns) to avoid coloured edges of the inserted signals
on the screen.
10. Luminance signal input
The input signal should have a peak-to-peak amplitude of
0,45 V (peak white to sync) to obtain a black-white output
signal to 5 V at nominal contrast. It must be a.c.-coupled to
the input by a capacitor of about 22 nF. The signal is
clamped at the input to an internal reference voltage.
A 1 kW luminance delay line can be applied because the
luminance input impedance is made very high.
Consequently the charging and discharging currents of the
coupling capacitor are very small and do not influence the
signal level at the input noticeably. Additionally the
coupling capacitor value may be small.
11. Brightness control
The black level of the RGB outputs can be set by the
voltage on this pin (see Fig.5). The black level can be set
higher than 4 V however the available output signal
amplitude is reduced (see pin 7). Brightness control also
operates on the black level of the inserted signals.
12, 14, 16. RGB outputs
The output circuits for red, green and blue are identical.
Output signals are 5,25 V (R, G and B) at nominal input
signals and control settings. The black levels of the three
outputs have the same value. The blanking level at the
outputs is 2,1 V. The peak white level is limited to 9,3 V.
When this level exceeded the output signal amplitude is
reduced via the contrast control (see pin 7).
13, 15, 17. Inputs for external RGB signals
The external signals must be a.c.-coupled to the inputs via
a coupling capacitor of about 100 nF. Source impedance
should not exceed 150 W. The input signal required for
a 5 V peak-to-peak output signal is 1 V peak-to-peak.
At the RGB outputs the black level of the inserted signal is
identical to that of normal RGB signals. When these inputs
are not used the coupling capacitors have to be connected
to the negative supply.
18, 19, 20. Black level clamp capacitors
The black level clamp capacitors for the three channels are
connected to these pins. The value of each capacitor
should be about 100 nF.
21, 22. Inputs (B-Y) and (R-Y) demodulators
The input signal is automatically fixed to the required level
by means of the burst phase detector and A.C.C.
generator which are connected to pin 21 and pin 22. As the
burst (applied differentially to those pins) is kept constant
by the A.C.C., the colour difference signals automatically
have the correct value.
23, 24. Burst phase detect
At these pins the output of the burst phase detector is
filtered and controls the reference oscillator. An adequate
catching range is obtained with the time constants given in
the application circuit (see Fig.6).
25, 26. Reference oscillator
The frequency of the oscillator is adjusted by the variable
capacitor C1. For frequency adjustment interconnect pin
21 and pin 22. The frequency can be measured by
connecting a suitable frequency counter to pin 25.
28. Output of the chroma amplifier
Both burst and chroma signals are available at the output.
The burst-to-chroma ratio at the output is identical to that
at the input for nominal control settings. The burst signal is
not affected by the controls. The amplitude of the input
signal to the demodulator is kept constant by the A.C.C.
Therefore the output signal at pin 28 will depend on the
signal loss in the delay line.
Synchronization With TDA3576B.12V 70mA sync combination with transmitter identification and vertical 625 divider system
- Power supply on the bottom of the cabinet (SOPS Supply).
PHILIPS CHASSIS K40 Switched-mode self oscillating supply voltage circuit:POWER SUPPLY (SOPS - Self Oscillating Power Supply)
1. A switched-mode self-oscillating supply voltage circuit for converting an input voltage into an output d.c. voltage which is substantially independent of variations of the input voltage and/or of a load connected to the terminals of the output voltage, comprising a transformer having a primary and a feedback winding, a first controllable switch connected in series with the primary winding, the series arrangement thus formed being coupled between terminals for the input voltage, a second controllable switch coupled via a turn-off capacitor to the control electrode of the first switch to turn it off, means coupling the feedback winding to said control electrode, a transformer winding being coupled via a rectifier to an output capacitor having terminals which supply the output voltage, an output voltage-dependent control voltage being present on a control electrode of the second switch for controlling the conduction period of the first switch, the circuit being switchable between an operating state and a stand-by state in which relative to the operating state the supply energy supplied to the load is considerably reduced, a starting network connected to a terminal for the input voltage, means for adjusting the control voltage in the stand-by state to a value at which the first controllable switch is cut-off, a connection which carries current during the conduction period for the second controllable switch being provided between the starting network and said second switch, and means providing a connection between the starting network and the control electrode of the first switch, which connection does not carry current in the stand-by state.
3. A supply voltage circuit as claimed in claim 2, characterized in that the second controllable switch comprises a thyristor having a main current path included in the control electrode connection of the first controllable switch, said thyristor having a first control gate electrode for adjusting the turn-off instant of the first switch and a second control electrode to which the starting network and the resistor are connected.
Such a supply voltage circuit is disclosed in German Patent Application No. 2,651,196. With this prior art circuit supply energy can be applied in the operating state to the different portions of a television receiver. In the stand-by state the majority of the output voltages of the circuit are so low that the receiver is substantially in the switched-off condition. In the prior art circuit the starting network is formed by a resistor connected to the unstabilized input voltage and through which on turn-on of the circuit a current flows via the feedback winding to the control electrode of the first controllable switch, which is a switching transistor, and brings it to and maintains it in the conductive state, as a result of which the circuit can start.
The invention has for its object to provide an improved circuit of the same type in which in the stand-by state the supply energy applied to the load is reduced to zero. The prior art circuit cannot be improved in this respect without the use of mechanical switches, for example relays. According to the invention, the switched-mode self-oscillating supply voltage circuit does not comprise such relays and is characterized in that it further comprises means for adjusting the control voltage in the stand-by state to a value at which the first controllable switch is cut-off. A connection which carries current during the conduction period of the second controllable switch is provided between the starting network and said second switch while a connection present between the starting network and the control electrode of the first switch does not carry current in the stand-by state.
The invention is based on the recognition that the prior art supply voltage circuit cannot oscillate, so that the energy supplied by it is zero, if the control voltage obtains a value as referred to, while the starting network is connected in such a manner that in the stand-by state no current can flow through it to the control electrode of the first controllable switch.
It should be noted that in the said German Patent Application the starting network is in the form of a resistor which is connected to an unstabilized input d.c. voltage. It is, however, known, for example, from German Patent Specification No. 2,417,628 to employ for this purpose a rectifier network connected to an a.c. voltage from which the said input d.c. voltage is derived by rectification.
The invention will now be further described by way of example with reference to the accompanying drawing, which shows a basic circuit diagram of a switched-mode self-oscillating supply voltage circuit.
The self-oscillating supply circuit shown in the FIGURE comprises a npn-switching transistor Tr1 having its collector connected to the primary winding L1 of a transformer T, while the emitter is connected to ground via a small resistor R1, for example 1.5 Ohm. Resistor R1 is decoupled for the high frequencies by means of a 150 nF capacitor C1. One end of winding L1 is connected to a conductor which carries an unstabilized input d.c. voltage V B of, for example, 300 V. Voltage V B has a negative rail connected to ground and is derived from the electric power supply by rectification. One end of a feedback winding L2 is connected to the base of transistor Tr1 via the parallel arrangement of a small inductance L3 and a damping resistor R2. A terminal of a 47 μF capacitor C2 is connected to the junction of the elements L2, L3 and R2. The series arrangement of a diode D1 and a 2.2 Ohm-limiting resistor R3 is arranged between the other terminal of capacitor C2 and the other end of winding L2 and the series arrangement of a resistor R4 of 12 Ohm and a diode D2 is arranged between the same end of winding L2 and the emitter of transistor Tr1. A 150 nF capacitor C3 is connected in parallel with diode D2. The anode of diode D1 is connected to that end of winding L2 which is not connected to capacitor C2, while the anode of diode D2 is connected to the emitter of transistor Tr1. In the FIGURE the winding sense of windings L1 and L2 is indicated by means of dots.
Because of the feedback the described circuit oscillates independently as soon as the steady state is achieved. It will be described hereinafter how this state is obtained. During the time transistor Tr1 conducts the current flowing through the resistor R1 increases linearly. The resistor R4 then partly determines the base current of transistor Tr1. Capacitor C4 and resistor R7 form a voltage source the voltage of which is subtracted from the voltage drop across resistor R1. As soon as the voltage on the base of transistor Tr3 is equal to approximately 0.7 V this transistor becomes conductive, as a result of which the thyristor formed by transistors Tr2 and Tr3 becomes rapidly conductive and remains so. Across capacitor C2 there is a negative voltage by means of which transistor Tr1 is turned off. The inverse base current thereof flows through thyristor Tr2, Tr3. This causes charge to be withdrawn from capacitor C2, while the charge carriers stored in transistor Tr1 are removed with the aid of inductance L3. As soon as the collector current of transistor Tr1 has been turned off, the voltage across winding L2 reverses its polarity, which current recharges the capacitor. Now the voltage at the junction of components C2, R3 and R5 is negative, causing thyristor Tr2, Tr3 to extinguish.
In parallel with winding L1 there is the series network of a 2.2 nF tuning capacitor C8 and a 100 Ohm limiting resistor R8. The anode of a diode D6 is connected to the junction of components R8 and C8, while the cathode is connected to the other terminal of resistor R8. Winding L1 and capacitor C8 form a resonant circuit across which an oscillation is produced after windings L4, L5 and L6 have become currentless. At a later instant the current through circuit L1, C8 reverses its direction. As a result thereof a current is generated in winding L2 which flows via diode D2 and resistor R4 to the base of transistor Tr1 and makes this transistor conductive and maintains it in this state. The dissipation in resistor R8 is reduced by means of diode D6. A clamping network formed by the parallel arrangement of a 22 kOhm resistor R9 and a 120 nF capacitor C9 is arranged in series with a diode D7. This whole assembly is in parallel with winding L1 and cuts-off parasitic oscillations which would be produced during the period of time in which transistor Tr1 is non-conductive. The output voltages of the supply circuit are kept substantially constant in spite of variations of voltage V B and/or the loads, thanks to a control of the turning-on instant of thyrisistor Tr2, Tr3. For this purpose the emitter of a light-sensitive transistor Tr4 is connected to the base of transistor Tr3. The collector of transistor Tr4 is connected via a resistor R10 to the conductor which carries the voltage V B and to a Zener diode Z1 which has a positive voltage of approximately 7.5 V, while the base is unconnected. The other end of diode Z1 is connected to ground. A light-emitting diode D8, whose cathode is connected to the collector of an npn-transistor Tr5, is optically coupled to transistor Tr4. By means of a potentiometer R11 the base of transistor Tr5 can be adjusted to a d.c. voltage which is derived from the voltage V 0 of approximately 130 V across capacitor C6. The anode of diode D8 is connected to a d.c. voltage V 1 of approximately 13 V. A resistor R12 is also connected to voltage V 1 , the other end of the resistor being connected to the emitter of transistor Tr5, to the cathode of a Zener diode Z2 which has a voltage of approximately 7.5 V and to a smoothing capacitor C10. The other ends of diode Z2 and capacitor C10 are connected to ground. Voltage V1 can be generated by means of a transformer connected to the electric AC supply and a rectifier, which are not shown for the sake of simplicity, more specifically for a remote control to which constantly supply energy is always applied, even when the majority of the components of the receiver in what is referred to as the stand-by state are not supplied with supply energy.
A portion of voltage V 0 is compared with the voltage of diode Z2 by means of transistor Tr5. The measured difference determines the collector current of transistor Tr5 and consequently the emitter current of transistor Tr4. This emitter current produces across resistor R6 a voltage drop whose polarity is the opposite of the polarity of the voltage source formed by resistor R7 and capacitor C4. Under the influence of this voltage drop the turn-on instant of thyristor Tr2, Tr3 is controlled as a function of voltage V 0 . If, for example, voltage V 0 tends to decrease owing to an increasing load thereon and/or in response to a decrease in voltage V B , then the collector current of transistor Tr5 decreases and consequently also the said voltage drop. Thyristor Tr2, Tr3 is turned on at a later instant than would otherwise be the case, causing transistor Tr1 to be cut-off at a later instant. The final value of the collector current of this transistor is consequently higher. Consequently, the ratio of the time interval in which transistor Tr1 is conductive to the entire period, commonly referred to as the duty cycle, increases, while the frequency decreases.
The circuit comprises a 1 MOhm starting resistor R16, one end of which is connected to the base of transistor Tr2 and the other end to the conductor which carries the voltage V B . Upon turn-on of the circuit current flows through resistors R16 and R5 and through capacitor C2, which has as yet no charge, to the base of transistor Tr1. The voltage drop thus produced across resistor R5 keeps transistor Tr2, and consequently also transistor Tr3, in the non-conductive state, while transistor Tr1 is made conductive and is maintained so by this current. Current also flows through winding L2. In this manner the circuit can start as energy is built up in transformer T.
ponds to the worst dimensioning case in which the stand-by state is initiated while transistor Tr1 is in the non-conductive state.
If thyristor Tr2, Tr3 conducts, either in the operating state or in the stand-by state, current flows through resistor R16 via the collector emitter path of transistor Tr3 to ground. This current is too small to have any appreciable influence on the behaviour of the circuit. When thyristor Tr2, Tr3 does not conduct, the voltage on the left hand terminal of capacitor C2 is equal to approximately 1 V, while the voltage across the capacitor is approximately -4 V. So transistor Tr1 remains in the non-conductive state and a premature turn-on thereof cannot occur. If in the operating state transistor Tr1 conducts while thyristor Tr2, Tr3 is cut-off, then the current flows through resistor R16 in the same manner as it flows during the start to the base of transistor Tr1, but has relatively little influence as the base current caused by the energy stored in winding L2 is many times larger. If both transistor Tr1 and thyristor Tr2, Tr3 are non-conductive, then the current through resistor R16 flows through components R5, C2, L2, R4, C3 and R1. In this stand-by state capacitor C2 has indeed substantially no negative charge any longer but, in spite thereof, transistor Tr1 cannot become conductive since no current flows to its base. It will furthermore be noted that the circuit is protected in the event that thyristor Tr2, Tr3 has an interruption. Namely, in such a case the circuit cannot start.
In the foregoing a circuit is described which may be considered to be a switched-mode supply voltage circuit of the parallel ("flyback") type. It will be obvious that the invention may alternatively be used in supply voltage circuits of a different type, for example converters of the type commonly referred to as up-converters. It will also be obvious that transistor Tr1 may be replaced by an equivalent switch, for example a gate-turn-off switch.
Interface Unit (for SCART Socket signal interfacing )
- Sound AMPL. with TDA2040V 8222 280
- TRD Unit (Tuning control + drive) with SAB3035 (PHILIPS)
SAB3035 COMPUTER INTERFACE FOR TUNING AND CONTROL (CITAC)
The SAB3035 provides closed-loop digital tuning of TV receivers, with or without a.f.c., as required. lt
also controls up to 8 analogue functions, 4 general purpose I/O ports and 4 high-current outputs for
tuner band selection.
The IC is used in conjunction with a microcomputer from the MAB84OO family and is controlled via a two-wire, bidirectional I2 C bus.
Combined analogue and digital circuitry minimizes the number of additional interfacing components
Frequency measurement with resolution of 50 KHz
Selectable prescaler divisor of 64 or 256
32 V tuning voltage amplifier
4 high-current outputs for direct band selection
8 static digital to analogue converters (DACSI for control of analogue functions
Four general purpose input/output (l/O) ports
Tuning with control of speed and direction
Tuning with or without a.f.c.
Single-pin, 4 MHZ on-chip oscillator
I2 C bus slave transceiver
The SAB3035 is a monolithic computer interface which provides tuning and control functions and
operates in conjunction with a microcomputer via an I2 C bus.
This is performed using frequency-locked loop digital control. Data corresponding to the required tuner
frequency is stored in a 15-bit frequency buffer. The actual tuner frequency, divided by a factor of 256
(or by 64) by a prescaler, is applied via a gate to a 15-bit frequency counter. This input (FDIV) is
measured over a period controlled by a time reference counter and is compared with the contents of the frequency buffer. The result of the comparison is used to control the tuning voltage so that the tuner frequency equals the contents of the frequency buffer multiplied by 50 kHz within a programmable tuning window (TUW).
The system cycles over a period of 6,4 ms (or 2,56 ms), controlled by the time reference counter which is clocked by an on-chip 4 lVlHz reference oscillator. Regulation of the tuning voltage is performed by a charge pump frequency-locked loop system. The charge IT flowing into the tuning voltage amplifier is controlled by the tuning counter, 3-bit DAC and the charge pump circuit. The charge IT is linear with the frequency deviation Af in steps of 50 l
Testing Flyback Transformer
Nowadays, more and more monitor comes in with flyback transformers problems.
Testing flyback transformer are not difficult if you carefully follow the
instruction. In many cases, the flyback transformer can become short
circuit after using not more than 2 years. This is partly due to bad design
and low quality materials used during manufactures flyback transformer.
The question is what kind of problems can be found in a flyback transformer
and how to test and when to replace it. Here is an explanation that will help
you to identify many flyback transformer problems.
There are nine common problems can be found in a flyback transformer.
a) A shorted turned in the primary winding.
b) An open or shorted internal capacitor in secondary section.
c) Flyback Transformer becomes bulged or cracked.
d) External arcing to ground.
e) Internal arcing between windings.
f) Shorted internal high voltage diode in secondary winding.
g) Breakdown in focus / screen voltage divider causing blur display.
h) Flyback Transformer breakdown at full operating voltage (breakdown when under load).
i) Short circuit between primary and secondary winding.
Testing flyback transformer will be base on (a) and (b) since problem
(c) is visible while problem (d) and (e) can be detected by hearing the arcing
sound generated by the flyback transformer. Problem (f) can be checked with multimeter
set to the highest range measured from anode to ABL pin while (g) can be solved by
adding a new monitor blur buster (For 14' & 15' monitor only.) Problem (h) can only be
tested by substituting a known good similar Flyback Transformer. Different monitor have
different type of flyback transformer design. Problem (i) can be checked using an
ohm meter measuring between primary and secondary winding. A shorted turned or open
in secondary winding is very uncommon.
What type of symptoms will appear if there is a shorted turned in primary winding?
a) No display (No high voltage).
b) Power blink.
c) B+ voltage drop.
d) Horizontal output transistor will get very hot and later become shorted.
e) Along B+ line components will spoilt. Example:- secondary diode UF5404 and B+ FET IRF630.
f) Sometimes it will cause the power section to blow.
What type of symptoms will appear if a capacitor is open or shorted in a flyback transformer?
a. No display (No high voltage).
b. B+ voltage drop.
c. Secondary diode (UF5404) will burned or shorted.
d. Horizontal output transistor will get shorted.
e. Power blink.
f. Sometimes power section will blow, for example: Raffles 15 inch monitor.
g. Power section shut down for example: Compaq V55, Samtron 4bi monitor.
h. Sometimes the automatic brightness limiter (ABL) circuitry components will get burned.
This circuit is usually located beside the flyback transformer. For example: LG520si
a. High voltage shut down.
b. Monitor will have ‘tic - tic’ sound. Sometimes the capacitor may measure O.K. but
break down when under full operating voltage.
c. Horizontal output transistor will blow in a few hours or days after you have replaced it.
d. Sometimes it will cause intermittent "no display".
e. Distorted display i.e., the display will go in and out.
f. It will cause horizontal output transistor to become shorted and blow the power section.
How to check if a primary winding is good or bad in a Flyback Transformer?
a) By using a flyback/LOPT tester, this instrument identifies faults in primary winding by
doing a ‘ring’ test.
b) It can test the winding even with only one shorted turned.
c) This meter is handy and easy to use.
d) Just simply connect the probe to primary winding.
e) The readout is a clear ‘bar graph’ display which show you if the flyback transformer
primary winding is good or shorted.
f) The LOPT Tester also can be used to check the CRT YOKE coil, B+ coil and switch mode power transformer winding.
NOTE: Measuring the resistance winding of a flyback transformer, yoke coil, B+ coil and
SMPS winding using a multimeter can MISLEAD a technician into believing that a shorted
winding is good. This can waste his precious time and time is money.
How to diagnose if the internal capacitor is open or shorted?
By using a normal analog multimeter and a digital capacitance meter. A good capacitor have the range from 1.5 nanofarad to 3 nanofarad.*
1) First set your multimeter to X10K range.
2) Place your probe to anode and cold ground.
3) You must remove the anode cap in order to get a precise reading.
4) Cold ground means the monitor chassis ground.
5) If the needle of the multimeter shows a low ohms reading, this mean the internal capacitor
6) If the needle does not move at all, this doesn’t mean that the capacitor is O.K.
7) You have to confirm this by using a digital capacitance meter which you can easily get one
from local distributor.
8) If the reading from the digital capacitance meter shows 2.7nf, this mean the capacitor is
within range (O.K.).
9) And if the reading showed 0.3nf, this mean the capacitor is open.
10) You have three options if the capacitor is open or shorted.
- Install a new flyback transformer or
- Send the flyback transformer for refurbishing or
- Send the monitor back to customers after spending many hours and much effort on it.
* However certain monitors may have the value of 4.5nf, 6nf and 7.2nf.
Note: Sometimes the internal capacitor pin is connected to circuits (feedback) instead of ground.
Tv rca flyback transformer circuits usually do not have a internal capacitor in it.
If you have a flyback diagram and circuits which you can get it from the net, that would be an advantage to easily understand how to check them.
HR DIEMEN TV FLYBACK TRAFO HR6040 FOR MODELS BELOW WITH PHILIPS CHASSIS K40:
Analogue replacement FBT:KN-381804, F3818, 140.10246, 003390003, 031562, 10810246, 13836070, 13836072, 14010246, 14010269, 16CT4218, 17701MH, 20C051, 22C051, 22C052, 22CS3740, 22CS4360, 22CS4363, 22CS4460, 22CS4560, 22CS4850, 22CS4860, 22CS4861, 22CS5240, 22CS5242, 22CS5250, 22CS5350, 22CS5351, 22CS5355, 22CS5445, 22CS5447, 22CS5735, 22CS5739, 22CS5744, 22CS5745, 22CS5748, 22CS5750, 22CS5751, 22CS5755, 22CS5758, 26CD4895, 26CS4376, 26CS4377, 26CS4378, 26CS4379, 26CS4385, 26CS4386, 26CS4387, 26CS4390, 26CS4391, 26CS4392, 26CS4393, 26CS4396, 26CS4490, 26CS4590, 26CS4880, 26CS4895, 26CS5270, 26CS5272, 26CS5275, 26CS5280, 26CS5380, 26CS5382, 26CS5383, 26CS5385, 26CS5387, 26CS5390, 26CS5395, 26CS5475, 26CS5573, 26CS5577, 26CS5578, 26CS5770, 26CS5774, 26CS5775, 26CS5777, 26CS5780, 26CS5781, 26CS5785, 26CS5787, 26CS5790, 26CS5793, 26CS5795, 26CS5799, 26CS6573, 27CS657302, 27CS6590, 27CS6895, 36070, 36071, 36072, 36073, 36074, 36075, 36076, 36077, 36078, 36079, 37CS5600, 40001M, 4398, 4612080, 56KS4508, 56KS4509, 56KS5402, 56KS5418, 56KS5447, 56KS5457, 56KS5487, 66KS4808, 66KS5617, 66KS5702, 66KS5787, 66KS5917, BACH, BEETHOVEN, BELLINI, BREGENZSTEREO, CHASISK40, CHOPIN, DONATELLO, DONIZETTI, EXPERT, F3818, GIOTTO, GOJA, GOYA, GUARDI, INTERFUNK8349, INTERFUNK8399, INTERFUNK8499, INTERFUNK8599, K40, KN35018N, KN3818, KREFELD, LIPPI, LOT111, LT279P, MAGNASCO, MATCHLINEMONIT, MATCHLINERECEI, ME/540300, ME540300, MICHELANGELO, MORANDI, PHILETTAROYAL, PICASSO, PIRANESI, PUCCINI, REMBRANDT, RO146, ROSSINI, RUBENS, STRAUSS, SUPERSCREEN, TIEPOLO, TIZIANO, TR146, TRR246, TTR246, TURNER, V6720, V6721, V6820, V6821, V6830, V6850, V6851, VANGOGH, VERONESE, VIVALDI.
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Ambatipudi, Ravindra, Improving Transient Response of Opto-Isolated Converters.
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Power Integrations, Inc.'S Disclosure of Asserted Claims and Preliminary Infringement Contentions, Power Integrations, Inc. v. System General Corporation & System General USA, United States District Court, Northern District of California, San Francisco Division, Case No. C04 2581 JSW, Apr. 15, 2005.
Power Integrations, Inc.'S Revised Disclosure of Asserted Claims and Preliminary Infringement Contentions, Power Integrations, Inc. v. System General Corporation & System General USA, United States District Court, Northern District of California, San Francisco Division, Case No. C04 2581 JSW, May 24, 2005.
Defendants System General Corporation and System General USA's Preliminary Invalidity Contentions, Power Integrations, Inc. v. System General Corporation& System General USA, United States District Court, Northern District of California, San Francisco Division, Case No. C04 2581 JSW, May 27, 2005.
Fourth Joint Status Report, Power Integrations, Inc. v. System General Corporation& System General USA, United States District Court, Northern District of California, San Francisco Division, Case No. C04 2581 JSW, Jul. 5, 2006.
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Respondent System General Corporation's Petition for Review of the Final Intial Determination, In the Matter of Certain Power Supply Controllers and Products Containing the Same, United States International Trade Commission, Washington, DC 20436, Before the Honorable Paul J. Luckern, Administrative Law Judge, Inv. No. 337-TA-541, May 26, 2006.
Complainant Power Integration, Inc.'s Opposition to Respondent System General Corp.'s Petition for Review of the Final Intial Determination, In the Matter of Certain Power Supply Controllers and Products Containing the Same, United States International Trade Commission, Washington, DC 20436, Before the Honorable Paul J. Luckern, Administrative Law Judge, Inv. No. 337-TA-541, Jun. 5, 2006.
Response of the Office of Unfair Import Investigations to Respondent System General Corp.'s Petition for Review of the Final Intial Determination, In the Matter of Certain Power Supply Controllers and Products Containing the Same, United States International Trade Commission, Washington, DC 20436, Before the Honorable Paul J. Luckern, Administrative Law Judge, Inv. No. 337-TA-541, Jun. 5, 2006.
Notice of Commission Determination Not to Review a Final Initial Determination of Violation of Section 337; Schedule for Filing Written Submissions on Remedy, The Public Interest, and Bonding, In the Matter of Certain Power Supply Controllers and Products Containing the Same, United States International Trade Commission, Washington, DC 20436, Before the Honorable Paul J. Luckern, Administrative Law Judge, Inv. No. 337-TA-541, Jun. 30, 2006.
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Complainant Power Integrations, Inc.'s Posthearing Statement (Fully-Redacted), In the Matter of Certain Power Supply Controllers and Products Containing Same, United States International Trade Commission, Washington, DC 20436, Before the Honorable Paul J. Luckern, Administrative Law Judge, Inv. No. 337-TA-541, Feb. 10, 2006.
Respondent System General Corporation's Post-Hearing Brief (Fully-Redacted), In the Matter of Certain Power Supply Controllers and Products Containing Same, United States International Trade Commission, Washington, DC 20436, Before the Honorable Paul J. Luckern, Administrative Law Judge, Inv. No. 337-TA-541, Feb. 10, 2006.
Post-Hearing Brief of the Commission Investigative Staff (Fully-Redacted), In the Matter of Certain Power Supply Controllers and Products Containing Same, United States International Trade Commission, Washington, DC 20436, Before the Honorable Paul J. Luckern, Administrative Law Judge, Inv. No. 337-TA-541, Feb. 14, 2006.
Complainant Power Integrations, Inc.'s Posthearing Reply Statement (Fully-Redacted), In the Matter of Certain Power Supply Controllers and Products Containing Same, United States International Trade Commission, Washington, DC 20436, Before the Honorable Paul J. Luckern, Administrative Law Judge, Inv. No. 337-TA-541, Feb. 24, 2006.
Respondent System General Corporation's Post-Hearing Reply Brief (Fully-Redacted), In the Matter of Certain Power Supply Controllers and Products Containing Same, United States International Trade Commission, Washington, DC 20436, Before the Honorable Paul J. Luckern, Administrative Law Judge, Inv. No. 337-TA-541, Feb. 24, 2006.
United States Court of Appeals for the Federal Circuit 2007-1082, Judgement, System General Corp. v. International Trade Commission and Power Integrations, Inc., On Appeal from the United States International Trade Commission, In Case No. 337-TA-541, Before the Honorable Pauline Newman, Circuit Judge, the Honorable Raymond C. Clevenger, III, Senior Circuit Judge, and Timothy B. Dyk, Circuit Judge, Nov. 19, 2007.
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