The PHILIPS CHASSIS K11 is introducing the InLine PHILIPS 20AX CRT TUBE FAMILY.
This chassis was fitted in many models series phasing out the CRT DELTA TUBES FAMILY of previous era and models.
The chassis structural aspect is similar to CHASSIS K9 dual panel with deflections / power stages on the left big board (Large signal panel) and all signal stages on the smaller board on the right side (small signal board).
The concept and the development are different due to CRT TUBE 20AX TYPE.
This chassis is showing another time the awesome engineering of such tellyes.
This model is equipped with one of the first ULTRASONIC REMOTE CONTROL RECEIVER device in PHILIPS models , to feature remote control capability.
The version shown here in collection is the nT which was using a KASKADE EHT MULTIPLIER in the EHT section, other version like the nD were using for first time a DST Transformer and when i have time will show even this.
The PHILIPS K11 was replaced by the PHILIPS K12 MONOCARRIER.
The chassis K11 is higly sophisticated and complex but it has an unique fashinating structure and design which expands his technology in a way of simplicity which is today, long time, lost and forgotten (forever).
PHILIPS 26C566 /38Z (PHILIPS K11) CHASSIS K11 (20AX) E/W CORRECTION Circuit arrangement in an image display apparatus for (horizontal) line deflection
Line deflection circuit in which the deflection coil is east-west modulated. In order to cancel an east-west dependent horizontal linearity defect the inductance value of the linearity correction coil is made independent of the field frequency, for example by means of a compensating current. In an embodiment this current is supplied by the shunt coil of the east-west modulator.
1. Circuit arrangement for use with a line deflection coil, said circuit comprising a generator means adapted to be coupled to said coil for producing a sawtooth line-deflection current through said line deflection coil, said deflection current having a field-frequency component current, a horizontal linearity correction coil adapted to be coupled in series with said deflection coil and including an inductor having a bias-magnetized core, and means for making the inductance value of the linearity correction coil substantially independent of the field frequency component current. 2. Circuit arrangement as claimed in claim 1, wherein said making means includes a current supply source means for producing a compensating line-frequency sawtooth current through a winding of the linearity correction coil, the amplitude of the compensating current having a field-frequency variation. 3. Circuit arrangement as claimed in claim 2, wherein the direction of curvature of the field-frequency envelope of the compensating current is opposite to the direction of curvature of the field-frequency component current of the line deflection current, whereby the magnetic fields produced in the core of the correction coil by the two currents have the same direction. 4. Circuit arrangement as claimed in claim 2, wherein the direction of curvature of the field-frequency envelope of the compensating current is the same as the direction of curvature of the field-frequency component current of the line deflection current, whereby the magnetic fields produced in the core of the correction coil by the two currents have opposite directions. 5. Circuit arrangement as claimed in claim 2, wherein said correction coil further comprises an additional winding disposed on the core, said additional winding being coupled to said supply source means to receive the compensating current. 6. Circuit arrangement as claimed in claim 5, further comprising modulator means for modulating the line deflection current with said field frequency component, said modulator including a compensation coil coupled in series with said additional winding. 7. Horizontal linearity correction coil comprising a core made of a magnetic material and bias-magnetized by at least one permanent magnet, and an additional winding disposed on the core. 8. Image display apparatus including a circuit arrangement as claimed in claim 1.
The invention relates to a circuit arrangement in an image display apparatus for (horizontal) line deflection, which apparatus also includes a circuit arrangement for (vertical) field deflection, provided with a generator for generating a sawtooth line-frequency deflecting current through a line deflection coil and with a modulator for field-frequency modulation of this current, the deflection coil being connected in series with a linearity correction coil in the form of an inductor having a bias-magnetized core.
By means of the linearity correction coil the linearity error due to the ohmic resistance of the deflection circuit is corrected. The sign of the bias magnetisation is chosen so that it is cancelled by the deflection current at the beginning of the deflection interval, so that the inductance of the correction coil is a maximum, whereas the voltage drop across the deflection coil then is a minimum. This voltage drop is adjustable by adjustment of the starting inductance of the correction coil. During the deflection interval the core gradually becomes saturated so that the inductance of, and the voltage drop across, the correction coil decrease. Thus the linearity error can be cancelled exactly at the beginning of the interval, that is to say on the left on the screen of the image display tube, and with a certain approximation at other locations.
In image display tubes using a large deflection angle, raster distortion, which generally is pincushion-shaped, of the image displayed occurs. This distortion can be removed in the horizontal direction, the so-called east-west direction, by means of field-frequency modulation of the line deflection current, the envelope in the case of pincushion-shaped distortion being substantially parabolic so that the amplitude of the line deflection current is a maximum at the middle of the field deflection interval.
It was found in practice that the said two corrections are not independent of one another, that is to say the adjustment of the east-west modulation affects horizontal linearity. As long as the modulation depth is not excessive, a satisfactory compromise can be found. However, in display tubes having a deflection angle of 110° and particularly in colour display tubes in which the deflection coils have a converging effect also, it is difficult to find such a compromise. A tube of this type is described in "Philips Research Reports," volume Feb. 14, 1959, pages 65 to 97; the distribution of the deflection field is such that throughout the display screen the landing points of the electron beams coincide without the need for a converging device. Owing to this field distribution, however, the pin-cushion-shaped distortion in the image displayed in the east-west direction is greater than in comparable display tubes of another type. Hence there must be east-west modulation of the line deflection current to a greater depth. It is true that under these conditions horizontal linearity can correctly be adjusted over a given horizontal strip after the east-west modulation has been adjusted correctly, i.e., for a rectangular image, but it is found that in other parts of the display screen a serious linearity error remains. When vertical straight lines are displayed as straight lines in the right-hand part of the screen, they are displayed as curved lines in the left-hand part.
It is an object of the present invention to remove the said defect so that horizontal linearity can satisfactorily be adjusted throughout the screen, and for this purpose the circuit arrangement according to the invention is characterized in that it includes means by which the inductance of the linearity correction coil is made substantially independent of the field frequency.
The invention is based on the recognition that the defect to be removed is due to a field-frequency variation of the said inductance because the latter is current-dependent. According to a further recognition of the invention the circuit arrangement is characterized in that it includes a current supply source for producing a compensating line-frequency sawtooth current through a winding of the linearity correction coil, the amplitude of the current being field-frequency modulated. The circuit arrangement according to the invention may further be characterized in that an additional winding is provided on the core of the linearity correction coil and is traversed by the compensating current. A circuit arrangement in which the modulator for modulating the line deflection current includes a compensation or bridge coil may according to the invention be characterized in that the additional winding is connected in series with the said coil.
The invention also relates to a linearity correction coil for use in a line deflection circuit having a core which is made of a magnetic material and is bias magnetized by at least one permanent magnet, which coil is characterized in that an additional winding is provided on the core.
Embodiments of the invention will now be described by way of example, with reference to the accompanying diagrammatic drawings, in which
FIG. 1 is the circuit diagram of a known circuit arrangement for line deflection in which the line deflection current is east-west modulated,
FIG. 2 shows the distorted image which is displayed on the screen when the circuit arrangement of FIG. 1,
FIG. 3 is a graph explaining the observed defect, and
FIGS. 4 and 7 show embodiments of the circuit arrangement according to the invention by which this defect can be cancelled.
FIG. 1 is a greatl simplified circuit diagram of a line deflection circuit of an image display apparatus, not shown further. The circuit includes the series combination of a line deflection coil L y , a linearity correction coil L and a trace capacitor C t , which series combination is traversed by the line deflection current i y . The collector of an npn switching transistor T r and one end of a choke coil L 1 are connected to a junction point A of a diode D, a capacitor C r and the said series combination. The other end of the choke coil is connected to the positive terminal of a supply voltage source which supplies a substantially constant direct voltage V b and to the negative terminal of which the emitter of transistor Tr is connected. This negative terminal may be connected to earth. The other junction point B of elements D and C r and of the series combination of elements C t , L y and L is connected to one terminal of a modulation source M for east-west correction which has its other terminal connected to earth. Diode D has the pass direction shown in the FIG.
To the base of transistor Tr line-frequency switching pulses are supplied. In known manner the said series combination is connected to the supply voltage source during the deflection interval (the trace time), diode D and transistor Tr conducting alternately. During the retrace time these elements are both cut off. Under these conditions the current i y is a sawtooth current. The coil L, which has a saturable ferrite core which is bias-magnetized by means of at least one permanent magnet, serves to correct the linearity of the current i y during the trace time, whilst the capacitance of the capacitor C t is chosen so that the currenct i y is subjected to what is generally referred to as S correction. During the retrace time, at point A pulses are produced the amplitude of which is much higher than that of the voltage V b and would be constant in the absence of modulation source M. Information from the field deflection circuit, not shown, of the image display apparatus and line retrace pulses, the latter for example by means of a transformer, are supplied in known manner to modulation source M. Amplitude-modulated line retrace pulses having a field-frequency parabolic envelope, as indicated in the FIG., are produced at point B. During the line trace time the voltage at point B is zero. Thus the current i y is given the desired field-frequency modulated form which is also shown in FIG. 1.
The amplitude of the envelope in point B at the beginning and at the end of the field trace time and the amplitude of this envelope at the middle of the said time can both be adjusted so that the image displayed on the display screen of the display tube (not shown) has the correct substantially rectangular form. If, however, the required modulation depth is comparatively large, a linearity error of the line deflection is produced which cannot be removed by means of the correction coil L.
FIG. 2 shows the image of a pattern of vertical straight lines as it is displayed on the screen with the correction coil L adjusted so that horizontal linearity is satisfactory along and near the central horizontal line. In FIG. 2 the defect is exaggerated. It is found that horizontal linearity is defective in other areas of the screen so that the vertical lines are displayed correctly in the right-hand half of the screen but as curves in the left-hand path, the defect increasing as the line is farther to the left.
This phenomenon can be explained with reference to FIG. 3. In this FIG. the inductance L of the linearity correction coil is plotted as a function of the magnetic field strength H. In the absence of current, H has a value H 0 owing to the bias magnetization. If an approximately linear sawtooth current i (t) as shown in the bottom left-hand part of FIG. 3 flows through the coil, the field strength H varies proportionally about the value H 0 , for the mean value of the current is zero. Because the curve of L is not linear, the variation L(t) of L, which is shown in the top right-hand part, is not a linear function of time. The resulting curve may be regarded as composed of a linear component and a substantially parabolic component which is to be taken into account when choosing the capacitance of capacitor C t .
Because owing to the east-west modulation the amplitude of current i(t) varies, the amplitude of L(t) also varies. This implies a field-frequency variation of L which is non-linear. This variation is undesirable. In the case of a small variation of the amplitude of current i(t) the variation of L(t) can be more or less neglected, but this is no longer possible when the amplitude of current i(t) varies greatly owing to the east-west modulation. L(t) varies according to different curves. FIG. 3 shows two of such curves and also illustrates the fact that the undesirable variation of L(t) is greatest at the beginning of the trace time and smallest at the end thereof.
FIG. 4 shows a circuit arrangement in which the defect described can be corrected. On the core of the correction coil L of the circuit of FIG. 1 an additional winding L 2 is provided. Winding L 2 is connected to a current source which produces a compensating current i 2 which has a line-frequency sawtooth variation and a field-frequency amplitude modulation. The envelope here also is parabolic, however, with a shape opposite to that of deflection current i y , that is to say having a minimum at the middle of the field trace time. The direction of current i 2 and the winding sense of winding L 2 relative to that of coil L are chosen so that the magnetic field produced in the core by winding L 2 has the same direction as the field produced by coil L. Hence the two field strengths are added. The amplitude of current i 2 and the turns number of winding L 2 can be chosen so that current i y flows through inductances the total value of which is not dependent upon the field frequency. The curve L(t) of FIG. 3 remains substantially unchanged. Consequently the undesirable field-frequency modulation is removed without variation of the bias magnetization, which would have been varied if current i 2 were a field-frequency current. Obviously the same result can be achieved by a choice such of the direction of current i 2 and of the winding sense of winding L 2 that the two field strengths are subtracted one from the other, whilst the curvature of the envelope of current i 2 has the same direction as that of the envelope of current i y .
The current source of FIG. 4 may be formed in known manner by means of a modulator in which a line-frequency sawtooth signal is field-frequency modulated, the envelope being parabolic. FIG. 5 shows a circuit arrangement in which current i 2 is produced by the modulation source which provides the east-west correction. In FIG. 5, the source M of FIG. 1 comprises a diode D', a coil L' and two capacitors C' r and C' t , which elements constitute a network of the same structure as the network formed by elements D, L y , C r and C t . The capacitor C' t is shunted by a modulation source V m which supplies a field-frequency parabolic voltage having a minimum at the middle of the field trace time.
With the exception of the linearity correction means to be described hereinafter, the circuit arrangement of FIG. 5 was described in more detail in U.S. Pat. No. 3,906,305. Hence it will be sufficient to mention that the capacitances of capacitors C r and C' r and of a capacitor C 1 connected between junction point A and earth and the inductance of coil L' are chosen so that the three sawtooth currents flowing through L y , L' and L 1 have the same retrace time. The capacitances of capacitors C t and C' t , which are large, are ignored. When voltage V b is constant, current i y is subjected to the desired east-west modulation having the form shown in FIG. 1.
Coil L y is connected in series with correction coil L, and winding L 2 is connected in series with coil L'. FIG. 5 shows that the current flowing through winding L 2 has the same waveform as the current i 2 of FIG. 4, for its envelope has the same shape as the voltage supplied by source V m . By a suitable choice of the number of turns of winding L 2 it can be ensured that the linearity correction remains the same for every line during the field trace time.
Modified embodiments of the circuit arrangement of FIG. 5 can also be used. FIG. 6 shows such a modified embodiment in which the capacitive voltage divider C r , C' r of FIG. 5 is replaced by an inductive voltage divider by means of a tapping on coil L 1 . A capacitor C 2 is included between the tapping and the junction point of diodes D and D', whilst capacitor C' t here forms part of two networks C t , L y and C' t , L' traversed by a sawtooth current. In FIG. 6 modulation source V m is connected via a choke coil L 3 to the junction point of D, D', C 2 and C' t . One end of winding L 2 is connected to the junction point of capacitor C' t and the coil L, whilst the other end is connected to earth via coil L'. The capacitances of capacitors C 1 and C 2 and the location of the tapping on coil L 1 are chosen so that the sawtooth currents flowing through L y , and L' and L 1 have the same retrace time, whilst the field-frequency linearity defect of FIg. 2 is cancelled by correctly proportioning winding L 2 .
Other east-west modulators are known in which the step of FIGS. 5 and 6 can be used. An example is the modulator described in the publication by Philips, Electronic Components and Materials: "110° Colour television receiver with A66-140X standard-neck picture tube and DT 1062 multisection saddle yoke," May 1971, pages 19 and 20, which modulator also comprises two diodes and a compensation coil L', which are arranged in a slightly different manner. In another example the east-west modulator and the line deflection generator are included in a bridge circuit whilst they are decoupled from one another by means of a bridge coil which has the same function as coil L' in FIGS. 5 and 6. In these circuit arrangements coil L and winding L 2 may be arranged in the same manner as in FIG. 6. The same applies to an east-west modulator using a transductor the operating winding of which is in series with the deflection coil.
In the abovedescribed embodiments of the circuit arrangement according to the invention the compensating current i 1 is provided by transformer action. In the embodiment of FIG. 7 the current source which supplies the current i 2 is connected in parallel with correction coil L, i.e., without an auxiliary winding. In this embodiment the east-west modulation is achieved not by means of a modulator, but by means of the fact that the supply voltage V b is the super-position of a field-frequency parabolic voltage on the direct voltage. In this known manner the supply source also is the modulator.
It will be seen that in the embodiments of FIGS. 4, 5 and 6 current i 2 counteracts the east-west modulation of deflection current i y . It was found in practice, however, that this counteraction is slight.
PHILIPS 26C566 /38Z (PHILIPS K11) CHASSIS K11 (20AX) NORTH SOUTH (NORD / SUD) CORRECTION CIRCUIT ARRANGEMENT FOR CORRECTING THE DEFLECTION OF AT LEAST ONE ELECTRON BEAM IN A TELEVISION PICTURE TUBE BY MEANS OF A TRANSDUCTOR :
A circuit arrangement for raster correction in a television picture tube by means of a transductor whose power winding is connected in parallel with at least a portion of the line deflection coils, the line deflection generator having a low internal impedance. In order to increase this impedance a mainly inductive impedance is connected in series with the generator. In a picture tube employing at least two electron beams the series impedance may include the convergence circuit. As a result the convergence in the corners of the picture screen is also improved. The linearity control circuit may likewise form part of the series impedance.
1. A deflection circuit for a cathode ray tube comprising a transistor horizontal deflection generator; a horizontal deflection coil parallel coupled to said generator; means for pincushion correction of said tube comprising a saturable reactor having a control winding adapted to receive a vertical deflection signal and a power winding parallel coupled to at least a portion of said deflection coil; and means for increasing the effectiveness of said correction means comprising an impedance element external to said generator having a substantially inductive reactance series coupled between said generator and said coil. 2. A circuit as claimed in claim 1 wherein said generator comprises a transformer having a tap and said power winding has a first end coupled to said coil and a second end coupled to said tap. 3. A circuit as claimed in claim 1 wherein said impedance element comprises means for controlling the linearity of the beam deflection. 4. A deflection circuit for a cathode ray tube having at least two electron beams comprising a transistor horizontal deflection generator; a horizontal deflection coil parallel coupled to said generator; means for pincushion correction of said tube comprising a saturable reactor having a control winding adapted to receive a vertical deflection signal and a power winding parallel coupled to at least a portion of said deflection coil; means for increasing the effectiveness of said correction means comprising an Impedance element external to said generator having a substantially inductive reactance series coupled between said generator and said coil; and means for dynamically converging said beams comprising a convergence circuit coupled to said horizontal generator and to said transductor. 5. A circuit as claimed in claim 4 wherein said generator comprises a transformer having a tap and said power winding has a first end coupled to said coil and a second end coupled to said tap. 6. A circuit as claimed in claim 4 wherein said impedance element comprises means for controlling the linearity of the beam deflection.
A circuit arrangement for raster correction with the aid of a transductor is described, for example, in U.S. Pat. No. 3,444,422. In this patent the power winding of a transductor is connected in parallel with the horizontal deflection coils while the control winding receives a signal of field frequency so that the current of line frequency which flows through the deflection coils is modulated at the field-frequency (East-West correction), whereas the vertical deflection current is modulated at the line frequency (North-South correction). However, in this known arrangement there is the difficulty that the transductor can exert little influence on the horizontal deflection current if the internal impedance of the deflection generator is low because the transductor then only constitutes an additional load on the generator. This is the case when the deflection generator includes a valve with feedback -- or a switch formed with one or more transistors. In order to be able to use a transductor arrangement also in such a case the circuit arrangement according to the invention is characterized in that a mainly inductive impedance is connected in series between the said parallel arrangement and the deflection generator.
Due to the step according to the invention the internal impedance of the deflection generator is increased and the different components of the circuit remain mainly inductive so that the deflection current is more or less linear when the voltage provided by the deflection generator during the line scan period is substantially constant. The series impedance may be, for example, a fixed coil. However, the invention is furthermore based on the recognition of the fact that the increase in the internal resistance of the horizontal deflection generator may not only be obtained by a constant impedance, but other arrangements envisaging other improvements of the deflection may be used for this purpose. In that case even special improvements may be obtained as will be apparent hereinafter and possible small non-linearities of the additionally used arrangements have no detrimental results.
It is true that in known convergence circuits in picture tubes employing a plurality of electron beams a satisfactory improvement is obtained for the central horizontal and vertical lines of a picture tube of the shadow mask type. However, it is found that convergence errors may subsist in the corners of the picture. Known circuit arrangements which correct these second-order errors are often complicated and expensive. In the circuit arrangement according to the invention a satisfactory compensation of such convergence errors is possible in a simple manner if the series impedance which is arranged between the horizontal deflection generator and the deflection coils includes the convergence circuit. In this manner the sum of the deflection current and of the current derived for the field correction and modulated by the transductor flows through the convergence circuit so that the desired additional convergence correction in the corners of the written raster is obtained.
In order that the invention may be readily carried into effect a few embodiments thereof will now be described in detail by way of example with reference to the accompanying diagrammatic drawings in which:
FIG. 1 shows a circuit arrangement in which the transductor is connected in parallel with the deflection coils, while in
FIG. 2 the transductor is only fed by part of the voltage applied to the deflection coils.
FIG. 1 shows two line-output transistors 1 and 2 which are arranged in series. The emitter of transistor 2 is connected to ground through a winding 3 while the collector of transistor 1 is connected through a winding 4 and a small series impedance 5, preferably a resistor, to the positive terminal of a supply source V b whose negative terminal is connected to ground.
Windings 3 and 4 are wound together with an EHT-winding 6 on the same transformer core 7. The ends of windings 3 and 4 remote from each other are connected through the capacitor 10 for the S-correction to the deflection-unit consisting of two windings 8 and 9 arranged, for example, in parallel. The base of transistors 1 and 2 receive pulses of line frequency in a manner not shown in FIG. 1 so that these transistors are cut off during the flyback period. During the scan period, a substantially constant voltage is applied to the deflection unit. Consequently a more or less sawtooth-shaped current flows through windings 8 and 9. The bipartite power winding 11 of a transductor ensuring the raster correction is connected in parallel with this deflection unit 8, 9. The control winding 12 of said transductor, and a converting capacitor 13 in parallel therewith form part of the circuit for the vertical deflection through terminals 14 and 15. An adjustable coil 16 with which the raster correction can be adjusted exactly is connected in series with winding 12.
Windings 3 and 4 have the same number of turns so that pulses of the same amplitude and reversed polarity are produced at the emitter of transistor 2 and at the collector of transistor 1. As a result a disturbing radiation of these pulses is reduced. Furthermore, transistor types are chosen in this Example for transistors 1 and 2 whose collector-base diodes may function as efficiency diodes. All this has been described in U.S. Pat. No. 3,504,224.
According to the invention the convergence circuit 17 is arranged through a separation transformer 20 between the end of winding 3 remote from winding 4 and the horizontal deflection coils 8, 9. Furthermore, this current branch includes the linearity control circuit 21 which comprises the parallel arrangement of a resistor and a coil whose inductance is adjustable, for example, by means of premagnetization of the core of the coil. A current, which is the sum of the current for the deflection coils 8, 9 and of the current for the power winding 11 of the transductor, flows through the primary winding of transformer 20. This primary current is transformed to the secondary circuit of transformer 20 so that a current flows through convergence circuit 17.
In known arrangements the convergence current is only influenced by the deflection current itself. It has been found that in this case the convergence correction is not sufficient in the corners of the picture. At these areas, where the deflection in both directions is at a maximum, a greater intensity of the convergence current is required. This is especially the case in picture tubes having a great deflection angle and according to the invention this is achieved in that the current which is derived from the power winding 11 of the transductor for the raster correction is also applied to the convergence circuit. This current flows from the horizontal deflection generator constituted by windings 3 and 4 through the primary winding of transformer 20 to power winding 11 of the transductor. The transductor current is in fact at a minimum in the center of the picture and increases towards the edges and particularly towards the corners. Thus the convergence current varies in the desired manner. According to the invention the desired improvements of the convergence correction and simultaneously the likewise desired increase in the internal resistance of the horizontal deflection generator is consequently obtained without a considerable increase in the number of required circuit elements and without disturbing the normal operation of the circuit arrangement. Due to transformer 20 a terminal of convergence circuit 17 may be connected to ground so that the convergence can be adjusted safely. If necessary, a suitable impedance transformation may also be obtained with the aid of transformer 20.
The linearity control circuit 21 may alternatively be connected in series with the said branch which includes transformer 20. As a result the internal resistance of the horizontal deflection generator for the line frequency is further increased without the field correction and the convergence correction being disturbingly influenced.
FIG. 2 shows a modification of the circuit arrangement according to the invention in which the deflection current is not changed relative to that of FIG. 1. The end of power winding 11 of the transductor shown on the upper side of FIG. 1 is connected to ground in FIG. 2. In addition convergence circuit 17 is included between winding 3 and ground so that separation transformer 20 may be omitted. If as a first approximation the impedances 5 and 17 are assumed to be negligibly small relative to the other impedance of the circuit arrangement, power winding 11 may be considered to be connected to a tap on the deflection generator 3, 4. Consequently, only approximately half the voltage of the deflection generator is applied to transductor winding 11 which winding must therefore be proportioned in such a manner that it can convey a current which is approximately twice as large as that of FIG. 1. This larger current also flows through convergence circuit 17 which, with the omission of separation transformer 20, is favorable for the convergence in the corners of the picture screen.
In FIG. 2 the emitter of transistor 2 is connected to ground i.e., the said tap on the deflection generator. During the scan period the series arrangement of supply source V b and windings 3 and 4 FIG. 1 is substantially short-circuited by transistors 1 and 2. In order that these transistors in the circuit arrangement according to FIG. 2 operate under the same circumstances as those in FIG. 1, an additional winding 24 must be wound on core 7 between windings 4 and 6, winding 24 having the same number of turns as winding 3, and the collector of transistor 1 must be connected to the junction of windings 6 and 24.
The end of power winding 11 connected to ground in FIG. 2 may alternatively be connected for the desired adjustment of the corner convergence to a different tap on the transformer, that is to say, on winding 3 or 4.
Resistor 5 serves in known manner mainly as a safety resistor so that in case of an inadmissible load of the EHT, for example, as a result of flash-over in the picture tube, the supply voltage for transistors 1 and 2 is reduced so that overload of these transistors is avoided.
Colour television display apparatus incorporating a television display tube
1. Colour television display apparatus incorporating a television display tube having a display screen and two deflection coils for the deflection in two directions of electron beams which are generated in the tube substantially in one plane, a first direction of deflection being substantially parallel to the said plane whilst the second direction of deflection is substantially at right angles to the first direction, the field generated by the deflection coil for deflection in the first direction having a distribution in which its meridional image plane substantially coincides with the screen whilst the field generated by the deflection coil for deflection in the second direction has a distribution in which its sagittal image plane substantially coincides with the screen, the deflection errors due to comma and anisotropic astigmatism being substantially equal to zero, whilst at least one deflection coil is divided into two substantially equal coil halves, characterized in that in order to correct for tolerance angular errors in the orientation of the plane in which the electron beams are generated relative to the first direction of deflection the split deflection coil generates a magnetic quadripolar field the polar axes of which substantially coincide with the directions of deflection and the field strength of which is a substantially quadratic function of the instantaneous strength of the deflection current flowing through at least one deflection coil, and means for clamping the peak of said quadratic field. 2. Apparatus as claimed in claim 1, characterized in that a substantially parabolic correction current which is adjustable in amplitude and in polarity flows in the same direction as the deflection current in one coil half and in the opposite direction in the other coil half and is zero at the middle of the trace interval of the deflection current. 3. Apparatus as claimed in claim 2, in which one direction of deflection is horizontal and the other is vertical, characterized in that a line-frequency correcting current flows through the coil halves of the deflection coil for horizontal deflection and a field-frequency correction current flows through the coil halves of the deflection coil for vertical deflection. 4. Apparatus as claimed in claim 2, characterized in that a sawtooth current supplied by the deflection current generator which produces the deflection current flows through a potentiometer the setting of the slider on which determines the adjustment of the polarity and of the amplitude of the correcting current. 5. Apparatus as claimed in claim 4, in which the deflection current is of field frequency, characterized in that the setting of the slider on the potentiometer also renders symmetrical the deflection fields generated by the coil halves. 6. A display apparatus as claimed in claim 1 wherein said split coil field strength is substantially the sum of quadratic functions of the current flowing through both coils. 7. A color television deflection system for a television display tube having a display screen, said system comprising two deflection coils for the deflection in two directions of electron beams which are generated in the tube substantially in one plane, said first direction of deflection being substantially parallel to the said plane, the second direction of deflection being substantially at right angles to the first direction, the field generated by the deflection coil for deflection in the first direction having a distribution in which its meridional image plane substantially coincides with the screen, the field generated by the deflection coil for deflection in the second direction having a distribution in which its sagittal image plane substantially coincides with the screen, the deflection errors due to comma and anisotropic astigmatism being substantially equal to zero, at least one deflection coil comprising two substantially equal coil halves, means for correcting for tolernace angular errors in the orientation of the plane in which the electron beams are generated relative to the first direction of deflection comprising means for providing that the split deflection coil generates a magnetic quadripolar field the polar axes of which substantially coincide with the directions of deflection and the field strength of which is a substantially quadratic function of the instantaneous strength of the deflection current flowing through at least one deflection coil, and means for clamping the peak of said quadratic field. 8. A deflection system as claimed in claim 7 wherein said split coil field strength is substantially the sum of quadratic functions of the current flowing through both coils.
Such an apparatus is described by J. Haantjes and G. J. Lubben in "Philips Research Reports", Volume 14, February 1959, pages 65-97 and in U.S. Pat. No. 2,886,125. In this apparatus the landing points of the electron beams on the display screen coincide everywhere, in other words the various beams, which generally are three in number, which intersect the deflection plane along a straight line are imaged as points on the screen. It is assumed that both the construction of the device or devices which generate the beams, for example three cathodes, and the distribution of the deflection fields exactly satisfy the requirements derived in the said paper. In practice, however errors are produced which are due to tolerances so that the images of the beams on the screen are not points but lines which are substantially parallel to the second direction, i.e. convergence errors, for when a point is referred to what is actually meant is that each electron beam strikes a phosphor dot or stripe on the screen to cause it to luminesce in a given colour, the landing points being associated so as to be perceived as a single point. This is no longer the case if the aforementioned straight line, which is the projection of the plane of the three cathodes in the deflection plane, does not exactly coincide with the first direction of deflection but is at an angle thereto. This error is a tolerance error, i.e. it is small, and may be due to a slight misplacement of the cathodes and/or to a slightly incorrect field distribution within the display tube and hence to tolerances in the construction of the deflection coils.
If the first direction of deflection is horizontal and the second one is vertical, the said error entails a convergence error in the vertical direction. The aforementioned straight line in the deflection plane can be made to coincide with the horizontal direction of deflection by rotation. Attempts have been made to cancel the convergence errors due to this rotation by means of a coil which is axially arranged on the neck of the display tube and through which an adjustable direct current flows. The effect of this coil is comparable to that of a focussing coil; it exerts a force on the travelling electrons which causes their paths to be helical, so that some compensation is obtained. It has been found, however, that this solution has the following disadvantages: the residual errors in the corners are increased; the effect on the horizontal and vertical directions are different, so that satisfactory adjustment in both directions is difficult to realise; depending upon the axial position an undesirable influence may occur at the centre of the screen which in turn can be eliminated by the means, for example permanent magnets, provided for static convergence, requiring iterative and hence time-consuming trimming. Furthermore the coil is an expensive component.
The present invention is based on the recognition that the aforementioned convergence errors due to tolerance errors in the construction of the display tube and/or of the deflection coils can be eliminated by means of simple circuits without the need for additional components to be mounted on the neck of the display tube whilst avoiding the aforementioned disadvantages. For this purpose the apparatus according to the invention is characterized in that to correct for tolerance angular errors in the orientation of the plane in which the electron beams are generated relative to the first direction of deflection the split deflection coil generates a magnetic quadripolar field the polar axes of which substantially coincide with the directions of deflection and the field strength of which is a substantially quadratic function of the instantaneous strength of the deflection current flowing through either deflection coil or the sum of both quadratic functions.
It should be mentioned that it is known to use a split deflection coil to generate a quadripolar field the polar axes of which substantially coincide with the directions of deflection. This is described in U. S. Pat. No. 3,440,483 in which, however, the field strength of the quadripolar field is a function of the product of the values of the two deflection currents so that deflection errors due to anisotropic astigmatism can be corrected. In contradistinction thereto the present application described an apparatus having substantially no anistropic astigmatism whilst the quadripolar field generated according to the invention has a field strength which depends upon the value of either deflection current or upon the sum of the squares of the two deflection currents. For the sake of clarity it should be mentioned that in the apparatus according to the said U.S. Patent, in the absence of the correction quadripolar field described, the image of a beam on the screen is a tilted ellipse, whereas in the present application the corresponding image when not corrected is a vertical line.
The known apparatus has some isotropic astigmatism so that the vertical focal lines, i.e. the Meridional focal lines of the horizontal deflection plane and the sagittal focal lines of the vertical deflection plane, coincide with the display screen. Since the imaginary ribbon-shaped beam produced by the three beams together has substantially no dimension in the vertical direction, its image on the screen is a point. In these circumstances the term "isotropic astigmatism" as used herein in actual fact is to be understood to mean that the coefficients which determine the isotropic astigmatism differ from the desired values. Consequently the cross-sectional area on the screen of the imaginary thick beam of circular cross-section in the deflection plane (see FIG. 2 of the said paper in which, however, the three beams are generated in a vertical plane) does not degenerate into a straight line but takes the form of an ellipse the axes of which are parallel to the directions of deflection. Means for correcting such undesirable isotropic astigmatism is described in U.S. patent application Ser. No. 447,564 filed March 4, 1974. In this means a correcting quadripolar field which varies with the square of the strength of either deflection current is generated in the deflection region. However, the axes of said quadripolar field lie substantially along the diagonal between the axes of the deflection directions and the field is generated by separate windings and not by the deflection coil or coils. It should be noted that the apparatus according to the invention also may be subject to this defect which in this case may be corrected in the manner described in the said U.S. patent application. For the sake of simplicity this will be disregarded hereinafter, that is to say the deflection coil will be assumed to have the correct degree of isotropic astigmatism, causing the landing points of the beams on the screen to coincide in one point everywhere but for the abovementioned tolerance error.
In order that the invention may be more readily understood, embodiments thereof will now be described by way of example with reference to the accompanying diagrammatic drawings, in which:
FIG. 1 is a sectional view of a colour television display tube subject to the defect to be corrected,
FIG. 2 shows schematically the ensuring convergence error on the display screen of the tube,
FIGS. 3, 4, 5 and 7 are circuit diagrams of embodiments of correction circuits, and
FIG. 6 is a wave form obtained in the circuit of FIG. 5.
FIG. 1 is a simplified elevation of a cross-section of a colour television display tube 1 taken on the deflection plane at right angles to the axis of the tube in a direction opposite to the direction of propagation of the electron beams, the deflection coils being omitted for simplicity. Three electron beams L, C and R are generated in one plane, the beam C substantially coinciding with the axis of the tube 1 and the beams L and R being located to the left and to the right respectively of the beam C. If the construction of the devices, for example cathodes, which generate the beams and the field distribution within the tube 1 were exactly as desired, the points of intersection of the beams L, C and R with the deflection plane would be a straight line coinciding for example with the X axis, which coincides with the direction of horizontal deflection, the Y axis coinciding with the direction of vertical deflection. However, owing to tolerances the points of intersection lie on a straight line D which is at an angle α to the X axis which it intersects in C.
The paper mentioned in the second paragraph of this application shows that an imaginary thick beam may be considered the cross-section S of which with the plane of deflection is a circle. The line section LCR of FIG. 1 is a diameter of this circle. If the horizontal deflection field has a distribution in which the meridional image plane substantially coincides with the display screen of the tube 1 whilst the vertical deflection field has a distribution in which the sagittal image plane substantially coincides with the screen, and if moreover the deflection errors due to comma and both anisotropic and isotropic astigmatism are substantially equal to zero, all the points on and within the circle S are imaged on vertical line everwhere on the screen. It is supposed that the correct degree of isotropic astigmatism is actually obtained. Otherwise the image of the circle S would be an ellipse the axes of which are parallel to the X and Y axes, i.e. there would be a horizontal convergence error.
In these circumstances the beams L, C and R of FIG. 3 are imaged on the screen 2 of the tube 1 along vertical lines, some of which are shown (in exaggerated form) in FIG. 2, with the exception of the image at the midpoint of the screen, i.e. without deflection, where they coincide. In the ideal case in which the beams L and R of FIG. 1 would lie on the X axis, i.e. with α = 0, in each triplet L', C', R' in FIG. 2 the points L' and R' would coincide with the point C'. Consequently the error angle α results in a vertical convergence error on the screen. In FIG. 1 the beam L lies above the X axis and the beam R beneath the X axis. Because the beams cross within the tube, the points L' and R' in FIG. 2 always lie beneath and above the point C' respectively.
According to the invention a magnetic correction quadripolar field is generated the polar axes of which substantially coincide with the X and Y axes and four lines of force of which are shown in FIG. 1. The quadripolar field does not influence the beam C which is located at the centre of the deflection plane. The beams L and R are subject to forces F L and F R respectively which are superposed on the forces exerted by the deflection fields. FIG. 1 shows that as a result the angle α is effectively reduced to substantially zero so that the convergence error of FIG. 2 is cancelled.
Such a quadripolar field is obtainable by causing an additional current, the difference current, to flow through a deflection coil divided in two coil halves in a manner such that the said current is added to the deflection current in one coil half and subtracted from it in the other coil half. FIG. 2 shows that the convergence errors on the left-hand and right-hand halves of the screen 2 have the same sign and that they have the same sign in the upper and lower halves. Hence it is desirable for the value of the difference current to vary substantially as the square of each deflection. Because initially the value and polarity of the angle α are unknown, the current must be adjustable both in amplitude and in polarity. At the middle of the line and field trace intervals the angle must be zero. For this purpose either one or both deflection coils may be used.
Because the images L', C', R' in FIG. 2 are vertical, i.e. are not tilted, the convergence error to be corrected is to be considered as an isotropic astigmatic deflection error. Hence the line-frequency component of the difference current must be a function of horizontal deflection only and its field-frequency component must be a function of vertical deflection only. Thus it is simpler, but not necessary, to cause the line-frequency component of the difference current to flow through the split deflection coil for horizontal deflection and its field-frequency component to flow through the split deflection coil for vertical deflection.
FIG. 3 shows a simple circuit for generating a line-frequency difference current which satisfies the said requirements. A line deflection current generator 3 at one terminal supplies a line-frequency sawtooth current i H to line deflection coil halves 4' and 4", which in this embodiment are connected in parallel for the current i H . Adjustable coils 5' and 5" of low inductance are connected in series with the coil halves 4' and 4" respectively. The coils 5' and 5" may be adjusted jointly and oppositely so as to eliminate in knwon manner any asymmetry of the deflection fields generated by coil halves 4' and 4". The ends of the coils 5' and 5" not connected to the coil halves 4' and 4" respectively are connected to one another via a potentiometer 6 the slider on which is connected to the other terminal of the generator 3. The resistance of, for example, 4.7 ohms of the potentiometer 6 is low compared with the impedance of the coil halves 4' and 4" for the line repetition frequency. Thus a sawtooth voltage the polarity and amplitude of which depend upon the position of the slider is produced across the potentiometer 6. As a first approximation this voltage may be considered as being produced by a voltage source of low internal impedance. The coil halves 4' and 4" pass a current which is proportioned to the integral of the voltage across the potentiometer 6 and consequently is the required parabolic correction difference current i KH . In one coil half it flows in the same direction as the current i H /2 and in the other coil half it flows in the opposite direction. For this purpose it is required that the position of the slider on the potentiometer 6 should differ from the electric midpoint thereof.
The potentiometer 6 is shunted by the series combination of two resistors 7' and 7" the junction point of which is connected to the anode of a diode 8 the cathode of which is connected to the slider on the potentiometer 6. The diode 8 and the resistors 7' and 7" ensure that the peak of the parabola will be at zero. In actual fact the diode 8 produces a direct current which compensates for the sagging of the parabola, provided that the resistances of the resistors 7' and 7" are equal and have the correct value, for example 8.2 ohms. This direct current also is a difference current and since the diode 8 is connected to the slider on the potentiometer 6 the reversal of its polarity is automatically effected together with that of the parabolic component.
A disadvantage of the circuit of FIG. 3 may be that the obtainable amplitude of the current i KH is limited because the permissible value of the potentiometer 6 is limited, for a comparatively large value of this potentiometer will increase dissipation and give rise to a linearity error of the deflection current whilst the current i KH will no longer be parabolic but will also include higher-order components. The amplitude i KH may be increased without increasing the resistance of the potentiometer 6 by coupling the latter to the remainder of the circuit by means of a transformer. This may be achieved by an autotransformer, as is shown in FIG. 4. Two windings 19' and 19" which are bifilarly wound on the same core and have the polarities shown are connected in series between the ends of the coils 5' and 5" not connected to the coil halves 4' and 4" respectively. The potentiometer 6 is connected between two tappings on the windings 19' and 19" which are symmetrical with respect to the junction point thereof and the potentiometer slider is connected to said junction point via the series combination of the diode 8 and a resistor 7.
In the circuit shown in FIG. 4 the operation of the balancing coils 5' and 5" is not disturbed, provided that the overall inductance value of the windings 19' and 19" between the junction point of the winding 19' and the coil 5' and the junction point between the winding 19" and the coil 5" is small compared with the inductance value of the coil halves 4' and 4" and the coils 5' and 5" measured between the same points. In a practical embodiment of the circuit of FIG. 4 the latter value is 3.55 mH and the former value is 1.25 mH. This means that the effect of the balancing coils 5' and 5" is reduced by only about one third. The tappings on the windings 19' and 19" are provided at the midpoints thereof, the value of the resistor 6 is about 3.3 ohms and that of the resistor 7 about 0.5 ohm. It should be noted that the resistance of the windings 19' and 19" should not be too small, for otherwise the direct component of the difference current would be short-circuited.
FIG. 5 shows a simple circuit for producing a field-frequency difference current in field-frequency deflection coil halves 9' and 9". Since these coil halves are predominantly resistive for the field repetition frequency, the circuits shown in FIGS. 3 and 4 cannot be used. A field deflection current generator 10 supplies a fieldfrequency sawtooth current i V to coil halves 9' and 9" which are connected in series in this embodiment. The series combination of a diode 11', a potentiometer 12' and a second diode 13' and the series combination of a third diode 11", a potentiometer 12" and a fourth diode 13" are connected in parallel with the series combination of the said coil halves, the said four diodes having the polarities shown in FIG. 5. An isolating resistor 14' is connected between the slider on the potentiometer 12' and the junction point of the coil halves 9' and 9", and an isolating resistor 14" is connected between the slider of the potentiometer 12" and the said junction point, the values of the isolating resistors being high relative to the impedance of the coil halves, for example about 100 ohms.
During one half of the field trace interval the current i V flows in the direction shown. Diodes 11' and 13' are conducting whereas diodes 11" and 13" are cutoff. Across the potentiometer 12' a sawtooth voltage is produced so that, if the position of the slider of the potentiometer 12' is different from the electric midpoint of the potentiometer, a sawtooth correction difference current i' KV flows through the coil half 9', for example in a direction opposite to that of the current i V , whilst the coil half 9" passes a sawtooth correction difference current i" KV in the same direction as the current i V , the currents i' KV and i" KV being substantially equal. It should be noted that a difference current, in this embodiment i" KV , flows through a diode, in this embodiment 13', from the cathode to the anode. However, because the elements 9', 9", 11', 12', 13' and 14' form a Wheatstone bridge comprising resistors, the diodes cannot be cut off.
During the other half of the trace interval current i V flows in the other direction. The diodes 11" and 13" are conducting and the diodes 11' and 13' are cut off. Sawtooth difference currents are produced which are derived from the slider on potentiometer 12". In FIG. 6 the variation of the extreme value i KVmax of the difference currents is shown as a function of time, T denoting the field trace interval. At the middle of the interval T these currents are zero, because the current i V and hence the voltage across the potentiometer 12' or 12" respectively are zero. Owing to the voltage drop across the diode the difference currents are zero for a certain time before and after the middle of the interval T. The resulting curves may be regarded as approximate parabolas, for practice has shown that the residual convergence error is negligibly small. Because the difference currents produced are sawtooth currents, the potentiometers 12' and 12" ensure also that the deflection fields generated by the coil halves 9' and 9" are symmetrical. An advantage of the circuit of FIG. 5 is that the adjustments of the upper half and of the lower half are independent of one another, which conduces to clarity. In the embodiment described both potentiometers have a resistance of about 330 ohms.
It will be appreciated that the quadripolar field generated will only be capable of correcting for the vertical convergence error if the angle α is very small. The error introduced by the incorrect position of the line D is compensated for by the quadripolar field according to the invention, it is true, however, at large values of the angle α this field in turn introduces new errors, especially in the corners of the screen. Practice has shown that an angle of from 2° to 3° still can be corrected.
Hereinbefore no statement has been made about the construction of the deflection coils. If they are in the form of saddle coils, no special steps are required. If, however, they are wound toroidally, a step as described in U.S. patent application Ser. No. 390,701 filed August 23, 1973 must be used which consists in the introduction of the difference currents into the deflection coil halves via tappings. In this case the simple circuits of FIGS. 3, 4 and 5 are to be replaced by circuits in which the parabolic difference currents are generated in a different manner, for example by separate generators.
In the embodiments described the coil halves 4' and 4" for horizontal deflection are connected in parallel for the line deflection current i H , whereas the coil halves 9' and 9" for vertical deflection are connected in series for the field deflection current i V . Obviously this is not of importance for the invention and the coil halves may be connected in a different manner. FIG. 7 shows an embodiment in which the coil halves 4' and 4" are connected in series for the current i H . In this embodiment two diodes 8' and 8" are required. It will further be appreciated that the invention may also be applied if the electron beams are generated in a plane of substantially vertical orientation, in which case the convergence error to be corrected is horizontal.
ULTRASONIC REMOTE CONTROL RECEIVER PHILIPS CHASSIS K11.
PHILIPS 26C566 /38Z (PHILIPS K11) CHASSIS K11 (20AX) Tube Philips 20 AX A66-500X.
Includes on separated PCB the Ultrasonic Remote Control Receiver Devices Assy's With Amplifier and Discriminator and LOGIC.
(NOTE that PHILIPS has developed his own circuit, and these descriptions are for educational purphoses)
An ultrasonic remote control receiver wherein an incoming ultrasonic signal is converted to square wave pulses of the same frequency by a Schmitt trigger circuit; digital circuits are thereafter used to count pulses resulting from the incoming signal over a predetermined period of time; a decoder activates one of a plurality of outputs in dependance to the number of pulses counted, provision is made to prevent interference signals from producing undesired control outputs.
1. An ultrasonic remote control receiver for applying a control signal to a selected one of a plurality of control channels in response to and dependent on the frequency of a received ultrasonic signal comprising:
2. An ultrasonic remote control receiver comprising:
3. An ultrasonic remote control receiver comprising:
4. The ultrasonic remote control receiver as defined in claim 3, wherein said means producing square pulses is a Schmitt trigger circuit and said means providing a signal input to said sequence controller is a retriggerable monostable multivibrator.
5. An ultrasonic remote control receiver comprising:
6. An ultrasonic remote control receiver comprising:
7. An ultrasonic remote control receiver as defined in claim 6 further comprising a monostable multivibrator between the output of said Schmitt trigger circuit and the remaining elements of said receiver.
8. An ultrasonic remote control receiver as defined in claim 6 further comprising a bistable multivibrator between the output of said Schmitt trigger circuit and the remaing elements of said receiver.
9. The ultrasonic remote control receiver as defined in claim 7 wherein the hold period of said monostable multivibrator is slightly less than one half the period of said square wave pulses from said Schmitt trigger circuit.
To obtain the simplest possible transmitter construction in ultrasonic remote control, modulation of the emitted ultrasonic frequencies is not employed; to control different operations different frequencies are emitted which must be recognized in the receiver and evaluated for carrying out the different functions associated therewith. Presently, to recognize the different frequencies, use is made of resonant circuits, each of which contains one or more coils tuned in each case together with a capacitor to one of the useful frequencies.
These hitherto known receivers have numerous disadvantages. Thus, for example, before starting operation of the receiver a time-consuming alignment procedure must be carried out with which the resonant frequencies of the individual resonant circuits are set. Since it is inevitable that with time the resonant circuits become detuned, it may be necessary to repeat the alignment procedure.
A further disadvantage is that the known receivers cannot be made by integrated techniques because the coils used therein are not suitable for such techniques.
The problem underlying the invention is thus to provide an ultrasonic remote control receiver of the type mentioned at above which is extremely simple to set and in addition can be made by integrated techniques.
To solve this problem, according to the invention an ultrasonic remote control receiver of the type mentioned above contains a counter for counting the useful frequency oscillations received during a fixed measuring time, a sequence control device which determines the measuring time and which is started on receipt of a useful frequency, and a decoder comprising several outputs which is connected to the outputs of the counter, said decoder emitting a control signal at the output associated with the count reached at the end of the measuring time.
In the receiver constructed according to the invention the frequency emitted by the transmitter is identified by counting the oscillations received during a measuring time. The evaluation of the count reached at the end of the measuring time takes place in a decoder which emits a control signal at a certain output according to the count. The measuring time is fixed by a sequence control device which is set in operation on receipt of useful frequency signals.
In such a receiver the only quantity which has to be exactly fixed is the measuring time; it is therefore no longer necessary to align components to certain frequencies. Since no coils are required, the novel receiver can also be made up of integrated circuits.
A further development of the invention resides in that an interference identifying device is provided which on receipt of interference frequencies differing from the useful frequencies interrupts the operation of the sequence control device.
Hitherto known ultrasonic remote control receivers respond to any oscillation received if the frequency thereof has a value which excites a resonant circuit in the receiver. There is no way of distinguishing between oscillations received from the remote control transmitter and from interference sources.
Interfering ultrasonic oscillations may be due to many different causes. For example, noises such as hand clapping, rattling of short keys such as safety keys, operating cigarette lighters, rattling of crockery and the like cover a frequency spectrum reaching from the audio frequency range far into the ultrasonic region. The ultrasonic components may have the effect of simulating a useful frequency and cause an erroneous function in the receiver.
The interference identifying device according to the further development is constructed in such a manner that it recognizes oscillations having frequencies deviating from the useful frequencies and as a result of this recognition switches off the sequence control device. This switching off prevents the counter state reached from being passed to the decoder and consequently the latter cannot emit an erroneous control signal.
With this further development of the ultrasonic remote control receiver the operation of equipment such as radio and television sets is made extremely reliable and interference-free. During the operation of such a set it is no longer possible for the remote control to become operative, triggered by interference noises, eliminating for example the possibility of unintentional program or volume changes.
Examples of embodiment of the invention are illustrated in the drawings, wherein:
FIG. 1 shows a block circuit diagram of a remote control receiver according to the invention;
FIG. 2 is a diagram explaining the mode of operation of the circuit according to FIG. 1;
FIG. 3 shows another embodiment of the invention;
FIG. 4 is a diagram explaining the mode of operation of the circuit according to FIG. 3;
FIG. 5 is a diagram illustrating interference frequency identification in the circuit according to FIG. 3;
FIG. 6 shows a block circuit diagram of another embodiment of part of the circuit according to FIG. 3;
FIG. 7 is a diagram explaining the mode of operation of the embodiment according to FIG. 6;
FIG. 8 is a block circuit diagram of a further embodiment of a part of the circuit according to FIG. and, an
FIG. 9 is a diagram explaining the mode of operation of the embodiment according to FIG. 8.
The ultrasonic remote control receiver shown in FIG. 1 comprises an input 1 which is connected to an ultrasonic microphone intended to receive ultrasonic signals coming from a remote control transmitter. For each function to be performed by the receiver the remote control transmitter emits one of several unmodulated different useful frequencies which are spaced from each other a constant channel spacing Δ f and which all lie within a useful frequency band.
To obtain a signal which is as free as possible from noise at the input 1, a band filter and a limiting amplifier are preferably incorporated between the ultrasonic microphone and the input 1. The band filter may be made up of two active filters whose resonant frequencies are offset with respect to each other so that a pass band curve in the useful frequency band is obtained which is as flat as possible.
The input 1 leads to a Schmitt trigger 2 which converts the electrical signal applied thereto with the frequency of the ultrasonic signal to a sequence of rectangular pulses. The output 3 of the Schmitt trigger 2 is connected to the input 6 of a frequency divider 7 which is in operation for the duration of a control pulse applied to its control input 8 and divides the recurrence frequency of the pulses supplied thereto at the input 6 thereof in a constant division ratio. The output 9 of the frequency divider 7 is connected to the input 10 of a counter 11 which counts the pulses coming from the frequency divider 7. The counter 11 is a four-stage binary counter whose stage outputs are connected to the inputs of a store (register) 12 which is so constructed that on application of a control pulse to the input 12 thereof it takes on the counter state in the counter 11 and stores said counter state until the next pulse at the input 13. The stage outputs of the store 12 are fed to the inputs of a decoder 14 which decodes the counter state contained in the store 12 in such a manner that a control signal is emitted at that one of its outputs D0 to D9 which is associated with the decoded counter state.
The output 3 of the Schmitt trigger 2 is also connected to the input 4 of a monoflop 5 which is brought into its operating state by each pulse at the output 3 of the Schmitt trigger. It returns from this operating state to its quiescent state after expiration of a hold time depending on its intrinsic time constant if it does not receive a new pulse prior to expiration of this hold time. It is held in the operating state by each pulse received during the hold time until it finally flops back into the quiescent state when the interval between two successive pulses is greater than its hold time.
The output 15 of the monoflop circuit 5 is connected to the input 16 of a sequence control device 17 which is set in operation by the signal emitted in the operating state of the monoflop 5. Supplied to the sequence control device by 17 via a Schmitt trigger 18 at a control input 19 are pulses having a recurrence frequency derived from oscillations of the same frequency, for example, twice the mains frequency of 100 c/s, applied to the input 20. The sequence control device 17 is so constructed that in a cyclically recurring sequence in time with the pulses supplied to it at the input 19 it emits pulses at the outputs 21, 22 and 23 whose duration is equal to the period of the oscillation applied to the input 20. The output 21 of the sequence control device 17 is connected to the control input 8 of the frequency divider 7, the output 22 is connected to the control input 13 of the store 12 and the output 23 thereof is connected to the reset input 24 of the counter 11.
The mode of operation of the circuit of FIG. 1 will now be explained with the aid of the diagram of FIG. 2 which shows the variation with time of the signals at the output 3 of the Schmitt trigger 2 and at the inputs 16 and 19 as well as the outputs 21, 22 and 23 of the sequence control device 17.
It will be assumed that a useful frequency oscillation is being received at the input 1. The Schmitt trigger 2 then emits at the output 3 rectangular pulses whose recurrence frequency is equal to the frequency of said useful frequency oscillation. The first pulse emitted by the Schmitt trigger 2 puts the monoflop 5 into its operating state. The hold time of the monoflop 5 is so dimensioned that for all useful frequencies occurring it is longer than the recurrence period of the rectangular pulses emitted at the output 3. The monoflop 5 therefore remains in its operating state for as long as the useful frequency oscillation is applied to the input 1 and supplies to the control input 16 of the sequence control device 17 a control signal throughout this time.
Due to the control signal applied to the input 16 the sequence control device 17 emits at its outputs 21, 22 and 23 in time with the pulses supplied to it via the Schmitt trigger 18 at the input 19 mutually offset control pulse sequences, the duration of the control pulses being equal to the time interval of the leading edges of the pulses supplied at the input 19 and thus equal to the period of the oscillation applied to the input 20 and the pulse sequences being offset with respect to each other by one pulse duration. The control pulses emitted by the sequence control device 17 perform the following functions:
a. The first control pulse appearing at the output 21 sets in operation for its duration via the input 8 the frequency divider 7 so that the latter divides the recurrence frequency of the pulses supplied thereto from the Schmitt trigger 2 and thus the frequency of the useful frequency oscillations received in a constant ratio and passes counting pulses to the input 10 of the counter 11 with a correspondingly reduced recurrence frequency.
b. Via the input 13 the second pulse occurring at the output 22 causes the store 12 to take on and to store the count of the counter 11 reached at the end of the first control pulse.
c. The third control pulse appearing at the output 23 resets the counter 11 via the reset input 24.
COntrol pulse sequences continue to be emitted for as long as the monoflop 5 remains in its operating state.
Since the stage outputs of the store 12 are permanently connected to the inputs of the decoder 14, the store content is continuously being decoded. The decoder 14 therefore emits a control signal at the output which is associated with the count contained in the store.
During each group of three offset control pulses of the three control pulse sequences emitted by the sequence control device 17, the counter 11 receives counting pulses from the frequency divider 8 only for the duration of the control pulse of the first control pulse sequence emitted at the output 21. The duration of this control pulse thus determines the measuring time during which the oscillations of the useful frequency signal received are counted. Since the duration of the control pulses emitted by the sequence control device 17 is however equal to the period of the oscillation applied to the input 20, the measuring time is fixed by the period of said oscillation.
The frequency divider 7 is connected in front of the counter 11 so that a small capacity of the counter 11 is sufficient to obtain a clear indication of the received frequency even when the measuring time is so long that a large number of periods of the useful frequency oscillation is received during the measuring time. This is for example, the case when the oscillation supplied to the input 20 has twice the mains frequency. Since the frequency divider 7 divides the frequency of the useful frequency oscillations received in the constant ratio k, the counter 11 need count only the oscillations having a correspondingly reduced frequency. If the division ratio k of the divider 7 is so set that it is equal to the product of the measuring time t and channel spacing Δ f, only a frequency which differs by at least the channel spacing Δ f from a previously received frequency will change the count of the counter 11.
The purpose of the monoflop 5 is to prevent interference frequencies supplied to the input 1 from producing at one of the outputs D0 to D9 of the decoder 14 a control signal which could lead to an erroneous function of the equipment being controlled. The interference sources usually encountered emit a frequency spectrum whose components lie predominantly in the audio region, i.e., below the ultrasonic region. If the hold time of the monoflop 5 is set to a value slightly greater than the period of the smallest useful frequency but smaller than the period of the highest interference frequency occurring, the monoflop 5 returns to its quiescent state before the end of the period of an interference frequency. Since in this state no signal is supplied to the control input 16 of the sequence control device 17, the latter is put out of operation and consequently the received signal cannot be evaluated because the count of the counter 11 is not transferred to the store 12 and thus no decoding takes place.
To facilitate understanding of the invention, the function of the circuit of FIG. 1 will now be explained numerically by way of example. The channel spacing Δ f will be taken as 1,200 c/s so that for a frequency of 100 c/s of the oscillation applied to the input 20 and thus a measuring time of 10 ms a division ratio of the frequency divider 7 of k = t . Δf = 12 results. It will further be assumed that ten different channel frequencies are to be evaluated; the counter 11 is therefore so connected that it has a capacity of 10. With these values, during the measuring time the counter 11 runs through several count cycles. This means that for the received frequency during the measuring time the counter 11 reaches its maximum count several times and then starts counting again from the beginning. The count reached at the end of the measuring time is however still a clear indication of the received useful frequency provided the number of useful frequencies having a channel spacing Δf is at the most equal to the counter capacity Z. The relationship between the useful frequency f received and the count reached at the end of each measuring time t while this useful frequency is being received is expressed by the following equation:
f = (k/t) . (n . Z + m + 0.5)
f = useful frequency received in c/s
t = measuring time in seconds
k = division ratio of the frequency divider 7
Z = capacity of the counter 11
n = number of count cycles passed through (integral)
m = count
The term 0.5 in brackets is a correction factor which ensures that a new count is reached whenever the received frequency differs at least by half the channel spacing Δf from the channel center frequency of the neighboring channel. With a channel spacing Δ of 1,200 c/s, a measuring time t of 10 ms, a division ratio k of the frequency divider 7 of 12, a capacity Z of the counter 11 of 10 and an input frequency f of 33 k c/s, the count 7 is for example reached after two complete count cycles. This is because the input frequency of 33 k c/s is first divided by 12 by the frequency divider 7 so that pulses having a recurrence frequency of 2.750 k c/s reach the input 10 of the counter 11. Since the frequency divider 7 emits counting pulses only during the measuring time of 10 ms, during said time only 27.5 pulses reach the input 10 of the counter 11. For this number of pulses the counter thus runs through two complete cycles and finally stops at the count 7. Similarly, for an input frequency of 39 k c/s the counter stops at the count 2 after passing through three complete counter cycles. With the numerical values given up to 10 different frequencies may be received without any ambiguity occurring in the evaluation.
FIG. 3 illustrates a further embodiment of an ultrasonic remote control receiver which differs from the embodiment described above primarily in that to fix the measuring time it is not necessary to supply a reference frequency. In the illustration of FIG. 3 the same reference numerals as in FIG. 1 are used for identical circuit components. The part of the circuit enclosed in the dashed line represents the sequence control device 17' which emits at its outputs 21', 22', 23' control signals which have substantially the same functions as the control signals emitted at the outputs 21, 22 and 23 of the sequence control device 17 of FIG. 1.
The useful frequency signal received is again supplied to the input 1. The input 1 is connected to the input of the Schmitt trigger 2 which again converts the input useful frequency oscillations into a sequence of pulses whose recurrence frequency is equal to the input useful frequency. The output 3 of the Schmitt trigger 2 is connected to the input B1 of a monoflop 25 which is contained in the sequence control device 17' and which is so constructed that it is switched to its operating state by a pulse received at the input B1 but during its hold time cannot be tripped again by any further pulse. The output 3 of the Schmitt trigger 2 is also connected to the input 26 of an AND gate 27 whose other input 28 is connected to that output 21' of the sequence control device 17' which is directly connected to the output Q1 of the monoflop 25. The output Q1 of the monoflop 25 which emits the signal complementary to the signal at the output Q1 is connected to the input B2 of a further monoflop 29 whose output Q2 is connected to the input A1 of the monoflop 25. The input 10 of the counter 11 is connected to the output of the AND gate 27. The stage outputs of the counter 11 are connected to the inputs of a gate circuit 30 which on receipt of a control pulse at its input 31 transfers the count contained in the counter 11 to the decoder 14 connected to its outputs. In the decoder 14 the count is then decoded in the manner already explained in conjunction with FIG. 1 so that a control signal is emitted at the output corresponding to the transferred count.
The output 3 of the Schmitt trigger 2 is further connected to the input 32 of an AND gate 33 which is contained in the sequence control circuit 17' and the other input 34 of which is connected to the output of a NOR gate 35. The output Q1 of the monoflop 25 is directly connected to one input 36 of the NOR gate 35 and is connected to the other input 37 via a delay member 38 and an inverter 39.
The output of the AND gate 33 represents the output 22' of the sequence control circuit 17' which is directly connected to the control input 31 of the gate circuit 30. In addition, the output of the AND gate 33 is directly connected to one input 40 of a NOR gate 41 and to the other input 42 thereof via a delay member 43 and an inverter 44. The output of the NOR gate 41 represents the output 23' of the sequence control circuit 17', to which output the reset input 24 of the counter 11 is connected.
The mode of operation of the circuit of FIG. 3 is explained in FIG. 4. Since the measuring time in the arrangement of FIG. 3 is substantially shorter than in the arrangement of FIG. 1, the time scale in FIG. 4 has been enlarged compared with FIG. 2 in order to clarify the illustration. When useful frequency oscillations are supplied to the input 1 of the receiver, pulses whose recurrence frequency is equal to the useful frequency appear at the output 3 of the Schmitt trigger 2. It will be assumed that the presence of a pulse corresponds to the logical signal value 1 whereas a pulse space represents the logical signal value 0. The leading edge of the first pulse at the output 3 puts the monoflop 25 into its operating state in which it emits the signal value 1 for the duration of its hold time at its output Q1, resulting in the control pulse at the output 21', which passes to the input 28 of the AND gate 27. Since the other input 26 of the AND gate 27 is directly connected to the output 3 of the Schmitt trigger 2, for the duration of each pulse at the output 3 the signal value 1 is also applied to the input 26 of the AND gate 27. Thus, the pulses occurring at the output 3 of the Schmitt trigger 2 are transferred for the duration of the control pulse at the output 21', i.e. during the hold time of the monoflop 25, as count pulses to the counter 11 and counted by the latter. The hold time of the monoflop 25 thus determines the measuring time; the capacity of the counter 11 must be greater than the number of pulses received during the measuring time for the greatest useful frequency. The count of the counter 11 reached at the end of the measuring time is then a clear indication of the received useful frequency.
When the monoflop 25 flops back into the quiescent state at the end of its hold time, it applies the signal value 0 via its output Q1 to the input 28 of the AND gate 27 so that no further count pulses can enter the counter 11. At the same time there appears at the output Q1 of the monoflop 25 the signal value 1 which at the input B2 puts the monoflop 29 into the operating state. In this state the monoflop 29 emits at its output Q2 the signal value 1 which blocks the monoflop 25 via the input A1 for the duration of the hold time of the monoflop 29 in such a manner that it cannot be switched into the operating state by pulses at the input B1. This is necessary to enable the sequence control device 17' to have sufficient time to generate the control pulses appearing at the outputs 22' and 23' for the transfer of the count or resetting of the counter.
With the return of the monoflop 25 to its quiescent state, the signal value 0 passes to the input 26 of the NOR gate 35 directly connected to the output Q1. During the operating state of the monoflop 25 the signal value 0 is applied with a delay determined by the delay member 38 via the inverter 39 to the input 37 of the NOR gate 35, said signal value 0 being replaced by the signal value 1 only after the delay time of the delay member 38 and not simultaneously with the flop back of the monoflop 25. Thus, for the duration of this delay time the signal value 0 is applied to both inputs 36 and 37 of the NOR gate 35 and consequently for this period of time the signal value 1 appears at the output of the NOR gate 35. The circuits 35, 38, 39 thus effect the generation of a short pulse which immediately follows the return of the monoflop 25 and the duration of which is determined by the delay of the delay member 38. This pulse is applied to the input 34 of the AND gate 33 (FIG. 4). The same effect could obviously alternatively be obtained with a monoflop which is tripped by the signal at the output Q1 changing from the value 1 to the value 0.
Now, if during this time a pulse is emitted at the output 3 of the Schmitt trigger 2, i.e., a signal value 1 is at the input 32 of the AND gate 33, said gate supplies to the control input 31 of the gate circuit 30 a control pulse for the duration of the delay of the delay member 38. This control pulse opens the gate circuit so that it allows the count reached at the end of the hold time of the monoflop 25 to pass to the decoder 14. The latter then emits a control signal at the output associated with this count. The signal value 1 present at the output of the AND gate 33 during the delay of the delay member 38 also passes directly to the input 40 of the NOR gate 41, at the other input 42 of which the signal value 0 is applied for the duration of the same pulse but with a delay determined by the delay member 43. Thus, in a manner similar to the circuits 35, 38, 39 the circuits 41, 43, 44 produce a short pulse which immediately follows the end of the output pulse of the AND gate 33 and appears at the output 23' of the sequence control circuit and is applied to the reset input 24 of the counter 11 (FIG. 4). This pulse resets the counter 11.
The hold time of the monoflop 29 is so set that it flops back into its quiescent state again only when the transfer process from the counter to the decoder via the gate circuit and the resetting of the counter has been effected. When the monoflop 29 returns to its quiescent state, it emits at its output Q2 the signal value 0 which brings the monoflop 25 via the input A1 thereof into such a condition that it can again be brought into its operating state by a pulse at the output 3 of the Schmitt trigger 2. In this manner the measuring and evaluating periods can be repeated for as long as useful frequency oscillations are supplied to the input 1.
In the circuit according to FIG. 3, interference frequencies are suppressed by setting a certain hold time of the monoflop 25. It is apparent from the above description of the function that the transfer of the count of the counter 11 to the decoder 14 takes place immediately following the end of the hold time of the monoflop 25, i.e., immediately following the end of the measuring time. However, a control signal initiating the transfer can be applied by the AND gate 33 to the control input 31 of the gate circuit 30 only when simultaneously with the end of the measuring time a pulse, i.e., the signal value 1, is present at the output 3 of the Schmitt trigger 2. Now, if the hold time of the monoflop 25 is made equal to the reciprocal of the channel spacing Δf, this coincidence at the AND gate 33 at the end of the measuring time occurs only when quite definite frequencies are applied to the input 1; these frequencies lie only within frequency bands which in the example described here, in which the output pulses of the Schmitt trigger 2 have a pulse duty factor of 1:2, have the width of half a channel spacing. These frequency bands each contain one of the useful frequencies. Between these frequency bands there are gaps having the width of half the channel frequency and frequencies falling in these gaps do not produce coincidence at the AND gate 33 and consequently cannot be evaluated by transfer of the count of the counter 11 to the decoder 14. Thus, frequency windows are formed over the entire frequency range which can occur at the input 1 and only frequencies lying within these windows are treated by the circuit according to FIG. 3 as useful frequencies. All intermediate frequencies are recognized as interference frequencies and excluded from evaluation.
If the measuring time is made exactly equal to the reciprocal of the channel spacing the frequency bands in which evaluation takes place are such that the rated frequencies of the signals transmitted by the transmitter are disposed at the lower end of the frequency bands. Thus, in this case only frequencies starting from a rated frequency in each case and extending up to the frequency in the center between two channels would be evaluated as useful frequencies. Since the frequency of the signals emitted by the transmitter can however also fluctuate below the rated frequency, it is desirable to place the frequency bands in which evaluation takes place so that the rated frequencies lie substantially in the center of the bands. To achieve this, the hold time of the monoflop 25 and thus the measuring time is lengthened by a quarter of the reciprocal of the maximum rated frequency. Although with this setting only the maximum rated frequency lies exactly in the center of the corresponding frequency band, the other rated frequencies still lie within the corresponding frequency bands and consequently the frequencies of the useful signals can also deviate from the rated frequency downwardly without preventing evaluation. The frequency gaps including the frequencies treated as interference frequencies then lie in each case substantially in the center between two rated frequencies.
To facilitate understanding of the type of interference identification just outlined attention is drawn to FIG. 5; the latter shows at Q1 the output signal of the monoflop 25 determining the measuring time, at 3-F1, 3-F2, 3-F3 the pulse sequences appearing at the output 3 of the Schmitt trigger 2 for three different useful frequencies F1, F2, F3 and at 3-FS the pulse sequence which appears at the output 3 when an interference frequency FS is received which lies between the useful frequencies F2 and F3. It is apparent from this diagram that at the end of the measuring time a pulse is present at the output 3 of the Schmitt trigger only when useful frequencies are being received and that when an interference frequency is applied there is a pulse space at the end of the measuring time. Thus, at the AND gate 33 the presence of a pulse at the end of the measuring time is employed as criterion for the receipt of a useful frequency. It is also apparent from FIG. 5 that with the useful frequency F1 the counter 11 counts 4 pulses, with the useful frequency F2 up to 5 pulses and with the useful frequency F3 6 pulses.
Isolated short interference pulses which could reach the input 1 of the circuit of FIG. 3 between two useful pulses and undesirably increase the count may be made ineffective by inserting a flip-flop circuit 45 between the output 3 of the Schmitt trigger 2 and the rest of the circuit as illustrated in FIG. 6. The mode of operation of this flip-flop circuit 45 will be explained with the aid of FIG. 7, which shows the signals at the output 3 of the Schmitt trigger 2 and at the output 3a of the flip-flop circuit 45 firstly without interference and secondly with interference. The flip-flop circuit 45 is tripped by the leading edge of each output pulse of the Schmitt trigger 2. If a short interference pulse is received, the flip-flop circuit 45 supplies at its output 3a the signal value 0 for example on receipt of the useful pulse preceding the interference pulse, the signal value 1 on receipt of the interference pulse and the signal value 0 on receipt of the next useful pulse. If no interference pulse had occurred, the flip-flop circuit would not have been switched to the signal value 1 at the output until receipt of the next useful pulse. The flip-flop circuit thus effects on receipt of an interference pulse (and in general on receipt of an odd number of interference pulses) between two useful pulses a reversal of the signal values so that at the end of the measuring time coincidence is not reached at the gate 33 although a useful frequency was received. Without the flip-flop circuit 45 the count would be transferred, although because of the interference pulse received it would not correspond to the useful frequency received.
The embodiment of FIG. 3 differs from the embodiment of FIG. 1 also in that instead of the store (register) 12 the gate circuit 30 is used that allow the count to be evaluated to pass briefly only once in a measuring and evaluating time. Thus, at the output of the decoder 14, instead of a uniform signal as in the case of the embodiment of FIG. 1, a series of pulses appears with the spacing of the control signals at the input 31 of the gate circuit 30. The use of a gate circuit instead of a store is suitable in applications where the equipment to be controlled must be actuated with control pulses and not with a uniform signal.
The immunity to interference may be further increased if in accordance with FIG. 8 a further monoflop 46 which cannot be triggered again during its hold time is inserted between the output 3 of the Schmitt trigger 2 (or the output 3a of the flip-flop circuit 45 of FIG. 6) and the remainder of the circuit. This hold time is set to half the period of the highest useful frequency. This modification is effective against a particular type of interferences, i.e., cases where an amplitude break occurs within an oscillation at the input 1 of the Schmitt trigger 2; this break would lead at the output 3 of the Schmitt trigger to the emission of two pulses instead of the single pulse per oscillation emitted in the normal case. These two pulses give the same effect as the receipt of a frequency which is twice as high and consequently without the additional monoflop 46 erroneous evaluations could arise. However, the monoflop 46 prevents the two pulses from becoming separately effective because it always emits pulses having the duration of its hold time; short double pulses which can arise due to amplitude breaks in the received signal thus cannot have any effect. FIG. 9 shows the action of the monoflop 46 when an amplitude break occurs at the input 1 of the Schmitt trigger 2 which produces a double pulse at the output 3 of the Schmitt trigger. As is apparent, the pulses at the output 3b of the monoflop 46 are not affected by this double pulse.
One embodiment of the remote control receiver may also reside in that a sequence control counter fed by the pulses at the output of the Schmitt trigger 18 is used for the sequence control device 17 of FIG. 1; the stage outputs of said counter are connected to a decoder which is so designed that it activates one after the other one of its outputs for each count. Thus, for example, this decoder may have 10 outputs which are activated successively in each counting period of the sequence control counter. Since in accordance with the description of the example of embodiment of FIG. 1 a total of three control signals are required for the evaluation of the frequency received, the output signals at the fourth, fifth and seventh outputs may be used respectively for activating the frequency divider 7, opening the store 12 and resetting the counter 11. Since in this case the evaluation of the received frequency by the control pulses emitted from the output of the decoder of the sequence control device does not begin until the decoder emits a signal at its fourth output, there is an evaluation delay which has the advantage that short interference pulses produce no response in the receiver.
The advantageous formation of frequency band windows are used in the embodiment of FIG. 3 can also be applied in the embodiment of FIG. 1 if instead of the retriggerable monoflop 5 a monoflop is used which has no dead time and which is not retriggerable again during its hold time which as in the monoflop 35 of FIG. 3 is made equal to the reciprocal of the channel spacing Δ f. This monoflop thus always flops back into its quiescent state when there is a pulse pause at its input at the end of its hold time whereas it is returned to its operating state practically without dead time by a pulse applied to its input at the end of the hold time. Since a pulse at the input of the monoflop at the end of its hold time however occurs only for frequencies lying within the frequency bands mentioned in connection with the description of FIG. 3, only frequencies which lie within the frequency bands can be treated as useful frequencies. For all intermediate frequencies, the monoflop returns to its quiescent state in which it interrupts the sequence control device and thus prevents evaluation of said frequencies. For the same reasons as in the circuit of FIG. 3, in this case as well the hold time of the monoflop should be lengthened by a quarter of the reciprocal of the highest useful frequency.
The ultrasonic remote control receiver described above can be used not only to control television sets, radio sets and the like but is particularly suitable also for industrial use in which high immunity to interference is very important. It may, for example, be used for remote control of cranes on large building sites, where there are a great number of different interference sources. The ultrasonic remote control receiver according to the above description is so immune to interference that it operates satisfactorily even under the difficult conditions encountered in the aforementioned use.
The following table provides examples of integrated circuits from Texas Instruments Incorporated which may be used in the foregoing invention.
______________________________________ Schmitt-triggers 2 and 18 SNX 49713 Monoflops 25, 29 and 46 SN 74121 Monoflop 5 SN 74122 Frequency divider 7 SN 7492 Counter 11 SN 7490 Store 12 SN 7475 Control 17 SN 7476 Gate 30 SN 7432 Decoder 14 SN 7442 ______________________________________
Receiver tuning circuit PHILIPS CHASSIS K11
A receiver tuning circuit in which without operation of extra switches a change-over can be made from tuning by means of a continuously varying tuning voltage to tuning by means of one of a number of adjusted tuning voltages by using a capacitor controlled by an automatic tuning correction current source circuit for obtaining said voltage, and an automatic switch for applying the desired tuning voltages to this capacitor.
1. A receiver tuning circuit comprising a tuning section having a tuning input, a capacitor means coupled to said tuning input for applying a tuning voltage thereto, a controllable current source coupled to said capacitor, a tuning correction signal detector means coupled between said tuning section and said current source for applying an automatic tuning correction signal to said capacitor means through said current source, and means for immediately tuning said tuning section to a selected frequency independently of the previous voltage on said capacitor comprising a first switch coupled to said capacitor and an operating device means for controlling said switch for an operating period, said operating device including a memory means for storing the last adjusted state of said operating device, at least one potentiometer and a generator means for effecting that a signal from said potentiometer is applied to said capacitor through said switch upon operation of said operating device, and said first switch including a first time constant circuit means coupled to said generator for maintaining said switch in an on position for a selected period of time independent of said operating period.
2. A receiver tuning circuit as claimed in claim 1, wherein said switch comprises a current source which can be influenced by an operating signal, said source being coupled to two parallel branches the first of which includes a transistor having an emitter coupled to said current source, a base coupled to an input of the switch, and a collector, a current mirror circuit having an input coupled to said collector and an output, the second branch including a pair of series connected diodes coupled to the current source and to said output of the current mirror circuit, and output of the switch being coupled to the pair of diodes.
3. A circuit as claimed in claim 1, further comprising a manually operable second switch means for obtaining a continuous coupling between said potentiometer means and said capacitor.
4. A circuit as claimed in claim 1 further comprising a supply circuit means for obtaining a desired tuning voltage, said memory means being independent of said supply circuit, said first switch further comprising a second time constant circuit means coupled to said supply circuit means for temporarily applying a tuning voltage determined by the potentiometer to said capacitor when the supply voltage is switched on.
5. A receiver tuning circuit comprising a tuning section having a tuning input, a capacitor means coupled to said tuning input for applying a tuning voltage thereto, a controllable current source coupled to said capacitor, a tuning correction signal detector means coupled between said tuning section and said current source for applying an automatic tuning correction signal to said capacitor through said current source, means coupled to said capacitor for immediately tuning said tuning section to a selected frequency independently of the previous voltage on said capacitor, a memory means coupled to said immediate tuning means for storing a tuning voltage corresponding to a selected frequency and signal amplitude detector means coupled to said immediate tuning means for effecting that said tuning voltage stored in said memory means is applied to said capacitor through said immediate tuning means when said signal amplitude goes below a selected value.
The invention relates to a receiver tuning circuit having a tuning section tunable by means of a tuning voltage obtained from a capacitor whose charge can be changed by means of a current source circuit controllable by at least an automatic tuning correction signal, while furthermore a desired tuning voltage can temporarily be applied to said capacitor with the aid of a switch controllable by an operating device so as to make it possible to immediately tune to a desired frequency independently of the previous charge condition of said capacitor.
A tuning circuit of the kind described above is known from German Offenlegungsschrift No. 2,025,369 in which the said capacitor is optionally connected to a tuning potentiometer by means of a push-button switch for applying a voltage determined by said potentiometer to said capacitor as long as the push-button switch is operated, whereafter a tuning frequency thus selected is corrected with the automatic tuning correction signal through the current source circuit and the charge of said capacitor.
It is an object of the invention to enhance the comfort of operation of such a tuning circuit.
To this end a tuning circuit of the kind described in the preamble is characterized in that the operating device includes a memory for storing the last adjusted state of said operating device, and a signal generator which upon operation of the operating device applies a signal to an output thereof, which output is coupled to a time-constant circuit coupled to said switch for maintaining said switch switched on for a period determined by the time-constant circuit independently of the operating duration of the operating device.
Due to the step according to the invention it is possible at any moment to ascertain, by means of the state of the memory, the last operating action of the operating device, maintaining the advantage of a temporary tuning voltage supply to the capacitor so that subsequently other functions such as, for example, a tuning correction device or a search tuning device can become active on said capacitor through the current source circuit.
The invention will now be described with reference to the drawing.
FIG. 1 shows by way of a block-schematic diagram a receiver tuning circuit according to the invention;
FIG. 2 shows by way of a principle circuit diagram a possible embodiment of part of the receiver tuning circuit according to the invention.
In FIG. 1 a tuning section 1 has an input 3 to which a received RF signal is applied and an output 5 from which an IF signal is obtained. This IF signal is applied to an input 7 of an IF amplifier 9 and derived in an amplified form from an output 11 thereof and applied to an input 13 of a tuning correction signal detector 15 and an input 17 of a signal amplitude detector 19.
Furthermore, the tuning section 1 has an input 21 which receives a tuning voltage from a capacitor 23. The charge of the capacitor 23 can be changed with the aid of a current source circuit 25 for which purpose an output 27 thereof is connected to the capacitor 23 whose other end is connected to ground.
An input 29 of the current source circuit 25 is controlled by a tuning correction signal originating from an output 31 of the tuning correction signal detector 15. This correction signal can be rendered inactive with the aid of a switch-off device 33 incorporated in the connection between the output 31 and the input 29, and with the aid of signals applied to an input 35 or 37 thereof.
For this purpose the input 35 of the switch-off device 33 is connected to an output 39 of a station finder 41 two outputs 43, 45 of which are connected to inputs 47, 49 of the current source circuit 25. Thus, the station finder 41 can continuously bring about a charge or discharge of the capacitor 23 when the automatic tuning correction is switched off so that the tuning section 1 is continuously detuned. When a station is found, a signal is produced at the output 31 of the tuning correction signal detector 15, which signal causes stop signal at an input 55 of the station finder through a polarity correction circuit 51 and a delay circuit 53, and this for a certain period, for example, 1.5 seconds so that station finding is temporarily discontinued and the automatic tuning correction is activated. As a result, tuning is effected immediately and correctly at the frequency of the received station. If this station is not desired, further station finding can be continued after 1.5 seconds.
The capacitor 23 providing the tuning voltage for the tuning section 1 may be controlled not only by the current source circuit 25, but also by an output 57 of a switch 59 an input 61 of which is connected to an output 63 of an operating device 65.
A voltage originating from one of a plurality of tuning potentiometers 67, 69, 71 can be temporarily applied to the capacitor 23 with the aid of the operating device 65. When the device 65 is operated a signal is obtained to that end from a signal generator 73. This signal is applied through an output 75 of the operating device to an input 77 of a time-constant circuit 79. The time constant circuit 79 is coupled to the switch 59 and closes it for a certain time so that the capacitor 23 assumes the desired voltage of a selected potentiometer 67, 69 or 71.
The operating device 65 has a memory which is symbolically shown in the figure as a block 81. This memory 81 ensures that it can always be seen which potentiometers 67, 69 or 71 is interconnected to the output 63 of the operating device 65, while due to the action of the time-constant circuit 79 the voltage originating from this potentiometer is not continuously present at the capacitor 23. The said memory 81 may be either a mechanical or an electrical memory. When using a mechanical memory, the signal generator 73 may be an AFC switch which is present on many operating devices. When using an electrical memory, as is common practice with touch controls in the operating device 65, any change of state of this memory may be converted in a simple manner into a signal applied to the output 75.
The switch 59 has an output 83 which applies a signal to the input 37 of the switch-off device 33. This signal renders the automatic tuning correction inactive as long as the switch 59 is closed, as is the case when a tuning voltage is applied to the capacitor 23 with the aid of the operating device 65. The tuning correction is active again immediately when the switch 59 is open so that tuning is effected immediately and correctly when a selected station is received.
To be able to adjust the potentiometers 67, 69 or 71, easily, a switch 85, which can be operated manually, is connected to a further input 87 of the switch 59 which can be maintained closed with the aid of the manually operated switch 85 as long as is desired for adjustment.
Coupled to the switch 59 is a further time-constant circuit 89 which has an input 91 connected to an output 93 of a supply circuit 95. Thus, whenever the receiver is switched on, the switch 59 is maintained closed for some time so that firstly the station to which the operating device 65 is adjusted is tuned to, even if the station finder 41 were switched on. In that case the operating device 65 must have, for example, a mechanical memory 81, which is independent of the supply voltage, in order to maintain its adjustment also when the supply voltage is switched off.
Furthermore, the switch 59 has an input 97 which is connected to an output 99 of the signal amplitude detector 19. When the signal received by the receiver becomes too weak, the switch 59 can be closed via this path so that tuning to a frequency selected by the operating device 65 is maintained and is stil present when the received signal becomes stronger again. A further possibility, which may be particularly attractive for motorcar radios, is to incorporate a switch which can be operated in this manner between the capacitor and an output of a memory which can be coupled to that capacitor. When the field strength is sufficient, this memory may be written in with the voltage on the capacitor and when the field strength is insufficient, an output of this memory may be coupled to the capacitor for transferring the memory voltage to the capacitor. This memory may be, for example, a motor adjusting a potentiometer and operated with the aid of a control system. When the supply voltage drops out, the last adjusted state of the potentiometer is maintained.
The described tuning circuit may immediately change over from, for example, a search tuning state to a state tuned to a desired station without operating extra switches and only by operating the relevant operating members.
It will be evident that the switch tuning may be omitted, if desired.
FIG. 2 shows a possible embodiment of the switch 59 and, coupled thereto, the time-constant circuits 79 and 89 of the receiver tuning circuit of FIG. 1. The inputs and outputs have the same reference numerals as the corresponding inputs and outputs in FIG. 1.
The input 61 of the switch 59 is connected to the base of a npn transistor 201. The emitter of this transistor 201 is connected through a diode 203 to the collector of an npn transistor 205 arranged as a current source whose emitter is connected to the output 83 and is furthermore connected to ground through a resistor 207.
The collector of the transistor 201 is connected through a diode 209 to the input 91 to which the supply voltage is applied. The diode 209 shunts the base-emitter path of a pnp transistor 211 which together with the diode 209 constitutes a current mirror circuit. The collector of the transistor 211 allows a current to flow through a series arrangement of two diodes 215, 217, which current has substantially the same intensity as the current flowing through the diode 203. Furthermore, the diode 217 is connected to the collector of the transistor 205, while the junction of the collector of the transistor 205 and the diode 215 is connected to the output 57.
The base of the transistor 205 is connected to a tap on a potential divider 219, 221 between the supply voltage and ground. This potential divider will raise the voltage at the base of the transistor 205 to such an extent that it produces a current, which is further determined by the emitter resistor 207, equally distributed over the collector branches with the diode 203 and the transistor 201 and with the diodes 217 and 215, respectively. When the circuit is designed in a integrated form, it can be achieved in a simple manner that the output 57 will always assume the same voltage as the input 61. Since the output 57 is connected to the capacitor 23, both a discharge and a charge of this capacitor 23 is possible. Charging is effected through the transistor 211 and discharging is effected through the diodes 215, 217. The circuit is independent of temperature influences. The diode 203 and consequently the diode 217 are provided to prevent a too large voltage difference at the base-emitter junction of the transistor 201.
The current source 205 can be turned off by connecting the base of transistor 205 to ground with the aid of a npn transistor 223 connected across the resistor 221. This is effected when the base of this transistor receives a voltage from a potential divider comprising three resistors 225, 227, 229. However, when the base of the transistor 223 receives a low voltage through the input 87 or the input 97, the transistors 223 is cut off and the transistor 205 conducts so that the switch 59 is closed.
The voltage at the base of the transistor 223 remains low for some time after switching on the supply voltage because a capacitor 231, which is connected to the junction between the resistors 225 and 227, must firstly be charged. Thus, the switch 59 is closed during that period.
Furthermore, the voltage at the base of the transistor 223 may be decreased by discharging the capacitor 231 through a resistor 233 to the input 77 when this input is earthed for a moment during operating device 65. The voltage at the capacitor 231 will subsequently increase in accordance with a certain time constant and after a certain time the transistor 223 conducts again and the switch 59, which was closed when the transistor 223 was cut off, will be open again.
The input 97 is interconnected to the input 87 so that the transistor 223 is also cut off and the switch 59 starts to conduct when the voltage at the input 97 becomes low upon a drop-out of a transmitter signal.
The switch 59 in this embodiment also acts as an amplifier so that the adjustments of the tuning potentiometer 67, 69 or 71 do not have any influence on the rate at which the charge of the capacitor 23 is changed.
CONTACTLESS TOUCH SENSOR PROGRAM CHANGE KEYBOARD CIRCUIT ARRANGEMENT FOR ESTABLISHING A CONSTANT POTENTIAL OF THE CHASSIS OF AN ELECTRICAL DEVICE WITH RELATION TO GROUND :
Circuit arrangement for establishing a reference potential of a chassis of an electrical device such as a radio and/or TV receiver, such device being provided with at least one contactless touching switch operating under the AC voltage principle. The device is switched by touching a unipole touching field in a contactless manner so as to establish connection to a grounded network pole. The circuit arrangement includes in combination an electronic blocking switch and a unidirectional rectifier which separates such switch from the network during the blocking phase.
1. A circuit arrangement for establishing, at the chassis of an electrical device powered by a grounded AC supply network, a reference potential with relation to ground, said device having at least one contactless touching switch operating on the AC voltage principle, the switch being operated by touching a unipole touching field in a contactless manner, said arrangement comprising an electronic switch for selectively blocking the circuit of the device from the supply network, a half-wave rectifier including a pair of diodes individually connected in series-aiding relation between the terminals of the supply network and the terminals of the device for separating the electronic blocking switch from the supply network during a blocking phase defined by a prescribed half period of the AC cycle, and a pair of condensers individually connected in parallel with the respective diodes. 2. A circuit arrangement according to claim 1, wherein the capacitances of the two condensers are of equal magnitude.
In electronic devices, for example TV and radio receivers, there are used in ever increasing numbers electronic touching switches for switching and adjusting the functions of the device. In one known embodiment of this type of touching switch, which operates on a DC voltage principle, the function of the electronic device, is contactlessly switched by touching a unipole touching field, the switching being carried out by means of an alternating current voltage. When using such a unipole touching electrode, one takes advantage of the fact that the AC current circuit is generally unipolarly grounded. In order to close the circuit by touching the touching surface via the body of the operator to ground, it is necessary to provide an AC voltage on the touching field. In one special known embodiment there is employed a known bridge current rectifier for the current supply. This type of arrangement has the drawback that the chassis of the device changes its polarity relative to the grounded network pole with the network frequency. With such construction considerable difficulties appear when connecting measuring instruments to the device, such difficulties possibly eventually leading to the destruction of individual parts of the electronic device.
In order to avoid these drawbacks, the present invention provides a normal combination of a unidirectional rectifier with an electronic blocking switch that separates the chassis of the electronic device from the network during the blocking phase. In accordance with the present invention, the polarity of the chassis of the electronic device does not periodically change, because the electronic device is practically separated from the network during the blocking phase of the unidirectional rectifier by means of the electronic blocking switch.
In a further embodiment of the invention a further rectifier is connected in series with the unidirectional rectifier in the connection between the circuit and the negative pole of the chassis. Such further rectifier is preferably a diode which is switched in the transfer direction of the unidirectional rectifier. According to another feature of the invention there are provided condensers, a respective condenser being connected parallel with each of the rectifiers. Preferably the two condensers have equal capacitances. Because of the use of such condensers, which are required because of high frequency reasons, during the blocking phase there is conducted to the chassis of the electronic device an AC voltage proportional to the order of capacitances of the condensers. Thus there is placed upon the touching field in a desired manner an AC voltage, and there is thereby assured a secure functioning of the adjustment of the device when such touching occurs.
In the embodiment of the invention employing two rectifiers there is the further advantage that over a bridging over of the minus conduit of the rectifier that is connected between the network and the negative pole of the chassis connection, no injuries can be caused by a measuring instrument in the electronic device itself and in the circuit arrangement connected thereto.
In the accompanying drawing:
The sole FIGURE of the drawing is a circuit diagram of a preferred embodiment of the invention.
In the illustrated embodiment the current supply part of the device, shown at the left, is connected via connecting terminals A and B to an AC voltage source, the terminal B being grounded at 8. The current supply part consists of a unidirectional rectifier in the form of a diode 1 with its anode connected to the terminal I, the cathode of diode 1 being connected to one input terminal 9 of an electronic device 2. In the device 2 there is also arranged a sensor circuit 3, shown here mainly as a block, circuit 3 being shown as including a pnp input transistor the emitter of which is connected to an output terminal 11 of the device 2. The collector of such transistor is connected to the other output terminal 12 of the device 2. The base of the transistor is connected by a wire 13 to a unipolar touching field 4 which may be in the form of a simple metal plate instead of the pnp transistor shown, the sensor circuit itself may consist of a standard integrating circuit which controls, among other things, the periodic sequential switching during the touching time of the touching field 4. All of the circuits of the electronic device 2 are isolated in a known manner from the chassis potential. Between the network terminal B and the negative pole 10 of the chassis there is arranged in the direction opposite that of diode 1 a further diode 5, the anode of diode 5 being connected to the terminal 10, and the cathode of diode 5 being connected to the terminal B of the current supply. To provide for HF type bridging of the diodes 1 and 5 there are arranged condensers 6 and 7 respectively, which are connected in parallel with such diodes.
The invention functions by reason of the fact that in an AC network separate devices radiate electromagnetic waves which produce freely traveling fields in the body of the person who is operating and/or adjusting the device, thereby producing an alternating current through his body to ground, as indicated by the - line at the right of the circuit diagram. If now the person operating the device touches the switching field 4, then the pnp type input transistor of the sensor circuit 3, which is placed on a definite reference potential (for example 12 Volts) and is connected with the negative halfwave of the AC voltage potential, is made conductive. There is thereby released a control command in the sequential switching, for example, for switching the electronic device to the next receiving channel. It is understood that the most suitable connection is formed between ground and the touching field 4 by means of a wire. By the use of such wires it would be assured that in all cases the base of the transistor in circuit 3 is connected to ground. This would, however, not permit anyone to operate the switch without the use of an auxiliary means such as a wire. It will be assumed that the touching almost always results directly via the almost isolated human body. For this reason the AC current fields are necessary, because otherwise there cannot always be provided a ground contact. Thus this connection is established via the body resistance of the person carrying out the touching of the switch.
The positive half wave of the alternating current travels to the terminal 9 of the electronic device 2 after such current has been rectified and smoothed by the devices 1, 6. Such positive halfwave is also conducted to the sensor circuit 3. The thus formed current circuit is closed by way of the chassis of the electronic device 3, the diode 5, and the terminal B. When there is a negative halfwave of the alternating current delivered by the current supply, both diodes 1 and 5 remain closed so that the chassis of the device 2 remains separated from the network during the blocking phase. Nevertheless, by means of condensers 6 and 7 the chassis is placed in a definite network potential, which depends on the relationship of the order of magnitude of the two condensers 6 and 7. When the capacitances of such condensers are equal, there is placed upon the chassis of the device 2 the constant reference potential, and simultaneously there is present via the sensor circuit 3 the required AC voltage at the touching field 4 for adjusting the function or functions of the device 2 upon the touching of the touching field 4.
The reference character 15 indicates a terminal or point at which the potential of the chassis of the device 2 may be measured. As above explained, the diode 5 causes the potential of the chassis at 15 to be separated from the network ground when a negative AC halfwave arrives. It will be noted that the return conduit of the circuit is held at a fixed chassis potential. The input transistor of the sensor circuit 3 remains, however, locked because it is subjected to a DC current of about 12 volts. If now, by means of touching the touching field 4, the chassis potential is connected to ground, then the transistor switches through and releases a switching function.
If the connecting terminals AB of the current source are exchanged, as by changing the plug, then there is still secured the condition that the chassis of the device is separated from the network ground via the diode, in this case the diode 1. The reference potential of the chassis consequently remains constant and the changing AC fields which are superimposed on the condensers can produce in the touching human body an AC current voltage due to the fields which are radiated by the device.
A suitable sensor which may be employed for the circuit 3 herein may be a sensor known as the "SAS 560 Tastatur IS," manufactured and sold by Siemens AG.
It is to be understood that the present invention is not limited to the illustrated environment. They can also be used in electronic blocking switch including a Thyristor circuit, which in the same manner separates the electronic device during the blocking phase from the network rectifier. With such Thyristor circuit the drawbacks described in the introductory portion of the specification of known circuit arrangements are also avoided.
Although the invention is illustrated and described with reference to a plurality of preferred embodiments thereof, it is to be expressly understood that it is in no way limited to the disclosure of such a plurality of preferred embodiments, but is capable of numerous modifications within the scope of the appended claims.
List of sets known to have the K11 chassis (made from approximately 1975-1978)
= means that models are most likely the same or very similar, but the styling can be different in some cases. Information was amongst others taken from the Philips model number survey 2003, 3122 785 14570.
A side note for those who have noticed the K10 chassis is missing from the line up. Rumour has, that this was a K9 variant with another tube, probably Trinitron, that didn’t make it beyond the prototype stage. Instead, Philips decided to use the 20AX tube and named the chassis K11. This chassis was designated K9i in some countries, most notable Germany. The differences between the K9 and K11 chassis were probably thought of as minor as the K11 chassis was basically an improved version of the K9 chassis with some minor (evolutionary) updates, another tube and as a result less complicated convergence circuits.
Factory location Krefeld (KR)
It seems very strange that only one German model is mentioned. Quite possibly the person who compiled the official Philips model number survey got confused by the K9i nomenclature. As a result of that, the D26C865 mentioned in the K9 overview might actually be a K11 set. Other German K11 sets probably exist.
Factory location Norrköping (NF)
Factory location Martinsville
Other brands (Erres, possibly Schneider (F), ..)
Erres branded sets mostly used the prefix RS
The suffix KSK instead of K might indicate a Swedish model. I haven’t actually seen it on a set in person.
22545K = 22C545
As a rule, the model number below is prefixed by letters indicating the brand name as
follows (not all brands may be used, others may exist):
AR = Aristona
SA = Siera
RA = Radiola
DX = Dux
CT = Conserton?
The infix KSK instead of K might indicate a Swedish model. I haven’t actually seen it on a set in person.
56K545 = 22C545
56K549 = 22C549