A very singular chassis technology:
The LOEWE F875 SENSOTRONIC CHASSIS 543/490 is realized with the F800 LOEWE Chassis serie and in the Line / Horizontal deflection output like a COLOR TELEVISION CHASSIS with EHT output with Tripler and Horizontal output with thyristor tech.
Furthermore this chassis was introducing the improved LOEWE IF UNIT based around TELEFUNKEN TDA440.
LOEWE-OPTA Start-delay overload protection arrangement for a thyristor sweep circuit:An input switching thyristor of a deflection circuit is selectively excited at a sweep rate by a transistorized driver stage. A separate protective thyristor is disposed in a DC power source that supplies operating bias to the driver stage and to the switching thyristor via a supply capacitor. Initially, the supply capacitor is partially charged through a high-ohmic shunt resistor that budges the protective thyristor, whose control electrode is coupled to the output of the switching thyristor. A portion of the resulting voltage across the partially charged supply capacitor is applied via a voltage divider and a Zener diode to the base of a threshold-operated transistor that normally blocks the driver stage. When the supply capacitor has charged to a predetermined minimum value necessary to assume turn-on of the protective thyristor, the threshold-operated transistor is triggered to permit the driver stage to operate the switching thyristor at the sweep rate. The resulting turn-on of the protective thyristor permits the rapid completion of charge of the supply capacitor to its normal operating voltage, and thereafter provides full overload protection for the deflection circuit.
1. In a communications receiver wherein (a) a first input switching thyristor of a deflection circuit in the receiver is pulsed at a sweep rate by a driver stage when the latter is excited, (b) a supply capacitor is chargeable through a second protective thyristor by-passed by a high-impedance shunt to provide operating bias to both the driver stage and the switching thyristor, and, (c) the output of the switching thyristor is coupled to the control electrode of the protective thyristor, the improvement which comprises: 2. The improvement as defined in claim 1, in which the gating means comprises a first transistor having its collector emitter path connected in series with the driver stage, and in which the coupling means comprises, in combination, a voltage divider having its input terminals connected across the supply capacitor, and means for applying the output of the voltage divider to the base of the first transistor. 3. The improvement as defined in claim 2, in which the applying means comprises a Zener diode. 4. The improvement as defined in claim 2, in which the driver comprises a second transistor, and in which the first and second transistors are of opposite conductivity type.
One conventional horizontal sweep circuit for a communications receiver has an input switching thyristor which is excitable to trigger the flyback phase at the deflection transformer. For this purpose, the control electrode of the switching thyristor is supplied with pulses at a sweep rate via a transistorized driver stage, whose base is coupled to the output of a horizontal oscillator. A common DC power source coupled to AC mains provides operating bias to the switching thyristor and to the driver stage via an output supply capacitor.
In order to provide overload protection of the deflection circuit it has been proposed to arrange a second protective thyristor in shunt with a high-ohmic bypass resistor in the DC power sound between the main rectifier and the supply capacitor. The voltage at the output of the switching thyristor (determined by the magnitude of the operating bias) is fed back in integrated form to the control electrode of the protective thyristor to maintain the latter in a conductive stage as required. In the presence of an overload in the deflection circuit, the output voltage of the switching thyristor drops to a value insufficient to trigger the protective thyristor on, thereby effecting the removal of operating bias from the deflection circuit.
While this arrangement is generally satisfactory for protection purposes once the deflection circuit is fully operative, its design is such that during start-up of the deflection circuit the driver stage is effectively controlled independently of the DC power source. As a result, start-up of the deflection circuit can occur before the voltage amplitude at the output of the supply capacitor (and thereby at the output of the switching thyristor) has reached a value necessary to assume reliable turnon of the protective thyristor.
SUMMARY OF THE INVENTION
The present invention provides an arrangement for assuring the reliable turn-on of the protective thyristor in the DC source during the start-up of the deflection circuit. In an illustrative embodiment the collector-emitter path of a normally disabled, threshold-operated transistor gate is serially connected with the transistorized driver stage to normally prevent excitation of the latter by the supply capacitor of the DC source. Consequently, the driver stage is initially prevented from starting up the deflection circuit. The base of the threshold transistor is excited by a voltage divider coupled across the supply capacitor.
The rectified current in the DC source initially charges the supply capacitor through the high-ohmic shunt resistor that bridges the normally off protective thyristor. When the voltage across the supply capacitor has reached a predetermined minimum value necessary to assume turn-on of the protective thyristor, the transistor gate is triggered on and the driver stage is excited to supply pulses to the switching thyristor. The rectified current of the power supply can now rapidly complete the charging of the supply capacitor to its full operating voltage through the now-conductive protective thyristor.
BRIEF DESCRIPTION OF THE DRAWING
The invention is further set forth in the following detailed description taken in conjunction with the appended drawing, in which:
FIG. 1 -- is a combined block and schematic diagram of a start-delay deflection circuit protection arrangement in accordance with the invention; and
FIG. 2 -- is a schematic diagram showing in more detail a threshold circuit useful in the arrangement of FIG. 1.
Referring now to the drawing, FIG. 1 illustrates a position of a communication receiver 49 having a deflection circuit 50, wherein a deflection transformer 12 applies a sweep voltage to the horizontal plates of a cathode ray tube (not shown). The deflection circuit includes an input switching thyristor 7 which may be employed, i.e. to establish the flyback phase of the transformer 12. The transconductive path of the thyristor 7 is shunted by a diode 8 of the opposite polarity. The output of the thyristor 7 is coupled through a series inductor 9 and a storage capacitor 10 to a forward sweep 11, which is schematically shown as having a forward sweep thyristor 51 shunted by an oppositely poled diode 52. The output of the circuit 11 is coupled to a primary winding 12A of the transformer 12, which is returned to ground via a capacitor 13.
In the normal operation of the deflection circuit 50, the control electrode of the switching thyristor 7 is supplied with pulses at the sweep rate from a driver stage 23 embodied by a PNP transistor. The base of the transistor 23 is coupled via an isolating capacitor 27 to the output of a horizontal oscillator 28. The required pulses for exciting the switching thyristor 7 are supplied to its control electrode via the collector of the transistor 23 through capacitor 31. The base of the transistor 23 is returned to ground through a resistor 29.
Operating bias for the transistor 23 and the deflection circuit 50 is supplied from a capacitor 6 disposed at the output of a DC power source 56, whose input terminals coupled to conventional AC mains are shown. In particular, the output of the capacitor 6 is coupled via a voltage divider 25, 26 to the emitter of the driver transistor 23; such emitter is returned to ground via capacitor 24. The output of the capacitor 6 is also coupled through a coil 14 to the deflection circuit 50 for excitation of the transconductive path of the thyristor 7.
The power supply 56 includes a diode 1 whose rectified output is effective to change the capacitor via a resistor 5 and the parallel combination of a normally disabled protective thyristor 2 and a high-impedance shunt resistor 3. In a conventional manner, the control electrode of the thyristor 2 is coupled via resistor 15 and diode 16 to the output of the switching thyristor 7, whose voltage amplitude is determined by the potential on the capacitor 6. Such output voltage is integrated by a capacitor 17 to yield an excitation potential for the thyristor 2 that is greater than its turn-on potential, so that thyristor 2 is ideally rendered conductive during alternate half cycles of the AC means. The capacitor 17 is returned to ground via a capacitor 4.
Theoretically, the protective thyristor 2 is switched to its "on" condition as soon as the driver transistor 23 applies excitation pulses to the control electrode of the transistor 7. In practice, the horizontal oscillator 28 that controls the transistor 23 is independent of the power supply 56, and there is no assurance that the voltage developed across the capacitor 6 and topped off the output of the switching thyristor 7 during turn-on of the deflection circuit will be sufficient to ignite the protective thyristor 2. In accordance with the invention, facilities are provided for preventing turn-on of the deflection circuit 50 until the voltage across the capacitor 6 has reached a predetermined minimum value necessary to assure the turn-on of the protective thyristor 2. In the embodiment shown, such facilities include a normally disabled threshold-operated NPN transistor 21 whose emitter is returned to ground and whose collector is coupled to the collector of the driver transistor 23 through a resistor 22. The base of the transistor 21 is coupled to a center tap 20 of a voltage divider 18, 19 connected in parallel with the supply capacitor 6. The parameters of the voltage divider 18, 19 and the threshold level of the transistor 21 are preferably chosen so that the transistor 21 is triggered on when the voltage at the tap 20 corresponds to the voltage across the capacitor 6 necessary to assume turn-on of the protective thyristor 2.
In operation, when the power supply 56 is first made operative, the rectified current from diode 1 partially charges the capacitor 6 through the high-impedance shunt 3. When the voltage across the capacitor 6 has risen to the point where the potential at the center tap 20 exceeds the threshold voltage of the transistor 21, the latter conducts and permits operating bias from the capacitor 6 to be applied to the driver transistor 23. The latter accordingly conducts to excite the control electrode of the switching thyristor 7 to effect turn-on of the deflection circuit 50.
Under the circumstances just described, the delayed start-up of the deflection circuit 50 is always accompanied by the ignition of the protective thyristor 2. As soon as the latter is made conductive, the rectified current from the diode 1 can rapidly complete the charging of the capacitor 6 to its normal operating voltage. The deflection circuit 50 thereafter operates in its normal, fully protected mode.
In order to increase the stability of the thresold level of the transistor 21, a Zener diode 30 (FIG. 2) may be disposed between the center tap 20 and the base of the transistor 21, as shown. Thus, triggering of the transistor 21 will occur only when the potential at the center tap 20 exceeds the breakdown level of the diode 30.
In the foregoing, the invention has been described in connection with a preferred arrangement thereof. Many variations and modifications will now occur to those skilled in the art. It is accordingly desired that the scope of the appended claims not be limited to the specific disclosure herein contained.
LOEWE-OPTA Energy stabilization in a horizontal deflection circuit for a television receiver:
A thyratron-like gating device governs the current supplied to an inductance in energy-exchanging relation with the storage capacitor in a reactive horizontal sweep circuit. The gating device is triggered by a control pulse that is delayed with respect to the start of the flyback portion of the sweep cycle by an amount proportional to the magnitude of the voltage stored on the capacitor at the start of flyback. The control pulse is generated upon the coincidence of a first saw-tooth voltage triggered at the start of flyback and a second voltage derived from the magnitude of the pulse generated in a secondary winding of the deflection transformer at the start of flyback. Suitable gain control facilities limit the magnitude of the second voltage to a value that assures generation of the control pulse within the flyback interval.
1. In a deflection circuit having a repetitive sweep cycle including a first relatively short flyback interval and a second relatively long forward sweep interval wherein the circuit comprises, in combination, a first storage capacitor coupled between a supply inductor and the primary winding of a deflection transformer, first means for coupling the input of the inductor to a source of operating voltage whereby an oscillatory energy interchange occurs between the operating voltage source and the output of the deflection circuit via the inductance and the first storage capacitor, and a first thyratron-like gating means associated with the inductor, the switching time of the first gating means being varied in dependence on changes in the quantity of energy in such oscillatory energy interchange, the improvement wherein the first gating means is serially connected between the source of operating voltage and the input of the inductor, and wherein the circuit further comprises, in combination, first means for generating, during each first interval, an output control pulse of substantially fixed duration whose time of occurrence relative to the start of the associated first interval is proportional to the voltage stored on the first capacitor at the start of such first interval, and second means for coupling the output of the first generating means to the control electrode of the first gating means, whereby the time of occurrence of the output control pulse is automatically adjusted to maintain a constant dynamic current-voltage ratio at the output of the deflection transformer. 2. A circuit as defined in claim 1, in which the first generating means comprises, in combination, comparator means having first and second inputs and an output adapted to exhibit a pulse upon a coincidence of voltage amplitudes applied to the first and second inputs, second means triggered at the start of the first interval for generating a first sawtooth voltage whose magnitude reaches a first value at the conclusion of the first interval, third means for generating a second voltage proportional to the magnitude of the voltage stored on the first capacitor at the start of the first interval, third means for coupling the output of the second generating means to the first input of the comparator means, and fourth means for coupling the output of the third generating means to the second input of the comparator means. 3. A circuit as defined in claim 2, in which the third generating means further comprises, in combination, a first secondary winding of the deflection transformer, and means for rectifying the output of the first secondary winding. 4. A circuit as defined in claim 2, in which the third generating means further comprises gain control means for limiting the maximum value of the second voltage to the first value. 5. A circuit as defined in claim 3, in which the third generating means comprises, in combination, second and third gating means, second and third capacitors, means for coupling the output of the rectifying means across the second and third capacitors, respectively, means for producing a third voltage proportional to the difference between the voltage across the second capacitor and a reference value, fifth means for coupling the third voltage to the control electrode of the second gating means so that such second gating means conducts when the third voltage is within a prescribed range, sixth means for coupling the voltage across the third capacitor to the control electrode of the third gating means, and seventh means for coupling the output of the second gating means to the control electrode of the third gating means, the third gating means being made conductive upon the conduction of the second gating means. 6. A circuit as defined in claim 5, in which the deflection transformer includes a second secondary winding, and in which the circuit further comprises, in combination, a common terminal for exciting the second and third generating means, a diode for coupling the second secondary winding to one terminal of the diode, eighth means for coupling the other terminal of the diode to the common terminal, and ninth means for coupling the third capacitor to the common terminal, the diode being rendered non-conductive to decouple the second secondary winding from the common terminal in the presence of voltage across the third capacitor.
Triggered reactive-type deflection circuits are frequently employed to effect the horizontal sweep portion of a television raster. The deflection circuit operates into the primary winding of a deflection transformer, one secondary winding of which may be coupled to the horizontal deflection plates of the picture tube.
The deflection circuit commonly includes a storage capacitor connected in oscillatory energy-exchanging relation with an input supply inductance, which in turn is coupled to a source of operating voltage. A pair of normally unoperated switching devices (typically thyristors) are connected on the output and input sides, respectively, of the capacitor. Such switching devices are separately triggerable to initiate the flyback and forward sweep portions, respectively, of the deflection circuit.
Certain deflection circuit arrangements of this type employ an additional secondary winding on the deflection transformer to trigger a regulating circuit that maintains a constant relation between dynamic deflection current changes (caused, e.g. by beam current variations in the picture tube) and the resulting changes in the voltage developed by the deflection circuit during the oscillations of energy between the operating voltage source and the deflection circuit load.
In one proposed arrangement of this type, facilities are provided for deriving, from a flyback pulse picked up by the additional secondary winding, a voltage proportional to the voltage on the storage capacitor at the start of the flyback interval. Such derived voltage is employed to vary the inductance of a regulating element coupled to the input of the deflection circuit. Such variation of the element inductance correspondingly varies the oscillatory energy exchange of the deflection circuit.
This type of control scheme has several main disadvantages. The -variable regulating element, besides being expensive to instrument, is bulky and occupies a large space in the television receiver. Moreover, this type of control scheme tends (except when operating at a particular load point such as maximum load current) to overcompensate for dynamic energy changes resulting in the application of an excess amount of correcting energy to the deflection circuit. Such excess is recycled between the capacitor in the deflection circuit and the power supply capacitor of the operating voltage source, and leads to relatively high power losses in the circuit.
SUMMARY OF THE INVENTION
The present invention provides an improved arrangement for stabilizing the energy conditions in a deflection circuit having a thyratron-like switch associated with an input storage inductance, while avoiding the above-mentioned disadvantages. In one embodiment, such thyratron-like switch (hereafter first gating device) is interposed in series between the operating voltage source and the input of the supply inductance. The first gating device is triggered at a point during the flyback interval determined by the magnitude of the flyback pulse picked off the additional secondary winding and thereby the voltage stored on the capacitor at the start of such flyback interval.
With this arrangement, the decrease in the amplitude of the flyback pulse which initially accompanies an increase in the deflection load current causes a triggering of the first gating device earlier in the flyback interval, thereby resulting in a higher current developed by the supply inductor at the end of the flyback interval. The energy increment represented by such developed current flow is transferred directly to the capacitor to increase the voltage stored thereacross during the succeeding forward sweep interval, thereby avoiding recycling losses, the necessity of "dumping" the excess energy stored in the input inductance, and the requirement of blocking diodes and similar circuitry. Since in this way the above-mentioned increase in load current effects a corresponding increase in the deflection circuit voltage, their ratio is maintained constant as desired.
A feature of the invention is the provision of facilities for converting variations in the amplitude of the flyback pulse to corresponding variations in the times of occurrence of the control pulses for the first gating device. An auxiliary saw-tooth voltage triggered at the start of the flyback interval is applied to one input of a comparator, e.g. a transistorized phase-shifting stage. To the other input of the comparator circuit is coupled a second voltage proportional to the amplitude of the flyback pulse. This second voltage is advantageously developed by a gain-controlled amplifier arranged so that the highest amplitude of the second voltage corresponds to the voltage level attained by the saw-tooth voltage at the end of the flyback interval.
Excitation of the saw-tooth generator, the comparator, and the gain-control amplifier may be accomplished over a common excitation path by either the second voltage or by a third voltage obtained from an additional auxiliary winding of the deflection transformer through a series diode. The presence of the second voltage disables such series diode to decouple the third voltage from the excitation path.
BRIEF DESCRIPTION OF THE DRAWING
The invention will be further set forth in the following detailed description taken in conjunction with the appended drawing, in which:
FIG. 1 is a combined block and schematic diagram of a regulated deflection circuit employing a thyratron-like gating device in accordance with the invention;
FIG. 2a is a composite graph showing several voltage and current wave forms in the arrangement of FIG. 1;
FIG. 2b and 2c are relative plots of the times of occurrence of trigger pulses respectively generated at the moment of triggering and at the start of the flyback interval of the gating device of FIG. 1;
FIG. 2d is a composite graph representing the production, by the arrangement of FIG. 1, of a thyratron control pulse at a given time during the flyback interval; and
FIG. 3 is a detailed schematic diagram of the circuit of FIG. 1.
Referring now to FIG. 1 of the drawing, there is pictorially represented a portion of a TV receiver including a picture tube 101. Several of the required operating voltages of the tube 101 and of the remainder of the receiver are supplied conventionally via a plurality of secondary windings of a transformer 8. Such windings include, e.g. a winding 10 for applying high voltage to the tube 101, a second winding 102 for exciting the horizontal deflection plates of the tube 101, and a third winding 103 for supplying filament current to the tube 101.
A primary winding 7 of the transformer 8 is driven by a triggered, substantially reactive horizontal deflection circuit 110. The circuit 110 establishes the horizontal portion of the television raster by generating a repetitive sweep voltage for application to the horizontal plates of the tube 101 via winding 102. Each cycle of such sweep voltage has a relatively short flyback portion starting at time T O (FIG. 2a) and a relatively long forward sweep portion starting at time T 3 .
The circuit 110 includes a main storage capacitor 6 coupled at its output to the primary winding 7, which is returned to ground via a capacitor 9. The storage capacitor 6 is coupled at its input to a main supply inductance 12 via a coil 5. The storage inductance 12 is coupled to a source of operating voltage V B as described below.
The transconductive path of a first switching thyristor 3 is coupled between the output of capacitor 6 and ground, and is shunted by a diode 4 of opposite polarity. The transconductive path of a second switching thyristor 1 is coupled between the junction of inductors 5 and 12 and ground, and is likewise shunted by a diode 2 of opposite polarity.
The thyristor 1 is triggered at the time T O of each sweep cycle to initiate the flyback interval. For this purpose, a suitable ignition generator 15 synchronized by a conventional horizontal sync pulse generator 14 is coupled to the control electrode of the thyristor 1. The generator 15 may be embodied by the horizontal oscillator 15 described, e.g., in U.S. Pat. No. 3,767,960 issued to P. R. Ahrens on Oct. 23, 1973.
The thyristor 3 is triggered at the instant T 3 in each sweep cycle to terminate such flyback portion and initiate the forward sweep portion. This is accomplished by coupling an auxiliary winding 111 of the supply inductance 12 to the control electrode of the thyristor 3 over a coupling circuit 112.
In order to stabilize the load conditions of the circuit 110 in the presence of dynamic variations of load energy (e.g., via beam current variations in the tube 101), the deflection circuit is provided with facilities for maintaining the ratio of the dynamic load current variation to the variation in voltage developed by the deflection circuit at a sensibly constant value, e.g. 0.5. Such facilities include an additional secondary winding 11 of the transformer 8 designed to pick off, at the time T O representing the start of the flyback interval, a pulse developed by the transformer 8 in response to the discharge current of the storage capacitor 6 through the now-triggered thyristor switch 1. Such discharge current is depicted as curve 121 in FIG. 2a. The amplitude of the pulse developed by the winding 11 is proportional to the voltage stored across the capacitor 6 at the time T O ; the capacitor voltage curve is represented as curve 122 in FIG. 2a.
In accordance with the invention, such regulating facilities for the deflection circuit 110 further include a gating device 13 including a thyratron-like switch 20 (illustratively a thyristor) as its active element. The thyristor 20 is triggered, as indicated below, only during the flyback interval T O - T 3 . The time of occurrence T X of the triggering instant relative to the time T O is made proportional to the amplitude of the flyback pulse picked off the auxiliary winding 11. (The relative times of occurrence of typical pulses for triggering the thyristor 20 and the flyback-initiating thyristor 1, respectively, are depicted in FIGS. 2b and 2c.)
The triggering of the thyristor 20 serves to couple the operating voltage V B to the inductance 12 for the remaining portion (T X - T 3 ) of the flyback interval. During the portion T X - T 3 , the energy supplied to the deflection circuit is determined in accordance with the following current developed in the inductor 12: ##EQU1## where i LS is the inductor current (represented as curve 123 in FIG. 2a), and K is a constant related to the inductance of the inductor 12.
As indicated further in FIG. 2a, the energy increment represented by the inductor current i LS is transferred to the storage capacitance 6 during the portion T 3 -T Y of the next succeeding forward sweep interval. Such transferred energy is represented by a corresponding increment of the voltage stored across the capacitor 6. Such voltage increment, in turn, leads to a corresponding increment in the amplitude of the flyback pulse picked off winding 11 at the start of the next succeeding flyback interval.
As a result of this sequence of operations, an instantaneous increase in the power extracted by the load on the deflection circuit (represented e.g. by an increase in beam current of the tube 101) will initially cause a drop in the power consumed by the deflection circuit itself and thereby a corresponding reduction in the voltage stored in the capacitor 6. The instant regulating arrangement for the deflection circuit responds to the resulting reduction in the amplitude of the next flyback pulse developed by the winding 11 to advance the time T X of triggering of the thyristor 20 so that the latter conducts for a longer portion of the flyback interval. This causes an increase in the current through the inductance 12 at the time T 3 . The increase in energy applied to the deflection circuit represented by such increased current is thereafter directly transferred, during the next interval T 3 - T Y , to the storage capacitor 6 as an increase in its stored voltage. Thus, the assumed increase in load current is effective to cause a corresponding voltage increase in the deflection circuit so that the ratio of the current change to the voltage change remains sensibly constant.
It will be appreciated that a corresponding effect will be obtained when the energy balance between the deflection circuit and the load is disturbed for any other reason, e.g. a decrease in load energy consumption (leading to a retardation of the triggering time T X ), or a change in the level of the operating voltage V B . In all such cases, the energy unbalance is exactly compensated without the lossy recycling of energy typical of the priorly proposed arrangements.
One suitable arrangement for converting the amplitude of the flyback pulses from the winding 11 into suitably timed control pulses for the thyristor 20 is indicated at 130 in FIG. 1. The horizontal oscillator 14 triggers, in synchronism with the ignition generator 15, a sawtooth generator 16 which provides a first linearly increasing voltage [represented by curve 131 in FIG. 2d ]. The output of the sawtooth genertor 16 is coupled to a first input of a comparator 17. A second voltage [represented by a level 132 in FIG. 2d], proportional to the amplitude of the preceding flyback pulse on winding 11, is coupled via a gain-controlled amplifier 18 to a second input of the comparator 17. As represented in FIG. 2d, the comparator exhibits an output pulse 133 at the instant T X that the first sawtooth voltage reaches the level of the second voltage. Such output pulse is suitably shaped if desired in a pulse shaper 19 and is then applied to the control electrode of the thyristor 20.
In order to assure triggering of the thyristor 20 via the comparator 17 during the flyback interval T 0 - T 3 , the controlled amplifier 18 is provided with facilities for limiting the maximum value of the second voltage to a value corresponding to the voltage level of the first sawtooth wave from the generator 16 at the time T 3 .
The converting arrangement 130, together with facilities for triggering the other thyristors 1 and 3 in the deflection circuit 110, are shown in more detail in FIG. 3. The gain-controlled amplifier 18 includes an input switching transistor 33 and an output coupling transistor 36; the latter is effective when conductive to effect the application of the second voltage to the second input of the comparator 17. In particular, the output of the winding 11 is rectified via diodes 29 and 39 and individually stored across capacitors 30 and 38, the latter via capacitor 40 and resistor 41. The output of the capacitor 38 is coupled via resistor 37 to the base of the transistor 36. The output of a voltage divider 31, whose input is coupled across capacitor 30, is coupled via Zener diode 32 to the base of switching transistor 33. The conduction condition of the transistor 33 is established, e.g. when the voltage across the capacitor 30 is in a prescribed range relative to the stabilized voltage level of the Zener diode.
The output of transistor 33 is coupled via resistor 35 to the base of transistor 36, which is triggered on when the switching transistor 33 is conductive to establish the second voltage level across a capacitor 43. By suitably selecting the voltage level at the voltage divider 31, such second voltage may be maintained within the range necessary to assure triggering of the thyristor 20 within the flyback interval as indicated before.
The comparator 17 is embodied as a phase shifter having a transistor 44. A capacitor 45 is connected from the base of transistor 44 to ground to establish the first comparator input. The above-mentioned capacitor 43, which stores the second voltage level, is connected from the emitter of transistor 44 to ground. The output of the sawtooth generator 16, whose active element is a transistor 49 and which otherwise is provided with conventional R-C and flyback circuitry, is coupled across the capacitor 45. The output of the transistor 44 is coupled to the control electrode of the thyristor 20 via pulse shaper 19 (including stages 48, 57 and 60) and a transformer 21.
Excitation of the sawtooth generator 16, the comparator circuit 17, the pulse shaper 19 and the gain-controlled amplifier 18 is provided over two separate paths: The winding 103 of the transformer 8 via a diode 63, and the portion of the rectified flyback pulse voltage developed across capacitor 38. This latter voltage serves, when present, to diasble the diode 63, so that only one of the two excitation paths is active at any given time.
In the foregoing, the invention has been described in connection with a preferred arrangement thereof. Many variations and modifications will now occur to those skilled in the art. It is accordingly desired that the scope of the appended claims not be limited to the specific disclosure herein contained.
LOEWE F875 SENSOTRONIC CHASSIS 543/490: Electron beam deflection circuit including thyristors Further Discussion and deepening of knowledge, Thyristor horizontal output circuits General concept :
2. A deflection circuit as claimed in claim 1, wherein said amount of additional current is greater than or equal to 5 per cent of the peak-to-peak value of the current flowing through the deflection winding.
3. A deflection circuit as claimed in claim 1, wherein said means for drawing a substantial amount of additional current through said first switching means comprises a resistor connected in parallel to said first capacitor.
4. A deflection circuit as claimed in claim 1, wherein said means for drawing an additional current is formed by connecting said first and second energy sources in series so that the current charging said reactive circuit means forms the said additional current.
5. A deflection circuit as claimed in claim 1, further including a series combination of an autotransformer winding and a second high-value capacitor, said combination being connected in parallel to said first switching means, wherein said autotransformer comprises an intermediate tap located between its terminals respectively connected to said first switching means and to said second capacitor, said tap delivering, during said trace portion, a suitable DC supply voltage lower than the voltage across said second capacitor; and wherein said means for drawing a substantial amount of additional current comprises a load to be fed by said supply voltage and having one terminal connected to ground; and further controllable switching means controlled to conduct during at least part of said trace portion and to remain cut off during said retrace portion, said further switching means being connected between said tap and the other terminal of said load.
The present invention constitutes an improvement in the circuit described in U.S. Pat. No. 3,449,623 filed on Sept. 6, 1966, this circuit being described in greater detail below with reference to FIGS. 1 and 2 of the accompanying drawings. A deflection circuit of this type comprises a first thyristor switch which allows the conenction of the horizontal deflection winding to a constant voltage source during the time interval used for the transmisstion of the picture signal and for applying this signal to the grid of the cathode ray tube (this interval will be termed the "trace portion" of the scan), and a second thyristor switch which provides the forced commutation of the first one by applying to it a reverse current of equal amplitude to that which passes through it from the said voltage source and thus to initiate the retrace during the horizontal blanking interval.
A undirectional reverse blocking triode type thyristor or silicon controlled rectifier (SCR), such as that used in the aformentioned circuit, requires a certain turn-off time between the instant at which the anode current ceases and the instant at which a positive bias may be applied to it without turning it on, due to the fact that there is still a high concentration of free carriers in the vicinity of the middle junction, this concentration being reduced by a process of recombination independently from the reverse polarity applied to the thyristor. This turn-off time of the thyristor is a function of a number of parameters such as the junction temperature, the DC current level, the decay time of the direct current, the peak level of the reverse current applied, the amplitude of the reverse anode to cathode voltage, the external impedance of the gate electrode, and so on, certain of these varying considerably from one thyristor to another.
In horizontal deflection circuits for television receivers, the flyback or retrace time is limited to approximately 20 percent of the horizontal scan period, the retrace time being in the case of the CCIR standard of 625 lines, approximately 12 microseconds and, in the case of the French standard of 819 lines, approximately 9 microseconds. During this relatively short interval, the thyristor has to be rendered non-conducting and the electron beam has to be returned to the origin of the scan. The first thyristor is blocked by means of a series resonant LC circuit which is subject to a certain number of restrictions (limitations as to the component values employed) due to the fact that, inter alia, it simultaneously determines the turn-off time of the circuit which blocks the thyristor and it forms part of the series resonant circuit which is to carry out the retrace. To obtain proper operation of the deflection circuit of the aforementioned Patent, especially when used for the French standard of 819 lines per image, the values of the components used have to subject to very close tolerances (approximately 2%), which results in high costs.
The improved deflection circuit, object of the present invention, allows the lengthening of the turn-off time of the circuit for turning the scan thyristor off, without altering the values of the LC circuit, which are determined by other criteria, and without impairing the operation of the circuit.
According to the invention, there is provided an electron beam deflection circuit for a cathode ray tube with electromagentic deflection by means of a sawtooth current waveform having a trace portion and a retrace portion, said circuit comprising: a deflection winding; a first source of electrical energy formed by a first capacitor; first controllable switching means comprising a parallel combination of a first thyristor and a first diode, connected together to conduct in opposite directions, for connecting said winding to said first source during said trace portion when said first switching means is turned on; a second source of electrical energy including a first inductive energy storage means coupled to a voltage supply; reactive circuit means including a combination of inductive and capacitive reactances for storing the energy supplied by the said second source; a second controllable switching means, substantially identical with the first one, for completing a circuit including said reactive circuit means and said first switching means, when turned on, so as to pass through said first thyristor an oscillatory current in the opposite direction to that which passes through it from said first source and to turn it off after these two currents cancel out, the oscillatory current then flowing through said first diode for an interval termed the circuit turn-off time which has to be greater than the turn-off time of said first thyristor; and means for drawing duing at least a part of said trace portion a substantial amount of additional current from said first switching means in the direction of conduction of said first diode, whereby said circuit turn-off time is lengthened in proportion to the amount of said additional current, without altering the values of the reactances in the reactive circuit by shifting the waveform of the current flowing through said first switching means towards the negative by an amount equal to that of said additional current.
A further object of the invention consists in using the supplementary current in the recovery diode of the first switching means to produce a DC voltage which may be used as a power supply for the vertical deflection circuit of the television receiver, for example.
The invention will be better understood and other features and advantages thereof will become apparent from the following description and the accompanying drawings, given by way of example, and in which:
FIG. 1 is a schematic circuit diagram partially in bloc diagram form of a prior art deflection circuit according to the aforementioned Patent;
FIG. 2 shows waveforms of currents and voltages generated at various points in the circuit of FIG. 1;
FIG. 3 is a schematic diagram of a deflection circuit according to the invention which allows the principle of the improvement to be explained;
FIG. 4 is a diagram of the waveforms of the current through the first switching means 4, 5 of the circuit of FIG. 3;
FIG. 5 is a circuit diagram of another embodiment of the circuit according to the invention;
FIG. 6 is a schematic representation of the preferred embodiment of the circuit according to the invention; and
FIG. 7 shows voltage waveforms at various points of the high voltage autotransformer 21 of FIG. 6.
In all these Figures the same reference numerals refer to the same components.
FIG. 1 shows the horizontal deflection circuit described and claimed in the U.S. Pat. No. 3,449,623 mentioned above, which comprises a first source of electrical energy in the shape of a first capacitor 2 having a high capacitance C 2 for supplying a substantially constant voltage Uc 2 across its terminals. A first terminal of the first capacitor 2 is connected to ground, whilst its second terminal which supplies a positive voltage is connected to one of the terminals of a horizontal deflection winding shown as a first inductance 1. A first switching means 3, consisting of a first reverse blocking triode thyristor 4 (SCR) and a first recovery diode 5 in parallel, the two being interconnected to conduct current in opposite directions, is connected in parallel with the series combination formed by the deflection winding 1 and the first capacitor 2. The assembly of components 1, 2, 4 and 5 forms the final stage of the horizontal deflection circuit in a television receiver using electromagnetic delfection.
The deflection circuit also includes a drive stage for this final stage which here controls the turning off of the first thyristor 4 to produce the retrace or fly-back portion of the scan during the line-blanking intervals i.e. while the picture signal is not transmitted. This driver stage comprises a second voltage source in the shape of a DC power supply 6 which delivers a constant high voltage E. The negative terminal of the power supply 6 is connected to ground and its positive terminal to one of the terminals of a second inductance 7 of relatively high value, which draws a substantially lineraly varying current from the power supply 6 to avoid its overloading. The other terminal of the second inductance 7 is connected, on the one hand, to the junction of the deflection winding 1 and the first switching means 3 by means of a second inductance 8 and a second capacitor 9 in series and, on the other hand, to one of the terminals of a second controllable bi-directionally conducting switching means 10, similar to the first one 3, including a parallel combination of a second thyristor 11 and a second recovery diode 12 also arranged to conduct in opposite directions.
The respective values of the third inductance 8 (L 8 ) and of the second capacitor 9 (C 9 ) are principally selected so that, on the one hand, one half-cycle of oscillation of the first series resonant circuit L 8 - C 9 , (i.e. π √ L 8 . C 9 ) is longer than the turn-off time of the first thyristor 4, but still is as short as possible since this time interval determines the speed of the commutation of the thyristor 4, and, on the other hand, one half-cycle of oscillation of another series resonant circuit formed by L 1 , L 8 and C 9 , i.e. π √ (L 1 + L 8 ) . C 9 , is substantially equal to the required retrace time interval (i.e. shorter than the horizontal blanking interval).
The gate (control electrode) of the second thyristor 11 is coupled to the output of the horizontal oscillator 13 of the television receiver by means of a first pulse transformer 14 and a first pulse shaping circuit 15 so that it is fed short triggering pulses which are to turn it on.
The gate of the first thyristor 4 fed with signals of a substantially rectangular waveform which are negative during the horizontal blanking intervals, is coupled to a winding 16 by means of a second pulse shaping circuit 17, the winding 16 being magnetically coupled to the second inductance 7 to make up the secondary winding of a transformer of which the inductance 7 forms the primary winding. It will be noted here that it is also possible to couple the secondary winding 16 magnetically to a primary winding connected to a suitable output (not shown) of the horizontal oscillator 13.
The operation of a circuit of this type will be explained below with reference to FIG. 2 which shows the waveforms at various points in the circuit of FIG. 1 during approximately one line period.
FIG. 2 is not to scale since one line period (t 7 - t 0 ) is equal to 64 microseconds in the case of 625 lines and 49 microseconds in the case of 819 lines, while the durations of the respective horizontal blanking intervals are approximately 12 and 9.5 microseconds.
Waveform A shows the form of the current i L1 passing through deflection winding 1, this current having a sawtooth waveform substantially linear from t 0 to t 3 and from t 5 to t 7 , and crossing zero at time instants t 0 and t 7 , and reaching values of + I 1m and - I 1m , at time instants t 3 and t 5 respectively, these being its maximum positive and negative amplitudes.
During the second half of the trace portion of the horizontal deflection cycle, that is to say from t 0 to t 3 , the thyristor 4 of the first switching means 3 is conductive and makes the high value capacitor 2 discharge through the deflector winding 1, which has a high inductance, so that current i L1 increases linearly.
A few microseconds (5 to 8 μ s) before the end of the trace portion, i.e. at time instant t 1 , the trigger of the second thyristor 11 receives a short voltage pulse V G11 which causes it to turn on as its anode is at this instant at a positive potential with respect to ground, which is due to the charging of the second capacitor 9 through inductances 7 and 8 by the voltage E from the power supply 6.
When thyristor 11 is made conductive at time t 1 , on the one hand, inductance 7 is connected between ground and the voltage source 6 and a linearly increasing current flows through it and, on the other hand, the reactive circuit 8, 9 forms a loop through the second and first switching means 10 and 3, thus forming a resonant circuit which draws an oscillatory current i 8 ,9 of frequency ##EQU1##
This oscillatory current i 8 ,9 will pass through the first switching means 3, i.e. thyristor 4 and diode 5, in the opposite direction to that of current i L1 . Since the frequency f 1 is high, current i 8 ,9 will increase more rapidly than i L1 and will reach the same level at time t 2 , that is to say i 8 ,9 (t 2 ) = -i L1 (t 2 ) and these currents will cancel out in the thyristor 4 in accordance with the well known principle of forced commutation. After time instant t 2 , current i 8 ,9 continues to increase more rapidly than i L1 , but the difference between them (i 8 ,9 - i L1 ) passes the diode 5 (see wave form B) until it becomes zero at time instant t 3 which is the turn off time instant of the first switching means 3, at which the retrace begins.
The interval between the time instant t 2 and t 3 , i.e. (t 3 -t 2 ), during which diode 5 is conductive and the thyristor is reverse biased will be termed in what follows the circuit turn-off time and it should be greater than the turn-off time of the thyristor 4 itself since the latter will subsequently become foward biased (i.e. from t 3 to t 5 ) by the retrace or flyback pulse (see waveform E) which should not trigger it.
At time instant t 3 , the switching means 3 is opened (i 4 and i 5 are both zero -- see waveforms B and C) and the reactive circuit 8, 9 forms a loop through capacitor 2 and the deflection coil 1 and thus a series resonant circuit including (L 1 + L 8 ) and C 9 , C 2 being of high value and representing a short circuit for the flyback frequency ##EQU2## thus obtained.
The retrace which stated at time t 3 takes place during one half-cycle of the resonant circuit formed by reactances L 1 , L 8 and C 9 , i.e. during the interval between t 3 and t 5 . In the middle of this interval i.e. at time instant t 4 , both i L1 (waveform A) and i 8 ,9 (waveform D) pass through zero and change their sign, whereas the voltage at the terminals of the first switching means 3 (V 3 , waveform E) passes through a maximum. Thus, from t 4 onwards, thyristor 11 will be reverse biased and diode 12 will conduct the current from the resonant circuit 1, 8 and 9 in order to turn the second thyristor 11 off.
At time instant t 5 , when current i L1 has reached - I 1m and when voltage v 3 falls to zero, diode 5 of the first switching means 3 becomes conductive and the trace portion of scan begins.
Current i 8 ,9 nevertheless continues to flow in the resonant circuit 8, 9 through diodes 5 and 12, which causes a break to appear in waveform D at t 5 , and a negative peak to appear in waveform D and a positive one in waveform B in the interval between t 5 and t 6 , these being principally due to the distributed capacities of coil 1 or to an eventual capacitor (not shown) connected in parallel to the first switching means 3.
At time instant t 6 , diode 12 of the second switching means 10 ceases to conduct after having allowed thyristor 11 time to become turned off completely.
The level of current i 8 ,9 at time instant t 5 (i.e. I c ) as well as the negative peak I D12 in i 8 ,9 and the positive peak I D5 in i 5 depend on the values of L 8 and C 9 in the same way as does the turn-off time of the circuit (t 3 - t 2 ). If, for example, L 8 and C 9 , are increased I D5 increases towards zero and this could cause diode 5 to be cut off in an undesirable fashion. I c also increases towards zero, which is liable to cause diode 12 to be blocked and thyristor 11 to trigger prematurely.
From the foregoing it can be clearly seen that the choice of values for L 8 and C 9 is subject to four limitations which prevent the values from being increased to lengthen the turn-off time of the driver circuit of first switching thyristor 4 so as to forestall its spurious triggering.
Waveform F shows the voltage v G4 obtained at the gate of thyristor 4 from the secondary winding 16 coupled to the inductor 7. This voltage is positive from t 0 to t 1 and from t 6 to t 7 and is negative between t 2 and t 6 i.e. while the second switching means 10 is conducting.
The present invention makes the lengthening of the turn-off time of thyristor 4 possible without altering the parameters of the circuit such as inductance 8 and capacitor 9.
In the circuit shown in FIG. 3, which illustrates the principle of the present invention, means are added to the circuit in FIG. 1 which enable the turn-off time to be lengthened by connecting a load to diode 5 so as to increase the current which flows through it during the time that it is conductive. These means are here formed by a resistor 18 connected in parallel with a capacitor 20 (which replaces capacitor 2) which is of a higher capacitance so that, in practice, it holds its charge during at least one half of the line period. FIG. 4, which shows the waveform of the current in the first switching means 3 for a circuit as shown in FIG. 3, makes it possible to explain how this lenthening of the turn-off time is achieved.
In FIG. 4, the broken lines show the waveform of the current in the first switch device 3 in the circuit of FIG. 1, this waveform being produced by adding waveforms B and C of FIG. 2. The current i 4 above the axis flows through thyristor 4 and current i 5 below the axis flows through diode 5. When the capacitance C 20 of the capacitor in series with the deflector coil is increased to some tens of microfarads (C 2 having been of the order of 1 μ F) and when there is connected in parallel with capacitor 20 a resistor 18 the value of which is calculated to draw a strong current I R18 from capacitor 20, that is to say a current at least equal to 0,1 I m (I m being of the order of some tens of amperes), current I R18 is added to that i 5 which flows through diode 5 without in any way altering the linearity of the trace portion nor the oscillatory commutation of thyristor 4 which is brought about by the resonant circuit L 8 , C 9 .
The fact of loading capacitor C 20 by means of a resistor 18 thus has the effect of permanently displacing the waveform of the current in the negative direction by I R18 . Thus, during the trace portion of the scan, the transfer of the current from the diode 5 to the thyristor 4 begins at time t 10 instead of t 0 , that is to say with a delay proportional to I R18 . The effect of the triggering pulse delivered by the horizontal oscillator (13 FIG. 1) to the second thyristor 11 at time instant t 1 , will be to start the commutation process of the first thyristor 4 when the current it draws is less by I R18 than that i 4 (t 1 ) which it would have been drawing had there been no resistor 18. Because of this, the turn-off time of the thyristor 4 proper, which as has been mentioned increases with the maximum current level passing throught it, is slightly reduced. Moreover, because the oscillatory current i 8 ,9 (FIG. 2) from circuit L 8 , C 9 which flows through thyristor 4 in the opposite direction is unchanged, it reaches a value equal to that of the current i L1 (FIG. 1) flowing in the coil 1 in a shorter time, that is to say at time t 12 . Diode 5 will thus take the oscillatory current i 8 ,9 (FIG. 2) over in advance with respect ro time instant t 2 and will conduct it until it reaches zero value at a time instant t 13 later than t 3 , the amounts of advance (t 2 - t 12 ) and delay (t 13 - t 3 ) being practically equal.
It can thus be seen in FIG. 4 that the circuit turn-off time T R of a circuit according to the invention and illustrated by FIG. 3 is distinctly longer than that T r of the circuit in FIG. 1. This increase in the turn-off time (T R - T r ) depends on the current I R18 and increases therewith.
It should be noted at this point that the current I R18 produces a voltage drop at the terminals of the resistor the only effect of which is to heat up the resistor since the level of this voltage (40 to 60 volts) does not necessarily have a suitable value to be used as a voltage supply for other circuits in an existing transistorised television receiver.
In accordance with one embodiment of the invention, illustrated in FIG. 5, an application is proposed for the additional current which is to be drawn through diode 5. In FIG. 5, the positive terminal of capacitor 20 is connected by a conductor 19 to the negative pole of the power supply 6 and the voltage at the terminals of capacitor 20 is thus added to that E from the source 6.
In the preferred embodiment of the present invention, which is shown in FIG. 6, it is possible to cause a supplementary current of a desired value to flow through the first diode 5 while obtaining a voltage which has a suitable value for use in another circuit in the television receiver.
If the voltage at the terminals of capacitor 20 in FIG. 3 is not a usable value, it is possible to connect in parallel with the series circuit comprising the deflector coil 1 and the capacitor 2 in FIG. 1, i.e. in parallel with the terminals of the first switching means 3, a series combination of an autotransformer 21 and a high value capacitor 22 (comparable with capacitor 20 in FIGS. 3 and 5). The autotransformer 21 has a tap 23 is suitably positioned between the terminal connected to capacitor 22 at the tap 24 connected to the first switching means 3. This autotransformer 21 may be formed by the one conventionally used for supplying a very high voltage to the cathode ray tube, as described for example in U.S. Pat. No. 3,452,244; such a transformer comprises a voltage step-up winding between taps 24 and 25, which latter is connected to a high voltage rectifier (not shown).
The waveform of the voltage at the various points in the autotransformer is shown in FIG. 7, in which waveform A shows the voltage at the terminals of capacitor 22, waveform B the voltage at tap 24 and waveform C the voltage at tap 23 of the autotransformer 21.
The voltage V c22 at the terminals of capacitor 22 varies slightly about a mean value V cm . It is increasing while diode 5 is conducting and decreasing during the conduction of the thyristor 4.
The voltage v 24 at tap 24 follows substantially the same curve as waveform E in FIG. 2, that is to say that during the retrace time interval from t 13 to t 5 to a positive pulse called the flyback pulse is produced and, during the time interval while the first switching means 3 is conducting, the voltage is zero. The mean valve of the voltage v 24 at tap 24 of the auto-transformer 21 is equal to the mean value V cm of the voltage at the terminals of capacitors 2 and 22.
Thus, there is obtained at tap 23 a waveform which is made up, during the retrace portion, of a positive pulse whose maximum amplitude is less than that of v 24 at tap 24 and, during the trace portion, of a substantially constant positive voltage, the level V of which is less than the mean value V cm of the voltage v c22 at the terminals of capacitor 22. By moving tap 23 towards terminals 24 the amplitude of the pulse during fly-back increases while voltage V falls and conversely by moving tap 23 towards capacitor 22 voltage V increases and the amplitude of the pulse drops.
In more exact terms, the voltage V at tap 23 is such that the means value of v 23 is equal to V cm . It has thus been shown that by choosing carefully the position of tape 23, a voltage V may be obtained during the trace portion of the scan, which may be of any value between V cm and zero.
This voltage V is thus obtained by periodically controlled rectification during the trace portion of the scan. For this purpose an electronic switch is used to periodically connect the tap 23 of trnasformer winding 21 to a load. This switch is made up of a power transistor 26 whose collector is connected to tap 23 and the emitter to a parallel combination formed by a high value filtering capacitor 27 and the load which it is desired to supply, which is represented by a resistor 28. The base of the transistor 26 receives a control voltage to block it during retrace and to unblock it during the whole or part of the trace period. A control voltage of this type may be obtained from a second winding 29 magnetically coupled to the inductance 7 of the deflection circuit and it may be transmitted to the base of transistor 26 by means of a coupling capacitor 30 and a resistor 31 connected between the base and the emitter of transistor 26.
It may easily be seen that the DC collector/emitter current in transistor 26 flows through the first diode 5 of the first switching means 3 via a resistor 28 and the part of the winding of auto-transformer 21 located between taps 23 and 24.
Experience has shown that a circuit as shown in FIG. 6 can supply 24 volts with a current of 2 amperes to the vertical deflection circuit of the same television set, the voltage at the terminals of capacitor 22 being from 50 to 60 volts.
It should be mentioned that, when the circuit which forms the load of the controlled rectifier 26, 27 does not draw enough current to sufficiently lengthen the circuit turn-off time T R , an additional resistor (not shown) may be connected between the emitter of transistor 26 and ground or in parallel to capacitor 22, which resistor will draw the additional current required.
INTEGRAL THYRISTOR-RECTIFIER DEVICEA semiconductor switching device comprising a silicon controlled rectifier (SCR) and a diode rectifier integrally connected in parallel with the SCR in a single semiconductor body. The device is of the NPNP or PNPN type, having gate, cathode, and anode electrodes. A portion of each intermediate N and P region makes ohmic contact to the respective anode or cathode electrode of the SCR. In addition, each intermediate region includes a highly conductive edge portion. These portions are spaced from the adjacent external regions by relatively low conductive portions, and limit the conduction of the diode rectifier to the periphery of the device. A profile of gold recombination centers further electrically isolates the central SCR portion from the peripheral diode portion.
Gating circuit for television SCR deflection system AND REGULATION / stabilization of horizontal deflection NETWORK CIRCUIT with Transductor reactor / Reverse thyristor energy recovery circuit.GRUNDIG SUPER COLOR 6236/30 SERIE 30F22 CHASSIS 29301-114.68(07)
In a television deflection system employing a first SCR for coupling a deflection winding across a source of energy during a trace interval of each deflection cycle and a second SCR for replenishing energy to the source of energy during a commutation interval of each deflection cycle, a gating circuit for triggering the first SCR. The gating circuit employs a voltage divider coupled in parallel with the second SCR which develops gating signals proportional to the voltage across the second SCR.
1. In a television deflection system in which a first switching means couples a deflection winding across a source of energy during a trace interval of each deflection cycle and a second switching means replenishes energy to said source of energy during a commutation interval of each deflection cycle, a gating circuit for said first switching means, comprising:
capacitive voltage divider means coupled in parallel with said second switching means for developing gating signals proportional to the voltage across said second switching means; and
means for coupling said voltage divider means to said first switching means to provide for conduction of said first switching means in response to said gating signals.
2. A gating circuit according to claim 1 wherein said voltage divider includes first and second capacitors coupled in series and providing said gating signals at the common terminal of said capacitors. 3. A gating circuit according to claim 2 wherein said first and second capacitors are proportional in value to provide for the desired magnitude of gating signals. 4. A gating circuit according to claim 3 wherein said means for coupling said voltage divider means to said first switching means includes an inductor. 5. A gating circuit according to claim 4 wherein said inductor and said first and second capacitors comprise a resonant circuit having a resonant frequency chosen to shape said gating signal to improve switching of said first switching means.
This invention relates to a gating circuit for controlling a switching device employed in a deflection circuit of a television receiver.
Various deflection system designs have been utilized in television receivers. One design employing two bidirectional conducting switches and utilizing SCR's (thyristors) as part of the switches is disclosed in U.S. Pat. No. 3,452,244. In this type deflection system, a first SCR is
One type regulator system design alters the amount of energy stored in a commutating capacitor coupled between the first and second SCR's during the commutating interval. A regulator design of this type may employ a regulating SCR and diode for coupling the input reactor to the source of B+. With this type regulator a notch, the width of which depends upon the regulation requirements, is created in the current supplied through the reactor and which notch shows up in the voltage waveform developed on the separate winding or tap of the input reactor which provides the gating voltage for the first SCR. The presence of the notch, even though de-emphasized by a waveshaping circuit coupling the gating voltage to the first SCR, causes erratic control of the first SCR.
SUMMARY OF THE INVENTION
In accordance with one embodiment of the invention, a gating circuit of a television deflection system employing a first switching means for coupling a deflection winding across a source of energy during a trace interval of each deflection cycle and a second switching means for replenishing energy to said source of energy during a commutation interval of each deflection cycle includes a voltage divider means coupled in parallel with the second switching means for developing gating signals proportional to the voltage across the second switching means. The voltage divider means are coupled to the first switching means to provide for conduction of the first switching means in response to the gating signals.
A more detailed description of a preferred embodiment of the invention is given in the following description and accompanying drawing of which:
FIG. 1 is a schematic diagram, partially in block form, of a prior art SCR deflection system;
FIG. 2 is a schematic diagram, partially in block form, of an SCR deflection system of the type shown in FIG. 1 including a gating circuit embodying the invention;
FIG. 3 is a schematic diagram, partially in block form, of one type of a regulator system which employs an SCR as a control device and which is suitable for use with the SCR deflection system of FIG.2;
FIG. 4 is a schematic diagram, partially in block form, of another type of a regulator system suitable for use with the deflection circuit of FIG. 2; and
FIG. 5 is a schematic diagram, partially in block form, of still another type of a regulator system suitable for use with the SCR deflection system of FIG. 2.
DESCRIPTION OF THE PREFERRED EMBODIMENT
FIG. 1 is a schematic diagram, partially in block form, of a prior art deflection system of the retrace driven type similar to that disclosed in U.S. Pat. No. 3,452,244. This system includes a commutating switch 12, comprising a silicon controlled rectifier (SCR) 14 and an oppositely poled damper diode 16. The commutating switch 12 is coupled between a winding 18a of an input choke 18 and ground. The other terminal of winding 18a is coupled to a source of direct current voltage (B+) by means of a regulator network 20 which controls the energy stored in the deflection circuit 10 when the commutating switch is off, during an interval T3 to T0' as shown in curve 21 which is a plot of the voltage level at the anode of SCR 14 during the deflection cycle. A damping network comprising a series combination of a resistor 22 and a capacitor 23 is coupled in parallel with commutating switch 12 and serves to reduce any ringing effects produced by the switching of commutating switch 12. Commutating switch 12 is coupled through a commutating coil 24, a commutating capacitor 25 and a trace switch 26 to ground. Trace switch 26 comprises an SCR 28 and an oppositely poled damper diode 30. An auxiliary capacitor 32 is coupled between the junction of coil 24 and capacitor 25 and ground. A series combination of a horizontal deflection winding 34 and an S-shaping capacitor 36 are coupled in parallel with trace switch 26. Also, a series combination of a primary winding 38a of a horizontal output transformer 38 and a DC blocking capacitor 40 are coupled in parallel with trace switch 26.
A secondary of high voltage winding 38b of transformer 38 produces relatively large amplitude flyback pulses during the retrace interval of each deflection cycle. This interval exists between T1 and T2 of curve 41 which is a plot of the current through windings 34 and 38a during the deflection cycle. These flyback pulses are applied to a high voltage multiplier (not shown) or other suitable means for producing direct current high voltage for use as the ultor voltage of a kinescope (not shown).
An auxiliary winding 38c of transformer 38 is coupled to a high voltage sensing and control circuit 42 which transforms the level of flyback pulses into a pulse width modulated signal. The control circuit 42 is coupled to the regulator network 20.
A horizontal oscillator 44 is coupled to the gate electrode of commutating SCR 14 and produces a pulse during each deflection cycle slightly before the end of the trace interval at T0 of curve 21 to turn on SCR 14 to initiate the commutating interval. The commutating interval occurs between T0 and T3 of curve 21. A resonant waveshaping network 46 comprising a series combination of a capacitor 48 and an inductor 50 coupled between a winding 18b of input choke 18 and the gate electrode of trace SCR 28 and a damping resistor 52 coupled between the junction of capacitor 48 and inductor 50 and ground shapes the signal developed at winding 18b (i.e. voltage waveform 53) to form a gating signal voltage waveform 55 to enable SCR 28 for conduction during the second half of the trace interval occurring between T2 and T1' of curve 41.
The regulator network 20, when of a type to be described in conjunction with FIG. 3, operates in such a manner that current through winding 18a of input choke 18 during an interval between T4 and T5 (region A) of curves 21, 53 and 55 is interrupted for a period of time the duration of which is determined by the signal produced by the high voltage sensing and control circuit 42. During the interruption of current through winding 18a a zero voltage level is developed by winding 18b as shown in interval T4 to T5 of curve 53. The resonant waveshaping circuit 46 produces the shaped waveform 55 which undesirably retains a slump in region A corresponding to the notch A of waveform 53. The slump in waveform 55 applied to SCR 28 occurs in a region where the anode of SCR 28 becomes positive and where SCR 28 must be switched on to maintain a uniform production of the current waveshape in the horizontal deflection winding 34 as shown in curve 41. The less positive amplitude current occurring at region A of waveform 55 may result in insufficient gating current for SCR 28 and may cause erratic performance resulting in an unsatisfactory raster.
FIG. 2 is a schematic diagram, partially in block form, of a deflection system 60 embodying the invention. Those elements which perform the same function in FIG. 2 as in FIG. 1 are labeled with the same reference numerals. FIG. 2 differs from FIG. 1 essentially in that the signal to enable SCR 28 derived from sampling a portion of the voltage across commutating switch 12 rather than a voltage developed by winding 18b which is a function of the voltage across winding 18a of input choke 18 as in FIG. 1. This change eliminates the slump in the enabling signal during the interval T4 to T5 as shown in curve 64 since the voltage across the commutating switch 12 is not adversely effected by the regulator network 20 operation.
A series combination of resistor 22, capacitor 23 and a capacitor 62 is coupled in parallel with commutating switch 12, one terminal of capacitor 62 being coupled to ground. The junction of capacitors 23 and 62 is coupled to the gate electrode of SCR 28 by means of the inductor 50. The resistor 52 is coupled in parallel with capacitor 62.
Capacitors 23 and 62 form a capacitance voltage divider which provides a suitable portion of the voltage across commutating switch 12 for gating SCR 28 via inductor 50. The magnitude of the voltage at the junction of capacitors 23 and 62 is typically 25 to 35 volts. It can, therefore, be seen that the ratio of values of capacitors 23 and 62 will vary depending on the B+ voltage utilized to energize the deflection system. Capacitors 23 and 62 and inductor 50 form a resonant circuit tuned in a manner which provides for peaking of the curve 64 between T4 and T5. This peaking effect further enhances gating of SCR 28 between T4 and T5.
Since the waveshape of the voltage across commutating switch 12 (curve 21) is relatively independent of the type of regulator system employed in conjunction with the deflection system, the curve 64 also is independent of the type of regulator system.
When commutating switch 12 switches off during the interval T3 to T0' curve 21, the voltage across capacitor 62 increases and the voltage at the gate electrode of SCR 28 increases as shown in curve 64. As will be noted, no slump of curve 64 occurs between T3 and T5 because there is no interruption of the voltage across commutating switch 12.
FIG. 3 is a schematic diagram, partially in block form, of one type of a regulator system which may be used in conjunction with the invention. B+ is supplied through a regulator network 20 which comprises an SCR 66 and an oppositely poled diode 68. The diode is poled to provide for conduction of current from B+ to the horizontal deflection circuit 60 via winding 18a of input choke 18. Current flows through the diode during the period T3 to T4 of curve 21 FIG. 1 after which current tries to flow through the SCR 66 from the horizontal deflection circuit to B+ since the commutating capacitor 25 is charged to a voltage higher than B+.
The horizontal deflection circuit 60 produces a flyback pulse in winding 38a of the flyback transformer 38 which is coupled to winding 38c. The magnitude of the pulse on winding 38c determines how long the signal required to switch SCR 66 on is delayed after T4 curve 21 FIG. 1. If the flyback pulse is greater than desirable, the SCR 66 turns on sooner than if the flyback pulse is less than desirable and provides a discharge path for current in commutating capacitor 25 back to the B+ supply. In this manner a relatively constant amplitude flyback pulse is maintained.
FIG. 4 is a schematic diagram, partially in block form, of another well-known type of a regulator system which may be used in conjunction with the invention shown in FIG. 2. B+ is coupled through winding 18a of input choke 18 and through a series combination of windings 70a and 70b of a saturable reactor 70 and a parallel combination of a diode 72 and a resistor 74 to the horizontal deflection circuit 60. Diode 72 is poled to conduct current from the horizontal deflection circuit 60 to B+.
Flyback pulse variations are obtained from winding 38c of the horizontal output transformer 38 and applied to a voltage divider comprising resistors 76, 78 and 80 of the high voltage sensing and control circuit 42. A portion of the pulse produced by winding 38c is selected by the position of the wiper terminal on potentiometer 78 and coupled to the base electrode of a transistor 82 by means of a zener diode 84. The emitter electrode of transistor 82 is grounded and a DC stabilization resistor 85 is coupled in parallel with the base-emitter junction of transistor 82. When the pulse magnitude on winding 38c exceeds a level which results in forward biasing the base-emitter junction of transistor 82, current flows from B+ through a resistor 86, a winding 70c of saturable reactor 70 and transistor 82 to ground. Due to the exponential increase of current in winding 70c during the period of conduction of transistor 82, the duration of conduction of transistor 82 determines the magnitude of current flowing in winding 70c and thus the total inductance of windings 70a and 70b. The current in winding 70c is sustained during the remaining deflection period by means of a diode 88 coupled in parallel with winding 70c and poled not to conduct current from B+ to the collector electrode of transistor 82. A capacitor 90 coupled to the cathode of diode 88 provides a bypass for B+. Windings 70a and 70b are in parallel with input reactor 18a and thereby affect the total input inductance of the deflection circuit and thereby controls the transfer of energy to the deflection circuit. The dotted waveforms shown in conjunction with a curve 21' indicate variations from a nominal waveform provided at the input of horizontal deflection circuit 60 by the windings 70a and 70b.
FIG. 5 is a schematic diagram of yet another type of a regulator system which may be used in conjunction with the invention. B+ is coupled through a winding 92a and a winding 92b of a saturable reactor to the horizontal deflection circuit 60. Windings 92a and 92b are used to replace the input choke 18 shown in FIGS. 1 and 2 while also providing for a regulating function corresponding to that provided by regulating network 20.
Flyback pulse variations are obtained from winding 38c and applied to the high voltage sensing and control circuit 42 as in FIG. 4. Current flows from B+ through resistor 86, a winding 92c and transistor 82 to ground. As in FIG. 4 the duration of the conduction of transistor 82 determines the energy stored in winding 92c and thus the total inductance of windings 92a and 92b which control the amount of energy transferred to the deflection circuit during each horizontal deflection cycle. The variations in waveforms of curve 21', shown in conjunction with FIG. 4, are also provided at the input of horizontal deflection circuit 60 by windings 92a and 92b.
LOEWE F875 SENSOTRONIC CHASSIS F800 (543/490) CONTACTLESS TOUCH SENSOR PROGRAM CHANGE KEYBOARD CIRCUIT ARRANGEMENT FOR ESTABLISHING A CONSTANT POTENTIAL OF THE CHASSIS OF AN ELECTRICAL DEVICE WITH RELATION TO GROUND :
Circuit arrangement for establishing a reference potential of a chassis of an electrical device such as a radio and/or TV receiver, such device being provided with at least one contactless touching switch operating under the AC voltage principle. The device is switched by touching a unipole touching field in a contactless manner so as to establish connection to a grounded network pole. The circuit arrangement includes in combination an electronic blocking switch and a unidirectional rectifier which separates such switch from the network during the blocking phase.
1. A circuit arrangement for establishing, at the chassis of an electrical device powered by a grounded AC supply network, a reference potential with relation to ground, said device having at least one contactless touching switch operating on the AC voltage principle, the switch being operated by touching a unipole touching field in a contactless manner, said arrangement comprising an electronic switch for selectively blocking the circuit of the device from the supply network, a half-wave rectifier including a pair of diodes individually connected in series-aiding relation between the terminals of the supply network and the terminals of the device for separating the electronic blocking switch from the supply network during a blocking phase defined by a prescribed half period of the AC cycle, and a pair of condensers individually connected in parallel with the respective diodes. 2. A circuit arrangement according to claim 1, wherein the capacitances of the two condensers are of equal magnitude.
In electronic devices, for example TV and radio receivers, there are used in ever increasing numbers electronic touching switches for switching and adjusting the functions of the device. In one known embodiment of this type of touching switch, which operates on a DC voltage principle, the function of the electronic device, is contactlessly switched by touching a unipole touching field, the switching being carried out by means of an alternating current voltage. When using such a unipole touching electrode, one takes advantage of the fact that the AC current circuit is generally unipolarly grounded. In order to close the circuit by touching the touching surface via the body of the operator to ground, it is necessary to provide an AC voltage on the touching field. In one special known embodiment there is employed a known bridge current rectifier for the current supply. This type of arrangement has the drawback that the chassis of the device changes its polarity relative to the grounded network pole with the network frequency. With such construction considerable difficulties appear when connecting measuring instruments to the device, such difficulties possibly eventually leading to the destruction of individual parts of the electronic device.
In order to avoid these drawbacks, the present invention provides a normal combination of a unidirectional rectifier with an electronic blocking switch that separates the chassis of the electronic device from the network during the blocking phase. In accordance with the present invention, the polarity of the chassis of the electronic device does not periodically change, because the electronic device is practically separated from the network during the blocking phase of the unidirectional rectifier by means of the electronic blocking switch.
In a further embodiment of the invention a further rectifier is connected in series with the unidirectional rectifier in the connection between the circuit and the negative pole of the chassis. Such further rectifier is preferably a diode which is switched in the transfer direction of the unidirectional rectifier. According to another feature of the invention there are provided condensers, a respective condenser being connected parallel with each of the rectifiers. Preferably the two condensers have equal capacitances. Because of the use of such condensers, which are required because of high frequency reasons, during the blocking phase there is conducted to the chassis of the electronic device an AC voltage proportional to the order of capacitances of the condensers. Thus there is placed upon the touching field in a desired manner an AC voltage, and there is thereby assured a secure functioning of the adjustment of the device when such touching occurs.
In the embodiment of the invention employing two rectifiers there is the further advantage that over a bridging over of the minus conduit of the rectifier that is connected between the network and the negative pole of the chassis connection, no injuries can be caused by a measuring instrument in the electronic device itself and in the circuit arrangement connected thereto.
In the accompanying drawing:
The sole FIGURE of the drawing is a circuit diagram of a preferred embodiment of the invention.
In the illustrated embodiment the current supply part of the device, shown at the left, is connected via connecting terminals A and B to an AC voltage source, the terminal B being grounded at 8. The current supply part consists of a unidirectional rectifier in the form of a diode 1 with its anode connected to the terminal I, the cathode of diode 1 being connected to one input terminal 9 of an electronic device 2. In the device 2 there is also arranged a sensor circuit 3, shown here mainly as a block, circuit 3 being shown as including a pnp input transistor the emitter of which is connected to an output terminal 11 of the device 2. The collector of such transistor is connected to the other output terminal 12 of the device 2. The base of the transistor is connected by a wire 13 to a unipolar touching field 4 which may be in the form of a simple metal plate instead of the pnp transistor shown, the sensor circuit itself may consist of a standard integrating circuit which controls, among other things, the periodic sequential switching during the touching time of the touching field 4. All of the circuits of the electronic device 2 are isolated in a known manner from the chassis potential. Between the network terminal B and the negative pole 10 of the chassis there is arranged in the direction opposite that of diode 1 a further diode 5, the anode of diode 5 being connected to the terminal 10, and the cathode of diode 5 being connected to the terminal B of the current supply. To provide for HF type bridging of the diodes 1 and 5 there are arranged condensers 6 and 7 respectively, which are connected in parallel with such diodes.
The invention functions by reason of the fact that in an AC network separate devices radiate electromagnetic waves which produce freely traveling fields in the body of the person who is operating and/or adjusting the device, thereby producing an alternating current through his body to ground, as indicated by the - line at the right of the circuit diagram. If now the person operating the device touches the switching field 4, then the pnp type input transistor of the sensor circuit 3, which is placed on a definite reference potential (for example 12 Volts) and is connected with the negative halfwave of the AC voltage potential, is made conductive. There is thereby released a control command in the sequential switching, for example, for switching the electronic device to the next receiving channel. It is understood that the most suitable connection is formed between ground and the touching field 4 by means of a wire. By the use of such wires it would be assured that in all cases the base of the transistor in circuit 3 is connected to ground. This would, however, not permit anyone to operate the switch without the use of an auxiliary means such as a wire. It will be assumed that the touching almost always results directly via the almost isolated human body. For this reason the AC current fields are necessary, because otherwise there cannot always be provided a ground contact. Thus this connection is established via the body resistance of the person carrying out the touching of the switch.
The positive half wave of the alternating current travels to the terminal 9 of the electronic device 2 after such current has been rectified and smoothed by the devices 1, 6. Such positive halfwave is also conducted to the sensor circuit 3. The thus formed current circuit is closed by way of the chassis of the electronic device 3, the diode 5, and the terminal B. When there is a negative halfwave of the alternating current delivered by the current supply, both diodes 1 and 5 remain closed so that the chassis of the device 2 remains separated from the network during the blocking phase. Nevertheless, by means of condensers 6 and 7 the chassis is placed in a definite network potential, which depends on the relationship of the order of magnitude of the two condensers 6 and 7. When the capacitances of such condensers are equal, there is placed upon the chassis of the device 2 the constant reference potential, and simultaneously there is present via the sensor circuit 3 the required AC voltage at the touching field 4 for adjusting the function or functions of the device 2 upon the touching of the touching field 4.
The reference character 15 indicates a terminal or point at which the potential of the chassis of the device 2 may be measured. As above explained, the diode 5 causes the potential of the chassis at 15 to be separated from the network ground when a negative AC halfwave arrives. It will be noted that the return conduit of the circuit is held at a fixed chassis potential. The input transistor of the sensor circuit 3 remains, however, locked because it is subjected to a DC current of about 12 volts. If now, by means of touching the touching field 4, the chassis potential is connected to ground, then the transistor switches through and releases a switching function.
If the connecting terminals AB of the current source are exchanged, as by changing the plug, then there is still secured the condition that the chassis of the device is separated from the network ground via the diode, in this case the diode 1. The reference potential of the chassis consequently remains constant and the changing AC fields which are superimposed on the condensers can produce in the touching human body an AC current voltage due to the fields which are radiated by the device.
A suitable sensor which may be employed for the circuit 3 herein may be a sensor known as the "SAS 560 Tastatur IS," manufactured and sold by Siemens AG.
It is to be understood that the present invention is not limited to the illustrated environment. They can also be used in electronic blocking switch including a Thyristor circuit, which in the same manner separates the electronic device during the blocking phase from the network rectifier. With such Thyristor circuit the drawbacks described in the introductory portion of the specification of known circuit arrangements are also avoided.
Although the invention is illustrated and described with reference to a plurality of preferred embodiments thereof, it is to be expressly understood that it is in no way limited to the disclosure of such a plurality of preferred embodiments, but is capable of numerous modifications within the scope of the appended claims.
1. Description of the Prior Art
Voltage multipliers of the type involved in the present invention are normally used in color television receivers to produce the high voltage for the picture tube anode. The known electrical voltage multipliers of this type are normally manually assembled from individual components and are soldered together. Such an arrangement is time consuming and it is not always possible to automate the soldering process in such a method of construction.
2. Field of the Invention
The field of art to which this invention pertains is solid state voltage multipliers and in particular to such voltage multipliers which are formed of an integral arrangement of layer type capacitors end contacted by the Schoop process.
SUMMARY OF THE INVENTION
It is an important feature of the present invention to provide an improved structure for a voltage multiplier.
It is another feature of the present invention to provide a voltage multiplier using a layer type capacitor arrangement.
It is a principle object of the present invention to provide an improved solid state voltage multiplier which is subject to easy production techniques.
It is another object of the present invention to provide a voltage multiplier formed of a layer type capacitor which is manufactured in such a way as to permit a series of diodes used in the multiplier to be readily easily soldered to the capacitor terminals in a single plane.
It is an additional object of the present invention to provide a voltage multiplier as described above wherein the capacitors of the multiplier consist of two layer type capacitors, each being slit in such a way as to form two sets of series connected individual capacitors.
It is also an object of this invention to provide a voltage multiplier as described above wherein the two sets of individual capacitors are arranged in side by side spaced relation.
It is a further object of the present invention to provide a voltage multiplier as described above wherein the two sets of serially connected individual capacitors are stacked one upon the other, and the diodes used in the multiplier are caused to span the body of the capacitor assembly and make electrical contact between the end faces thereof.
These and other features, advantages and objects of the present invention will be understood in greater detail from the following description and the associated drawings wherein reference numerals are utilized as to designate the preferred embodiments.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 2 shows a voltage multiplier in accordance with the present invention wherein two separate capacitor networks are used and arranged in side by side spaced relation.
FIG. 3 shows an arrangement having electrical characteristics similar to FIGS. 1 and 2 but wherein the capacitor networks or sets are stacked upon one another in a spaced saving arrangement.
FIG. 4 is a diagrammatic cross sectional view of the arrangement shown in FIG. 3.
FIG. 5 shows two capacitor networks which are separated by an insulating layer which extends beyond the Schoop layers.
DESCRIPTION OF THE PREFERRED EMBODIMENTS
The present invention relates to an electric voltage multiplier of the type used in television receivers to produce the high anode voltage for the picture tube. Such multipliers usually consist of two rows of serially connected capacitors with a plurality of diodes connecting the capacitor terminals on one row with the like terminals on the other row. The diodes are connected in series to pass a unidirectional pulse and are arranged in such a way as to charge each of the capacitors.
By virtue of the present invention, there is provided a smaller physical design for a voltage multiplier and a physical construction which is easily suited to automation processes.
This is accomplished in the present invention by providing the two rows of serially connected capacitors to be formed from end contacted layer capacitors which are integrally joined together. The metal coatings of each layer of the capacitors have free edges at opposite sides, and the capacitor layers are split by a series of slits which form individual capacitors connected electrically in series. The slits pass from one end face to a point just beyond the capacitor zone of the capacitor. In each case the individual capacitors are connected in series with the other capacitors by an inner coating which is not severed in making the slits.
Capacitor networks of the type described are known from the German text laid open to public inspection No. 1,764,861. It is not possible however to achieve a simple production technique for voltage multipliers by the use of the teachings of this patent. Only by means of the combination of features of the present invention such as using an inner series connection has it been possible to develop an arrangement where all contact surfaces lie on the same end face of the capacitor network. In this way it has been possible to achieve a physical construction which enables the taking advantage of the most automated production techniques. Furthermore the use of capacitor networks which are based upon the principle of layer capacitors provides the added advantage that it is possible to use silicon diodes which are considerably smaller than the selenium rectifiers heretofore used.
In one arrangement of the invention, the two capacitor networks are arranged next to one another and the diodes are arranged substantially parallel to one another except for one of the diodes at the end of the circuit. This enhances the possibility of using automating processes. For the same reason, it is advisable for the contact surfaces to lie in a single plane which is possible according to the present invention.
A particularly space saving embodiment of the invention is provided in which the two capacitor networks are arranged one above another in a stacked manner and are separated from one another by an insulating intermediate layer which may extend beyond the Schoop layer as shown in FIG. 5.
A construction which is particularly advantageous is one in which the two capacitor networks are congruent and the diodes are arranged on one side of the capacitor network and welded to the Schoop layers. In this case the diodes are arranged perpendicularly to the end contact layers.
A simple process for the production of voltage multipliers according to the present invention is such that the capacitor networks are arranged in side by side spaced relation and the contact surfaces are generally in a single plane. In this arrangement the diodes are arranged in a direction perpendicular to the longitudinal direction of the foils in the capacitor, and the diodes span the end faces of the structure. A diode arranged at the beginning or end of the unit can be positioned at an angle, while the other diodes are arranged in a generally parallel orientation.
The preferred embodiment of the invention which is particularly small designed voltage multipliers consist of stacking the two layer capacitors upon each other and separating them by an insulating layer which does not project over the ends. The slits which are then formed on the capacitor layers, divide the capacitors into individual elements, and the diodes are placed in position on the top of the arrangement in such a way that the terminals of the diodes connect the end faces of the capacitor layers.
Referring to the drawings for greater detail FIG. 1 shows a schematic of a voltage multiplier according to the present invention in which capacitors 1 form one serially connected set and capacitors 2 form a second serially connected set. The capacitor network 4 of FIG. 2 is formed from the capacitors 1 of FIG. 1. The capacitor network 5 of FIG. 2 is formed in accordance with the invention from the capacitors 2 of FIG. 1. In FIG. 1 the diodes 3 are connected from the terminals of the capacitors 1 to the terminals of capacitors 2 as shown to produce a series diode arrangement which charges each of the capacitors as is well understood in the art of voltage multipliers.
The contact surface 6 (FIG. 2) is grounded, while the contact surface 7 is connected to the pulse input. Contact surface 8 is the tap for the high voltage output and the contact surface 9 serves to contact the diodes to two of the capacitors of one capacitor network and the contact surface 22 serves to contact the last capacitor of the capacitor network 5 to two of the diodes.
As in FIG. 2, the diodes 3 are placed on the contact surfaces 7, 8, 22, and 9 and are electrically connected thereto by spot welding. Slots 10 are provided and filled with synthetic material in the course of encasing the arrangement so that they possess the requisite dielectric strength. During operation these slots are connected at least temporarily with the full voltage of a capacitor in the case of television, for example, the voltage across one of these slots may be 8.5 K.V.
In FIG. 3, an arrangement is shown where the capacitor networks 4 and 5 are congruent and are stacked one above the other. In this case, the outer flanks 11 of the capacitor networks 5 are not used. The two end faces of the capacitor networks 4 and 5 are entirely covered with Schoop layers. The diodes 3 are arranged generally in a parallel layout with respect to the slots 10. The diodes connect the opposite contact surfaces 13 and 14 of the Schoop layers.
As illustrated in FIG. 4, the contact surfaces 13 and 14 contact only the corresponding metal coatings 15 and 17 of one of the capacitor networks 4 and 5. The coating 16 of the capacitor network 4 contacts neither of the two Schoop layers 13 and 14. However, these coatings extend beyond the slot depth 19, so that after the completion of the capacitor network, electrically conductive arms remain outside the slots and are integrally joined to the blind coatings. Accordingly, the blind coatings 18 of the capacitor 5 project beyond the slot depth 20. Thus, it is only possible to contact such a capacitor network on one end side.
The diodes 3 can thus only be connected by welding their two terminals to in each case one contact surface 13 and 14 in accordance with FIG. 1. This construction also must be sealed in order to achieve the required dielectric strength.
The intermediate layer 21 does not project beyond the end faces of the capacitor network 4 and 5 and is covered by the Schoop layer. It prevents sparkovers in the region of the cut edges and breakdowns through the dielectric, since, particularly in the use of a multiple inner series connection, the dielectric does not possess a dielectric strength sufficient to support the voltage of the overall capacitor.
In FIG. 5 the two capacitor networks 4 and 5 are separated from each other by an insulating intermediate layer 21 which extends beyond the Schoop layers 13 and 14. The Schoop layers 13 and 14 are interrupted by this intermediate layer and the upper and lower part of the Schoop layers 13 and 14 can always be contacted and wired separately.
LOEWE F875 SENSOTRONIC CHASSIS F800 (543/490) ELECTRICAL COMPONENT PROTECTED AGAINST HIGH TENSION, PARTICULARLY FOR COLOR TELEVISION RECEIVERS AND METHOD OF ITS PRODUCTION:Voltage multipliers of the type involved normally used in color television receivers to produce the high voltage for the picture tube anode:
A grid-shaped electrical component is formed by molding a plurality of electrically interconnected capacitors and diodes, physically forming a grid structure, within a synthetic casting resin. The electrical components form the cores of the struts of the grid structure and each strut is provided with a plurality of indentations in the synthetic resin in the area of the walls of the components.
rallel to said one plane, the components extending in the oblique direction disposed to lie in one of said planes, said components comprising a plurality of capacitors disposed in the struts which extend in one grid direction and a plurality of rectifiers which are disposed in said struts which extend in the other grid direction and in the oblique grid direction, said capacitors forming serially connected rows of capacitors and said rectifiers connected to the ends of said capacitors, a first group of conductors disposed at one end of said grid structure and having the individual conductors thereof connected to the junctions of said electrical components and each including a portion extending substantially parallel to the respective capacitors and a portion extending through said grid structure generally perpendicular to the first-mentioned portions, and a second group of electrical conductors at the other end of said grid structure connected to the junctions of said electrical components thereat, said grid structure further including a base having mounting shoulders formed thereon, said grid structure together with said base and said conductors forming a self-supporting structure for permitting the passage of cooling air between said struts.
1. Field of the Invention
This invention relates to a grid-shaped electrical component having grid struts extending substantially in two main grid directions, and more particularly to grid-shaped components which are protected against high tension, particularly for color television receivers.
2. Description of the Prior Art
Heretofore, the prior art recognized a conventional expedient to provide a color television receiver with a line transformer to generate the high voltage required for the picture tube. This line transformer was advantageously utilized to provide high voltage pulses which were then rectified prior to being employed at the picture tube. Construction of the line transformer for protection against high voltages, as well as the associated rectifier arrangement, has proven difficult in situations wherein the transformer was required to deliver the full value of the required high voltage. This is particularly true in the case of color television receivers since a direct current voltage of approximately 25kv. is ordinarily required for proper operation of the picture tube.
In view of the foregoing it is therefore advisable to design the line transformer for a lower voltage and to generate a direct current voltage of the required magnitude by utilization of a multiplier cascaded with the line transformer. For example, it is significant that the line transformer delivers recoil pulses with an amplitude of 8.5kv., from which a DC voltage of 25kv. can be obtained in a multiplier cascade having 5 silenium rectifiers and four or five capacitors.
In electrical equipment technology, however, there is always the problem, as here in a multiplier cascade, to design components free from brush discharge and protected against high tension, so that neither adjacent components nor the operating personnel can be harmed. To this end, it is generally known in the art to combine such units which are exposed to high tension into a single component which can be built in or exchanged as a one unit component both during original manufacture of the apparatus and in case of maintenance or repair.
Arrangements of the type initially mentioned avoid a majority of the inconvenience experienced heretofore by components manufactured to meet the above conditions. Thus, for example, the corresponding components are frequently cast with plastic into one compact block. However, in addition to the problem of an effective heat dissipation there is the further difficulty that the wall strength of the grouting mass between the components is not sufficiently constant, or that the metallic connecting elements of the components or the wiring may appear on the surface and cause glow defects or flash-overs.
To avoid the latter drawbacks it is also an old expedient in the art to fixedly arrange the components of the unit on a base plate and to cast the entire unit in a beaker consisting of a material which is combined with the grouting mass. However, these solutions have the drawback that casting in a beaker is relatively expensive and that as a result of these steps the problem of cooling the component parts is rendered more difficult.
Although the component parts of the type initially mentioned have great advantages from the electrical point of view as well as with regard to the manufacturing cost, there is also an additional difficulty as a result of the size of the components thus produced. In many cases, particularly in the interior of electrical apparatus such as, for example, color television sets, the measurements of which one seeks to reduce through the use of integrated switching circuits, it is frequently undesirable or even quite impossible to accommodate the component parts.
Hence, it is an object of the invention to produce an electrical component protected against high tension which has the advantages of a component of a type mentioned initially, but which has smaller dimensions than heretofore known in employing the same individual components.
SUMMARY OF THE INVENTION
As shown in relatively long experiments, it is very important for an effective operation of the apparatus that the wall strength of the synthetic resin enveloping the component parts is substantially equally large throughout the entire apparatus. It has namely been shown that the important point is that the thickness of the layer of synthetic resin varies only relatively little throughout the apparatus, and in all respects the variation is not excessive, since otherwise there may be the danger that the sealing layer will crack upon cooling. The danger of a crack formation may also be responsible for the fact that with the component parts known heretofore and case in the form of a block with or without a beaker, only adverse results were obtained since the plastics suitable for the electrical components with respect to high tension do not lend themselves to casting into blocks.
A new type of apparatus which is protected against high tension in accordance with the principles of the present invention is particularly suitable for a high tension multiplier cascade in color television sets of the type mentioned initially which contains a plurality of rectifier elements and capacitors. A further development of the invention provides that the capacitors lie parallel one behind the other in an upper plane, that in the second plane lying therebeneath a portion of the rectifier elements is spaced at a distance from the other plane and lie perpendicular to the capacitors, and that in a diagonal line of each rectangle defined by two capacitors and two rectifiers there is arranged an additional rectifier in the second lower plane. For the electrical connection of the component part with the elements of a corresponding circuit, it is provided that electrical conductors are guided outwardly from adjacent ends of the rows of capacitors, the conductors first extending a distance within the synthetic casting resin substantially parallel to the capacitors at one end of the apparatus, and at the opposite end of the apparatus the electrical conductors extend from the rows of capacitors whose ends are connected with a rectifier element and guided outwardly therefrom through the envelope of synthetic resin.
BRIEF DESCRIPTION OF THE DRAWINGS
Other objects, features and advantages of the invention will become apparent from the following detailed description taken in conjunction with the accompanying drawings, in which:
FIG. 1 is a plan view, partially in section, illustrating apparatus constructed according to the invention; and
FIG. 2 is a sectional view taken along the line II--II of FIG. 1.
DESCRIPTION OF THE PREFERRED EMBODIMENT
FIG. 1 illustrates a high tension cascade constructed in accordance with the principles of the present invention to produce the accelerating voltage required for color television tubes amounting to approximately 25 kv. consisting of five silenium rectifiers 21, 31 and four capacitors 22, fifth capacitor not cast within the component part must be separately connected thereto. Therefore, this capacitor is not shown in the drawing. The reference numerals 25, 26 and 27 designate the electrical conductors which are guided outwardly through the synthetic resin 23. The electrical conductors 25 and 26 on the base side of the component part are first pulled a distance upwardly in the synthetic resin casing so that when they exit the component part (see FIG. 2) the conductors are already sufficiently separated from the base side which is generally connected with the base frame. The indentations 24 in the synthetic resin compound are shown somewhat exaggerated schematically, since the indentations generally end, at least partly, on the wall portions of the component parts which are protected against high tension. The indentations 24 are produced during the manufacture of the cascade in accordance with the invention, wherein the interconnected electrical components which form a loose grid are placed in a mold consisting of a material, such as polyethylene or polypropylene, which does not combine with the casting resin, for example, epoxy resin. Within this mold there are disposed spacing blocks which are preferably formed directly on the mold and which are arranged and dimensioned such that they assure a sufficient space between live metal parts of the electrical components and the inner wall of the mold when the loose grid is installed. Owing to the tolerances required for a simple insertion of the grid of component parts, not all component parts will rest against all spacing blocks provided for the centering thereof so that not all of the indentations 24 in the synthetic resin compound 23 reach as far as the high tension resistant wall of the component part accommodated within the corresponding strut. The fact is illustrated schematically in the drawing where some of the indentations do not reach quite as far as the component parts.
As apparent from FIG. 2, a plastic strip 28 is provided on the narrow side (base side) of the component part remote from the high tension conductor 27, which interconnects the parallel rows of capacitors and wherein recesses 29 are provided by means of which the component part can be screwed onto the apparatus or attached in a different manner.
When employed in television sets, particularly color television sets, component parts in accordance with the invention may be built into advantage in horizontal position in so-called knapsacks, where as a result of their grid-shaped construction, they not only do not impede the air circulation caused by heating of the other component parts of the set, but are at the same time effectively cooled as a result of the air currents flowing therethrough.
Changes and modifications may be made of the invention within the scope and spirit of the appended claims which define what is believed to be new and desired to have protected by Letters Patent.
1. A voltage multiplier comprising a plurality of series connected diodes and a plurality of capacitors, each capacitor being connected between opposite ends of a pair of said diodes and including an intermediate thermoplastic dielectric layer and a pair of metal foil electrode layers on opposite sides of said dielectric layer, said electrode and dielectric layers being rolled into a plurality of overlapping layers including said plurality of capacitors within a common roll, said diodes being connected to said electrode layers and being disposed about opposite outer sides of said roll. 2. The device of claim 1 including means applying a.c. and d.c. voltages to different respective groups of said capacitors. 3. The device of claim 1 wherein said plurality of diodes surround the sides and one end of said roll, the other end of said roll having external connections thereto, and a common thermoplastic cover encapsulating said diodes and capacitors, said external connections extending from one end of said cover. 4. The device of claim 1 including means applying a.c. and d.c. voltages to different respective groups of said capacitors within said common roll, said a.c. voltage capacitors being within the inner layers and said d.c. voltage capacitors being within the outer layers. 5. The device of claim 1 wherein said roll has external connections to the two ends thereof. 6. The device of claim 1 wherein said thermoplastic dielectric layer includes two outer layers of polystyrene and a layer of polyester therebetween. 7. The device of claim 1 wherein one electrode layer is common to two capacitors within said roll. 8. The device of claim 6 wherein said electrode layers are of aluminum.
1. Field of the Invention
The present invention relates to a voltage multiplier arrangement with diodes and capacitors wherein at least two capacitors are combined into a unit.
2. Description of the Prior Art
Voltage multiplier arrangements serve to produce high voltages and are particularly useful for the operation of television picture tubes. The diodes and capacitors are connected in separate series paths with a capacitor in parallel with two series diodes and are generally embedded in plastic for voltage protection.
In one known arrangement, the diodes and capacitors are disposed in a lattice configuration to keep the volume to a minimum. At each long side, two capacitors are arranged one behind another in the longitudinal direction; a diode is located at each short side and in the middle therebetween, and an additional diode is disposed in each diagonal direction. Thus, the length of this arrangement is essentially determined by the length of the capacitors, while the width is determined by the length of the diodes.
In another known arrangement, in order to meet the requirement for optimum utilization of the space available and for technical simplification, the ac-voltage-operated capacitors are connected in series and potted to form a unit, and the same is done with the dc-voltage-operated capacitors.
SUMMARY OF THE INVENTION
The primary object of the present invention is to provide a simplified voltage multiplier that occupies less space and insures the necessary voltage protection.
According to the invention a plurality of capacitors are combined in a unit which forms a capacitor roll. The space occupied by the potted elements is reduced by at least one-half that of known arrangements.
According to one feature of the invention, the capacitors are combined into two separate capacitor rolls and the diodes are disposed between said rolls.
In a variation of the invention, all capacitors are combined into one capacitor roll and surrounded by diodes on three sides.
The invention will now be explained in further detail with reference to the accompanying drawings.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a circuit diagram of a voltage tripler using the present novel arrangement;
FIG. 2 schematically shows the structure of the arrangement of FIG. 1;
FIG. 2a shows the actual physical arrangement of the elements of FIG. 2;
FIG. 3 is a circuit diagram of a voltage doubler as in the present invention;
FIG. 4 schematically shows the structure of the arrangement of FIG. 3;
FIG. 4a shows the physical arrangement of the elements of FIG. 4; and
FIG. 5 is a section through a portion of a laminated structure of a capacitor roll.
DESCRIPTION OF THE PREFERRED EMBODIMENT
In FIGS. 1 and 2, the diodes of the voltage tripler are designated by the reference numerals 1, 2, 3, 4, and 5, and the reference numerals 6 and 7 denote ac-voltage-operated capacitors which come into operation only during charging, while 8, 9, and 10 are dc-voltage-operated capacitors. The tripler is constructed in the form of a five-stage cascade circuit, the capacitor of the first stage, formed by the diode 1 and the capacitor 8, being grounded, and is used, for example, to generate the high voltage for operating color picture tubes. The capacitors 6 and 7 of the ac-voltage portion of the circuit are combined into a single capacitor roll 11, and the capacitors 8, 9, and 10 of the dc-voltage portion are combined in another capacitor roll 12. As shown in FIGS. 2 and 2a, the diodes 1, 2, 3, 4, and 5 are disposed centrally between capacitor rolls 11 and 12. The arrangement is potted in a case 13 filled with plastic. The ac-voltage input terminal 14, the dc-voltage output terminal 15, the ground terminal 16, and an additional terminal 17 are brought out on one side of the case 13. In addition, a terminal 18 is provided at a diode 19 connected to the ac-voltage input terminal 14.
The high voltages necessary for operating the black-and-white picture tube in a television receiver are generated with a voltage doubler. In FIGS. 3, 4, and 4a, the diodes are designated by the reference numerals 20, 21, and 22, while 23 denotes an ac-voltage-operated capacitor, and 24 and 25 the dc-voltage-operated capacitors of a three-stage cascade circuit. The alternating voltage to be doubled is applied to the terminal 28, and the dc voltage to be doubled appears at the terminal 29. The terminal 30 represents the ground terminal, to which is connected the capacitor of the first stage of the cascade, formed by the diode 20 and the capacitor 24. Both the ac-voltage-operated capacitor 23 and the dc-voltage-operated capacitors 24 and 25, are all combined into one capacitor roll 26. As shown in FIGS. 4 and 4a, the capacitor roll 26 is disposed centrally and surrounded by diodes 20, 21, and 22 on three sides. The arrangement is potted in a case 27 filled with plastic. The terminals 28, 29, 30 are brought out on the fourth, free side. The capacitor 25, shown with a broken line in FIG. 3, is dispensed with in the arrangement of FIG. 4. This is primarily replaced by the self-capacitance of the picture tube when the multiplier arrangement is connected into the television receiver.
The individual capacitors used in the present multiplier arrangements are designed as high-voltage capacitors having a laminated dielectric of polystyrene and polyester, and metal foil electrode plates. As shown in FIG. 5, the plates of a capacitor roll of this type are preferably aluminum foils 31, 32. Disposed between these foils are two polystyrene foils 33 with an intermediate polyester foil 34. The various terminals may be brought out at one or the other end of the roll or at both ends. At least two capacitor rolls of this laminated structure are wound over one another so that at least one multiple capacitor roll is obtained.
Since ac-voltage- and dc-voltage-operated capacitors may be combined into one capacitor roll, as shown in the embodiments of FIGS. 3 and 4, and as can also be done in the embodiments of FIGS. 1 and 2, the ac-voltage-operated capacitors are advantageously located inside within the roll. With such an arrangement, the corona discharge at the surface of the plastic-filled case 13 or 27 is greatly reduced.
In addition, the various possibilities of combining the capacitors, in conjunction with the diodes, make it possible to achieve particular input capacitances for voltage multipliers. For example, the input capacitance of the voltage multiplier arrangement is increased if at least one dc-voltage-operated and at least one ac-voltage-operated capacitor are combined into one roll.