BRIONVEGA COSMO 2 TVC 26 CHASSIS 509 INTERNAL VIEW.

BRIONVEGA COSMO2 TVC 26. Internal Chassis and Tube viewing.
Developed with philips components.














































































Supply, Line + EHT and Frame deflection + E/W Correction, Frequency Synthesizer Tuning System (Without uC) like Philips TRD, Signal Section Parts.

 

 

 

BRIONVEGA COSMO 2 TVC 26. Chassis detailed parts:
Front Local setup Commands keyboard, RGB Matrix (TDA2530) and RGB Amplifiers,
Luminance and Chrominance (TDA2560/3) (TDA2522), IF sound + Sound Amplifier,
Twin tuner (VHF + UHF) with prescaler unit, FRAME Deflection + E/W Correction in one Module, CRT Socket for PHILIPS A66-540X.


The BRIONVEGA CHASSIS 509.01 is developed and organized on 2 main sections and a central fitted power supply unit.


Right side Panel Line + EHT and Frame deflection + E/W Correction,

Left side panel Signal Section Parts.


SYNCHRONIZATION UNIT: 509.01.3561

FRAME DEFLECTION AND E/W CORRECTION UNIT:509.01.3701

SOUND I.F. + AMPL UNIT:509.01. (TDA4290 - TBA120U).

RGB AMPLIFIER UNIT:509.01.3941


The BRIONVEGA COSMO 2 TVC 26 CHASSIS 509 POWER SUPPLY Control UNIT is developed around the PHILIPS TDA2640.
"Television Switched-Mode Power Supply Using the TDA2640", Mullard Technical Communications, L. M. White, pp. 258-279, Jul. 1975.

A switched-mode power supply provided with a control stage and a switching stage coupled by means of a transformer. The collector of an additional transistor is connected to the transformer. In this manner the ratio of the collector current to the base current of the switching transistor can assume a predetermined value, for example a constant value whatever the value of the mains voltage applied to the power supply.

What is claimed is:

1. A control circuit for a switched-mode power supply, said power supply comprising a non-regulated rectified DC voltage source, a driver transistor, a first transformer having primary and secondary windings, an end of said primary being coupled to the collector-emitter path of said driver transistor, a switching transistor having a base coupled to said secondary, a second transformer having a primary winding coupled in series with said switching transistor, and a plurality of secondary windings, said control circuit comprising a first additional transistor having a collector coupled to the remaining end of the primary winding of the first transformer not connected to the driver-transistor and an emitter coupled to the non-regulated direct voltage source.

2. A control circuit as claimed in claim 1, further comprising a constant voltage source coupled to the base of the additional transistor.

3. A control circuit as claimed in claim 1, further comprising a constant current source, and a resistor coupled between the emitter of the additional transistor and the constant current source.

4. A control circuit as claimed in claim 3, wherein the constant current source comprises a second additional transistor, the two additional transistors being of complementary conductivity and their emitters being connected with each other through said resistor, the collector of the second additional transistor being coupled to the non-regulated rectified direct voltage source and the collector of the first additional transistor being coupled to the end of the primary winding of the first transformer not connected to the driver transistor.

5. A control circuit as claimed in claim 4, further comprising a resistor coupled in series with the collector circuit of said second additional transistor and the non-regulated rectified direct voltage source.

6. A control circuit as claimed in claim 5, further comprising a zener diode coupled between the base of the second additional transistor and the non-regulated voltage source.

7. A control circuit as claimed in claim 6, further comprising a resistance bridge coupled to the base of the first additional transistor and arranged between the two electrodes of the zener diode.

8. A control circuit as claimed in claim 7, wherein the driver transistor and the switching transistor do not conduct simultaneously, and the voltage between the two electrodes of the zener diode as well as the values of the resistors arranged between the said electrodes and of the resistor arranged between the emitters of the two additional transistors are chosen so that the first additional transistor is in the saturated state at the lowest value of the non-regulated voltage while it operates in the linear state at a higher value of said non-regulated voltage.

Description:

The present invention relates to a control circuit for a switched-mode power supply, particularly in a television receiver, said power supply comprising a rectified, non-regulated rectified DC voltage source, a driver transistor whose collector-emitter path is arranged in series with a primary winding of a first transformer, a secondary winding of the latter being coupled to the base of a switching transistor which is arranged in series with a primary winding of a second transformer having a plurality of secondary windings.
This type of switched-mode power supply is used more and more because of the numerous advantages it presents as regards energy efficiency, reliability, compactness, etc. However, as for the majority of the other types of power supplies, its operation on mains supplies of different voltages imposes the use of either a transformer with taps or switch-over from full wave rectification at the highest mains voltage to a voltage doubler rectification for the lowest mains voltage.

It is known that the specific qualities of a switched-mode power supply depend for a large part on the switching speed of the switching transistor at the moment at which the latter passes periodically from the conductive state to the blocking state; this speed is at its maximum when the switching transistor presents, at the turn-off moment, a certain ratio between the collector current and the base current IC/IB: if this ratio is too low, the delay in the recombination of the charges stored in the base increases the switching time; if it is too high there is the risk that the transistor is brought out of saturation before it is blocked, which results in its substantially immediate destruction. For the known switched-mode power supplies it is not possible to maintain a suitable IC/IB ratio in the presence of large variations of the non-regulated rectified DC voltage which result from the connection to the nominal mains voltages of, for example, 110 or 220 V; actually, if the variations in IB are substantially proportional to the variations in the non-regulated voltage, the same does not happen for those of the IC whose amplitude is less.
However, the importance of having a power supply which can operate without any switching on mains supplies of 110 or 220 V is evident: for the manufacturer it is cheaper to produce and the reliability is increased; while the user does not run the risk of incorrect manipulations, particularly when the power supply is destined for use in portable television sets.
One of the objects of the invention is to realize a control circuit which permits the switched-mode power supply to operate without switching in conditions which are substantially optimum and in the presence of mains voltage variations in the range of 90 to 250 Volts.
A further object of the invention is to ensure that said IC/IB ratio of the switching transistor has a predetermined and, more particularly a constant value at the turn-off moment whatever the value of the mains voltage applied to the power supply.
The control circuit according to the invention is characterized in that the end of the primary winding of the first transformer not connected to the driver transistor is connected to the collector of an additional transistor whose emitter is coupled with the non-regulated direct voltage source. Advantageously it is characterized in that the emitter of the additional transistor is connected to one end of a resistor, the other end of this resistor being connected to a constant current source, and that the constant current source is constituted by a second additional transistor, the two additional transistors being of complementary conductivity and their emitters being connected with each other through a resistor, whilst the collector of the second additional transistor is connected to one of the poles of the non-regulated rectified direct voltage source and the collector of the first additional transistor is connected to the end of the primary winding of the first transformer not connected to the driver transistor.
Whilst combining the action of a ballast transistor with that of a variable current generator, the circuit according to the invention thus maintains automatically a desired IC/IB ratio of the switching transistor whatever the value of the mains voltage applied to the power supply.

BRIONVEGA COSMO 2 TVC 26 CHASSIS 509 Digital phase locked loop tuning system / PLL FREQUENCY SYNTHESIZER:

A phase locked loop circuit for use in an automatic frequency synthesizing system. The system includes a programmer circuit which is responsive to a channel number input signal and generates a first digital control signal which is representative of the selected channel number and a second digital control signal which is representative of a predetermined group of channel numbers. A programmable divider is controlled by the programming circuit and generates a digital output signal which causes the phase locked loop circuit to generate a desired system output frequency corresponding to the selected channel number input signal. The phase locked loop circuit includes automatic fine tuning and manual fine tuning features.


1. A digital phase locked loop tuning system responsive to a local oscillator signal for producing a frequency synthesized digital output signal which is utilized to control the frequency of the local oscillator, the local oscillator having a plurality of frequencies associated therewith corresponding, respectively, to a plurality of selectable channels, each of the channels being allocated to one of at least two channel groups with each channel in a particular channel group being separated from an adjacent channel in the particular channel group by a predetermined frequency spacing of the local oscillator, comprising:

programming means responsive to an input signal representing a selected channel number of a particular channel group for generating a first digital control signal having a value corresponding to the selected channel number and for generating a second digital control signal representative of said particular channel group, said second digital control signal being a constant predetermined value for all of said channel numbers that are within said group; and
programmable divider means coupled to said programming means being responsive to said first, second digital control signals and the local oscillator signal, in a local oscillator mode, for generating the digital output signal which is representative of a desired frequency corresponding to said selected channel number, said programmable divider means including means for dividing the local oscillator signal by first and second factors, said first factor being related to the frequency separation between local oscillator signals by an integral number, the local oscillator signal being divided by said first factor during a first interval for a first number of periods of the output signal and being divided by said second factor for a second number of periods of the output signal, said first number of periods being related to the number of the channel selected, said second number being related to the channel group within which the selected channel lies.
2. Phase locked loop system according to claim 1, wherein said programming means including means coupled to said programming means for receiving an MFT signal and being responsive to said MFT signal for altering said first and second digital control signals, and said programmable divider means being responsive to said altered digital control signals for generating an altered system output frequency. 3. Phase locked loop system according to claim 2, wherein said programming means includes first terminal means coupled to said programming means for receiving an AFT control signal, and first logic means responsive to the input signal and the AFT control signal for generating the first digital control signal. 4. Phase locked loop system according to claim 3, wherein said programming means includes second logic means coupled to said first logic means and responsive to the AFT control signal for generating the second digital control signal. 5. Phase locked loop system according to claim 4, wherein said second logic means includes group decoder means coupled to said first logic means. 6. Phase locked loop circuit means according to claim 5, wherein said second logic means includes memory means coupled to said group decoder means and to said first terminal means. 7. Phase locked loop system according to claim 6, wherein said second logic means includes second terminal means for receiving an MFT signal, and up/down counter latch means coupled to said memory means and to said second terminal means for altering said first and second digital control signals in response to said MFT signal. 8. Phase locked loop system according to claim 7, wherein said second logic means includes adder means coupled to said up/down counter latch means to said memory means. 9. Phase locked loop system according to claim 3, wherein said first logic means includes channel number generator means coupled to said first terminal means and responsive to said input signal. 10. Phase locked loop system according to claim 9, wherein said channel number generator means includes first and second data selector means coupled to said first terminal means, and adder means coupled to said second data selector means and to said up/down counter latch means. 11. Phase locked loop system according to claim 1, wherein said means for dividing the local oscillator signal includes programmable counter means for generating a modulus control output signal, and variable modulus prescaler divider means coupled to and responsive to said programmable counter means, said variable modulus prescaler divider means dividing the local oscillator signal by said first and second factors. 12. Phase locked loop system according to claim 11, wherein said programmable counter means includes third data selector means coupled to receive said first and second digital control signals and said modulus control signal. 13. Phase locked loop system according to claim 12, wherein said programmable counter means includes a programmable counter coupled to said third data selector means and to said variable modulus prescaler divider means. 14. Phase locked loop system according to claim 13, wherein said programmable counter means includes look ahead circuit means coupled to said programmable counter, and divide by two circuit means coupled to said look ahead circuit means for generating said modulus control output signal. 15. Phase locked loop tuning system according to claim 1 including digital automatic fine tuning (AFT) means wherein:
said programmable divider means includes switching means responsive to an AFT control signal to inhibit the local oscillator signal to said programmable divider means and to provide an input signal thereto of a different frequency than the local oscillator signal; and
said programming means including logic means responsive to said AFT control signal for altering said first and second digital control signals to predetermined values to cause the phase locked loop tuning system to be operable in an automatic fine tuning mode.
16. Phase locked loop tuning system of claim 15 wherein said programmable divider means includes:
programmable counter means for generating first and second modulus control signals; and
dual modulus prescaler means responsive to said first modulus control signal for dividing the local oscillator signal in said local oscillator mode and said input signal of a different frequency in said automatic fine tuning mode by said first factor which is equal to the integer six and being responsive to said second modulus control signal for dividing said local oscillator signal and said input signal of a different frequency by said second factor which is equal to the integer five respectively.
17. Phase locked loop tuning system of claim 16 wherein said signal of a different frequency is an intermediate frequency signal provided by the tuning system and supplied to said switching means.

18. In a phase locked loop tuning system for receiving a channel number input signal and a local oscillator signal having groups of selectable frequencies wherein the frequency spacing between each adjacent local oscillator frequency within a single group is uniform, the improvement comprising programmable divider means for generating a digital output signal representative of a desired tuning system output frequency including variable modulus prescaler divider means having a prescaler division ratio being equal to P = S/Y' for dividing the local oscillator frequency by said prescaler division ratio during a first interval for a first number of periods of the digital output signal and for dividing the local oscillator frequency by a second prescaler division ratio during a second interval for a second number of periods, said second ratio being related to said first ratio, where S is the frequency spacing between each adjacent local oscillator frequency within a single group (i), Y_i =D_i -X_i S, where D_i is said desired tuning system output frequency within said selected group; X_i =D_i /S rounded off to the nearest integer; Y' is chosen such that Y_i /Y' is an integer and S/Y' is an integer and Y' is the smallest value of all values of Y_i. 19. In a receiver including a tuning apparatus for providing a plurality of local oscillator signals each corresponding to a respective one of a plurality of selectable channels, each of the channels being allocated to one of at least two channel groups wherein each channel is separated from an adjacent channel in the respective channel group by a predetermined frequency spacing, a phase locked loop tuning system for producing a frequency synthesized output signal for controlling the frequency of the local oscillator, comprising:
variable modulus divider means for selectively dividing the frequency of the local oscillator signal by first and second factors in response to a modulus control signal to provide an output signal, said first factor being related to the frequency separation between local oscillator signals by an integral number; and
programmable means for generating said modulus control signal to cause said variable modulus divider means to divide by said first factor during a first interval for a first number of periods of said output signal and to divide by said second factor during a second interval for a second number of periods of said output signal, said first number of periods being related to the number of the channel selected, said second number of periods being related to the channel group corresponding to the selected channel.
20. The phase locked loop tuning system of claim 19 wherein said programmable means includes:
programming means responsive to a selected channel input signal for producing first and second digital output signals, said first digital output signal being related to the selected channel number plus one of two constant values which are determined in accordance within which channel group the selected channel input signal lies, said second digital signal being a constant value for all selected channels within a channel group; and
programmable divider means responsive to said first and second digital output signals from said programming means for providing said variable modulus control signal and the frequency synthesized output signal.
21. The phase locked loop tuning system of claim 20 wherein said programming means includes automatic fine tuning (AFT) means responsive to a AFT control signal being applied thereto when the receiver is placed in an AFT mode wherein:
said variable modulus divider means is caused to receive a input signal different from the local oscillator signal;
said programming means being responsive to the AFT control signal for altering said first and second digital signals such that the receiver is finely tuned to the frequency of the received signal applied to the receiver.
22. The phase locked loop tuning system of claim 21 wherein said programming means includes means for receiving a manual fine tuning (MFT) signal for altering said first and second digital output signals, and said programmable divider means being responsive to said altered digital control signals for generating an altered output signal. 23. The phase locked loop tuning system of claim 19 wherein the one of said first and second factors is an even number and the other is an odd number. 24. The phase locked loop tuning system of claim 23 wherein said first factor is the integer six and said second factor is the integer five.

Description:

BACKGROUND OF THE INVENTION
This invention relates to digital tuning systems, and more particularly, to a simplified digital phase locked loop (PLL) tuning system incorporating unique digital automatic fine tuning and manual fine tuning schemes.
Since the appearance of varactor tuners for television, many tuning address schemes have evolved for controlling them. PLL techniques have maintained a performance advantage but have suffered a cost disadvantage due to complexity, the high frequencies involved, the need for automatic fine tuning and in some localities, the need for a manual fine tuning arrangement. With the advances that have taken place in semiconductor technology in the last several years, the high operating frequencies no longer present a significant problem.
Prior art PLL systems for use in television tuners have not yet been able to incorporate an automatic fine tuning feature, nor have they been able to incorporate a manual fine tuning system which would enable the PLL tuning system to be intentionally offset in predetermined increments. Television sets normally have an automatic fine tuning (AFT) feature, but this is normally incorporated as a separate circuit which is not directly incorporated into the television tuner.

An additional disadvantage of prior art PLL systems which are designed for use in a television tuner environment is that they are highly complex and relatively expensive. In order to convert the channel number input into the proper digital control signals for the PLL, a relatively large ROM having a capacity on the order of 82 words by 12 bits was required. The best prior art PLL tuning systems require two high speed programmable counters which greatly increase the system complexity. This together with the large ROM which the system required, greatly decreased the cost effectiveness of the system so that commercial manufacturers were able to use these prior art PLL systems only in their most expensive commercial television receivers.
Therefore, it is a feature of this invention to provide a digital PLL tuning system which incorporates design techniques that vastly simplify the complexity of the PLL while at the same time allowing the system to meet the latest needs of a television tuning system or any other PLL tuning system which is addressed by a channel number.
It is another feature of this invention to provide a digital PLL tuning system that has the ability to automatically tune nonprecise station frequencies and the ability to be manually fine tuned.
It is yet another feature of the present invention to provide a digital PLL tuning system having only a single high speed programmable counter and requiring a ROM capacity of only 5 words by 9 bits.
It is still another feature of this invention to provide a digital PLL tuning system which performs the automatic fine tuning feature by utilizing the PLL tuning system as a digital discriminator.
It is yet another feature of this invention to provide a digital PLL tuning system incorporating a manual fine tuning (MFT) arrangement which is capable of intentionally offsetting the local oscillator frequency of a TV tuner in one megahertz steps or of offsetting TV IF frequency in steps of 125 kilohertz.
SUMMARY OF THE INVENTION
The preferred embodiment of the present invention includes a phase locked loop circuit means for an automatic frequency synthesizing system. The phase locked loop circuit means includes programming means which is responsive to an input signal representing a selected channel number for generating a first digital control signal representative of the selected channel number and for generating a second digital control signal representative of a predetermined group of channel numbers. A programmable divider means is coupled to the first and second digital control signals and generates a digital output signal representative of a desired system output frequency corresponding to the selected channel number.
The phase locked loop circuit means further includes an automatic fine tuning feature for fine tuning the phase locked loop output frequency to the exact frequency of the received signal. The system further includes a manual fine tuning provision which allows the phase locked loop operating frequency to be intentionally offset in predetermined increments.



PHILIPS TDA2593 SYNCHRO AND HORIZONTAL DEFLECTION CONTROL FOR COLOR TV SET

DESCRIPTION
The TDA2593 is a circuit intended for the horizontal
deflectionof color TVsets, suppliedwith transistors
or SCR’S.

.LINE OSCILLATOR(two levels switching) .PHASE COMPARISON BETWEEN SYNCHRO-
PULSE AND OSCILLATOR VOLTAGE
Ø 1, ENABLED BY AN INTERNAL PULSE,
(better parasitic immunity) .PHASE COMPARISON BETWEEN THE FLYBACK
PULSES AND THE OSCILLATORVOLTAGE
Ø2 .COINCIDENCE DETECTOR PROVIDING A
LARGE HOLD-IN-RANGE .FILTER CHARACTERISTICS AND GATE
SWITCHING FOR VIDEO RECORDER APPLICATION
.NOISE GATED SYNCHRO SEPARATOR .FRAME PULSE SEPARATOR .BLANKING AND SAND CASTLE OUTPUT
PULSES .HORIZONTAL POWER STAGE PHASE LAGGING
CIRCUIT .SWITCHING OF CONTROL OUTPUT PULSE
WIDTH .SEPARATED SUPPLY VOLTAGE OUTPUT
STAGE ALLOWING DIRECT DRIVE OF
SCR’S CIRCUIT .SECURITY CIRCUIT MAKES THE OUTPUT
PULSE SUPPRESSED WHEN LOW SUPPLY
VOLTAGE.

A phase-lock loop circuit of a video display apparatus is synchronized by a horizontal synchronizing input signal. The phase-lock loop circuit includes a frequency-to-voltage converter that is responsive to the synchronizing input signal for generating a control voltage indicative of the frequency of the synchronizing input signal. A voltage-to-current converter responsive to the control voltage generates a control current that is indicative of the frequency of the input signal. The control current varies the free running frequency of a controlled oscillator of the phase-lock loop circuit such that the free running frequency of the oscillator is directly related to the frequency of the input signal. The phase of the output signal of the oscillator is controlled by a second control current that is indicative of the phase difference between the oscillator output signal and the synchronizing input signal. When the frequency of the input signal is lower than a predetermined frequency, the first control current is at a level that causes the free running frequency to be above a predetermined minimum frequency.

CRT Line Output Stage Operation Principle:

I'll examine the operation of the line output stage, whose basic job is to generate a sawtooth current in the line scan coils so that the beams are deflected horizontally across the picture tube's screen. The beams are deflected from the left-hand side to the right-hand side to give the forward line scan: this is followed by a rapid, blanked flyback to the left-hand side ready to trace out the next viewed line. Because of the way in which the flyback is achieved, the line output transformer generates various pulse voltages which are rectified to produce the e.h.t. required by the tube and other supplies. The line output stage is not just any sort of amplifier. The active device, almost always a transistor though valves, thyristors and gate -controlled switches have been used in the past, operates as a switch, the inductive components in the stage being mainly responsible for generating the sawtooth current waveform. Tuning is used to generate and control the flyback. The line drive waveform controls the output transistor's on/off switching and thus determines the timing of the cycle of operations, keeping them phase synchronised with the transmitted picture signal.
Basic Operation

Fig. 1 shows in most basic form the main elements in the line output stage, the active device (transistor) being shown as a switch. When the switch is closed, capacitor C and diode D are shorted out and the 150V supply is connected across coil L. Now it's a basic law of inductance that when a d.c. voltage is connected across a coil the current flowing through the coil builds up linearly from zero. Fig. 2(a) shows this as a positive -going ramp that starts at time t 1 , when the switch is closed. After about 26psec (t2), roughly the time required to deflect the beams from screen centre flows via the large -value capacitor CR, charging the tuning capacitor C with the result that the voltage at its 'upper' plate (the one connected to the coil) rises to a relatively high positive value. When all the energy in coil L has been transferred to capacitor C (time t3) the latter begins to discharge, passing the energy back the other way to L via CR which, as far as the circuit's a.c. operation is concerned, can be regarded as a short-circuit. At time t4 the capacitor has discharged, having transferred the energy back to the coil. This to-and-fro interchange of energy between L and C, which from the a.c. point of view are in parallel (CR representing a short-circuit), is the normal action of a tuned/resonant/oscillatory circuit. The resonant frequency is determined by the values of L and C. These are selected so that when time t4 is reached, i.e. after a half cycle of oscillation, the sawtooth current has passed through zero to a negative point on the ramp and the beams have been deflected to the left-hand side of the screen ready for the next active line scan. To complete the oscillatory cycle (the normal resonant circuit action) the voltage at the upper plate of capacitor C would have to move negatively with respect to chassis. It can't do so because of the presence of diode D, which is called the efficiency diode - we'll explain that in a minute. When the voltage at the cathode of D tries to swing negatively it conducts, i.e. switches on, providing a discharge path for the coil. Once again because of the inductance in the circuit there's a gradual, linear current discharge, the enegery being returned to the supply's reservoir capacitor CR. During this discharge, the beams are deflected back towards the centre of the screen (times t4 to t5). At this point the magnetic flux (energy) in L has been dissipated. C is still in its discharged state, being shorted out by diode D. So at time t5, with the beams at screen centre (zero deflection), the switch has to be closed so that the cycle of operation can be repeated. The action of diode D has, with the inductance in the circuit, provided half the scan power while in the process returning the energy (minus inevitable circuit losses) to the reservoir capacitor. No wonder it's called the efficiency diode. It's important to note that the beam flyback period t2 to t4 is governed by the time -constant of L and C, consisting of one half cycle of oscillation. To achieve a flyback time of 12μsec the duration of one cycle needs to be 24μsec: so the resonant frequency of L and C works out at 41.67kHz. Fig. 3 illustrates the four phases in the operation of the line output stage. Now the voltage developed across an inductor is propor- tional to the rate of change of the current flowing through it. Thus the voltage across L is relatively low during the forward scan period but correspondingly high during the flyback, when the current flow is faster because of the circuit resonance. The voltage developed at the positive plate of capacitor C is shown in Fig. 2(b), typically peaking at 1,200V. Both the line output transistor and the efficiency diode must be capable of withstanding this high reverse voltage. As we've seen, the circuit action is highly efficient as the energy stored in L is returned to the supply during the first half of the forward scan: indeed with 'perfect' components there would be no net demand on the power supply at all! In practice because of the resistance of the inductor and the losses in the diode, switch and capacitor the circuit takes out a little more than it puts back, while the practice of loading the transformer with rectifier circuits to provide power for other sections of the set increases the stage's current demand. To make up for these losses, the line output transistor is switched on slightly before instead of at the centre of the forward scan. In a practical circuit L is the primary winding of the line output transformer and the deflection coils are connected across it via a d.c. blocking capacitor, CB, as shown in Fig. 4. This coupling capacitor also provides scan -correction (often referred to as S -correction). Why is this required? If a linear deflection current was used to control the scanning with a relatively flat -faced picture tube the sides of the picture would be stretched out in comparison with the centre section. Hence S -correction: the value of the coupling capacitor is chosen so that it resonantes with the inductance of the scan coils at about 5kHz. This has the effect of adding a sinewave component to the sawtooth current, as shown in Fig. 5. Thus the deflection power is tailored to suit the length of the beam paths as the screen is scanned, correcting the horizontal linearity of the display. At the line scanning frequency the scan coils behave as an almost perfect inductor, but their small d.c. resistance is in series with the fixed voltage that should be present across the coil. It has the effect of introducing an asymmetric sensitivity loss during the forward scan. To counteract it a further component is added in series with the scan coils - an inductor with a saturable magnetic core, biased by a permanent magnet so that its inductance falls as the scan current increases. The voltage drop across this inductor, which is known as the linearity coil, varies in the opposite sense to that produced by the resistance of the coils, thus providing an equal -but -opposite cancellation effect. In some TV sets the permanent magnet can be adjusted to trim the linearity correction, though many modern sets use components with such tight tolerances that a sealed linearity -correction coil can be used. With some very small -screen sets the horizontal non -linearity effect is small enough to be ignored.

Practical Line Output Stage
Fig. 6 shows a relatively simple line output stage circuit used with a 90° -deflection tube. Tr5 is the line output transistor, which incorporates the efficiency diode in the same package. The primary winding of the line output trans- former T4 is the section between pins 2 and 10, C95 being the flyback tuning capacitor. Scan coil coupling and S - correction are provided by C94, the line linearity coil L14 being connected in series on the chassis side of the scan current path. L14 is damped by R110 to prevent it ringing when the line flyback pulse occurs - the effect of an undamped linearity coil is velocity modulation of the beams at the beginning of their sweeps, showing up as black -and - white vertical striations at the left-hand side of the screen. C92 is the reservoir capacitor, the h.t. feed being via 8105. 8106 and R109 feed pulses to the second phase -locked loop (APC2) in the sync chip - we dealt with this in last month's instalment. A second pulse feed from the same point goes to the colour decoder chip to provide line blanking, burst gating and PAL switch drive - this particular set doesn't use the sandcastle pulse approach.

Secondary Supplies

So much for the generation and control of the sawtooth scanning current. The rest of the components in this circuit are used to harness the energy in the transformer to provide power supplies for other sections of the receiver. The winding between pins 4 and 8 pulse energises the picture tube's heaters at 6.3V r.m.s. The other supplies make use of the transformer as the heart of a d.c.-to-d.c. converter system, by means of secondary windings that provide pulse feeds to diode/capacitor rectifier circuits. Small -value (0.680) resistors in the 25V and 200V supplies provide surge limiting and protection (by going open -circuit) in the event of a short-circuit in one of these supplies. The most significant supply is obtained from the diode - split winding that starts at pin 9. Although not shown in full detail it consists of several 'cells', each of which consists of an electrically isolated secondary winding, a built-in high - voltage rectifier diode and, as the reservoir capacitor, the carefully contrived capacitance that's present between adjacent, highly -insulated winding layers. These cells are connected in series to form a voltage -multiplier system capable of providing an e.h.t. supply for the tube's final anode of typically 24kV - it may be as high as 30kV in some designs. There's a built-in surge limiter resistor at the output end of the chain of cells. An important part of the e.h.t. multiplier system is the final reservoir capacitor that split chain provides about 8kV to a built-in potential -divider chain that contains two presets: the one at the top provides the supply for the tube's focus electrode while the one near the bottom provides its first anode supply of about 800V. The bottom of the diode -split chain (pin 9) is returned to chassis via a diode/capacitor/resistor network (not shown here). The voltage developed across this network is proportional to the total beam current, since this flows from the tube's cathodes via the e.h.t. connector and the diode -split chain to chassis. Above a certain threshold the voltage at pin 9 reduces the picture brightness and/or contrast via the colour decoder/matrixing chip, limiting the beam current and hence the dissipation in the tube's shadowmask to safe levels. The winding between pins 10 and 7 of the transformer produces 50-70V pulses that sit on the h.t. voltage present at pin 10. When rectified by D23 and C100 a 200V supply is provided for the RGB output stages that drive the tube's cathodes. Secondary winding 4-6 feeds D24 and C99 which provide a 25V supply for the field timebase. In some designs supplies for the audio output stage and the signal sections of the receiver are also obtained from the line output transformer: in this particular chassis they are obtained from the chopper transformer in the power supply instead. Incidentally there have been one or two designs, the Ferguson/philco TX10 chassis being a well-known example, where the e.h.t. is also obtained from the chopper transformer, the line output transformer then acting mainly as a load for the line output transistor. In earlier designs a separate diode - capacitor multiplier unit (tripler) was fed from a single line output transformer overwiding to provide the e.h.t.

Scan Rectification

The e.h.t., focus and 200V supplies derived from the transformer are relatively lightly loaded, i.e. no great current demand is placed on them. They can therefore be obtained by rectifying the pulses present during the flyback period (time t2 -t4 in Fig. 2), which is about twenty per cent of the scan cycle. Where the current demand is greater, e.g. in a supply for the field timebase or an audio output stage, the phasing of the relevant transformer winding is often arranged so that the rectifier diode conducts during the scan rather than the flyback period. Although the voltage available is much lower, it's present for a longer period (about eighty per cent of the scan/duty cycle). As a result the output regulation is much better. The relatively high peak reverse voltage has to be taken into account in the rectifier diode's specification.

EHT Regulation

The internal impedance of a diode -split e.h.t. supply is typically about 1MOhm. Thus with a total beam current of lmA, present when a bright picture is being displayed on a 22in. picture tube, the e.h.t. voltage will drop by about 1kV or five per cent. The result of this is some ballooning, i.e. increase in picture size. Compensation can be provided by reducing the line scanning power. Careful choice of the value of the resistor that feeds the line output transformer - R105 in Fig. 6 - gives automatic compensation in the horizontal direction, while deriving the supply for the field output stage from the line output transformer tends to cancel out the ballooning in the vertical plane. Various 'anti -breathing' arrangements are used in TV receiver design. Most operate via the diode -modulator circuit we'll come to shortly. With any line output stage circuit the picture width and e.h.t. voltage depend on the stage's h.t. supply, so this must be well regulated and set up correctly. In the circuit shown in Fig. 6 the h.t. voltage has to be 119V with a 20in. tube and 145V with a 22in. tube.


Pincushion Distortion

The raster produced on an almost -flat faced picture tube by constant -amplitude scan currents has pincushion distortion at all four sides. This is because of the disparity between the image plane and the screen's profile -  . As a general rule the deflection yokes used with modern 90° tubes have built-in correction for both NS (vertical) and EW (horizontal) pincushion distortion while 110° tubes (generally above 22in. screen size) have in -yoke correction for NS distortion but cannot fully compensate for the pincushion effect at the sides of the screen. Thus with these the line scan current has to be amplitude -modulated by a parabolic waveform at field frequency as shown in Fig. 7. With present-day tube designs a modulation depth of about seven per cent is required. the peak -to -peak scan current being typically 4.1A at the top and bottom of the screen and 4.4A towards the centre of the screen, where the deflection power is greatest. Amplitude modulation of the line scan current can be achieved by including a saturable -reactance transformer in series with the scan coils, but this is expensive. You could put a suitably -shaped ripple on the supply to the line output stage, but the parabola would be superimposed on any secondary supplies derived from the line output transformer. The most widely used solution is to employ a diode -modu- lator circuit, since this gives full control of the raster shape and scan amplitude while providing a constant load current and flyback time.

The Diode Modulator
Fig. 8 shows the essence of a diode -modulator arrange- ment. The efficiency diode is split in two, DI and D2, which perform the same clamping action as before. The flyback tuning capacitor is also split in two, Cl and C2: the upper one tunes the transformer and scan coils (L1) as before while the lower one tunes a bridge coil, L2, via C4 to the same flyback frequency of about 42kHz. C3 is the scan coupling capacitor, which corresponds with CB in Fig. 4. Modulation is achieved by using transistor Tr2, whose conduction governs the scan width, to vary the load across C4. When Tr2 is off, the scan energy is shared between the the two series LC combinations C3/L1 and L2/C4. The charge on C3 and C4 is in the ratio of about 7:1, the scan current being reduced in proportion. When Tr2 is fully conductive, C4 is effectively shorted out and acquires no charge. Thus a greater proportion of the energy is present in C3/L1 and the scan current and picture width are increased. By varying the conduction of Tr2 during the forward scan in a parabolic manner, EW pincushion correction is achieved. The basic picture width can be controlled by varying Tr2's standing bias. Choke L3 and the large -value capacitor C5 filter the line -frequency energy so that it doesn't reach Tr2. And because both sections of the load (L 1/C1 and L2/C2) are individually tuned to the flyback frequency the flyback time, and hence the e.h.t. and the other line output transformer -derived supplies, remain constant over the field period despite the line scan current variation. There are several different versions of the diode -modu- lator arrangement. Some tube/yoke combinations have a scan -geometry characteristic such that when the line scan current is modulated by a simple parabolic waveform as described above the raster has inner pincushion distortion as shown in Fig. 9.

 Because of this. the EW-correction system also has to modulate the S -correction. Fig. 10 shows, in skeleton circuit form. how this can be done. There are two coupling/S-correction capacitors. C3 and C3A. C3 is the usual S -correction capacitor, but C3A has an increasing influence as the diode modulator begins to have maximum effect towards the centre of the screen. Critical choice of the value of C3A ensures that the inner curved verticals shown in Fig. 9 are straightened out to give a raster completely free from geometric distortion. Although all diode modulators work on the same basic principle, in some designs a transformer is used in place of the bridge coil to give better impedance matching and balance. Fig. 11 shows such an arrangement, used by Bang and Olufsen. The EW correction waveform is applied to transformer T6. whose winding 1-2 takes the place of L2 in Figs. 8 and 10. This circuit also provides inner -pincushion distortion correction as just described, the supplementary S - correction capacitor being C36.

Diode Modulator Drive

The parabolic EW drive waveform required is easily obtained by feeding the field -scan sawtooth waveform to a double integrator. By adding a sawtooth component the shape of the parabolic waveform can be tilted in either direction to give keystone -distortion correction if required - this is not generally necessary with modern tube/yoke designs. These EW correction characteristics are adjustable by preset resistors or, in the case of bus -programmable sets, remote control commands to the deflection processor. Very often the EW modulator is used to correct the previously mentioned picture breathing effect: this is done by feeding to the EW modulator's control circuit a voltage that's proportional to beam current.
TDA2522 PAL TV CHROMA DEMODULATOR COMBINATION
FAIRCHILD LINEAR INTEGRATED CIRCUIT
GENERAL DESCRIPTION- The TDA2522 is a monolithic integrated circuit designed as
a synchronous demodulator for PAL color television receivers. It includes an 8,8 MHz
oscillator and divider to generate two 4.4 MHz reference signals and provides color difference outputs.
PACKAGE OUTLINE 9B

The TDA2522 is Intended to Interface directly with the TDA2560 with a minimum oF external components. The TDA2530 may be added if RGB drive is required. The TDA2522
is constructed using the Fairchild Planar* process.





TDA2560 LUMINANCE AND CHROMINANCE CONTROL COMBINATION
The TDA2560 is a monolithic integrated circuit for use in decoding systems of COLOR
television receivers. The circuit consists of a luminance and chrominance amplifier.
The luminance amplifier has a low input impedance so that matching of the luminance
delay line is very easy.
It also incorporates the following functions:
- d.c. contrast control;
- d.c. brightness control;
- black level clamp;
- blanking;
- additional video output with positive-going sync.
The chrominance amplifier comprises:
- gain controlled amplifier;
- chrominance gain control tracked with contrast control;
- separate d.c. saturation control:
- combined chroma and burst output, burst signal amplitude not affected by contrast and
saturation control;
- the delay line can be driven directly ‘by the IC.

APPLICATION INFORMATION (continued)
The function is quoted against the corresponding pin number
Balanced chrominance input signal (in conjunction with pin 2)
This is derived from the chrominance signal bandpass filter, designed to provide a
push-pull input. A signal amplitude of at least 4 mV peak-to-peak is required
between pins l and 2. The chrominance amplifier is stabilized by an external feedback
loop from the output (pin 6) to the input (pins I and 2). The required level at pins l
and 2 will be 3 V.
All figures for the chrominance signals are based on a colour bar signal with 75%
saturation: i.e. burst-to-chrominance ratio of input signal is 1 1 2.
Chrominance signal input (see pin 1)
A. C.C. input
A negative-going potential, starting at +l,2 V, gives a 40 dB range of a. c. c.
Maximum gain reduction is achieved at an input voltage of 500 mV.
Chrominance saturation control
A control range of +6 dB to >-14 dB is provided over a range of d. c. potential on
pin 4 from +2 to +4 V. The saturation control is a linear function of the control
voltage.
Negative supply (earth)
Chro minance signal output
For nominal settings of saturation and contrast controls (max. -6 dB for saturation,
and max. -3 dB for contrast) both the chroma' and burst are available at this pin, and
in the same ratio as at the input pins 1 and 2. The burst signal is not affected by the
saturation and contrast controls. The a.c. c. circuit of the TDA2522 will hold
constant the colour burst amplitude at the input of the TDA2522. As the PAL delay
line is situated here between the TDA256O and TDA2522 there may be some variation
of the nominal 1 V peak-to-peak burst output of the TDA2560, according to the
tolerances of the delay line. An external network is required from pin 6 of the
TDA256O to provide d. c. negative feedback in the chroma channel via pins I and 2.
Burst gating and clamping pulse input
A two-level pulse is required at this pin to be used for burst gate and black level
clamping. The black level clamp is activated when the pulse level is greater than
7 V. The timing of this interval should be such that no appreciable encroachment
occurs into the sync pulse on picture line periods during normal operation of the
receiver. The burst gate, which switches the gain of the chroma amplifier to
maximum, requires that the input pulse at pin 7 should be sufficiently wide, at least
8 ps, at the actuating level of 2,3 V.

+12 V power supply
Correct operation occurs within the range 10 to 14 V. All signal and control levels
have a linear dependency on supply voltage but, in any given receiver design, this
range may be restricted due to considerations of tracking between the power supply
variations and picture contrast and chroma levels.
Flyback blanking input waveform
This pin is used for blanking the luminance amplifier. When the input pulse exceeds
the +2, 5 Vlevel, the output signal is blanked to a level of about 0 V. When the input
exceeds a +6 V level, a fixed level of about 1, 5 V is inserted in the output. This
level can be used for clamping purposes.
Luminance sigal output
An emitter follower provides a low impedance output signal of 3 V black-to-white
amplitude at nominal contrast setting having a black level in the range 1 to 3 V. An
external emitter load resistor is not required.
The luminance amplitude available for nominal contrast may be modified according
to the resistor value from pin 13 to the +12 V supply. At an input bias current
114 of 0,25 mA during black level the amplifier is compensated so that no black
level shift more than 10 mV occurs at contrast control. When the input current
deviates from the quoted value the black level shift amounts to 100 mV/rnA.
Brightness control
The black level at the luminance output (pin 10) is identical to the control voltage
required at this pin, A range of black level from l to 3 V may be obtained.
Black level clamp capacitor
Luminance gain setting resistor
The gain of the luminance amplifier may be adjusted by selection of the resistor
value from pin 13 to +12 V. Nominal luminance output amplitude is then 3 V
black-to-white at pin 10 when this resistor is 2, 7 l

TDA2530 RGB MATRIX PREAMPLIFIER
The TDA2530 is an integrated RGB -matrix preamplifier for colour television receivers,
incorporating a matrix preamplifier for RGB cathode drive of the picture tube with
clamping circuits. The three channels have the same layout to ensure identical frequency
behaviour.
This integrated circuit has been designed to be driven from the TDA2522 Synchronous
demodulator and oscillator IC.











PHILIPS TDA2653A Vertical deflection circuit

DESCRIPTION
The TDA2653A is a monolithic integrated circuit for vertical deflection in large screen colour television receivers.
The circuit incorporates the following functions:
· Oscillator; switch capability for 50 Hz/60 Hz operation
· Synchronization circuit
· Blanking pulse generator with guard circuit
· Sawtooth generator with buffer stage
· Preamplifier with fed-out inputs
· Output stage with thermal and short-circuit protection
· Flyback generator
· Voltage stabilizers.

APPLICATION INFORMATION
The function is described against the corresponding pin number
1, 13. Oscillator
The oscillator frequency is determined by a potentiometer at pin 1 and a capacitor at pin 13.
2. Sync input/blanking output
Combination of sync input and blanking output. The oscillator has to be synchronized by a positive-going
pulse between 1 and 12 V. The integrated frequency detector delivers a switching level at pin 12.
The blanking pulse amplitude is 20 V with a load of 1 mA.
3. Sawtooth generator output
The sawtooth signal is fed via a buffer stage to pin 3. It delivers the signal which is used for linearity control,
and drive of the preamplifier. The sawtooth is applied via a shaping network to pin 11 (linearity) and via a
resistor to pin 4 (preamplifier).
4. Preamplifier input
The DC voltage is proportional to the output voltage (DC feedback). The AC voltage is proportional to the
sum of the buffered sawtooth voltage at pin 3 and the voltage, with opposite polarity, at the feedback
resistor (AC feedback).
5. Positive supply of output stage
This supply is obtained from the flyback generator. An electrolytic capacitor between pins 7 and 5, and a
diode between pins 5 and 9 have to be connected for proper operation of the flyback generator.
6. Output of class-B power stage
The vertical deflection coil is connected to this pin, via a series connection of a coupling capacitor and a
feedback resistor, to ground.
7. Flyback generator output
An electrolytic capacitor has to be connected between pins 7 and 5 to complete the flyback generator.
8. Negative supply (ground)
Negative supply of output stage and small signal part.
9. Positive supply
The supply voltage at this pin is used to supply the flyback generator, voltage stabilizer, blanking pulse
generator and buffer stage.
10. Reference voltage of preamplifier
External adjustment and decoupling of reference voltage of the preamplifier.
11. Sawtooth capacitor
This sawtooth capacitor has been split to realize linearity control.
12. 50 Hz/60 Hz switching level
This pin delivers a LOW voltage level for 50 Hz and a HIGH voltage level for 60 Hz. The amplitudes of the
sawtooth signals can be made equal for 50 Hz and 60 Hz with these levels.




TDA2545A TDA2546Quasi-split-sound circuit:

GENERAL DESCRIPTION The TDA2545A is a monolithic integrated circuit for quasi-split-sound processing in television receivers. Features · 3-stage gain controlled i.f. amplifier · A.G.C. circuit · Reference amplifier and limiter amplifier for vision carrier (V.C.) processing · Linear multiplier for quadrature demodulation.


TDA2541 IF AMPLIFIER WITH DEMODULATOR AND AFC
DESCRIPTION
The TDA2540 and 2541 are IF amplifier and A.M.
demodulator circuits for colour and black and white
televisionreceiversusingPNPorNPNtuners. They
are intended for reception of negative or positive
modulation CCIR standard.
They incorporate the following functions : .Gain controlled amplifier .Synchronous demodulator .White spot inverter .Video preamplifier with noise protection .Switchable AFC .AGC with noise gating .Tuner AGC output (NPN tuner for 2540)-(PNP
tuner for 2541) .VCR switch for video output inhibition (VCR
play back).














































BRIONVEGA COSMO 2 TVC 26. Chassis detailed viewing:
Line output transistor (RCA BU208A), PCB Power part (large signals) view, PCB Signal Part (small signals) view.
509.01.3492

BU208(A)

Silicon NPN
npn transistors,pnp transistors,transistors
Category: NPN Transistor, Transistor
MHz: <1 MHz
Amps: 5A
Volts: 1500V
HIGH VOLTAGE CAPABILITY
JEDEC TO-3 METAL CASE.

DESCRIPTION
The BU208A, BU508A and BU508AFI are
manufactured using Multiepitaxial Mesa
technology for cost-effective high performance
and use a Hollow Emitter structure to enhance
switching speeds.

APPLICATIONS:
* HORIZONTAL DEFLECTION FOR COLOUR TV With 110° or even 90° degree of deflection angle.

ABSOLUTE MAXIMUM RATINGS
Symbol Parameter Value Unit
VCES Collector-Emit ter Voltage (VBE = 0) 1500 V
VCEO Collector-Emit ter Voltage (IB = 0) 700 V
VEBO Emitter-Base Voltage (IC = 0) 10 V
IC Collector Current 8 A
ICM Collector Peak Current (tp < 5 ms) 15 A
TO - 3 TO - 218 ISOWATT218
Ptot Total Dissipation at Tc = 25 oC 150 125 50 W
Tstg Storage Temperature -65 to 175 -65 to 150 -65 to 150 oC
Tj Max. Operating Junction Temperature 175 150 150 °C.

BRIONVEGA COSMO 2 TVC 26 CHASSIS 509 Electronic digital clocks:

 This invention relates to an electronic digital clock having at least one time-programmed electrical output for triggering switching functions.

Electronic clocks providing a digital display are known, in which a constant-frequency signal is applied to a clock circuit. The constant-frequency signal can be generated by a crystal or can be derived from the electrical mains supply. The clock circuit generates coded signals, usually BCD signals, which are applied to a display logic circuit. The display logic circuit in turn drives for example a seven segment display. The switching sequence of the coded signals and the display logic circuit is determined by a multiplexer which is supplied with pulses from the clock circuit. In this fashion, it is possible to provide an opto-electronic digital time display, which may for example include a display of the day of the week.

Time switches are known which have a time-programmed electrical output in order to trigger switching functions. The time-programming is performed mechanically by a timer disc which at a preset time, which is determined mechanically, closes or opens switching contacts. In electronic digital equipment such an arrangement is not possible because no mechanical moving parts are present.

In electronic digital clocks it is, however, known to provide a time switching function. For this purpose, an electronic store is provided and, when the stored value corresponds with the time displayed by the clock, a signal triggering a switching function is generated. The store can consist of a decade counter pulsed by the clock circuit. A disadvantage of such an arrangement is the relatively high outlay in circuitry involved in the execution of just one switching function. A further disadvantage resides in the fact that in the event of any interruption in the current supply the store is erased and the information has to be restored into it.

This invention seeks to provide a clock in which at least the former disadvantage is reduced.

According to this invention there is provided an electronic digital clock having at least one time-programmed electrical output for triggering switching functions, the clock comprising a digital electrical time display, a clock circuit for producing and applying to said display coded signals representing time, a multiplexer, at least one coded fixed-value store, a sampling logic circuit and a comparator circuit, wherein coded values stored in the fixed-value store are sampled under the control of the multiplexer by the sampling logic circuit and are compared in the comparator circuit with said coded signals representing time, and wherein in the event of coincidence between the sampled values and the coded signals representing time an output signal is produced at said time-programmed electrical output.

It is possible in this fashion to provide a virtually arbitrary number of stores without increasing the outlay in circuitry as a consequence of the number of stores. By using fixed-value stores which comprise coding switches, it is also possible to contrive that even after an interruption in the current supply, the stored value previously written in is retained un-disturbed. A further advantage resides in the fact that the stored values adjusted can be read out at any time, i.e. it is possible at any desired time to determine which store is set to which value.