TDA5330T VHF, UHF and Hyperband mixer/oscillator for TV and VCR 3-band tuners
The TDA5330T is a monolithic integrated circuit that performs the band A, band B and band C mixer/oscillator functions
in TV and VCR tuners. This device gives the designer the capability to design an economical and physically small 3-band
tuner which will be capable of meeting the most stringent requirements e.g. FTZ or FCC. The tuner development time
can be drastically reduced by using this device.
· Balanced mixer with a common emitter input for band A
· Amplitude-controlled oscillator for band A
· Balanced mixer with common base input for band B and C
· Balanced oscillator for band B and C
· Local oscillator buffer output for external prescaler
· SAW filter preamplifier with an output impedance of 100 W
· Bandgap voltage stabilizer for oscillator stability
· Electronic bandswitch
TSA5511 1.3 GHz Bidirectional I2C-bus
· Complete 1.3 GHz single chip system
· Low power 5 V, 35 mA
· I2C-bus programming
· In-lock flag
· Varicap drive disable
· Low radiation
· Address selection for Picture-In-Picture (PIP), DBS
tuner (3 addresses)
· Analog-to-digital converter
· 8 bus controlled ports (5 for TSA5511T), 4 current
limited outputs (1 for TSA5511T), 4 open collector
· Power-down flag
· TV tuners
· VCR Tuners
The TSA5511 is a single chip PLL frequency synthesizer
designed for TV tuning systems. Control data is entered
via the I2C-bus; five serial bytes are required to address
the device, select the oscillator frequency, programme the
eight output ports and set the charge-pump current. Four
of these ports can also be used as input ports (three
general purpose I/O ports, one ADC). Digital information
concerning those ports can be read out of the TSA5511 on
the SDA line (one status byte) during a READ operation.
A flag is set when the loop is “in-lock” and is read during a
READ operation. The device has one fixed I2C-bus
address and 3 programmable addresses, programmed by
applying a specific voltage on Port 3. The phase
comparator operates at 7.8125 kHz when a 4 MHz crystal
TDA8341 Television IF amplifier and demodulator
The TDA8340;Q and TDA8341;Q are integrated IF
amplifier and demodulator circuits for colour or black/white
television receivers, the TDA8340;Q is for application with
n-p-n tuners and the TDA8341;Q for p-n-p tuners.
The TDA8340;Q and TDA8341;Q are pin-compatible
successors with improved performance to types
TDA2540/2541;Q and TDA3540/3541;Q.
· Full range gain-controlled wide-band IF amplifier
· Linear synchronous demodulator with excellent
· White spot inverter
· Wide-band video amplifier with noise protection
· AFC circuit with AFC on/off switching and
· Low impedance AFC output
· AGC circuit with noise gating
· Tuner AGC output for n-p-n tuners (TDA8340) or p-n-p
· External video switch for switching-off the video output
· Reduced sensitivity for high sound carriers
· Integrated filter to limit second harmonic IF signals
· Wide supply voltage range
· Requires few external components
TEA2164 SWITCH MODE POWER SUPPLY PRIMARY CIRCUIT
.POSITIVE AND NEGATIVE OUTPUT CURRENT
UP TO 1.2AAND – 1.7A .A TWO LEVEL COLLECTOR CURRENT LIMITATION
.COMPLETE TURN OFF AFTER LONG DURATION
OVERLOADS .UNDER AND OVER VOLTAGELOCK-OUT .SOFT START BY PROGRESSIVE CURRENT
LIMITATION .DOUBLE PULSE SUPPRESSION .BURST MODE OPERATION UNDER STANDBY
In amaster slave architecture, the TEA2164control
IC achieves the slave function. Primarily designed
for TV receivers and monitors applications, this
circuit provides an easy synchronizationand smart
solution for low power stand by operation.
Located at the primary side the TEA2164 Control
IC ensures :
- the power supply start-up
- the power supply control under stand-by conditions
- the process of the regulation signals sent by the
master circuit located at the secondary side
- directbasedrive of the bipolarswitching transistor
- the protection of the transistor and the power
supply under abnormal conditions.
II. GENERAL DESCRIPTION
In a master slave architecture, the TEA2164 Control
IC, located at the primary side of an off line
power supply achievesthe slave function ;whereas
the master circuit is located at the secondary side.
The link between both circuits is realized by a small
In the operation of the master-slave architecture,
four majors cases must be considered :
- normal operating
- power supply start-up
- abnormal conditions : off load, short circuit, ...
II.1. Normal Operating (master slave mode)
In this configuration, the master circuit generatesa
pulse widthmodulatedsignal issued from themonitoring
of the output voltage which needs the best
accuracy (in TV applications : the horizontal deflection
stagesupplyvoltage).Themaster circuit power
supply can be supplied by another output.
The PWM signal are sent towards the primary side
through small differentiating transformer. For the
TEA2164 positive pulses are transistor switchingon
commands ; and negative pulses are transistor
switching-offcommands (Figure 4). In this configuration,
only by synchronizing the master oscillator,
the switching transistor may be synchronized with
an external signal.
II.2. Stand-by Mode
In this configuration the master circuit no longer
sends PWM signals, the structure is not synchronized
; and the TEA2164 operates in burst mode.
The average power consumption at the secondary
side may be very low 1W 3 P 3 6W (as it is
consumed in TV set during stand by).
By action on the maximum duty cycle control, a
primary loop maintains a semi-regulation of the
output voltages.Voltage on feed-back is applied on
Burst period is externally programmedby capacitor
II.3. Power Supply Start-up
After the mains have been switched-on, the VCC
storage capacitor of the TEA2164 is charged
through a high value resistor connected to the
rectified high voltage.When Vcc reaches VCC start
threshold (9V typ), the TEA2164 starts operatingin
burst mode. Since available output power is low in
burst mode the output power consumption must
remain low before complete setting-up of output
voltage. In TV application it can be achieved by
maintaining the TV in stand-by mode during startup.
When VCC exceeds VCC max, an internal flip-flop
stops output conduction signals. The circuit will
start again after the capacitor C1 discharge ; it
means : after loss of synchronization or after Vcc
stop crossing (Figure 7).
In flyback converters, this function protects the
power supply against output voltage runaway.
MIVAR 17L1 BIFONICO CHASSIS TV3492 Synchronized switch-mode power supply:
In a switch mode power supply, a first switching transistor is coupled to a primary winding of an isolation transformer. A second switching transistor periodically applies a low impedance across a second winding of the transformer that is coupled to an oscillator for synchronizing the oscillator to the horizontal frequency. A third winding of the transformer is coupled via a switching diode to a capacitor of a control circuit for developing a DC control voltage in the capacitor that varies in accordance with a supply voltage B+. The control voltage is applied via the transformer to a pulse width modulator that is responsive to the oscillator output signal for producing a pulse-width modulated control signal. The control signal is applied to a mains coupled chopper transistor for generating and regulating the supply voltage B+ in accordance with the pulse width modulation of the control signal.
The invention relates to switch-mode power supplies.
Some television receivers have signal terminals for receiving, for example, external video input signals such as R, G and B input signals, that are to be developed relative to the common conductor of the receiver. Such signal terminals and the receiver common conductor may be coupled to corresponding signal terminals and common conductors of external devices, such as, for example, a VCR or a teletext decoder.
To simplify the coupling of signals between the external devices and the television receiver, the common conductors of the receiver and of the external devices are connected together so that all are at the same potential. The signal lines of each external device are coupled to the corresponding signal terminals of the receiver. In such an arrangement, the common conductor of each device, such as of the television receiver, may be held "floating", or conductively isolated, relative to the corresponding AC mains supply source that energizes the device. When the common conductor is held floating, a user touching a terminal that is at the potential of the common conductor will not suffer an electrical shock.
Therefore, it may be desirable to isolate the common conductor, or ground, of, for example, the television receiver from the potentials of the terminals of the AC mains supply source that provide power to the television receiver. Such isolation is typically achieved by a transformer. The isolated common conductor is sometimes referred to as a "cold" ground conductor.
In a typical switch mode power supply (SMPS) of a television receiver the AC mains supply voltage is coupled, for example, directly, and without using transformer coupling, to a bridge rectifier. An unregulated direct current (DC) input supply voltage is produced that is, for example, referenced to a common conductor, referred to as "hot" ground, and that is conductively isolated from the cold ground conductor. A pulse width modulator controls the duty cycle of a chopper transistor switch that applies the unregulated supply voltage across a primary winding of an isolating flyback transformer. A flyback voltage at a frequency that is determined by the modulator is developed at a secondary winding of the transformer and is rectified to produce a DC output supply voltage such as a voltage B+ that energizes a horizontal deflection circuit of the television receiver. The primary winding of the flyback transformer is, for example, conductively coupled to the hot ground conductor. The secondary winding of the flyback transformer and voltage B+ may be conductively isolated from the hot ground conductor by the hot-cold barrier formed by the transformer.
It may be desirable to synchronize the operation of the chopper transistor to horizontal scanning frequency for preventing the occurrence of an objectionable visual pattern in an image displayed in a display of the television receiver.
It may be further desirable to couple a horizontal synchronizing signal that is referenced to the cold ground to the pulse-width modulator that is referenced to the hot ground such that isolation is maintained.
A synchronized switch mode power supply, embodying an aspect of the invention, includes a transfromer having first and second windings. A first switching arrangement is coupled to the first winding for generating a first switching current in the first winding to periodically energize the second winding. A source of a synchronizing input signal at a frequency that is related to a deflection frequency is provided. A second switching arrangement responsive to the input signal and coupled to the second winding periodically applies a low impedance across the energized second winding that by transformer action produces a substantial increase in the first switching current. A periodic first control signal is generated. The increase in the first switching current is sensed to synchronize the first control signal to the input signal. An output supply voltage is generated from an input supply voltage in accordance with the first control signal.
MIVAR 17L1 BIFONICO CHASSIS TV3492 Switch-mode power supply with burst mode standby operation:
In a switch mode power supply, a first switching transistor is coupled to a primary winding of a transformer for generating pulses of a switching current. A secondary winding of the transformer is coupled via a switching diode to a capacitor of a control circuit for developing a control signal in the capacitor. The control signal is applied to a mains coupled chopper second transistor for generating and regulating supply voltages in accordance with pulse width modulation of the control signal. During standby operation, the first and second transistors operate in a burst mode that is repetitive at a frequency of the AC mains supply voltage such as 50 Hz. In the burst mode operation, during intervals in which pulses of the switching current occur, the pulse width and peak amplitude of the switching current pulses progressively increase in accordance with the waveform of the mains supply voltage to provide a soft start operation in the standby mode of operation within each burst group.
The invention relates to switch-mode power supplies.
In a typical switch mode power supply (SMPS) of a television receiver the AC mains supply voltage is coupled to a bridge rectifier. An unregulated direct current (DC) input supply voltage is produced. A pulse width modulator controls the duty cycle of a chopper transistor switch that applies the unregulated supply voltage across a primary winding of a flyback transformer. A flyback voltage at a frequency that is determined by the modulator is developed at a secondary winding of the transformer and is rectified to produce DC output supply voltages such as a voltage B+ that energizes a horizontal deflection circuit of the television receiver and a voltage that energizes a remote control unit.
During normal operation, the DC output supply voltages are regulated by the pulse width modulator in a negative feedback manner. During standby operation, the SMPS is required to generate the DC output supply voltage that energizes the remote control unit. However, most other stages of the television receiver are inoperative and do not draw supply currents. Consequently, the average value of the duty cycle of the chopper transistor may have to be substantially lower during standby than during normal operation.
Because of, for example, storage time limitation in the chopper transistor, it may not be possible to reduce the length of the conduction interval in a given cycle below a minimum level. Thus, in order to maintain the average value of the duty cycle low, it may be desirable to operate the chopper transistor in an intermittent or burst mode, during standby. During standby, a long dead time interval occurs between consecutively occurring burst mode operation intervals. Only during the burst mode operation interval switching operation occurs in the chopper transistor. The result is that each of the conduction intervals is of a sufficient length.
In accordance with an aspect of the invention, burst mode operation intervals are initiated and occur at a rate that is determined by a repetitive signal at the frequency of the AC mains supply voltage. For example, when the mains supply voltage is at 50 Hz, each burst mode operation interval, when switching cycles occur, may last 5 milliseconds and the dead time interval when no switching cycles occur, may last during the remainder portion or 15 milliseconds. Such arrangement that is triggered by a signal at the frequency of the mains supply voltage simplifies the design of the SMPS.
The burst mode operation intervals that occur in standby operation are synchronized to the 50 Hz signal. During each such interval, pulses of current are produced in transformers and inductances of the SMPS. The pulses of current occur in clusters that are repetitive at 50 Hz. The pulses of current occur at a frequency that is equal to the switching frequency of the chopper transistor within each burst mode operation interval. Such qurrent pulses might produce an objectionable sound during power-off or standby operation. The objectionable sound might be produced due to possible parasitic mechanical vibrations as a result of the pulse currents in, for example, the inductances and transformers of the SMPS.
In accordance with another aspect of the invention, the change in the AC mains supply voltage during each period causes the length of the conduction interval in consecutively occurring switching cycle during the burst mode operation interval to increase progressively. Such operation that occurs during each burst mode operation interval may be referred to as soft start operation. The soft start operation causes, for example, gradual charging of capacitors in the SMPS. Consequently, the parasitic mechanical vibrations are substantially reduced. Also, the frequency of the switching cycles within each burst mode operation interval is maintained above the audible range for further reducing the level of such audible noise during standby operation.
A switch mode power supply, embodying an aspect of the invention, for generating an output supply voltage during both a standby-mode of operation and during a run-mode of operation includes a source of AC mains input supply voltage. A control signal at a given frequency is generated. A switching arrangement energized by the input supply voltage and responsive to the first control signal produces a switching current during both the standby-mode of operation and the run-mode operation. The output supply voltage is generated from the switching current. An arrangement coupled to the switching arrangement and responsive to a standby-mode/run-mode control signal and to a signal at a frequency that is determined by a frequency of the AC mains input supply voltage controls the switching arrangement in a burst mode manner during the standby-mode of operation. During a burst interval, a plurality of switching cycles are performed and during an alternating dead time interval no switching cycles are performed. The two intervals alternate at a frequency that is determined by the frequency of the AC mains input supply voltage.
- Video chrominance and Luminance with TDA3562A
TDA3562A (Philips)PAL/NTSC ONE-CHIP DECODERDESCRIPTION
The TDA3562A is a monolithic IC designed as
decode PAL and/or NTSC colour television standards
and it combines all functions required for the
identification and demodulation of PAL and NTSC
.CHROMINANCE SIGNAL PROCESSOR
.LUMINANCE SIGNAL PROCESSING WITH
.HORIZONTAL AND VERTICAL BLANKING
.LINEAR TRANSMISSION OF INSERTED
.LINEAR CONTRAST AND BRIGHTNESS
CONTROL ACTING ON INSERTED AND MATRIXED
.AUTOMATIC CUT-OFF CONTROL
.NTSC HUE CONTROL
· A black-current stabilizer which
controls the black-currents of the
three electron-guns to a level low
enough to omit the black-level
· Contrast control of inserted RGB
· No black-level disturbance when
non-synchronized external RGB
signals are available on the inputs
· NTSC capability with hue control.
· Teletext/broadcast antiope
· Channel number display.
It follows that the
external switches and filters which
are required for the TDA3562A are
not required for the TDA3566A.
There is no difference between the
amplitudes of the colour output
signals in the PAL or NTSC mode.
· The clamp capacitor at pins 10, 20
and 21 in the black-level
stabilization loop can be reduced to
100 nF provided the stability of the
loop is maintained. Loop stability
depends on complete application.
The clamp capacitors receive a
pre-bias voltage to avoid coloured
background during switch-on.
· The crystal oscillator circuit has
been changed to prevent parasitic
oscillations on the third overtone of
the crystal. Consequently the
optimum tuning capacitance must
be reduced to 10 pF.
· The hue control has been improved
THE PHILIPS TDA3562A Circuit arrangement for the control of a picture tube :
1. Circuit arrangement for the control of at least one beam current in a picture tube by a picture comprising
a control loop which in one sampling interval obtains a measuring signal from the value of the beam current on the occurrence of a given reference level in the picture signal, stores a control signal derived therefrom until the next sampling interval and thereby adjusts the beam current to a value preset by a reference signal.
and a trigger circuit which suppresses auxiliary pulses used to generate the beam current after the picture tube has been started up and issues a switching signal for the purpose of closing the control loop during the sampling intervals and for releasing the control of the beam current by the picture signal after the measuring signal has exceeded the threshold value,
a change detection arrangement which delivers a change signal when the stored signal has assumed a largely constant value, and
a logic network which does not release the control of the beam current by the picture signal outside the sampling intervals until the change signal has also been issued after the switching signal.
2. Circuit arrangement as set forth in claim 1, in which the picture signal comprises several color signals for the control of a corresponding number of beam currents for the display of a color picture in the picture tube and the control loop stores a part measuring signal or a part control signal derived therefrom for each color signal, characterized in that the change detection arrangement includes a change detector for each color signal which delivers a part change signal when the relevant stored signal has assumed a largely constant value, and the logic network does not release the control of the beam currents by the color signals outside the sampling intervals until the part change signals have been delivered by all change detectors.
3. Circuit arrangement as set forth in claim 1, including a comparator arrangement which compares the measuring signal with the reference signal and derives the control signal from this comparison, characterized in that the change detection arrangement detects a change in the control signal with respect to time and issues the change signal when the control signal has assumed a largely constant value.
4. Circuit arrangement as set forth in claims 1, 2, 3 including a control signal memory which contains at least one capacitor, characterized in that the change detection arrangement delivers the change signal when a charge-reversing current of the capacitor occuring during the starting up of the picture tube falls below a limit value.
5. Circuit arrangement as set forth in claim 2, including a comparator arrangement which compares the measuring signal with the reference signal and derives the control signal from this comparison, characterized in that the change detection arrangement detects a change in the control signal with respect to time and issues the change signal when the control signal has assumed a largely constant value.
The invention relates to a circuit arrangement for the control of at least one beam current in a picture tube by a picture signal with a control loop which in one sampling interval obtains a measuring signal from the value of the beam current on the occurrence of a given reference level in the picture signal, stores a control signal derived therefrom until the next sampling interval and by this means adjusts the beam current to a value preset by a reference signal, and with a trigger circuit which suppresses auxiliary pulses used to generate the beam current after the picture tube is turned on and issues a switching signal for the purpose of closing the control loop during the sampling intervals and releasing the control of the beam current by the picture signal after the measuring signal has exceeded a threshold value.
Such a circuit arrangement has been described in Valvo Technische Information 820705 with regard to the integrated color decoder circuit PHILIPS TDA3562A and is used in this as a so-called cut-off point control. In the known circuit arrangement, such a cut-off point control provides automatic compensation of the so-called cut-off point of the picture tube, i.e. it regulates the beam current in the picture tube in such a way that for a given reference level in the picture signal the beam current has a constant value despite tolerances and changes with time (aging, thermal modifications) in the picture tube and the circuit arrangement, thereby ensuring correct picture reproduction.
Such a blocking point control is particularly advantageous for the operation of a picture tube for the display of color pictures because in this case there are several beam currents for different color components of the color picture which have to be in a fixed ratio with one another. If this ratio changes, for example, as the result of manufacturing tolerances or ageing processes, distortions of the colors occur in the reproduction of the color picture. The beam currents, therefore, have to be very accurately balanced. The said cut-off point control prevents expensive adjustment and maintenance time which is otherwise necessary.
Conventional picutre tubes are constructed as cathode-ray tubes with hot cathodes which require a certain time after being turned on for the hot cathodes to heat up. Not until a final operating temperature has been reached do these hot cathodes emit the desired beam currents to the full extent, while gradually rising beam currents occur in the time interval when the hot cathodes are heating up. The instantaneous values of these beam currents depend on the instantaneous temperatures of the hot cathodes and on the accelerating voltages for the picture tube which build up simultaneously with the heating process and are undefined until the end of the heating time. After the picture tube is turned on, these values initially produce a highly distorted picture until the beam currents have attained their final value. These picture distortions after the picture tube is turned on are even further intensified by the fact that the cut-off point control is not yet adjusted to the beam currents which flow after the heating time is over.
For the purpose of suppressing distorted pictures during the heating time of the hot cathodes, the known circuit arrangement has a turn-on delay element operating as a trigger circuit which, in essence, contains a bistable flip-flop. When the picture tube and the circuit arrangement controlling the beam currents flowing in it are turned on, the flip-flop is switched into a first state in which it interrupts the supply of the picture signal to the picture tube. Thus, during the heating time the beam currents are suppressed, and the picture tube does not yet display any picture. In sampling intervals which are provided subsequent to flybacks of the cathode beam into an initial position on the changeover from the display of one picture to the display of a subsequent picture and even within the changeover, that is outside the display of pictures, the picture tube is controlled for a short time in such a way that beam currents occur when the hot cathodes are sufficiently heated up and an accelerating voltage is resent. If these currents exceed a certain threshold value, the flip-flop circuit switches into a second state and releases the picture signal for the control of the beam currents and the cut-off point control.
It is found, however, that the picture displayed in the picture tube immediately after the switching over of the flip-flop is still not fault-free. Because, in fact, the beam currents are supported during the heating time of the hot cathodes, the cut-off point control cannot respond yet. This response of the cut-off point control takes place only after the beam currents are switched on, i.e. after the flip-flop is switched into the second state and therefore at a time in which the picture signal already controls the beam currents. In this way the response of the blocking point control makes its presence felt in the picture displayed.
With the known circuit arrangement the brightness of the picture gradually increases, during the response of the cut-off point control, from black to the final value.
This slow increase in the picture brightness after the tube is turned on is disturbing to the eyes of the viewer not only in the case of the black-and-white picture tubes with one hot cathode, but especially so in the case of colour picture tubes which usually have three hot cathodes. With a color picture tube, color purity errors can also occur in addition to the change in the picture brightness if, as a result of different speeds of response of the cut-off point control for the three beam currents, there are found to be intermittent variations from the interrelation between the beam currents required for a correct picture reproduction.
SUMMARY OF THE INVENTION
The aim of the invention is to create a circuit arrangement which suppresses the above-described disturbances of brightness and color of the displayed picture when the picture tube is being started.
The invention achieves this aim in that a circuit arrangement of the type mentioned in the preamble contains a change detection arrangement which emits a change signal when the stored signal has assumed an essentially constant value, and a logic network which does not release the control of the beam current by the picture signal until the change signal has also been emitted after the switching signal.
In the circuit arrangement according to the invention, therefore, the display of the picture is suppressed after the picture tube is turned on until the cut-off point control has responded. If the picture signal then starts to control the beam current, a perfect picture is displayed immediately. In this way, all the disturbances of the picture which affect the viewer's pleasure are suppressed. The circuit arrangement of the invention is of simple design and can be combined on one semiconductor wafer with the existing picture signal processing circuits and also, for example, with the known circuit arrangement for cut-off point control. Such an integrated circuit arrangement not only requires very little space on the semiconductor wafer, but also needs no additional external leads. Thus the circuit arrangement of the invention can be arranged, for example, in an integrated circuit which has precisely the same external connections as known integrated circuits. This means that an integrated circuit containing the circuit arrangement of the invention can be directly incorporated in existing equipment without the need for additional measures.
In one embodiment of the said circuit arrangement, in which the picture signal contains several color signals for the control of a corresponding number of beam currents for representing a color picture in the picture tube and, for each color signal, the control loop stores a part measuring signal or a part control signal derived from it, the change detection arrangement contains a change detector for each color signal which emits a part change signal when the relevant stored signal has assumed an essentially constant value, and the logic network does not release the control of the beam currents by the color signals outside the sampling intervals until the part change signals have been emitted from all change detectors.
In principle, therefore, such a circuit arrangement has three cut-off point controls for the three beam currents controlled by the individual color signals. To reduce the cost of the circuitry, the measuring stage is common to all the cut-off point controls, as in the known circuit arrangement. All three beam currents are then measured successively by this measuring stage. In this way, a part measuring signal or a part control signal derived from it is obtained for each beam current and is stored sesparately according to which of the beam currents it belongs. Changes in the part measuring signal or part control signal are detected for each beam current by one of the change detectors each time. Each of these change detectors issues a part change signal to the logic network. The latter does not release the control of the beam currents by the picture signal outside the sampling intervals until all the part change signals indicate that the part measuring signal or the part control signal, as the case may be, remains constant. This ensures that the cut-off point controls for the beam currents of all color signals have responded when the picture appears in the picture tube.
In a further embodiment of the circuit arrangement according to the invention with a comparator arrangement which compares the measuring signal with the reference signal and derives the control signal from this comparison, the change detection arrangement detects a change in the control signal with respect to time and issues the change signal when the control signal has assumed an essentially constant value. In the case of the representation of a color signal the comparator arrangement derives several part control signals, whose changes with time are detected by the change detectors, from a corresponding comparison of the part measuring signals with the reference signal. In this embodiment of the circuit arrangement of the invention, preference is given to storage of only the control signal or the part control signals for the purpose of controlling the beam currents.
In another embodiment of the circuit arrangement of the invention which includes a control signal memory which contains at least one capacitor in which a charge or voltage corresponding to the control signal is stored, the change detection arrangement issues the change signal when a charge-reversing current of the capacitor occurring during the turning on of the picture tube has fallen below a limit value and has thus at least largely decayed. Such a detection of the steady state of the cut-off point control is independent of the actual magnitude of the control signal and therefore independent of, for example, the level of the picture tube cut-off voltage, circuit tolerances or ageing processes in the circuit arrangement or the picture tube.
Detection of whether or not the charge-reversing current exceeds the limit value is performed preferentially by a current detector which is designed with a current mirror system which is arranged in a supply line to a capacitor acting as a control signal store. A current mirror arrangement of this kind supplies a current which coincides very precisely with the charging current of the capacitor. This current is then compared, preferably in a further device contained in the change detection arrangement, with a current representing a limit value or, after conversion into a voltage, with a voltage representing the limit value. The change signal is obtained from the result of this comparison.
On the other hand, digital memories may also be used as control signal memories, especially when the picture signal is supplied as a digital signal and the blocking point control is constructed as a digital control loop. In such a case, the comparator arrangement, the change detection arrangement and the trigger circuit are also designed as digital circuits. Then, the change detection arrangement advantageously forms the difference of the signals stored in the control signal memory in two successive sampling intervals and compares this with the limit value formed by a digital value. If the difference falls short of the limit value, the change signal is issued.
BRIEF DESCRIPTION OF THE DRAWINGS
An embodiment of the invention is described in greater detail below with the aid of the drawings in which:
FIG. 1 shows a block circuit diagram of the embodiment,
FIG. 2 shows a somewhat more detailed block circuit diagram of the embodiment,
FIG. 3 shows time-dependency diagrams of some signals occurring in the circuit diagram shown in FIG. 2, and
FIG. 4 shows a somewhat moredetailed block circuit diagram of a part of the circuit diagram shown in FIG. 2.
DETAILED DESCRIPTION OF THE INVENTION
FIG. 1 shows a block circuit diagram of a circuit arrangement to which a picture signal is fed via a first input 1 of a combinatorial stage 2. From the output 3 of the combinatorial stage 2 the picture signal is fed to the picture signal input of a controllable amplifier 5 which at an output 6 issues a current controlled by the picture signal. This current is fed via a measuring stage 7 to a hot cathode 8 in a picture tube 9 and forms therein a beam current of a cathode ray by means of which a picture defined by the picture signal is displayed on a fluorescent screen of the picture tube 9.
The measuring stage 7 measures the current fed to the hot cathode 8, i.e. the the beam current in the picture tube 9, and at a measuring signal output 10, issues a measuring signal corresponding to the magnitude of this current. This is fed to a measuring signal input 11 of a comparator arrangement 12 to which a reference signal is supplied at a reference signal input 13. In a preferably periodically recurring sampling interval during the occurrence of a given reference level in the picture signal, the comparator arrangement 12 forms a control signal from the value of the measuring signal fed to the measuring signal input 11 at this time, on the one hand, and the reference signal, on the other, by means of substraction and delivers this at a control signal output 14. From there the control signal is fed to an input 15 of a control signal memory 16 and is stored in the latter. The control signal is fed via an output 17 of the control signal memory 16 to a second input 18 of combinatorial stage 2 in which it is combined with the picture signal, e.g. added to it.
The combinatorial stage 2, the controllable amplifier 5, the measuring stage 7, the comparator arrangement 12 and the control signal memory 16 form a control loop with which the beam current is guided towards the reference signal in the sampling interval during the occurrence of the reference level in the picture signal. For the reference level, use is made in particular of a black level or a level with small, fixed distance from the black level, i.e. a value in the picture signal which produces a black or almost back picture area in the displayed picture in the picture tube. In this case the control loop, as described, forms a cut-off point control for the picture tube. If the reference level is away from the black level, the control loop is also designated as quasi-cut-off-point control.
The circuit arrangement as shown in FIG. 1 also has a trigger circuit 19 to which the measuring signal from the measuring signal output 10 of measuring stage 7 is fed at a measuring signal input 20. When the circuit arrangement and therefore the picture tube are turned on, the trigger circuit 19 is set in a first state in which by means of a first connection 21 it blocks the comparator arrangement 12 in such a way that the latter delivers no control signal or a control signal with the value zero at its control signal output 14. This prevents the control signal memory 16 from storing undefined values for the control signal at the moment of turning on or immediately thereafter.
The circuit arrangement shown in FIG. 1 also has a logic network 22 which is connected via a second connection 23, by means of which a switching signal is supplied, with the trigger circuit 10 and via a third connection 24 with the controllable amplifier 5. Like the trigger circuit 19, the logic network 22 also finds itself controlled, when the circuit arrangement is being turned on, by the switching signal in a first stage in which by way of the third connection 24 it blocks the controllable amplifier 5 with a blocking signal in such a way that no beam currents controlled by the picture signal can yet flow in the picture tube 9. Thus the picture tube 9 is blanked; no picture is displayed yet.
When picture tube 9 is turned on, the hot cathode 8 is still cold so that no beam current can flow anyhow. The hot cathode 8 is then heated up and, after a certain time, begins gradually to emit electrons as the result of which a cathode ray and therefore a beam current can form. However, during the heating up of the hot cathode 8, and because the cut-off point control has not yet responded, this would be undefined and is therefore suppressed by the controllable amplifier 5. Only in time intervals which are provided immediately subsequent to flybacks of the cathode rays into an initial position at the changeover from the display of one image to that of a subsequent image, but even before the start of the display of the subsequent image, the controllable amplifier 5 delivers a voltage in the form of an auxiliary pulse for a short time at its output 6, and when the hot cathode 8 in the picture tube 9 is heated up sufficiently, this voltage produces a beam current. The time interval for the delivery of this voltage is selected in such a way that a cathode ray produced by its does not produce a visible image in the picture tube 9, and coincides for example with the sampling interval.
The measuring stage 7 measures the short-time cathode current produced in the manner described and, at its measuring signal output 10, delivers a corresponding measuring signal which is passed via measuring signal output 20 to the trigger circuit 19. If the measuring signal exceeds a definite preset threshold value, the trigger circuit 19 is switched into a second state in which it releases the comparator arrangement 12 via the first connection 12 and, by means of the second connection 23, uses the switching signal to also bring the logic network 22 into a second state. The comparator arrangement 12 now evaluates the measuring signal supplied to it via the measuring signal input 11, i.e. it forms the control signal as the difference between the measuring signal and the reference signal supplied via the reference signal input 13. The control signal is transferred via the control signal output 14 and the input 15 into the control signal memory 16. It is subsequently fed via the output 17 of the control signal memory 16 to the second input 18 of the combinatorial stage 2 and is there combined with the picture signal at the first input 1, e.g. is superimposed on it by addition. This superimposed picture signal is fed to the picture signal input 4 of the controllable amplifier 5 via the output 3 of the combinatorial stage 2.
In the second state of the logic network 22 the controllable amplifier 5 is switched via the third connection 24 by the blocking signal in such a way that the picture signal controls the beam currents only during the sampling intervals and that, for the rest, no image appears yet in the picture tube. The cut-off point control now gebins to respond, i.e. the value of the control signal is changed by the control loop comprising the combinatorial stage 2, the controllable amplifier 5, the measuring stage 7, the comparator arrangement 12 and the control signal memory 16 until such time as the beam current in the picture tube 9 at the blocking point or at a fixed level with respect to it is adjusted to a value preset by the reference signal. For this purpose the sampling interval, in which the picture signal controls the beam current via the controllable amplifier 5 is selected in such a way that within it the picture signal just assumes a value corresponding to the cut-off point or to a fixed level with respect to it.
During the response of the cut-off point control the control signal fed to the control signal memory 16 changes continuously. Between the control signal output 14 of the comparator arrangement 12 and the input 15 of the control signal memory 16 is inserted a changed detection arrangement 25 which detects the variations of the control signal. When the cut-off point control has responded, i.e. the control signal has assumed a constant value, the change detection arrangement 25 delivers a change signal at an output 26 which indicates that the steady stage of the cut-off point control is achieved and the said signal is fed to a change signal input 27 of the logic network 22. The logic network then switches into a third state in which via the third connection 24 it enables the controllable amplifier 5 in such a way that the beam currents are now controlled without restriction by the picture signal. Thus a correctly represented picture appears in the picture tube 9.
A shadow-like representation of individual constituents of the circuit arrangement in FIG. 1 is used to indicate a modification by which this circuit arrangement is equipped for the representation of color pictures in the picture tube 9. For example, three color signals are fed in this case as the picture signal via the input 1 to the combinatorial stage 2. Accordingly, the input 1 is shown in triplicate, and the combinatorial stage 2 has a logic element, e.g. an adder, for example of these color signals. The controllable amplifier 5 now has three amplifier stages, one for each of the color signals, and the picture tube now contains three hot cathodes 8 instead of one so that three independent cathode rays are available for the three color signals.
However, to simplify the circuit arrangement and to save on components, only one measuring stage 7 is provided which measures all three beam currents successively. Also, the comparator arrangement 12 forms part control signals from the successively arriving part measuring signals for the individual beam currents with the reference signal, and these part control signals are allocated to the individual color signals and passed on to three storage units which are contained in the control signal memory 16. From there, the part control signals are sent via the second input 18 of the combinatorial stage 2 to the assigned logic elements.
The circuit arrangement thus forms three independently acting control loops for the cut-off point control of the individual color signals, in which case only the measuring stage 7 and to some extent at least the comparator arrangement 12 are common to these control loops.
The change detection arrangement 25 now has three change detectors each of which detects the changes with time of the part control signals relating to a color signal. Then via the output 26 each of these change detectors delivers a part change signal to the change signal input 27 of the logic network 22. These part change signals occur independently of one another when the relevent control loop has responded. The logic network 22 evaluates all three part change signals and does not switch into its third stage until all part change signals indicate a steady state of the control loops. Only then, in fact, is it ensured that all the color signals from the beam currents controlled by them are correctly reproduced in the picture tube, and thus no distortions of the displayed image, especially no color purity errors, occur. The color picture displayed then immediately has the correct brightness and color on its appearance when the picture tube is turned on.
FIG. 2 shows a somewhat more detailed block circuit diagram of an embodiment of a circuit arrangement equipped for the processing of a picture signal containing three colour signals. Three color signals for the representation of the colors red, green and blue are fed to this circuit arrangement via three input terminals 101, 102, 103. A red color signal is fed via the first input terminal 101 to a first adder 201, a green colour signal is fed via the second input terminal to a second adder 202, and a blue colour signal is fed via the third input terminal 103 to a third adder 203. From outputs 301, 302 and 303 of the adders 201, 202, 203 the color signals are fed to amplifier stages 501, 502 and 503 respectively. Each of the amplifier stages contains a switchable amplifier 511, 512 and 513, an output amplifier 521, 522 and 523 as well as a measuring transistor 531, 532 and 533 respectively. The emitters of these measuring transistors 531, 532, 533 are each connected to a hot cathode 801, 802, 803 of the picture tube 9 and deliver the cathode currents, whereas the collectors of measuring transistors 521, 532, 533 are connected to one another and to a first terminal 701 of a measuring resistor 702 the second terminal of which 703 is connected to earth. The current gain of the measuring transistors 531, 532 and 533 is so great that their collector currents coincide almost with the cathode currents. By measuring the voltage drop produced by the cathode currents at the measuring resistor 802 it is then possible to measure the cathode currents and therefore the beam currents in the picture tube 9 with great accuracy.
The falling voltage at the measuring resistor 702 is fed as a measuring signal to an input 121 of a buffer amplifier 120 with a gain factor of one, at the output 122 of which the unchanged measuring signal is therefore available at low impedance. From there it is fed to a first terminal 131 of a reference voltage source 130 which is connected with its second terminal 132 to inverting inputs 111, 112 and 113 of three differential amplifiers 123, 124, 125 respectively. The differential amplifiers 123, 124, 125 also each have a non-inverting input 114, 115, and 116 respectively. These are connected to each other at a junction 117, to earth via a leakage current storage capacitor 126 and to the output 122 of the buffer amplifier 120 via decoupling resistor 118 and a leakage current sampling switch 119. In addition, the input 121 of the buffer amplifier 120 can be connected to earth via a short-circuiting switch 127.
The operation of the comparator arrangement 12 which consists mainly of the buffer amplifier 120, the reference voltage source 130 and differential amplifiers 123, 124, 125 will be explained below with the aid of the pulse diagrams in FIG. 3. FIG. 3a shows a horizontal blanking signal for a television signal which, as the picture signal, controls the beam currents in the picture tube 9. In this diagram, H represents horizontal blanking pulses which follow one another in the picture signal at the time interval of one line duration and by means of which the beam currents are switched off during line flyback between the display of the individual picture lines in the picture tube. FIG. 3b shows a vertical blanking pulse V by means of which the beam currents are switched off during the change ober from the display of one picture to the display of the next picture. FIG. 3c shows a measuring signal control pulse VH which is formed from a vertical blanking pulse lengthened by three line duration.
The short-circuiting switch 127 is now controlled in such a way that it is non-conducting only throughout the duration of the measuring signal control pulse VH and during the remaining time short-circuits the input 121 of the buffer amplifier 120 to earth. This means that a measuring signal only reaches the comparator arrangement 12 during frame change so that the parts of the picture signal which control the beam currents producing the picture in the picture tube exert no influence on comparator arrangement 12 and therefore on the blocking point control.
Throughout the duration of the measuring signal control pulse VH, the measuring signal from output 122, reduced by a reference voltage issued by the reference voltage source 130 between its first 131 and its second terminal 132, is present at the inverting inputs 111, 112, 113 of differential amplifiers 123, 124, 125. If the differential amplifiers 123, 124, 125 were not present, this difference would be fed directly as part control signals to the control signal storage capacitors 161, 162, 162. The differential amplifiers 123, 124, 125 amplify the difference and thus form the control amplifiers of the control loops.
The comparator arrangement 12 further contains a device for compensation of the influence of any leakage currents occurring in the picture tube 9. For this purpose, a voltage to which the leakage current storage capacitor 126 is charged is fed to the non-inverting inputs 114, 115, 116 of the three differential amplifiers 123, 124 and 125. The charging is performed by the measuring signal from output 122 of the buffer amplifier 120 via the decoupling resistor 118 and the leakage current sampling switch 119 which is closed only within the period of the vertical blanking pulse V, and in certain cases only during part of the latter. Within this time the beam currents are, in fact, totally switched off by the picture signal so that in certain cases only a leakage current flows through the measuring resistor 702. Consequently, throughout the duration of the vertical blanking pulse V the measuring signal corresponds to this leakage current. Because the leakage current also flows during the remaining time, even outside the duration of the vertical blanking pulse the measuring signal contains a component originating from the leakage current which therefore is also contained in the voltage fed to the inverting inputs 111, 112, 113 of differential amplifiers 123, 124, 125 and is subtracted out in the differential amplifiers 123, 124, 125.
The part control signal is fed from output 141 of differential amplifier 123 by the first control signal sampling switch 154 to the first terminal 151 of the first control signal storage capacitor 161 during the period of a storage pulse L1 and is stored in the said capacitor. Similarly, the part control signal from output 143 of differential amplifier 125 is fed to the third control signal storage capacitor 163 during the period of a storage pulse L2 and the part control signal from output 142 of differential amplifier 124 is fed to the second control signal storage capacitor 162 during a storage pulse L3. The storage pulses L1, L2 and L3 are illustrated in FIGS. 3d, e and f. They lie in sequence in one of the three line periods by which the measuring signal control pulse VH is longer than the vertical blanking pulse V. These three line periods form the sampling interval for the measuring signal or the part measuring signals, as the case may be. During the remaining periods the outputs, 141, 152, 143 of the differential amplifiers 123, 124, 125 are isolated from the control signal storage capacitors 161, 162, 163 so that no interference can be transmitted from there and any distortion of the stored part control signals caused thereby is eliminated. For the duration of storage pulses L1, L2 and L3 the color signals at the input terminals 101, 102, 103 are at their reference level i.e. in the present embodiment at a level, corresponding to the blocking point or at a fixed level with respect to it so that the control loops can adjust to this level.
The switchable amplifiers 511, 512, and 513 each receive at each input 241, 242, 243 a blanking signal BL1, BL2, BL3 respectively, the curves of which are shown in FIGS. 3g, h, i. These blanking signals interrupt the supply of the color signals during line flybacks and frame change, i.e. during the period of the measuring signal control pulse VH, and thus the beam currents in these time intervals are switched off. Naturally, the red color signal is let through during the first line period after the end of the vertical blanking pulse V, the blue color signal during the second line period after the end of the vertical blanking pulse V and the green color signal during the third line period after the end of the vertical blanking pulse V by the switchable amplifiers 511, 512, 513 respectively so that they can control the beam currents. Blanking signals BL1, BL2 and BL3 also provide for interruptions in the frame change blanking pulse, which corresponds to the measuring signal control pulse, in the corresponding time intervals. In these time intervals the beam currents are measured and part control signals are determined from the part measuring signals and stored in the control signal storage capacitors 161, 162, 163.
The circuit arrangement shown in FIG. 2 further contains a trigger circuit 19 to which a supply voltage is fed via a supply terminal 190. Via a reset input 191 a voltage is also supplied to the trigger circuit 19 from a third terminal 133 of the reference voltage source 130. When the circuit arrangement is turned on, this voltage is designed so as to be delayed with respect to the supply voltage so that when the circuit arrangement is brought into operation the interplay of the two voltages produces a switch-on reset signal such that a low-value voltage pulse occurs at the reset input 191 during turn on, which means that the trigger circuit 19 is set in its first state. The reset input 191 can also be connected to another circuit of any configuration which generates a switch-on reset signal when the picture tube is turned on.
The trigger circuit 19 is further connected via a second connection 23 to a logic network 22 which, when the circuit arrangement is turned on, is also set into a first state via the second connection 23. In this first state the logic network 22 delivers a blocking signal at a blocking output 240 which is fed to the three switchable amplifiers 511, 512, 513. By this means the supply of the color signals to the output amplifiers 521, 522, 523 is interrupted completely so that no beam currents can be generated by these. No picture is therefore displayed.
An insertion signal EL which extends over the three line periods by which the measuring signal control pulse VH is longer than the vertical blanking pulse V, i.e. over the sampling interval, is also fed via a line 233 to the trigger circuit 19 and the logic network 22. As long as the trigger circuit 19 is in its first state, this insertion pulse EL is issued via a control output 192 from the trigger circuit 19 and fed to the pulse generator 244. During the period of the insertion pulse EL this generator produces a voltage pulse of a definite magnitude and passes this to output amplfiiers 521, 522, 523 as an auxiliary pulse via switching diodes 245, 246, 247. By this means the beam currents are switched on for a short time so as to receive a measuring signal despite the disconnected color signals as soon as at least one of the hot cathodes 801, 802, 803 delivers a beam current.
In its first state the trigger circuit 19 also delivers a signal via a control line 211, and this signal is used to switch the outputs 141, 142, 143 of the differential amplifiers 123, 124, 125 to earth potential or practically to earth potential. This suppresses effects of voltages at the inputs 111 to 116 of the differential amplifiers 123, 124, 125, especially effects of the reference voltage source 130 which may in some cases initiate incorrect charging of the control signal storage capacitors 161, 162, 163.
The measuring signal produced by means of the pulse generator 244 at the input 121 of the buffer amplifier 120 is also fed to the trigger circuit 19 via a measuring signal input 20. If it exceeds a preset threshold value, the trigger circuit 19 switched into its second state. The logic network 22 is then also switched into its second state via the second connection 23. The differential amplifiers 123, 124, 125, too, are triggered by the signal along the control line 211 into issuing a control signal defined by the difference in the voltages at its inputs 111 to 116. The pulse generator 244 is blocked by the control output 192. The blocking signal issued from the blocking output 240 of the logic network 22 now turns on the switchable amplifiers 511, 512, 513 in the time intervals defined by the storage pulses L1, L2, L3 in such a way that in these time intervals the color signals can produce beam currents to form a measuring signal by which the control loops respond. However, the display of the picture is still suppressed. The control signal storage capacitors 161, 162, 163 are charged up in this process. In the leads to the first terminals 151, 152, 153 there are change detectors 251, 252, 253 which detect the changes of the charging currents of the control signal storage capacitors 161, 162, 163 and at their outputs 261, 262, 263 in each case deliver a part change signal when the charging current of the control signal storage capacitor in question has decayed and thus the relevant control loop has responded. The part change signals are fed to three terminals 271, 272, 273 of the change signal input 27 of the logic network 22.
When part change signals are present from all change detectors 251, 252, 253, when therefore all control loops have responded, the logic network 22 switches from its second to its third state. The blocking signal from the blocking output 240 is now completely disconnected such that the switchable amplifiers 511, 512, 513 are now switched only by the blanking signals BL1, BL2, BL3. The colour signals are then switched through to the output amplifiers 521, 522, 523 and the picture is displayed in the picture tube.
FIG. 4 shows an embodiment for a trigger circuit 19 and a logic network 22 of the circuit arrangements as shown in FIGS. 1 or 2. The trigger circuit 19 contains a flip-flop circuit formed from two NAND-gates 194, 195 to which the switch-on reset signal, by which the trigger circuit 19 is returned to its first stage, is fed via the reset input 191. All the elements of the circuit arrangement in FIG. 4 are shown in positive logic. Thus, a short-time low voltage at the reset input 191 immediately after the circuit arrangement is started up is used to set the flip-flop circuit 194, 195 in such a way that a high voltage occurs at the output of the second NAND gate 194 and a low voltage at the output of the second NAND gate 195. The low voltage at the output of the second NAND gate 195 blocks differential amplifiers 123, 124, 125 via the control line 211 in the manner described.
The insertion pulse EL is fed via the line 233 to the trigger circuit 19, is combined via an AND gate 196 with the signal from the output of the first NAND gate 194 and is delivered at the control output 192 for the purpose of controlling the pulse generator 244.
The signals from the outputs of the NAND-gates 194, 195 are fed via a first line 231 and a second line 232 of the second connection 23 as a switching signal to the logic network 22. The first line 231 is connected to reset inputs R of three part change signal memories 221, 222, 223 in the form of bistable flip-flop circuits which when the circuit arrangement is started up are reset via the first line 231 in such a way that they carry a low voltage at their outputs Q. The second line 232 of the second connection 23 leads via three AND gates 224, 225, 226 to setting inputs S of the three part change signal memories 221, 222, 223. By means of the AND gates 224, 225, 226 the signal on the second line 232 of the second connection 23 is combined each time with one of the part change signals supplied via the terminals 271, 272, 273. The signals from the outputs Q of the part change signal memories 221, 222, 223 are combined by means of a collecting gate 227 in the form of an NAND gate and are held ready at its output 228.
The measuring signal is fed to the trigger circuit 19 via the measuring signal input 20 and passed to a first input 197 of a threshold detector 198 to which at a second input a threshold value, in the form of a threshold voltage for example, produced by a threshold generator 199 is also supplied. When the voltage at the first input 197 of the threshold detector 198 is smaller than the voltage delivered by the threshold generator 199, the threshold detector 198 delivers a high voltage at its output 200. When, on the other hand, the voltage at the first input 197 is greater than the voltage of the threshold generator 199, the voltage at the output 200 jumps to a low value. This voltage is supplied as the setting signal of the flip-flop circuit 194, 195, reverses the latter and thereby switches the trigger circuit 19 into its second state when the voltage at the first input 197 exceeds the voltage of the threshold generator 199.
Between the output 200 and the flip-flop circuit 194, 195 in the circuit arrangement shown in FIG. 4 there is inserted an inquiry gate 181 in the form of an OR gate to which an inquiry pulse is fed via an inquiry input 193 of the trigger circuit 19. This ensures that the flip-flop circuit 194, 195 is switched over only at a time fixed by the inquiry pulse--in the present case a negative voltage pulse--and not at any other times due to disturbances. As such an inquiry pulse it is possible to use, for example, a pulse which occurs in the second line period after the end of the vertical blanking pulse V, i.e. one which largely corresponds to the storage pulse L2.
After the switching over of the flip-flop circuit 194, 195 corresponding to the setting of the trigger circuit 19 into the second state, appropriately modified signals are supplied via the control line 211 and the output 192 for the purpose of controlling the pulse generator 244 and the differential amplifiers 123, 124, 125. Modified voltages also appear on the lines 231, 232 of the second connection 23, and these voltages release the part change signal memories 221, 222, 223 such that they can each be set when the part change signals reach the terminals 271, 272, 273.
In certain cases, a further flip-flop circuit 234 is inserted in the lines 231, 232 to delay the signals passing along these lines; this is reset via the first line 231 when the circuit arrangement is started up and thus it also resets the part change signal memories 221, 222, 223. However, after the trigger circuit 19 is switched into the second state the further flip-flop circuit 234 is not set via the second line 232 of the second connection 23 until a release pulse arrives via a release input 235 and another AND gate 236, for example a period of approximately the interval of two vertical blanking pulses V after the switching of the trigger circuit 19 into the second state. In this way it is possible to bridge a period of time in which no defined signal values are present at the terminals 271, 272, 273.
The signal at the output 228 of the collecting gate 227 changes its state when the last of the three part change signals has also arrived and has set the last of the three part change signal memories. The signal is then combined via a gate arrangement 229 of two NAND gates and one AND gate with the insertion pulse EL of line 223 and with the signal on the second line 232 of the second connection 23 or from the output Q of the further flip-flop circuit 234 to the blocking signal delivered at the blocking output 24 which is fed to the switchable amplifiers 511, 512, 513.
The circuit arrangement described is designed in such a way that the trigger circuit 19 remains in its second state and logic network 22 remains in its third state even if charging currents reappear at the difference signal storage cpacitors 161, 162, 163 due to disturbances during the operation of the circuit arrangement. The cutoff point control then makes readjustments without the displayed picture being disturbed.
In the circuit arrangement shown in FIG. 2, the green color signal can also be let through during the second line period after the end of the vertical blanking pulse V and the blue color signal during the third line period after the end of the vertical blanking pulse V by the switchable amplifiers 511, 512, 513 for the purpose of controlling the beam currents. The storage pulses L2 and L3 at the control signal sampling switches 155 and 156 and the second and third blanking signals BL2 and BL3 at the blanking inputs 242 and 243 are then to be interchanged. The resulting insertion signals A2 and A3 as shown in FIGS. 3m and n are also interchanged then accordingly.
In FIG. 2 a dashed line is used to indicate which components of the circuit arrangement can be combined advantageously to form an integrated circuit. The first terminals 151, 152, 153 of the difference signal storage capacitors 161, 162, 163, one terminal 128 of leakage current storage capacitor 126, three terminals 524, 525, 526 in the leads to the output amplifiers 521, 522, 523 as well as a line connection 704 between the first terminal 701 of the measuring resistor 702 and the input 121 of the buffer amplifier 120 will then form the connecting contacts of this integrated circuit
- Deflection power processing with TEA2029C
The TEA2029C is a complete (horizontal and vertical)
deflection processor with secondary to primary
SMPS control for color TV sets.
DEFLECTION .CERAMIC 500kHz RESONATOR FREQUENCY
REFERENCE .NO LINE AND FRAME OSCILLATOR ADJUSTMENT
.DUAL PLL FOR LINE DEFLECTION .HIGH PERFORMANCE SYNCHRONIZATION .SUPER SANDCASTLE OUTPUT .VIDEO IDENTIFICATION CIRCUIT .AUTOMATIC 50/60Hz STANDARD IDENTIFICATION
.EXCELLENT INTERLACING CONTROL .SPECIALPATENTED FRAME SYNCHRO DEVICE
FOR VCR OPERATION .FRAME SAW-TOOTH GENERATOR .FRAME PHASE MODULATOR FOR THYRISTOR
SMPS CONTROL .ERROR AMPLIFIER AND PHASE MODULATOR
.SYNCHRONIZATION WITH HORIZONTAL
DEFLECTION .SECURITY CIRCUIT AND START UP PROCESSOR
.OUTPUT PULSES ARE SENT TO THE PRIMARY
SMPS IC (TEA2261) THROUGH A
LOW COST TRANSFORMER
This integrated circuit uses I2L bipolar technology
and combines analog signal processing with digital
Timing signals are obtainedfrom a voltage-controlled
oscillator (VCO) operatingat 500KHzby means
of a cheap ceramic resonator. This avoids the
frequency adjustment normally required with line
and frame oscillators.
A chain of dividers and appropriate logic circuitry
produce very accurately defined sampling pulses
and the necessary timing signals.
The principal functions implemented are :
- Horizontal scanning processor.
- Frame scanning processor. Two applications are
- D Class : Power stage using an external
- B Class : Powerstageusing an externalpower
amplifier with fly-back generator
such as the TDA8170.
- Secondary switch mode power regulation.
The SMPS output synchronize a primary I.C.
(TEA2260/61)at the mains part.
This concept allows ACTIVE STANDBY facilities.
- Dual phase-locked loop horizontal scanning.
- High performance frameand line synchronization
with interlacing control.
- Video identification circuit.
- Super sandcastle.
- AGC key pulse output.
- Automatic 50-60Hz standard identification.
- VCR input for PLL time constant and frame synchro
- Frame saw-tooth generator and phase modulator.
- Switchingmode regulated power supplycomprising
error amplifier and phase modulator.
- Security circuit and start-up processor.
The circuit is supplied in a 28 pin DIP case.
VCC = 12V.
Line synchronization separator is clamped to
black level of input video signal with synchronization
pulse bottom level measurement.
The synchronization pulses are divided centrally
between the black level and the synchronization
pulse bottom level, to improve performance on
video signals in noise conditions.
Frame synchronization is fully integrated (no external
The frame timing identification logic permits automatic
adaptation to 50 - 60Hz standards or non-interlaced
An automatic synchronization window width system
- fast frame capture (6.7ms wide window),
- good noise immunity (0.4ms narrow window).
The internal generator starts the discharge of the
saw-tooth generator capacitor so that it is not disturbed
by line fly back effects.
Thanks to the logic control, the beginning of the
charge phase does not depend on any disturbing
effect of the line fly-back.
A 32ms timing is automatically applied on standardized
transmissions, for perfect interlacing.
In VCR mode, the discharge time is controlled by
an internal monostable independent of the line
frequency and gives a direct frame synchronization.
The horizontalscanningfrequencyis obtainedfrom
the 500kHz VCO.
The circuit uses two phase-locked loops (PLL) :
the first one controls the frequency, the second one
controls the relative phase of the synchronization
and line fly-back signals.
The frequency PLL has two switched time constants
to provide :
- capture with a short time constant,
- good noise immunity after capture with a long
The output pulse has a constant duration of 26ms,
independent of VCC and any delay in switching off
the scanning transistor.
The horizontal synchronization signal is sampled
by a 2ms pulse within the synchronization pulse.
The signal is integrated by an external capacitor.
The identification function provides three different
- 0V : no video identification
- 6V : 60Hz video identification
- 12V : 50Hz video identification
This information may be used for timing research
in the case of frequency or voltage synthetizer type
receivers, and for audio muting.
Super Sandcastle with 3 levels : burst, line flyback,
In the event of vertical scanning failure, the frame
blanking level goes high to protect the tube.
Frame blanking time (start with reset of Frame
divider) is 24 lines.
This provides for continuous use of the short time
constant of the first phase-locked loop (frequency).
In VCR mode, the frame synchronization window
widens out to a search window and there is no
delay of frame fly-back (direct synchronization).
FRAME SAW-TOOTH GENERATOR. The current
to charge the capacitoris automatically switched to
60Hz operation to maintain constant amplitude.
FRAME PHASE MODULATOR (WITH TWO DIFFERENTIAL
INPUTS). The output signal is a pulse
at the line frequency, pulse width modulatedby the
voltage at the differential pre-amplifier input.
This signal is used to control a thyristor which
provides the scanning current to the yoke. The
saw-tooth output is a low impedance,however, and
can therefore be used in class B operation with a
power amplifier circuit.
Switch Mode Power Supply (SMPS) Secondary
to Primary Regulation
This power supply uses a differential error amplifier
with an internal reference voltage of 1.26V and a
phase modulator operating at the line frequency.
The powertransistor is turnedoff bythe falling edge
of the horizontal saw-tooth.
The ”soft start” device imposes a very small conduction
angle on starting up, this angle progressively
increases to its nominal regulation value.
The maximum conductionangle may be monitored
by forcing a voltage on pin 15. This pin may also
be used for current limitation.
The outputpulse is sent to the primaryS.M.P.S. I.C.
(TEA2261) via a low cost synchro transformer.
Security Circuit and Start Up Processor
When the security input (pin 28) is at a voltage
exceeding 1.26V the three outputs are simultaneously
cut off until this voltagedrops below the 1.26V
threshold again. In this case the switch mode
power supply is restarted by the ”soft start” system.
If this cycle is repeated three times, the three
outputs are cut off definitively. To reset the safety
logic circuits, VCC must be zero volt.
This circuit eliminates the risk to switch off the TV
receiver in the event of a flash affecting the tube.
On starting up, the horizontal and vertical scanning
functions come into operation at VCC = 6V. The
power supply then comes into operation progressively.
On shutting down, the three functions are interrupted
simultaneously after the first line fly-back.
- Audio IC TDA8190.
The TDA8190is a completeTV soundchannel with
DC tone and volume controls plus separate VCR
input and output connections. Mounted in a Powerdip
16 + 2 + 2 package, the device delivers an
output power of 4W into 16W (d = 10%, Vs = 24V)
or 1.5W into 8W (d = 10%, Vs = 12V). Included in
the TDA8190 are : IF amplifier limiter, active lowpass
filter, AF pre-amplifier and power amplifier,
turn-off muting, mute circuit and thermal protection.
High output, high sensitivity, excellentAM rejection
and low distortion make the device suitable for use
in TVs of almost every type. Further, no screening
is necessary because the device is free of radiation
MIVAR 17L1 BIFONICO CHASSIS TV3492 SDA2516 EAROM (MIVAR SC10)
Features- Word-organized reprogrammable nonvolatile memory
in n-channel floating-gate technology (E2PROM)
- 128 ´ 8-bit organization
- Supply voltage 5 V
- Serial 2-line bus for data input and output (I2C Bus)
- Reprogramming mode, 10 ms erase/write cycle
- Reprogramming by means of on-chip control (without
- Check for end of programming process
- Data retention > 10 years
- More than 104 reprogramming cycles per address
- Compatible with SDA 2516. Exception:
Conditions for total erase and current consumption.
I2C Bus Interface
The I2C Bus is a bidirectional 2-line bus for the transfer of data between various integrated circuits.
It consists of a serial data line SDA and a serial clock line SCL. The data line requires an external
pull-up resistor to VCC (open drain output stage).
The possible operational states of the I2C Bus are shown in figure 1. In the quiescent state, both
lines SDA and SCL are high, i.e. the output stage of the data line is disabled. As long a SCL remains
"1", information changes on the data bus indicate the start or the end of data transfer between two
The transition on SDA from "1" to "0" is a start condition, the transition from "0" to "1" a stop
condition. During a data transfer the information on the data bus will only change while the clock line
SCL is "0". The information on SDA is valid as long as SCL is "1".
In conjunction with an I2C Bus system, the memory component can operate as a receiver and as a
transmitter (slave receiver or slave transmitter). Between a start and stop condition, information is
always transmitted in byte-organized form. Between the trailing edge of the eighth clock pulse and a ninth acknowledge clock pulse, the memory component sets the SDA line to low as a confirmation
of reception, if the chip select conditions have been met. During the output of data, the data output
of the memory is high in impedance during the ninth clock pulse (acknowledge master).
The signal timing required for the operation of the I2C Bus is summarized in figure 2.
Control Functions of the I2C Bus
The memory component is controlled by the controller (master) via the I2C Bus in two operating
modes: read-out cycle, and reprogramming cycle, including erase and write to a memory address.
In both operating modes, the controller, as transmitter, has to provide 3 bytes and an additional
acknowledge clock pulse to the bus after the start condition. During a memory read, at least nine
additional clock pulses are required to accept the data from the memory and the acknowledge
master, before the stop condition may follow. In the case of programming, the active programming
process is only started by the stop condition after data input (see figure 3).
The chip select word contains the 3 chip select bits CS0, CS1 and CS2, thus allowing 8 memory
chips to be connected in parallel. Chip select is achieved when the three control bits logically
correspond to the selected conditions at the select inputs.
Check for End of Programming or Abortion of Programming Process
If the chip is addressed during active reprogramming by entering CS/E, the programming process
is terminated. If, however, it is addressed by entering CS/A, the entry will be ignored. Only after
programming has been terminated will the chip respond to CS/A. This allows the user to check
whether the end of the programming process has been reached (see figure 3).
After the input of the first two control words CS/E and WA, the resetting of the start condition and the
input of a third control word CS/A, the memory is set ready to read. During acknowledge clock
nine, the memory information is transferred in parallel mode to the shift register. Subsequent to the
trailing edge of the acknowledge clock, the data output is low impedance and the first data bit can
be sampled, (see figure 4).
With every shift clock, an additional bit reaches the output. After reading a byte, the internal address
counter is automatically incremented when the master receiver switches the data line to “low” during
the ninth clock (acknowledge master). Any number of memory locations can thus be read one after
the other. At address 128, an overflow to address 0 is not initiated. With the stop condition, the data
output returns to high-impedance mode. The internal sequence control of the memory component
is reset from the read to the quiescent with the stop condition.
The reprogramming cycle of a memory word comprises an erase and a subsequent write process.
During erase, all eight bits of the selected word are set into "1" state. During write, "0" states are
generated according to the information in the internal data register, i.e. according to the third input
After the 27th and the last clock of the control word input, the active programming process is started
by the stop condition. The active reprogramming process is executed under onchip control.
The time required for reprogramming depends on component deviation and data patterns.
Therefore, with rated supply voltage, the erase/write process extends over max. 20 ms, or more
typically, 10 ms. In the case of data word input without write request (write request is defined as data
bit in data register set to “0”), the write process is suppressed and the programming time is
shortened. During a subsequent programming of an already erased memory address, the erase
process is suppressed again, so that the reprogramming time is also shortened.
TDA8170 TV VERTICAL DEFLECTION OUTPUT CIRCUIT:DESCRIPTION
The TDA8170 is a monolithic integrated circuit in
HEPTAWATTTM package. It is a high efficiency
power booster for direct driving of verticalwindings
of TV yokes. It is intended for use in Colour and B
&Wtelevision receivers as well as in monitorsand
The functions incorporated are :
The power dissipated in the circuit must be removed
by adding an external heatsink.
Thanks to the HEPTAWATTTM package attaching
the heatsink is very simple, a screwa compression
spring (clip) being sufficient. Betweenthe heatsink
andthe packageit isbetter to insert a layerof silicon
grease, to optimizethe thermal contact ; no electrical
isolation is needed between the two surfaces.
Symbol Parameter Value Unit
VS Supply Voltage (pin 2) 35 V
V5, V6 Flyback Peak Voltage 60 V
V3 Voltage at Pin 3 + Vs
V1, V7 Amplifier Input Voltage + Vs, – 0.5 V
Io Output Peak Current (non repetitive, t = 2 msec) 2.5 A
Io Output Peak Current at f = 50 or 60 Hz, t 3 10 msec 3 A
Io Output Peak Current at f = 50 or 60 Hz, t > 10 msec 2 A
I3 Pin 3 DC Current at V5 < V2 100 mA
I3 Pin 3 Peak to Peak Flyback Current at f= 50 or 60 Hz, tfly 31.5msec 3 A
Ptot Total Power Dissipation at Tcase = 90 °C 20 W
Tstg, Tj Storage and Junction Temperature – 40, +150 °C