PHILIPS I17T766 CHASSIS 3119 203 30664 INTERNAL VIEW.

The chassis is mainly based on discretes and of 2 asic's.

- TBA800

- TCA511B

The power supply unit is fitted on the right bottom side of the cabinet and the power transformer is fitted on the bottom part of the cabinet.

( Note one of my screwdrivers is even TRYING to be on the scene of the "Obsolete" show........as you can see it won't  stay away.............)

PHILIPS  I17T766  CHASSIS  3119 203 30664 B-W TELEVISION DIAGRAM AND DEFLECTION CIRCUIT:

A unidirectional conductive device is coupled from a base terminal to a collector terminal of a horizontal deflection output transistor in a television receiver and poled in a direction to prevent the transistor from saturating when it is driven into its conductive state during a portion of each deflection cycle. Biasing means is coupled to the diode to preselect the desired operating voltage of the transistor during its conduction period.

1. In a television receiver, a deflection circuit comprising: 2. A circuit as defined in claim 1 wherein said transformer is an auto-transformer and said second terminal is intermediate said first and 3. In a television receiver, a deflection circuit comprising:

Description:

The present invention relates to transistorized deflection circuits utilized in television receivers.

In present day transistor deflection circuits, for example, those used in the horizontal output stage of a television receiver; the output transistor is normally operated in a switching mode, that is, the transistor is driven into saturation during a trace interval of each deflection cycle and driven out of conduction during the retrace portion of each deflection cycle. By operating the transistor in its saturation region, average power losses are minimized. With saturated operation, however, the accumulation of minority carriers in the base region will effect a continuation in the flow of collector current after the trace interval during the initial portion of the retrace interval while the transistor is being driven into its non-conducting state. In addition to causing this undesirable delay time in turning off the transistor, losses occurring during this period may be localized in small areas commonly referred to as "hot spots." These losses are characterized in being regenerative and tend to cause second breakdown of the device. This effect is explained in greater detail in a paper authored by the present inventor and entitled "Thermal Regeneration in Power Dissipating Elements" which appeared in "The Electronic Engineer" publication in the January 1967 issue. Although operating the horizontal output transistor in its saturated region may reduce the average power dissipated in this device during its conduction interval, it increases the possibility of second breakdown during the turn-off time. With the advent of high voltage (1,500 volts) transistors, it is possible to develop the necessary output energy utilizing one of these transistors which can be operated in a non-saturated mode. The circuit of the present invention insures that the deflection output transistor will not be driven into saturation.

Certain low power transistor switching circuits, such as employed in computer applications, have utilized diodes in conjunction with resistive biasing means coupled between the base and collector terminals to prevent the transistor from saturating and thereby increase the maximum switching frequency of the circuit by reducing the turn-off time of the device.

In the solid state deflection art, however, it is desirable to reduce the turn-off time of the device not to increase the frequency of operation of the circuit, but rather to prevent second breakdown of the device as the relatively large inductive voltage pulse appears during the initial portion of the flyback interval, when current flowing through the deflection winding is interrupted to initiate the retrace portion of each deflection cycle.

The non-saturated operation of the deflection output transistor is achieved in circuits embodying the present invention by automatically holding the collector voltage above the saturation level by shunting excess base drive from the base to emitter junction into the collector circuit. Prior transistor deflection systems employ only the saturated operation of the deflection output device.

Circuits embodying the present invention include a deflection output transistor having a diode coupled between its base and collector terminals and poled to prevent the transistor from being driven into saturation during its conduction period of each deflection cycle.

The invention can be more fully understood by referring to the drawings together with the description below and the accompanying claims.

In the drawings:

FIG. 1 illustrates in block and schematic diagram form, a television receiver including a solid state deflection output stage embodying the present invention;

FIG. 2a is a waveform diagram of the voltage present at the collector terminal 55c of transistor 55 in FIG. 1;

FIG. 2b shows the drive current to terminal A in FIG. 1;

FIG. 2c is a waveform diagram of the current in diode 56 in FIG. 1;

FIG. 2d is a waveform diagram of the base current flowing in transistor 55 of FIG. 1;

FIG. 3 is a schematic diagram of an alternative embodiment of the present invention;

FIG. 4a is a waveform diagram of the voltage appearing at the terminal 366 in FIG. 3;

FIG. 4b is a waveform diagram of the drive current to terminal A in FIG. 3;

FIG. 4c is a waveform diagram of the current in diode 356 in FIG. 3; and

FIG. 4d is a waveform diagram of the base drive current to transistor 355 in FIG. 3.

Referring specifically to FIG. 1, an antenna 10 receives television signals and couples these signals to a tuner 12 which selects the desired radio frequency signals of a predetermined broadcast channel, amplifies these signals, and converts the amplified radio frequency signals to a lower intermediate frequency (I.F.). The tuner 12 is coupled to an I.F. amplifier 14 which amplifies the intermediate frequency signals. The I.F. amplifier 14 is coupled to a video detector 16 which derives video information from the I.F. signals. The video detector 16 is coupled to a video driver stage 18 which amplifies the video signals. The video driver stage 18 is coupled to a video output stage 20, an automatic gain control stage 25 and a synchronizing separator stage 42. An output signal from video driver stage 18 may also be coupled to a sound channel (not shown) to reproduce the audio portion of the transmitted television program. The video output stage 20 couples amplified video information to a control element, such as a cathode 28, of a kinescope 30.

The automatic gain control stage 25 operates in a conventional manner to provide gain control signals which are applied to a radio frequency amplifier included in tuner 12 and to the I.F. amplifier 14. Sync separator 42 separates the synchronization information from the video information and also separates the horizontal synchronizing information for the vertical synchronizing information. The vertical synchronizing pulses derived from sync separator 42 are applied to the vertical deflection system 44 which provides the required deflection current to a vertical deflection winding 43 associated with kinescope 30 by means of the interconnection Y--Y. The horizontal synchronizing pulses from sync separator 42 are applied to an automatic frequency control detector 45 which serves to synchronize a horizontal oscillator 46 with the horizontal synchronizing pulses. The horizontal oscillator stage 46 is coupled to a horizontal driver stage 48 which develops the required drive signal and may be coupled by means of an output transformer in stage 48 (not shown) to a transistorized horizontal output stage 50. The transformer secondary, coupled to terminal A, provides a direct current path for the drive current.

The horizontal output stage 50 includes an output transistor 55 having a base, a collector and an emitter terminal 55b, 55c and 55e, respectively. A resistor 52 and a capacitor 53 are coupled in parallel between the horizontal driver stage 48 and the base terminal 55b of transistor 55.

The output stage includes a unidirectional conductive device such as a diode 56 coupled between the base and collector terminals 55b and 55c of transistor 55. Stage 50 also includes a damper diode 57 coupled across transistor 55, a retrace capacitor 58 coupled across transistor 55 and the series combination of a horizontal deflection winding 59 and an S-shaping capacitor 60 also coupled across transistor 55. Output stage 50 also includes a flyback transformer 61 with a primary winding 61p coupled from a source of operating potential (B+) to the collector terminal 55c of transistor 55. A secondary winding 61s on transformer 61 develops high voltage pulses which are coupled to a high voltage rectifier 63 to provide the ultor voltage for application to a terminal 32 on kinescope 30. Flyback transformer 61 may also include additional windings (not shown) for providing, for example, keying pulses to the AGC stage 25.

The output stage 50 in FIG. 1 is a conventional shunt fed trace driven circuit with the exception of the diode 56 and the bias network including resistor 52 and capacitor 53. Beginning at the center of the trace interval of the deflection cycle, the yoke current is zero and capacitor 60 has a maximum charge. The drive signal applied to the base terminal 55b of transistor 55 turns this device on, thereby completing the conduction path for yoke current which includes capacitor 60, yoke 59 and the collector to emitter current path through transistor 55. During this portion of scan the yoke current is supplied by the charge on capacitor 60 and increases to a maximum value in one direction at which time scan retrace is initiated by driving transistor 55 out of conduction by applying an appropriate signal from the driver stage 48 to the base 55b of transistor 55. During the latter portion of the trace interval when the magnitude of the yoke current is increasing, the output transistor of prior circuits is normally driven into saturation and is in this conduction state at the instant retrace is initiated. During the first portion of retrace, the yoke current is at a maximum and resonates with the retrace capacitor 58 by charging capacitor 58 in a polarity to reverse bias the damper diode 57. As the yoke current decreases to zero, capacitor 58 has a maximum charge impressed upon it; and during the second portion of retrace, the capacitor (58) drives current through the yoke in a reverse direction until it is discharged and the voltage across it reverses sufficiently to forward bias damper diode 57. Diode 57 then conducts during this first portion of trace to complete the current path for yoke current which is, at this instant, at a maximum value in a direction in yoke 59 to charge capacitor 60 and is increasing toward zero. At the mid-point of trace the yoke current has reached zero and the cycle is completed by driving transistor 55 into conduction once again.

Turning now to the operation of the circuitry of FIG. 1 including the present invention, reference is made to the waveform diagrams of FIG. 2. The initial portion of trace is represented in FIG. 2 by the time period between t ₀ and t ₁ in the figure. It is recalled that during this period damper diode 57 is conducting. The voltage at collector terminal 55c of transistor 55 is represented by the voltage waveform (V _c ) in FIG. 2a and is equal to the forward voltage drop across diode 57 which is of the order of -0.7 volts. At some non-critical time before t ₁ , the horizontal driver 48 provides a drive current (I _A ), as is shown in FIG. 2b. This current flows through diode 56 as is illustrated in FIG. 2c, since the diode is forward biased. [The cathode of diode 56 is at the same voltage as collector terminal 55c (-0.07 volts) and the drive current produces a positive voltage at point A which is at the anode of diode 56.] As time t ₁ (the center of trace) is reached, damper diode 57 turns off allowing the collector voltage on transistor 55 to increase as shown in FIG. 2a. At the same time, a portion of the drive current flowing into terminal A is conducted by the now forward biased base to emitter junction of transistor 55 as is illustrated by the waveform of FIG. 2d. Transistor 55 is now conducting the increasing yoke current during the latter portion of scan represented by the period from t ₁ to t ₂ in FIG. 2. As the magnitude of the yoke current increases during the t ₁ to t ₂ interval, the base current in transistor 55 increases as shown in FIG. 2d. Diode 56 conducts as illustrated in FIG. 2c to shunt the remaining portion of the applied drive current at terminal A. It is noted that the sum of the currents shown in FIGS. 2c and 2d will equal the current shown in FIG. 2b. The values of resistor 52 and capacitor 53 can be selected to hold the transistor collector voltage at a preselected value sufficient to prevent saturation of the transistor 55. If, for example, the voltage across capacitor 53 is 5.3 volts, the voltage at terminal A with respect to ground will be approximately 6 volts (5.3 volts plus the forward voltage drop across the base-emitter junction of transistor 55). The collector voltage will then be approximately equal to the voltage at terminal A less the forward voltage drop across diode 56. It is desirable to choose values of resistor 52 and capacitor 53 to operate transistor 55 near but not into the saturation region of conduction during the latter portion of each trace interval.

At time t ₂ retrace is initiated by applying a relatively large negative drive signal as shown in FIG. 2b to the base terminal of transistor 55. During the retrace interval (t ₂ to t ₀ in FIG. 2), the collector voltage increases in a typical manner as illustrated in FIG. 2a. At time t ₀ the cycle is again repeated.

The circuit modification illustrated in FIG. 3 is another embodiment of the invention which reduces the change in voltage applied to the yoke 59 of FIG. 1 at time t ₁ . As shown in FIG. 2a, when diode 57 turns off and transistor 55 conducts, the voltage at the collector terminal 55c of transistor 55 changes by as much, for example, as 6 volts. This voltage change, which is coupled to the yoke 59, will vary the rate of change of yoke current during the center of trace and may, in certain circuits, cause an undesirable non-linearity in the scanning rate. As FIG. 4a illustrates, the circuit of FIG. 3 reduces this change in voltage at the mid-point of trace (t ₁ ).

Referring to FIG. 3, the circuit elements which correspond to those of FIG. 1 are prefaced by the numeral 3. In explaining FIG. 3, it is helpful to refer to the waveform diagrams of FIG. 4. Transformer 364 in FIG. 3 is a tightly coupled auto-transformer wherein the tap point 365 may be, for example, at the 5 percent point on the transformer. That is, the segment between terminals 365 and 366 contain 5 percent of the total number of windings on transformer 364. Transformer 364 may also include a secondary winding such as the high voltage winding which is not shown in the figure. In operation, as drive current is applied at sometime prior to t ₁ as is shown in FIG. 4b, damper diode 357 is conducting and the voltage at terminal 366 is therefore at approximately -0.7 volts. Drive current flowing into terminal A as represented in FIG. 4b will be conducted by diode 356 during this interval as indicated by the diode current waveform in FIG. 4c. At the middle portion of trace (t ₁ ), the damper diode turns off and voltage at terminal 366 is thereby allowed to go slightly positive (less than 0.7 volts). The collector voltage of transistor 355 is held at a value of approximately 5 volts (assuming, for example, the B+ voltage is equal to 100 volts and the collector is coupled to the tap 365 on transformer 364 at a 5 percent point). At this instant, the base to emitter junction will be forward biased and transistor 355 conducts. It is seen that the anode voltage of diode 356 is at approximately +0.7 volts and its cathode which is coupled to terminal 366 is at a less positive voltage. Diode 356 begins to conduct during the latter portion of trace as illustrated by the current waveform diagram shown in FIG. 4c.

During the latter portion of trace, the transistor tends to saturate and the collector voltage at terminal 355c tends to decrease. As this occurs, more current will flow from the B+ terminal through the upper portion of transformer 364. Due to the relatively tight coupling of the segments of transformer 364, terminal 366 experiences a decrease in voltage which controls the forward bias applied to diode 356 to shunt sufficient drive current to hold the transistor 355 out of saturation. The collector voltage of transistor 355 is thus held at some preselected value depending on the location of tap point 365 on transformer 364. Since transformer 364 is utilized, terminal 366 will remain at a low voltage during the latter portion of trace as shown in FIG. 4a, and diode 356 will be forward biased during the application of a positive drive signal to terminal A. As before, the base drive current will increase and diode 356 conduction will decrease generally as shown in FIGS. 4c and 4d during the latter portion of trace. At time t ₂ in FIG. 4, a negative drive pulse is applied to the circuit which initiates the retrace interval of the deflection cycle.

Although the specific embodiments of the invention are illustrated in the horizontal deflection output stage of a black and white television receiver, the invention has equal applicability to other deflection systems and may be utilized in a color television receiver.







TCA 511 TV HORIZONTAL AND VERTICAL PROCESSOR

The TCA 511 is a silicon monolithic integrated circuit in a 16—lead dual in—line plastic
package. It incorporates the following functions: high stability horizontal oscillator,
horizontal APC circuit with high noise immunity and large pull—in range, high stability
vertical oscillator and sawtooth generator.
lt is intended for driving TV horizontal and vertical transistorized output stages.



APPLICATION INFORMATION

Power Supply
The circuit can work with stabilized supply voltage having a value from 9 to 15 V.
A dropping resistor and a filter capacitor may be used to obtain the suipply from higher
voltages; however, the voltage on pins 3 and 4 must never exceed the maximum
permitted voltage.
Synchronization
Pins 2 and 6 can be DC driven if the reference level of the synchronization pulses is
less than 1 V. With reference levels greater than this value, a coupling capacitor must
be inserted in series with the input, and pins 2

and 6 must be connected to ground
via a resistor.
Vertical Oscillator
The capacitor connected to pin 1 must be selected with regard to the frequency
tolerance, to the thermal stability and to the capacitor's ageing.
The width of the output pulse, to be chosen according to the needs of the output
stages, is defined by the resistor connected between pin 1 and pin 16.
Vertical Output
The vertical output is taken from pin 14, which is a buffered output of the sawtooth
voltage generated at pin 15.
The output current from pin 14 is defined by an internal resistor in the integrated
circuit. if a greater current is needed, a resistor may be connected between pin 14
and pin 3.
The oscillator output pulse is available at pin 15 if the capacitor C9 is not connected. _
This configuration is used for driving output stages in which the sawtooth is generated
by Miller effect.
Horizontal Oscillator
The capacitor connected between pin 10 and ground must be selected with regard
to the frequency tolerance, 1:0 the thermal stability and to the capacit0r’s ageing.
In multistandard receivers, the oscillation frequency may be changed by switching the
value of the capacitor connected to pin 10.



 THE TBA800, TBA810 AUDIO integrated circuits:

AUDIO integrated circuits are being increasingly used in television chassis and certainly represent the simplest approach to improving the audio side of a TV set. A number of such i.c.s have appeared during the 70's.

Here describes the use of two fairly recent ones, the SGS-ATES TBA800 and TBA8I0S. Both devices can provide reasonably high outputs into a suitable loudspeaker-the TBA800 will give up to 5W and the TBA810S up to 7W.
The main difference between them being that the TBA800 is a somewhat higher voltage, lower current device. The TBA800 is used in the current Grundig and ASA 110° colour chassis while the Finlux 110' colour chassis uses a TBA810. In each of these chassis the audio i.c. is driven from a TBA120 intercarrier sound i.c. The TBA800 and TBA810S can also be used as the field output stage in 110' monochrome chassis with c.r.t.s of up to l7in. and as the field driver stage in larger screen monochrome sets.
The TBA800 is designed to provide up to 5W into a 16 Ohm load when operated from a 24V supply. It is encapsulated in the type cf quad -in -line case shown in Fig. I: the tabs at the centre are to assist in cooling the device and must be earthed. The TBA800 can be operated from power supply voltages up to the absolute maximum permissible value of 30V. It is best to regard 24V as being the upper limit however in order to provide an adequate safety margin and prevent possible damage during voltage surges. The minimum power supply voltage recommended by the manufacturers is 5V, but the power output is then less than 0-5W. The quiescent current taken by the TBA800 is typically 9mA from a 24V supply-no device of this type should draw more than 20mA. When an input signal is applied the current increases considerably- up to about 1.5A at full power. Two circuits for use with the TBA800 are shown in Figs. 2 and 3 and give comparable performance. The circuit shown in Fig. 2 is somewhat simpler but that
shown in Fig. 3 enables one side of the loudspeaker to be connected to chassis. The input resistance of the TBA800 is quite high (typically 5 MOhm) but a resistor must be connected between the input pin 8 and chassis otherwise the out- put stage will not operate with the correct bias. In the circuits shown the volume control VR1 provides this function: the bias current that flows through it is typically 1 microA (maximum 5 microA). The average voltage at the output pin 12 is half the supply potential. The loudspeaker must be capacitively coupled therefore and the low frequency response will be worse as this capacitor is decreased in value. The output coupling capacitor C4 in Fig. 2 also provides the bootstrap connection to pin 4. In Fig. 3 an additional capacitor (C9) is required for this purpose.
In both circuits the value of R1 controls the amount of feedback and thus the gain. The output signal is fed back to pin 6 via an internal 7 kOhm resistor. If R1 is reduced in value the gain will increase but the frequency response will be affected and the distortion will rise. With the component values shown the voltage gain of both circuits is typically 140 (43dB) which is quite adequate for most audio applications. R3 in Fig. 3 is necessary only if the power supply voltage is fairly low (less than about 14V).
C2 smooths the power supply input and C1 is connected between pin 1 and chassis to provide r.f. decoupling and help prevent instability. If mains hum is present on the supply line with the circuit shown in Fig. 3 capacitor C8 should be included between pin 7 and chassis. The circuits shown have a level frequency response (within ±3dB) between about 40Hz and 20kHz. If you wish to reduce the upper 3dB level to about 8kHz C5 can be increased to about 560pF. The total harmonic distortion provided by these circuits remains fairly constant at about 0.5% until the power output reaches 3W: it then rises rapidly with power level as shown in Fig. 4.
The TBA800 can be operated from a 13V supply to feed up to 2.5W into an 80 load or from a 17V supply to feed the same power into a 160 load without an additional heatsink. If more output power is required the cooling tabs must be connected to a heatsink. Two methods of mounting the TBA800 are shown in Figs. 5 and 6. In Fig. 5 the device is inserted into a circuit board and a heatsink is soldered to the same points as the tabs: this has the disadvantage that the heatsink extends above the board though on the other hand the whole board can be used for the construction of the circuit. In Fig. 6 the tabs are soldered directly to a suitable area of copper on the board: this method has the disadvantage that about two square inches of the board are not available for component mounting. It is generally best to make soldered connections to the pins of the device since this ensures good heat dissipation with minimum unwanted feedback. Observe the usual heat precautions when soldering. The pins can however be carefully bent so that they will fit into a 16 -pin dual -in -line socket.
The TBA810S has the same type of encapsulation as the TBA800 and the connections are also as shown in Fig. 1 except that there is no internal connection to pin 3. An alternative version, the TBA810AS, has two horizontal tabs with a hole in each (see Fig. 7) so that a heatsink can be bolted on. Some readers may find it easier to bolt a heatsink to a TBA810AS than to solder the TBA810S tabs. TBA810 devices can provide 7W of audio power to a 40 loudspeaker when operated from a I6V supply. Fig. 8 shows the change in maximum output power with different supply voltages. As a 4.5W output can be obtained with a 12V supply the TBA810 is much more suitable than the TBA800 for use with battery operated equipment. The TBA810 can provide output currents up to 2.5A.
Two circuits for use with TBA810 devices are shown in Figs. 9 and 10: they are very similar to the circuits shown in Figs. 2 and 3 though some of the capacitor values are larger because of the lower output impedance. The two circuits have comparable performance but that shown in Fig. 10 gives somewhat better results at low supply voltages (down to 4V). In either circuit R2 may be replaced with a 100k0 volume control. The bias current flowing in the pin 8 circuit is typically
0-4 microA and the input resistance 5M 0 (the value of R2 must be much less however to ensure correct bias.
 The gain decreases as the value of R1 is increased for the same reason as with the TBA800. The values of R1, C3 and C7 affect the high -frequency response. With the values shown the response is level within ±3dB from about 40Hz to nearly 20kHz. Fig. 11 shows values of C3 plotted against R1 where the frequency is 3dB down at 10kHz and 20kHz and C7 is five times C3. The output distortion with these circuits is about 0.3% for outputs up to 3W rising to about 1% at 4W, 3% at 5W and 9% at 6W with a 14.4V supply voltage. The voltage gain is typically 70 times (37dB). Although this value is half that obtained with the TBA800 the input voltage required to produce a given output power is about the same for both types. This is because a smaller output voltage is required to drive a 40 load at a certain power level than is required to drive a 160 load.

The TBA810S may be mounted in the same way as the TBA800. One way of mounting the TBA810AS is shown in Fig. 12. It is simpler however to bolt flat heatsinks to the tabs.
Devices of this type will be destroyed within a fraction of a second if the power supply is accidentally con- nected with reversed polarity. When experimenting therefore it is wise to include a diode in the positive power supply line to prevent any appreciable reverse current flowing in the event of incorrect power supply connection. The diode can be removed once the circuit has been finalised. The TBA800 is likely to be destroyed if the output is accidentally shorted to chassis. The TBA810S and TBA810AS however are protected from damage in the event of such a short-circuit even if this remains for a long time (but note that the earlier TBA8I0 and  TBA810A versions did not contain internal circuitry to provide this protection). The TBA800 is not protected against overheating but the TBA810S and TBA810AS incorporate a thermal shutdown circuit.
For this reason the heat- sinks used with the TBA810S and TBA810AS can have a smaller safety factor than those used with the TBA800. If the silicon chip in a TBA810S or TBA810AS becomes too hot the output power is temporarily reduced by the internal thermal shutdown circuit. As with all high -gain amplifiers great care should be taken to keep the input and output circuits well separated otherwise oscillation could occur. The de- coupling capacitors should be soldered close to the i.c. -especially the 0 1pF decoupling capacitor in the supply line (this should be close to pin I).
Field Output Circuit:

 Fig. 13 shows a suggested field output stage for monochrome receivers with 12-17in. 110° c.r.t.s using the TBA81OS. For safe working up to 50°C ambient temperature each tab of the device must be soldered to a square inch of copper on the board. The peak -to - peak scanning current is 1.5A, the power delivered to the scan coils 0.47W, power disspipation in the TBA810S 1 8W, scan signal amplitude 4.1V, flyback amplitude 5V and the maximum peak -to -peak current available in the coils 1.75

Power supply is realized with mains transformer and Linear transistorized power supply stabilizer, A DC power supply apparatus includes a rectifier circuit which rectifies an input commercial AC voltage. The rectifier output voltage is smoothed in a smoothing capacitor. Voltage stabilization is provided in the stabilizing circuits by the use of Zener diode circuits to provide biasing to control the collector-emitter paths of respective transistors.A linear regulator circuit according to an embodiment of the present invention has an input node receiving an unregulated voltage and an output node providing a regulated voltage. The linear regulator circuit includes a voltage regulator, a bias circuit, and a current control device.

In one embodiment, the current control device is implemented as an NPN bipolar junction transistor (BJT) having a collector electrode forming the input node of the linear regulator circuit, an emitter electrode coupled to the input of the voltage regulator, and a base electrode coupled to the second terminal of the bias circuit. A first capacitor may be coupled between the input and reference terminals of the voltage regulator and a second capacitor may be coupled between the output and reference terminals of the voltage regulator. The voltage regulator may be implemented as known to those skilled in the art, such as an LDO or non-LDO 3-terminal regulator or the like.
The bias circuit may include a bias device and a current source. The bias device has a first terminal coupled to the output terminal of the voltage regulator and a second terminal coupled to the control electrode of the current control device. The current source has an input coupled to the first current electrode of the current control device and an output coupled to the second terminal of the bias device. A capacitor may be coupled between the first and second terminals of the bias device.
In the bias device and current source embodiment, the bias device may be implemented as a Zener diode, one or more diodes coupled in series, at least one light emitting diode, or any other bias device which develops sufficient voltage while receiving current from the current source. The current source may be implemented with a PNP BJT having its collector electrode coupled to the second terminal of the bias device, at least one first resistor having a first end coupled to the emitter electrode of the PNP BJT and a second end, a Zener diode and a second resistor. The Zener diode has an anode coupled to the base electrode of the PNP BJT and a cathode coupled to the second end of the first resistor. The second resistor has a first end coupled to the anode of the Zener diode and a second end coupled to the reference terminal of the voltage regulator. A second Zener diode may be included having an anode coupled to the cathode of the first Zener diode and a cathode coupled to the first current electrode of the current control device.

A circuit is disclosed for improving operation of a linear regulator, having an input terminal, an output terminal, and a reference terminal. The circuit includes an input node, a transistor, a bias circuit, and first and second capacitors. The transistor has a first current electrode coupled to the input node, a second current electrode for coupling to the input terminal of the linear regulator, and a control electrode. The bias circuit has a first terminal for coupling to the output terminal of the linear regulator and a second terminal coupled to the control electrode of the transistor. The first capacitor is for coupling between the input and reference terminals of the linear regulator, and the second capacitor is for coupling between the output and reference terminals of the linear regulator. The bias circuit develops a voltage sufficient to drive the control terminal of the transistor and to operate the linear regulator. The bias circuit may be a battery, a bias device and a current source, a floating power supply, a charge pump, or any combination thereof. The transistor may be implemented as a BJT or FET or any other suitable current controlled device.


Power Supply: The examples chosen are taken from manufacturers' circuit diagrams and are usually simplified to emphasise the fundamental nature of the circuit. For each example the particular transistor properties that are exploited to achieve the desired performance are made clear. As a rough and ready classification the circuits are arranged in order of frequency: this part is devoted to circuits used at zero frequency, field frequency and audio frequencies. Series Regulator Circuit Portable television receivers are designed to operate from batteries (usually 12V car batteries) and from the a.c. mains. The receiver usually has an 11V supply line, and circuitry is required to ensure that the supply line is at this voltage whether the power source is a battery or the mains. The supply line also needs to have good regulation, i.e. a low output resistance, to ensure that the voltage remains constant in spite of variations in the mean current taken by some of the stages in the receiver. Fig. 1 shows a typical circuit of the power -supply arrangements. The mains transformer and bridge rectifier are designed to deliver about 16V. The battery can be assumed to give just over 12V. Both feed the regulator circuit Trl, Tr2, Tr3, which gives an 11V output and can be regarded as a three -stage direct -coupled amplifier. The first stage Tr 1 is required to give an output current proportional to the difference between two voltages, one being a constant voltage derived from the voltage reference diode D I (which is biased via R3 from the stabilised supply). The second voltage is obtained from a preset potential divider connected across the output of the unit, and is therefore a sample of the output voltage. In effect therefore Tr 1 compares the output voltage of the unit with a fixed voltage and gives an output current proportional to the difference between them. Clearly a field-effect transistor could do this, but the low input resistance of a bipolar transistor is no disadvantage and it can give a current output many times that of a field-effect transistor and is generally preferred therefore. The output current of the first stage is amplified by the two subsequent stages and then becomes the output current of the unit. Clearly therefore Tr2 and Tr3 should be current amplifiers and they normally take the form of emitter followers or common emitter stages (which have the same current gain). By adjusting the preset control we can alter the fraction of the output voltage' applied to the first stage and can thus set the output voltage of the unit at any desired value within a certain range. By making assumptions about the current gain of the transistors we can calculate the degree of regulation obtainable. For example, suppose the gain of Tr2 and Tr3 in cascade is 1,000, and that the current output demanded from the unit changes by 0.1A (for example due to the disconnection of part of the load). The corresponding change in Tr l's collector current is 0.1mA and, if the standing collector current of Tr 1 is 1mA, then its mutual conductance is approximately 4OmA/V and the base voltage must change by 2.5mV to bring about the required change in collector current. If the preset potential divider feeds one half of the output voltage to Tr l's base, then the change in output voltage must be 5mV. Thus an 0.1A change in output current brings about only 5mV change in output voltage: this represents an output resistance of only 0.0552.

 PHILIPS  I17T766  CHASSIS  3119 203 30664 Tuning unit with bandswitch for high frequency receivers

Abstract:

A tuning unit with a bandswitch for high frequency receivers having a potentiometer system for the control of capacity diodes is disclosed. The potentiometer system includes a plurality of parallelly disposed resistance paths on which wipers can be moved by means of screw tuning spindles mounted beside one another in a common housing made of an insulating material. The bandswitch is formed of metal wires and is associated with each tuning spindle. The tuning spindles are joined for rotation with sleeves simultaneously forming the operating knobs which are carried in apertures in the front plate and each have a flange engaging the back side of the front plate about the apertures. The flange is slightly larger than the cross section of the apertures and tapers conically away from the back side of the front plate.

1. Tuning unit with bandswitch for high frequency receivers having potentiometer means for the control of capacity diodes composed of a plurality of parallelly disposed resistance paths on which wipers are moved by means of screw tuning spindle means mounted beside one another in a common housing of insulating material, bandswitch means formed of metal wires associated with each tuning spindle means, said tuning spindle means being joined for rotation with sleeve means simultaneously forming operating knobs which are borne in apertures in the front plate and each sleeve means having an axial flange surface engaging the back side of the front plate about one aperture therein, said flange surface being slightly larger than the cross section of the apertures and tapering conically away from the back side of the front plate.

2. Tuning unit of claim 1 wherein the sleeve means are joined telescopically and coaxially with the tuning spindle means, and the flange surface engages the back side of the front plate when the sleeve means are in the state wherein they are pulled out of the front plate.

3. Tuning unit of claim 1 wherein the ends of the tuning spindle means which are opposite the front plate have each an annular groove into which a spring bracket engages whose bent end is supported against the housing and which has two diametrically disposed spring arms having opposite spring curvature, the said spring arms in each case contacting the opposite axial walls of the groove.

4. Tuning unit of claim 3 wherein the spring bracket rests with its bent end against the housing and the spring arms additionally engage a bracket formed on the housing or an intermediate bracket formed in one piece with the connection soldering lugs.

5. Tuning unit of claim 3 wherein the spring bracket is formed in one piece with the connection soldering lugs and has spring arms curved both in the same direction which engage an axial wall of the annular groove in the spindle and the opposite axial wall rests against a housing wall.

6. Tuning unit of claim 1 wherein the pointers associated with each potentiometer means lie on the one hand in windows associated with each tuning spindle means in the front plate, and on the other hand are rotatably mounted with their ends opposite the front plate in pivot pins on the housing, and the guiding pin of the spindle nuts carried in a longitudinally displaceable manner on each tuning spindle is provided with a slit disposed parallel to the longitudinal axis of the tuning spindle and slides with its peripheral surface resiliently within the slide tract of the pointer.

7. Tuning unit of claim 1 wherein the bandswitches are formed each of a displaceable metal rod which is in working engagement with stationary metal rods common to all bandswitches of a tuning unit, contacting each of them individually.

8. Tuning unit of claim 7 wherein the metal rods are metal wires.

9. Tuning unit of claim 7 wherein the metal rods are stamped metal parts.

10. Tuning unit of claim 7 wherein levers of insulating material are placed on the front ends of the displaceable metal rods and extend through windows which are provided with detents and which are associated with each tuning spindle in the housing front plate, while the opposite ends are held fixedly in the rearward end of the housing, and the displaceable metal rods individually make contact with contact cams on the stationary metal rods, these cams being in an offset array corresponding to the detents in the windows, the corresponding rods extending parallel to the front plate and parallel to one another behind the front plate.

11. Tuning unit of claim 7 wherein insulating material bridges or insulating material slide pieces are inserted between the contact cams of two adjacent, stationary metal rods and within the free space between two such parallel metal rods.

12. Tuning unit of claim 7 wherein the displaceable metal rods have, in the vicinity of their mountings on the housing, an articulation in the form of a vertically disposed flat portion.

Description:

BACKGROUND

The invention relates to a tuning unit with bandswitch for high frequency receivers, especially radio and television receivers, having a potentiometer system for the control of capacity diodes, the said potentiometer system consisting of a plurality of parallel resistance paths along which wiper contacts can be driven by means of screw spindles disposed adjacent one another in a common insulating material housing in which a bandswitch formed of metal rods is associated with each tuning spindle.
In these tuning units, the working voltages of the capacity diodes in the tuning circuits are recorded once a precise tuning to the desired frequency has been performed. A potentiometer tuning system has great advantages over the formerly used channel selectors operating with mechanically adjustable capacitors (tuning condensers) or mechanically adjustable inductances (variometers), mainly because it is not required to have such great precision in its tuning mechanism.
Tuning units with bandswitches formed of variable resistances and combined with interlocking pushbuttons controlling the supply of recorded working voltages to capacity diodes are known. Channel selection is accomplished by depressing the knobs, and the tuning or fine tuning are performed by turning the knobs. The resistances serving as voltage dividers in these tuning units are combined into a component unit such that they are in the form of a ladderlike pattern on a common insulating plate forming the cover of the housing in which the tuning spindles and wiper contacts corresponding to the variable resistances are housed. The number of resistances corresponds to the number of channels or frequencies which are to be recorded. The wiper contact picks up a voltage which, when applied to the capacity diodes determines their capacitance and hence the frequency of the corresponding oscillating circuit. The adjustment of the wipers is performed by turning the tuning spindle coupled to the tuning knob. By the depression of a button the electrical connection between a contact rod and a tuning spindle is brought about and thus the selected voltage is applied to the capacity diodes. Since the push buttons release one another, it is possible simply by depressing another button to tune to a different receiving frequency or a different channel, as the case may be.
To permit the switching of a number of channels in a certain tuning range, bandswitches for a plurality of tuning ranges, such as UHF and VHF for example, are often provided in the tuning units described above. In the pushbutton tuning unit of the above-named type, the bandswitch consists of a printed circuit board which is fastened on the housing of the tuning unit, and a switch lever which is preset by means of the pushbutton by turning, and is operated by depressing the pushbutton while at the same time selecting the channel.
Where this combination of knobs and pushbuttons is not possible, the selection of the range is accomplished by means of an additional lever which can be set over to select the range.
However, since such tuning units require too many riveting operations when they are assembled, tuning units were later created in which the individual parts in the voltage divider and pushbutton housing were loosely inserted and/or held in place by projections, lugs, hooks or tabs of resilient plastic. In spite of these initial improvements, the bandswitch, especially the one associated with the tuning units, was still technically intricate and very expensive.
THE INVENTION
It is the object of the invention, therefore, to create an additionally improved and simplified tuning unit containing a bandswitch of simple, space-saving and reliably operating design.
In accordance with the invention, this object is accomplished in a tuning unit with bandswitch of the kind described in the beginning by joining the tuning spindles for rotation with sleeves simultaneously forming the control knobs, which are mounted in apertures in the front plate of the housing and have each a flange engaging the back of the front plate around the aperture, the said flange being slightly larger than the aperture and tapering conically away from the back of the front plate.

In further development, the sleeves can be joined telescopically for rotation with the tuning spindles, and the flange is able to engage the back side of the front plate when the sleeve is in the position in which it is drawn out of the front plate. The sleeves constructed in this manner, whose portions projecting from the apertures in the front plate form the control knobs for the tuning spindles, permit easy assembly of the tuning unit and at the same time assure positive co-rotation of sleeves and spindles. The sleeves can be pushed from the front side of the front plate through the apertures onto the clutch surfaces of the spindles, this inward pushing being easily accomplished on account of the taper, and the dropping out of the sleeve being prevented by the flange engaging the back of the front plate. If the control knobs project only slightly out of the front plate, they can be operated from the outside by inserting a tool into them. With the telescoping type of coupling, however, it is possible to draw the sleeves or control knobs further outwardly so that they can be rotated by hand without the use of tools.
To provide constant assurance of the axial fixation of the tuning spindles, the tuning spindle ends farthest from the front plate can each be provided with an annular groove engaged by a spring bracket whose one leg is supported against the housing and whose other leg is forked to form two spring arms, each bent in the opposite direction and each engaging one of the two opposite walls of the annular groove. The tuning spindles are secured against axial displacement by this construction of the invention alone, without the need for further measures. This facilitates the joining of the sleeves or control knobs to the tuning spindle, because in this case there is no need for precise axial fixation and extreme dimensional accuracy.

Furthermore, the indicators associated with each potentiometer can be mounted in windows in the front plate which are associated with each tuning spindle or tuning knob for visual indication at the front, the other extremities farthest from the front plate being mounted for pivoting on pins set in the housing; the guiding pin on the spindle nut that is driven longitudinally on each tuning spindle can be provided with a slit disposed parallel to the long axis of the tuning spindles and can slide within the indicator slide lever slot, with its surface resiliently engaging the walls of said slot.
In an especially advantageous embodiment, the tuning unit can have bandswitches each formed of a displaceable metal rod which is in contacting engagement individually with stationary metal rods which are common to all of the bandswitches of a tuning unit. It contrast to the bandswitches known hitherto, which as a rule consist of a printed circuit board with switchable contacts thereon, this frequency bandswitch of the invention is of great simplicity, can be manufactured simply and inexpensively, and at the same time is very reliable in operation.
The displaceable and stationary metal rods of the bandswitches can be formed of metal wires or they can be of stamped sheet metal. Also, in further expansion of the concept of the invention, the stationary metal rods thus formed can be all entirely alike and merely offset from one another, thereby further simplifying the manufacture and stocking thereof.
To permit connection also to audiovisual apparatus, one or more of the stationary metal rods can be divided electrically into at least two parts each.
In a special development of this concept, lugs of insulating material can be mounted on the front ends of the displaceable metal wires, these lugs extending through windows in the front plate of the housing which are associated with each tuning spindle and are provided with detents, while the opposite ends can be held fixedly at the rear end of the housing, and the displaceable metal wires can make contact with contact humps on the stationary metal wires, the humps being offset from one another to correspond to the detents in the windows, and the stationary metal wires extending in back of the front plate, parallel to the latter and parallel to one another.
To increase switching reliability, bridges or sliding pieces made of insulating material can be inserted between the contact humps of adjacent stationary wires within the free space between two such parallel lying metal wires.
To achieve easy displacement of the displaceable metal wires despite the fixed end mounting on the housing, the displaceable metal wires, in further embodiment of the invention, can have each an articulation adjacent their end mountings, in the form of a vertically disposed flattened portion. This flat permits the metal wires to be deflected horizontally against a weak spring bias.
DESCRIPTION OF THE DRAWING
As an example of the embodiment of the invention, there is represented in the drawings a tuning unit with bandswitch for television receivers. In these drawings,
FIG. 1 is a front elevational view of a tuning unit with bandswitch,
FIG. 2 is a plan view showing the bandswitch of the tuning unit of FIG. 1,
FIG. 3 is a side elevational, cross-sectional view of the tuning unit of FIG. 1,
FIG. 4 is a rear elevational view of the tuning unit of FIG. 1,
FIG. 5 is a plan view showing the indicator means of the tuning unit of FIG. 1,
FIG. 6 shows the sleeve with the operating knob and tuning spindle,
FIG. 7 shows the telescoping manner in which the sleeve is joined to the tuning spindle,
FIG. 8 is a fragmentary view of the bandswitch,
FIG. 9 is another fragmentary view of the bandswitch, and
FIG. 10 shows how the tuning spindle is fixed in position.
DESCRIPTION
The method of representation used in the drawings is greatly simplified, for the purpose of better delineating the features of the invention. The tuning unit with bandswitch consists of an insulating material housing 1 with a front plate 2, which is closed by a cover plate 3 accommodating the resistance paths. The housing 1 is divided by parallel sidewalls 4 into chambers in which the tuning spindles 5 are disposed.
The embodiments is an 8-fold tuning unit having eight bandswitches assocated with each tuning spindle, and eight indicators.
Accordingly, there are eight apertures 6 in a central row, through which the operating knobs 7 of the sleeves 8 coupled with the tuning spindles 5 are passed. The operating knobs 7 have recessed surfaces 9 for turning with a turning tool. In a row extending parallel above the row of the apertures 6 there are eight windows 10, whose upper edge is provided with notches 11. Lugs 12 of insulating material extend through the windows 10 and engage the upper notches 11 and are joined behind the front plate to displaceable metal wires 13 of the bandswitch. In a row located beneath the row of apertures 6 another eight windows 14 are provided, through which the ends of the pointers of the indicators 15 protrude.
Now, the bandswitch consists in each case of a displaceable metal wire 13 which can be brought into working engagement with stationary metal wires 16, which are all of the same construction and are only disposed offset from one another. While the displaceable metal wire 13 extends substantially parallel to the longitudinal axis and thus at right angles to the front plate 2, the stationary, parallelly disposed metal wires 16 are parallel to the front plate 2 and are thus inserted at a right angle to the displaceable metal wire. A departure from parallelism or from the right angle, as the case may be, takes place substantially only when the displaceable metal wire 13 is deflected to the two outer notches. The rearward end 18 of the displaceable metal wire, which forms a vertical loop, is tightly inserted into a receiver 17. Just ahead of the loop 18, the metal wire 13 is provided with a vertically disposed portion 19 by a flattening on the metal wire 13. The movement, when the metal wire 13 is deflected into the desired notches or detents, takes place horizontally by the flexing of these portions 19. The stationary metal wires 16 are held tightly in their positions in projections 20 on the housing, or by lugs or the like. Since three switch actions are provided, that is, three ranges, for each tuning spindle, a bandswitch consists of one displaceable metal wire and three stationary metal wires 16, which are used for all switches.
To permit each bandswitch to have exactly three switching actions, each of the three stationary metal wires 16 has one contact hump 21 corresponding to one of the detents 11 in the windows 10 of the front plate 2. The contact humps 21 are thus located one next to the other as seen from the front plate 2. So that the displaceable metal wire 13 will always come into mechanical and electrical contact only with the desired contact hump, and prevent short circuits, insulating bridges 22 are installed between the adjacent metal wires 16, said insulating bridges being stationary.
If more or less than three switching actions are desired, all that need be done in the case of the bandswitch of the invention is to change the number of stationary metal rods or wires accordingly.
The sleeves 8 with the operating knob 7 have a flange 23 engaging the back of the front plate 2 and tapering back to the point where it joins the tuning spindle. This enables the sleeves to be pushed in, in the case of a housing that has already been manufactured with the tuning spindle installed, without creating the possibility that the sleeves 8 might escape after they have been inserted. The sleeves 8 are connected to the tuning spindles 5 usually by means of driving surfaces. If manual operation without tools is to be possible, rather than requiring a tool for the operation of the sleeves, the coupling of the sleeve 8 to the tuning spindle will be a telescoping coupling (see FIG. 7).
The actual firm axial fixation of the tuning spindle 5 is located on the rear end of the housing. Here the tuning spindle 5 has an annular groove 24 which is engaged by a spring by means of two diametrically disposed spring arms 25 and 26. The spring arms 25 and 26 have oppositely curved lugs and are supported on the housing at their terminal and marginal surfaces and their lugs engage opposite axial walls 27 and 28 of the annular groove 24.
Additional support is provided by the common, bent foot 29 of the spring arms 25 and 26 against the cover plate of the housing.
The indicator means of the tuning unit with bandswitch consists of a pointer 15 which is movable within the window 14, and a cam 30 which is a prolongation of the pointer 15. At its rearward end, the pointer is mounted rotatably in the housing on pin 31. Within the cam 30 slides a guiding pin 32 which is attached to the spindle nut or carriage 40. Upon the rotation of the tuning spindle, the spindle nut is longitudinally displaceable therewith. In order to achieve good guidance and hence precise indication, the guiding pin has a slit 33 extending parallel to the longitudinal axis of the tuning spindle 5, so that it will resiliently engage the cam 30 within the slot thereof.
The necessary soldering lugs are indicated at 34.
On the basis of the design of the tuning unit with bandswitch in accordance with the invention, a desired frequency range--UHF, for example--can be selected by deflecting a displaceable metal wire 13 into one of the detents 11 by means of the lug 12 mounted thereon. Within this range, a transmitter or channel can then be selected by turning the tuning spindle 5. The transmitter preselected in this manner can then be tuned in by means of a keyboard or by electronic recall from a keyboard which is not shown. The fine tuning of this tuned-in transmitter, as well as the selection of a different transmitter within the same frequency range, is accomplished by turning the tuning spindle 5.
All of the details explained in the above description and represented in the drawings are important to the invention.