SONY KV-1842E CHASSIS SCC-191A-A INTERNAL VIEW.

SONY KV-1842E  CHASSIS  SCC-191A-A

 This Chassis SCC-191A-A is developed on 2 main panels boards. - Deflections + Supply board on the central bottom. - Signal board on the left side vertical mounted.  Luminance and Chrominance combination ASIC VIF SIF - Chroma Processor Circuit - PAL Chrominance & Luminance Processor.

Above placed the commands for tunings and setups.

SONY KV-1842E  CHASSIS  SCC-191A-A  CONTACTLESS TOUCH SENSOR PROGRAM CHANGE KEYBOARD CIRCUIT ARRANGEMENT FOR ESTABLISHING A CONSTANT POTENTIAL OF THE CHASSIS OF AN ELECTRICAL DEVICE WITH RELATION TO GROUND :

Circuit arrangement for establishing a reference potential of a chassis of an electrical device such as a radio and/or TV receiver, such device being provided with at least one contactless touching switch operating under the AC voltage principle. The device is switched by touching a unipole touching field in a contactless manner so as to establish connection to a grounded network pole. The circuit arrangement includes in combination an electronic blocking switch and a unidirectional rectifier which separates such switch from the network during the blocking phase.


1. A circuit arrangement for establishing, at the chassis of an electrical device powered by a grounded AC supply network, a reference potential with relation to ground, said device having at least one contactless touching switch operating on the AC voltage principle, the switch being operated by touching a unipole touching field in a contactless manner, said arrangement comprising an electronic switch for selectively blocking the circuit of the device from the supply network, a half-wave rectifier including a pair of diodes individually connected in series-aiding relation between the terminals of the supply network and the terminals of the device for separating the electronic blocking switch from the supply network during a blocking phase defined by a prescribed half period of the AC cycle, and a pair of condensers individually connected in parallel with the respective diodes. 2. A circuit arrangement according to claim 1, wherein the capacitances of the two condensers are of equal magnitude.

Description:

This invention relates to a circuit arrangement for establishing a constant reference potential on the chassis of an electrical instrument such as a radio and/or a TV receiver. Such instrument includes at least one contactless touching switch operating under the AC voltage principle, whereby by touching a single pole touching field the contactless switch is operated.

In electronic devices, for example TV and radio receivers, there are used in ever increasing numbers electronic touching switches for switching and adjusting the functions of the device. In one known embodiment of this type of touching switch, which operates on a DC voltage principle, the function of the electronic device, is contactlessly switched by touching a unipole touching field, the switching being carried out by means of an alternating current voltage. When using such a unipole touching electrode, one takes advantage of the fact that the AC current circuit is generally unipolarly grounded. In order to close the circuit by touching the touching surface via the body of the operator to ground, it is necessary to provide an AC voltage on the touching field. In one special known embodiment there is employed a known bridge current rectifier for the current supply. This type of arrangement has the drawback that the chassis of the device changes its polarity relative to the grounded network pole with the network frequency. With such construction considerable difficulties appear when connecting measuring instruments to the device, such difficulties possibly eventually leading to the destruction of individual parts of the electronic device.

In order to avoid these drawbacks, the present invention provides a normal combination of a unidirectional rectifier with an electronic blocking switch that separates the chassis of the electronic device from the network during the blocking phase. In accordance with the present invention, the polarity of the chassis of the electronic device does not periodically change, because the electronic device is practically separated from the network during the blocking phase of the unidirectional rectifier by means of the electronic blocking switch.

In a further embodiment of the invention a further rectifier is connected in series with the unidirectional rectifier in the connection between the circuit and the negative pole of the chassis. Such further rectifier is preferably a diode which is switched in the transfer direction of the unidirectional rectifier. According to another feature of the invention there are provided condensers, a respective condenser being connected parallel with each of the rectifiers. Preferably the two condensers have equal capacitances. Because of the use of such condensers, which are required because of high frequency reasons, during the blocking phase there is conducted to the chassis of the electronic device an AC voltage proportional to the order of capacitances of the condensers. Thus there is placed upon the touching field in a desired manner an AC voltage, and there is thereby assured a secure functioning of the adjustment of the device when such touching occurs.

In the embodiment of the invention employing two rectifiers there is the further advantage that over a bridging over of the minus conduit of the rectifier that is connected between the network and the negative pole of the chassis connection, no injuries can be caused by a measuring instrument in the electronic device itself and in the circuit arrangement connected thereto.

In the accompanying drawing:

The sole FIGURE of the drawing is a circuit diagram of a preferred embodiment of the invention.

In the illustrated embodiment the current supply part of the device, shown at the left, is connected via connecting terminals A and B to an AC voltage source, the terminal B being grounded at 8. The current supply part consists of a unidirectional rectifier in the form of a diode 1 with its anode connected to the terminal I, the cathode of diode 1 being connected to one input terminal 9 of an electronic device 2. In the device 2 there is also arranged a sensor circuit 3, shown here mainly as a block, circuit 3 being shown as including a pnp input transistor the emitter of which is connected to an output terminal 11 of the device 2. The collector of such transistor is connected to the other output terminal 12 of the device 2. The base of the transistor is connected by a wire 13 to a unipolar touching field 4 which may be in the form of a simple metal plate instead of the pnp transistor shown, the sensor circuit itself may consist of a standard integrating circuit which controls, among other things, the periodic sequential switching during the touching time of the touching field 4. All of the circuits of the electronic device 2 are isolated in a known manner from the chassis potential. Between the network terminal B and the negative pole 10 of the chassis there is arranged in the direction opposite that of diode 1 a further diode 5, the anode of diode 5 being connected to the terminal 10, and the cathode of diode 5 being connected to the terminal B of the current supply. To provide for HF type bridging of the diodes 1 and 5 there are arranged condensers 6 and 7 respectively, which are connected in parallel with such diodes.

The invention functions by reason of the fact that in an AC network separate devices radiate electromagnetic waves which produce freely traveling fields in the body of the person who is operating and/or adjusting the device, thereby producing an alternating current through his body to ground, as indicated by the - line at the right of the circuit diagram. If now the person operating the device touches the switching field 4, then the pnp type input transistor of the sensor circuit 3, which is placed on a definite reference potential (for example 12 Volts) and is connected with the negative halfwave of the AC voltage potential, is made conductive. There is thereby released a control command in the sequential switching, for example, for switching the electronic device to the next receiving channel. It is understood that the most suitable connection is formed between ground and the touching field 4 by means of a wire. By the use of such wires it would be assured that in all cases the base of the transistor in circuit 3 is connected to ground. This would, however, not permit anyone to operate the switch without the use of an auxiliary means such as a wire. It will be assumed that the touching almost always results directly via the almost isolated human body. For this reason the AC current fields are necessary, because otherwise there cannot always be provided a ground contact. Thus this connection is established via the body resistance of the person carrying out the touching of the switch.

The positive half wave of the alternating current travels to the terminal 9 of the electronic device 2 after such current has been rectified and smoothed by the devices 1, 6. Such positive halfwave is also conducted to the sensor circuit 3. The thus formed current circuit is closed by way of the chassis of the electronic device 3, the diode 5, and the terminal B. When there is a negative halfwave of the alternating current delivered by the current supply, both diodes 1 and 5 remain closed so that the chassis of the device 2 remains separated from the network during the blocking phase. Nevertheless, by means of condensers 6 and 7 the chassis is placed in a definite network potential, which depends on the relationship of the order of magnitude of the two condensers 6 and 7. When the capacitances of such condensers are equal, there is placed upon the chassis of the device 2 the constant reference potential, and simultaneously there is present via the sensor circuit 3 the required AC voltage at the touching field 4 for adjusting the function or functions of the device 2 upon the touching of the touching field 4.

The reference character 15 indicates a terminal or point at which the potential of the chassis of the device 2 may be measured. As above explained, the diode 5 causes the potential of the chassis at 15 to be separated from the network ground when a negative AC halfwave arrives. It will be noted that the return conduit of the circuit is held at a fixed chassis potential. The input transistor of the sensor circuit 3 remains, however, locked because it is subjected to a DC current of about 12 volts. If now, by means of touching the touching field 4, the chassis potential is connected to ground, then the transistor switches through and releases a switching function.

If the connecting terminals AB of the current source are exchanged, as by changing the plug, then there is still secured the condition that the chassis of the device is separated from the network ground via the diode, in this case the diode 1. The reference potential of the chassis consequently remains constant and the changing AC fields which are superimposed on the condensers can produce in the touching human body an AC current voltage due to the fields which are radiated by the device.

A suitable sensor which may be employed for the circuit 3 herein may be a sensor known as the "SAS 560 Tastatur IS," manufactured and sold by Siemens AG.

It is to be understood that the present invention is not limited to the illustrated environment. They can also be used in electronic blocking switch including a Thyristor circuit, which in the same manner separates the electronic device during the blocking phase from the network rectifier. With such Thyristor circuit the drawbacks described in the introductory portion of the specification of known circuit arrangements are also avoided.

Although the invention is illustrated and described with reference to a plurality of preferred embodiments thereof, it is to be expressly understood that it is in no way limited to the disclosure of such a plurality of preferred embodiments, but is capable of numerous modifications within the scope of the appended claims.

(AFC) Automatic frequency control circuit:
An automatic frequency control (AFC) circuit is disclosed which comprises an oscillating circuit for generating repetitive pulses, a generator for generating comparison signals having a slope portion in response to the repetitive pulses, and a phase comparison circuit which compares the comparison signals, and synchronous (sync) signals and based on the comparison supplies control signals to the oscillating circuit. The AF circuit further comprises a limiting circuit connected between the comparison signal generator and the phase comparison circuit which limits the maximum and minimum levels of the comparison signals to predetermined levels, and thereby predeterminedly limits the control range of the AFC circuit.

1. An automatic frequency control signal generating circuit, comprising:
an oscillating circuit for generating repetitive pulses;
means for receiving said repetitive pulses and generating comparison signals having sloped portions and maximum and minimum levels in response to said repetitive pulses;
a source of reference signals;
phase comparison means having a first input terminal supplied with said comparison signals, a second input terminal supplied with said reference signals and an output terminal for supplying automatic frequency control signals to said oscillating circuit for controlling its frequency within a control range; and
limiting means connected between said comparison signal generating means and said phase comparison means for limiting said maximum level and said minimum level of said comparison signals to first and second predetermined levels respectively, thereby limiting said control range of said automatic frequency control signal generating circuit to a predetermined range.
2. An automatic frequency control signal generating circuit according to claim 1; wherein said comparison signal generating means comprises integrating means for generating saw-tooth wave signals as said comparison signals in response to said repetitive pulses. 3. An automatic frequency control signal generating circuit according to claim 1; wherein said limiting means comprises a series circuit including sources of first and second reference potentials, first and second diodes connected together and respectively connected to said first reference potential and said second reference potential, the connecting point of said first and second diodes being connected to an output terminal of said comparison signal generating means. 4. An automatic frequency control signal generating circuit for a television receiver, comprising;
a source of reference signals;
an oscillator for generating control pulses;
an output circuit for producing repetitive pulses in response to said control pulses;
an integrating circuit for generating saw-tooth wave signals having maximum and miniumum levels in response to said repetitive pulses from said output circuit;
a phase comparator for comparing the phase of said saw-tooth waves and the phase of said reference signals, and supplying output signals to said oscillator as automatic frequency control signals; and
an amplitude limiting circuit connected between said integrating circuit and said phase comparator, for limiting said maximum and minimum levels of said saw-tooth wave signals to first and second predetermined levels.

Description:

BACKGROUND OF THE INVENTION
1. Field of the Invention
The present invention relates generally to an automatic frequency control (AFC) circuit and more particularly to an AFC circuit, preferably is used in a television receiver, which has a predeterminedly limited control range.
2. Description of the Prior Art
In a prior art horizontal AFC circuit of a television receiver, as shown in FIG. 1, the output from a horizontal oscillator 1 which can be, for example, a control pulse is supplied to a horizontal output circuit 2 and a flyback pulse therefrom is fed to a saw-tooth wave generator 3. Though not shown, the horizontal output circuit 2 comprises a deflection circuit and a high voltage generating circuit. The saw-tooth wave signal from saw-tooth wave generator 3 is supplied, as a comparison signal, to a phase comparator 4 which is also supplied with a horizontal synchronizing (sync) signal H through a terminal 5. In phase comparator 4 the saw-tooth wave signal is compared with the horizontal sync signal H to detect the phase difference between the saw-tooth wave signal and the horizontal sync signal. The detected phase difference is applied to horizontal oscillator 1 as an AFC voltage e _c . As shown in FIG. 2, when the horizontal sync signal H is coincident with the center φ ₀ of the downward sloping portions of the comparison signal, that is, when the phase difference between the comparison and sync signals is 0 (zero), the synchronization is maintained for phase differences of which corresponds to the ends of the downward sloping portion of the comparison signal. For phase differences of ±φ _cm , the AFC output voltage e _c has a maximum value of ±e _cm .
If the AFC control sensitivity is taken as β, the maximum frequency range within which the f _cm oscillating frequencies are controlled, hereinafter referred to as the control range, is: f _cm =±2πβe _cm
When the value of e _cm in the above formula is need constant, regardless of frequency variation, then the value of f _cm is constant.
Generally, the comparison signal, which is supplied by saw-tooth wave generator 3, is provided by integrating the flyback pulse. Both the width of the flyback pulse, which is determined by an LC resonance of horizontal output circuit, and the inclination of the rising portion of the comparison signal, which is determined by an RC time constant of the sawtooth generator, are constant. Therefore, the value of e _cm varies with the frequency resulting in a variable value of f _cm .
For example, as shown throughout FIGS. 3A, 3B, and 3C, both the width of the downward sloping portion and the inclination of the rising portion of the comparison signals are constant. As a result, when the frequency becomes high (FIG. 3A) as compared with its reference state (FIG. 3B), the value of e _cm becomes small as represented by e' _cm and when the frequency becomes low (FIG. 3C) as compared with the reference state (FIG. 3B), the value of e _cm becomes large as represented by e" _cm .
Therefore, when the frequency of the horizontal output signal is high, the differences between the minimum and maximum control signal amplitudes, hereinafter referred to as the amplitude range become smaller resulting in a reduced control range f _cm . In contrast thereto, when the frequency of the horizontal oscillator signal is low, the amplitude range becomes larger resulting in an increase of control range f _cm .
Typically for a variable control range as described heretofore, the AFC circuit is designed with the smaller control range, corresponding to high oscillating frequencies, as a reference. Such a reference, however, results in the control range at lower oscillating frequencies becoming either too large or at least larger than necessary.
When the control range at the lower oscillating frequencies is too large, the amplitudes of the control signals, that is, the values of ±e _cm , are too high or low to be applied to horizontal oscillator 1 and thereby results in an unacceptable frequency correction. Such unacceptably high or low values of e _cm can occur, for example, when the sync signal disappears during switching of television channels resulting in the oscillating frequency becoming too low and thereby creating a voltage in the horizontal output circuit that is abnormally high. Therefore, it is necessary that the control range of the AFC circuit be made as small as possible for lower frequencies.
Further, if the control range is too large, the AFC circuitry may unnecessarily adjust the oscillating frequency when noise is present on a weakly received signal.
Accordingly, an AFC circuit should necessarily have as small a control range as possible. Such a small control range is possible by providing maximum and minimum values of e _cm irrespective of frequency variation.
OBJECTS AND SUMMARY OF THE INVENTION
Accordingly, it is an object of the present invention to provide an AFC circuit which avoids the drawbacks of the prior art.
More specifically, it is an object of the present invention to provide a new and improved AFC circuit whose control range is constant.
It is another object of the present invention to provide a new and improved AFC circuit which limits the maximum and minimum values of the amplitude range.
According to an aspect of the present invention, an AFC circuit comprises:
an oscillating circuit for generating repetitive pulses;
means for receiving said repetitive pulses and generating comparison signals having a slope portion and maximum and minimum levels in response to said repetitive pulses, said generating means having an output terminal;
a source of reference signals;
phase comparison means having a first input terminal supplied with said comparison signals, a second input terminal supplied with said reference signals and an output terminal for supplying said automatic frequency control signal to said oscillating circuit for controlling its frequency within a control range; and
limiting means connected between said comparison signal generating means and said phase comparison means for limiting said maximum level and said minimum level of said comparison signals to first and second predetermined levels respectively, thereby limiting said control range of said automatic frequency control signal generating circuit to a predetermined range.
The above, and other objects, features and advantages of the present invention will become apparent from the following description which is to be read in conjunction with the accompanying drawings, in which like reference numerals designate like elements and parts.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a block diagram showing a prior art AFC circuit;
FIG. 2 and FIGS. 3A to 3C are waveform diagrams used to explain the operation of the prior art circuit shown in FIG. 1;
FIG. 4 is an embodiment of saw-tooth wave generating and limiter circuitry in accordance with the present invention;
FIGS. 5A, 5B, and 5C illustrate respectively input and output waveforms of the saw-tooth wave generator and the output waveform of the limiter shown in FIG. 4;
FIG. 6 is an alternative embodiment of circuitry which replaces the circuitry of FIG. 4 and produces a comparison signal in accordance with the present invention; and
FIGS. 7A and 7B illustrate respectively an input waveform supplied to and an output waveform produced by the circuitry shown in FIG. 6.
DESCRIPTION OF THE PREFERRED EMBODIMENTS
The present invention will be hereinafter described with reference to the attached drawings.
FIG. 4 includes an input terminal 11 connected to horizontal output circuit 2 (not shown) and an input capacitor 12. Input capacitor 12 is connected to resistors 13 and 14. Capacitor 12 and resistors 13 and 14 form a bias circuit which biases a base of an NPN-type transistor 15. Transistor 15 has a grounded emitter and a collector connected through a resistor 16 to a power supply terminal 17 which is at a voltage V _cc and through a series connection of a resistor 18 and a capacitor 19 to ground. The connection point between resistor 18 and capacitor 19 is connected to power supply terminal 17 through a series connection of a capacitor 20 and a resistor 21. The connection point of capacitor 20 and resistor 21 is connected to power supply terminal 17 through a resistor 22 and a diode 23 wherein the cathode and anode of diode 23 are respectively connected to terminal 17 and resistor 22. The connection point of capacitor 20 and resistor 21 is also connected to ground through resistor 22 and a series connection of a diode 24 and a capacitor 25 which is grounded wherein the cathode and anode of diode 24 are respectively connected to the connection point of resistor 22 and diode 23 and to capacitor 25. The connection point of diode 24 and capacitor 25 is connected to the connection point of voltage dividing resistors 26 and 27. Resistors 26 and 27 are connected between terminal 17 and ground. The connection point between diodes 23 and 24 is connected to a base of an NPN-type transistor 28 whose collector is connected to power supply terminal 17 and whose emitter is grounded through a resistor 29 and connected to an output terminal 30. Output terminal 30 is connected to an input terminal of phase comparator 4 (not shown).
Diodes 23 and 24 form a limiter circuit and capacitor 25 and resistors 26 and 27 form a direct current (d.c.) voltage source having a voltage level of E.
When a repetitive pulse, such as a flyback pulse, as shown in FIG. 5A, is applied to input terminal 11, a saw-tooth wave (FIG. 5B) is generated at the connection point of capacitor 20 and resistor 22 which in turn is fed to the limiter circuit which produces a waveform as shown in FIG. 5C. That is, the value e _cm is limited to an upper level V _cc +V _D and a lower level E-V _D (where V _D is the forward voltage drop of diodes 23 and 24).
Since the value e _cm is within a fixed range, the control range is constant resulting in a desired control range which is fixed regardless of frequency. That is, the present invention provides a control range which is constant irrespective of the frequency and thereby avoids the possibility of an unnecessary expansion of the control range at low frequencies and the resulting erroneous operation caused by the expanded control range.
An alternative embodiment of the present invention is shown in FIG. 6 which includes input terminal 11 connected through capacitor 12 to the base of a PNP-type transistor 31, which is biased by the resistors 13 and 14. Transistor 31 has an emitter connected to power supply terminal 17 and collector grounded through a parallel circuit of a resistor 32 and a capacitor 33. The collector of transistor 31 is also connected through a series connection of a capacitor 34 and a resistor 35 to an emitter of an NPN-type transistor 36. A base of transistor 36 is connected to a voltage dividing point of resistors 37 and 38 and a collector thereof is connected to power supply terminal 17 through a resistor 39. An emitter of transistor 36 is grounded through a resistor 40. Output terminal 30 is connected to the collector of transistor 36.
A repetitive pulse, such as a flyback pulse as shown in FIG. 7A, is supplied from the horizontal output circuit 2 and is applied to the input terminal 11. During those periods of each cycle when there is no flyback pulse, transistor 31 turns ON which charges capacitor 33 resulting in voltage at the emitter of transistor 36 rising and thereby causing transistor 36 to turn OFF. During periods when transistor 36 is turned off, output terminal 30 is at a voltage level of V _cc as shown in FIG. 7B. During that portion of each cycle when a flyback pulse occurs, transistor 31 turns OFF, allowing capacitor 33 to discharge through resistor 32. As a result, the potential at the emitter of transistor 36 gradually lowers and transistor 36 turns ON resulting in the output voltage at output terminal 30 gradually lowering to ground potential. When the period of the flyback pulse terminates, once again transistors 31 turns ON, charging capacitor 33 immediately, turning transistor 36 OFF and charging the voltage at output terminal 30 to V _cc . That is, a signal is produced having a magnitude of V _cc when the flyback pulse is absent and gradually lowering to ground potential during the flyback pulse period.
Thus the comparison signal, e _cm , magnitude is restricted to a range between V _cc and ground potential and thereby provides a desired control range regardless of the frequency. More particularly, in either embodiment the present invention provides a predetermined control range.
The present invention, as described heretofore has used the downward sloping portion of the comparison signal during phase comparison with a reference signal. However, the present invention can be applied as well to circuitry which provides a comparison signal having a rising slope portion during flyback pulse periods wherein control signal e _cm is selected from the rising slope portion.
Although illustrative embodiments of this invention have been described in detail herein with reference to the accompanying drawings, it is to be understood that the invention is not limited to those precise embodiments, and that various changes and modifications may be effected therein by one skilled in the art without departing from the spirit and scope of the invention, as defined in the appended claims. 

 

SONY DST EHT FBT TRANSFORMER Bobbin structure for high voltage transformers EHT Output.

A coil bobbin for a fly-back transformer or the like having a bobbin proper. A plurality of partition members or flanges are formed on the bobbin proper with a slot between adjacent ones. At least first and second coil units are formed in the bobbin proper, each having several slots, formed between the flanges, and first and second high voltage coils are wound on the first and second coil units in opposite directions, respectively. A rectifying means is connected in series to the first and second coil units, and a cut-off portion or recess is provided on each of the partition members. In this case, a wire lead of the coil units passes from one slot to an adjacent slot through the cut-off portion which is formed as a delta groove, and one side of the delta groove is corresponded to the tangent direction to the winding direction.


1. A fly-back transformer comprising a coil bobbin comprising a plurality of parallel spaced discs with a first adjacent plurality of said disc formed with delta shaped slots having first edges which extend tangentially to a first winding direction and a first winding wound on said first adjacent plurality of said discs in said first winding direction, a second adjacent plurality of said discs formed with delta shaped slots having first edges which extend tangentially to a second winding direction opposite said first winding direction and a second winding wound on said second adjacent plurality of said discs in said second winding direction, a third adjacent plurality of said discs formed with delta shaped slots having first edges which extend tangentially to said first winding direction and a third winding wound on said third adjacent plurality of said discs in said first winding direction and said second plurality of adjacent discs mounted between said first and third plurality of adjacent discs. 2. A fly-back transformer according to claim 1 wherein adjacent ones of said first adjacent plurality of discs are mounted such that their delta shaped slots are orientated 180 degrees relative to each other. 3. A fly-back transformer according to claim 2 including a first winding turning partition mounted between said first and second adjacent plurality of discs and formed with grooves and notches for changing winding direction between said first and second windings and a second winding turning partition mounted between said second and third adjacent plurality of discs and formed with grooves and notches for changing the winding direction between said second and third windings. 4. A fly-back transformer according to claim 3 wherein said first and second winding turning partitions are formed with winding guiding slots for guiding the winding between the first, second and third adjacent plurality of discs. 5. A fly-back transformer according to claim 2 including a first rectifying means connected between one end of said first winding and one end of said second winding, and a second rectifying means connected between the second end of said second winding and one end of said third winding. 6. A fly-back transformer according to claim 5 wherein the second end of said first winding is grounded and a third rectifying means connected between the second end of said third winding and an output terminal.

Description:

BACKGROUND OF THE INVENTION
1. Field of the Invention
The present invention relates generally to a bobbin structure for high voltage transformers, and is directed more particularly to a bobbin structure for high voltage transformer suitable for automatically winding coils thereon.
2. Description of the Prior Art
In the art, when a wire lead is reversely wound on a bobbin separately at every winding block, a boss is provided at every winding block and the wire lead is wound on one block, then one end of the wire lead is tied to the boss where it will be cut off. The end of the wire lead is tied to another boss, and then the wire lead is wound in the opposite direction. Therefore, the prior art winding method requires complicated procedures and the winding of the wire lead cannot be rapidly done and also the winding can not be performed automatically. Further, the goods made by the prior art method are rather unsatisfactory and have a low yield.
OBJECTS AND SUMMARY OF THE INVENTION
Accordingly an object of the invention is to provide a coil bobbin for a fly-back transformer or the like by which a wire lead can be automatically wound on winding blocks of the coil bobbin even though the winding direction is different among the different winding blocks.
Another object of the invention is to provide a coil bobbin for a fly-back transformer or the like in which a bridge member and an inverse engaging device for transferring a wire lead from one wiring block to an adjacent wiring block of the coil bobbin and wiring the wire lead in opposite wiring directions between adjacent wiring blocks, and a guide member for positively guiding the wire lead are provided.
According to an aspect of the present invention, a coil bobbin for a fly-back transformer or the like is provided which comprises a plurality of partition members forming a plurality of slots, a first coil unit having several slots on which a first high voltage coil is wound in one winding direction, a second coil unit having several slots on which a second high voltage coil is wound in the other direction, a rectifying means connected in series to the first and second coil units, and a cut-off portion provided on each of the partition members, a wire lead passing from one slot to an adjacent slot through the cut-off portions, each of the cut-off portions being formed as a delta groove, and one side of the delta groove corresponding to a tangent to the winding direction.
The other objects, features and advantages of the present invention will become apparent from the following description taken in conjunction with the accompanying drawings through which the like reference numerals and letters designate the same elements and parts, respectively.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a schematic diagram showing the construction of a fly-back transformer;
FIG. 2 is a connection diagram showing an example of the electrical connection of the fly-back transformer shown in FIG. 1;
FIG. 3 is a schematic diagram showing an example of a device for automatically winding a wire lead of the fly-back transformer on its bobbin;
FIG. 4 is a perspective view showing an example of the coil bobbin according to the present invention;
FIG. 5 is a plan view of FIG. 4;
FIGS. 6 and 7 are views used for explaining recesses or cut-off portions shown in FIGS. 4 and 5; and FIGS. 8A and 8B cross-sectional views showing an example of the inverse engaging means according to the present invention.
DESCRIPTION OF THE PREFERRED EMBODIMENT
When the high voltage winding of a fly-back transformer used in a high voltage generating circuit of a television receiver is divided into plural ones and then wound on a bobbin, the divided windings (divided coils) are connected in series through a plurality of rectifying diodes.
When the winding is divided into, for example, three portions, such as divided coils La, Lb and Lc, they are wound on a bobbin proper 1 from, for example, left to right sequentially in this order as shown in FIG. 1. In this case, if the divided coils La and Lc are selected to have the same sense of turn and the middle coil Lb is selected to have the opposite sense of turn from the coils La and Lc, the distance between the terminal end of coil La and the start of coil Lb and the distance between the terminal end of coil Lb and the start of coil Lc can be got relatively long. Therefore, diodes Da and Db can be mounted by utilizing the space above the block on which the middle coil Lb is wound as shown in FIG. 1, so that it becomes useless to provide spaces for diodes between the divided coils La and Lb and between the divided coils Lb and Lc and hence the bobbin proper 1 can be made compact.
FIG. 2 is a connection diagram showing the connection of the above fly-back transformer. In FIG. 2, reference numeral 2 designates a primary winding (Primary coil) of the fly-back transformer, reference letter L designates its high voltage winding (secondary coil), including divided coils La, Lb and Lc, 3 an output terminal, and 4 a lead wire connected to the anode terminal of a cathode ray tube (not shown), respectively.
An example of the bobbin structure according to the invention, which is suitable to automatically wind coils, which are different in sense of turn in each winding block as shown in FIG. 1, on the bobbin, will be hereinafter described with reference to the drawings.
FIG. 3 is a diagram showing an automatic winding apparatus of a wire lead on a coil bobbin. If it is assumed that the wire lead is wound in the order of winding blocks A, B and C in FIG. 1 and the wire lead is wound on the block A with the bobbin proper 1 being rotated in the counter-clockwise direction as shown in FIG. 3, the relation between the bobbin proper 1 and the wire lead becomes as shown in FIG. 3. In this figure, reference numeral 6 designates a bobbin for feeding the wire lead.
Turning to FIG. 4, an example 10 of the bobbin structure or coil bobbin according to the present invention will be described now. In this example, the winding blocks A, B and C for the divided coils La, Lb and Lc are respectively divided into plural slots or sections by plural partition members or flanges 11, and a cut-off portion or recess 12 is formed on each of the flanges 11 through which the wire lead in one section is transferred to the following winding section.
As shown in FIG. 6, each recess 12 is so formed that its one side extends in the direction substantially coincident with the tangent to the circle of the bobbin proper 1 and its direction is selected in response to the sense of turn of the winding or wire lead. In this case, the direction of recess 12 means the direction of the opening of recess 12, and the direction of recess 12 is selected opposite to the sense of turn of the winding in the present invention.
Now, recesses 12A, which are formed in the winding block A, will be now described by way of example. The positions of recesses 12A formed on an even flange 11Ae and an odd flange 11A ₀ are different, for example, about 180° as shown in FIGS. 6A and 6B. Since the bobbin proper 1 is rotated in the counter-clockwise direction in the winding block A and hence the sense of turn of the wire lead is in the clockwise direction, the recess 12A is formed on the even flange 11Ae at the position shown in FIG. 6A. That is, the direction of recess 12A is inclined with respect to the rotating direction of bobbin proper 1 as shown in FIG. 6A. In this case, one side 13a of recess 12A is coincident with the tangent to the circle of bobbin proper 1, while the other side 13b of recess 12A is selected to have an oblique angle with respect to the side 13a so that the recess 12A has a predetermined opening angle.
The opening angle of recess 12A is important but the angle between the side 13a of recess 12A and the tangent to the circle of bobbin proper 1 is also important in the invention. When the wire lead is bridged or transferred from one section to the following section through the recess 12A, the wire lead in one section advances to the following section in contact with the side 13a of recess 12A since the bobbin proper 1 is rotated. In the invention, if the side 13a of recess 12A is selected to be extended in the direction coincident with the tangent to the circle of bobbin proper 1, the wire lead can smoothly advance from one section to the next section without being bent.
In the invention, since the middle divided coil Lb is wound opposite to the divided coil La, a recess 12B provided on each of flanges 11B of the winding block B is formed to have an opening opposite to that of recess 12A formed in the winding block A as shown in FIGS. 6C and 6D.
As shown in FIG. 5, terminal attaching recesses 14 are provided between the winding blocks A and B to which diodes are attached respectively. In the illustrated example of FIG. 5, a flange 15AB is formed between the flanges 11A ₀ and 11B ₀ of winding blocks A and B, and the recesses 14 are formed between the flanges 11A ₀ and 15AB and between 15AB and 11B ₀ at predetermined positions. Then, terminal plates 16, shown in FIG. 4, are inserted into the recesses 14 and then fixed there to, respectively. The terminal plates 16 are not shown in FIG. 5. Between the winding blocks B and C and between the blocks A and B, similar terminal attaching recesses 14 are formed, and terminal plates 16 are also inserted thereinto and then fixed thereto.
As described above, since the divided coil Lb is wound opposite to the divided coils La and Lc, it is necessary that the winding direction of the wire lead be changed when the wire lead goes from the block A to block B and also from the block B to block C, respectively.
Turning to FIG. 7, an example of the winding or wire lead guide means according to the present invention will be now described. In FIG. 7, there are mainly shown a bridge member for the wire lead and an inverse member or means for the wire lead which are provided between the winding blocks A and B. At first, a bridge means 20 and its guide means 21, which form the bridge member, will be described. The bridge means 20 is provided by forming a cut-out portion or recess in the middle flange 15AB located between the winding blocks A and B. In close relation to the bridge means or recess 20, the guide means 21 is provided on a bridge section X _A at the side of block A. This guide means 21 is formed as a guide piece which connects an edge portion 20a of recess 20 at the winding direction side to the flange 11A ₀ of block A in the oblique direction along the winding direction through the section X _A .
Next, an inverse engaging means 22 will be now described with reference to FIGS. 7 and 8. If the flange 11B ₀ of FIG. 7 is viewed from the right side, the inverse engaging means 22 can be shown in FIG. 8A. In this case, the tip end of one side 13a of recess 12B ₁ is formed as a projection which is extended outwards somewhat beyond the outer diameter of flange 11B ₀ . The inverse engaging means 22 may take any configuration but it is necessary that when the rotating direction of the bobbin proper 1 is changed to the clockwise direction, the wire lead can be engaged with the recess 12B ₁ or projection of one side 13a and then suitably transferred to the next station.
Another guide means 23 is provided on a bridge section X _B at the side of winding block B in close relation to the inverse engaging means 22. The guide means 23 is formed as a guide surface which is a projected surface from the bottom surface of section X _B and extended obliquely in the winding direction. This guide means or guide surface 23 is inclinded low into the means 22 and has an edge 23a which is continuously formed between the middle flange 15AB and the flange 11B ₀ .
In this case, it is possible that the guide means 21 and guide surface 23 are formed to be the same in construction. That is, both the guide means 21 and 23 can be made of either the guide piece, which crosses the winding section or guide surface projected upwards from the bottom surface of the winding section. It is sufficient if the guide means 21 and 23 are formed to smoothly transfer the wire lead from one section to the next section under the bobbin proper 1 being rotated.
Although not shown, in connection with the middle flange 15BC between the winding blocks B and C, there are provided similar bridge means 20, guide means 21, inverse engaging means 22 and another guide means 23, respectively. In this case, since the winding direction of the wire lead is reversed, the forming directions of the means are reverse but their construction is substantially the same as that of the former means. Therefore, their detailed description will be omitted.
According to the bobbin structure of the invention with the construction set forth above, the wire lead, which is transferred from the block A to the section X _A by the rotation of bobbin proper 1, is wound on the section X _B from the section X _A after being guided by the guide piece 21 to the recess 20 provided on the middle flange 15AB, and then transferred to the recess 22 provided on the flange 11B ₀ guide surface 23, bridged once to the first section of winding block B through the recess 22 (refer to dotted lines b in FIG. 7). Then, if the rotating direction of the bobbin proper 1 is reversed, the wire lead is engaged with the bottom of recess 22 (refer to solid lines b in FIG. 7). Thus, if the above reverse rotation of bobbin proper 1 is maintained, the wire lead is wound on the block B in the direction reverse to that of block A. When the wire lead is transferred from the block B to block C, the same effect as that above is achieved. Therefore, according to the present invention, the wire lead can be automatically and continuously wound on the bobbin proper 1.
After the single wire lead is continuously wound on blocks A, B and C of bobbin proper 1 as set forth above, the wire lead is cut at the substantially center of each of its bridging portions. Then, the cut ends of the wire lead are connected through diodes Da, Db and Dc at the terminal plates 16, respectively by solder.
In the present invention, the projection piece, which has the diameter greater than that of the flange 11B, is provided in the bridge recess 12 to form the inverse engaging means 22 as described above, so that when the winding direction is changed, the wire lead engages with the inverse engaging means 22 without errors when reversing the winding direction of the wire lead.
If the diameter of the projection piece of means 22 is selected, for example, to be the same as that of the flange 11B, it will not be certain that the wire lead engages with the means 22 because it depends upon the extra length of the wire lead and hence errors in winding cannot be positively avoided.
Further, in this invention, the bridge means is provided on the flange positioned at the bridging portion of the bobbin which has a number of dividing blocks separated by flanges, and the inverse engaging means is provided and also the guide means is provided at the former winding section to cooperate with the inverse engaging means. Therefore, the wire lead can be positively fed to the bridge means, and the transfer of the wire lead to the following winding section can be carried out smoothly.
Further, in this invention since one side of the recess 12 is selected coincident with the tangent of the outer circle of the bobbin proper 1 and also with the winding direction, the wire lead can be smoothly bridged to the following section. Due to the fact that the direction of recess 12 is changed in response to the winding direction, even if there is a block on which the wire lead is wound in the opposite direction to that of the other block, the wire lead can be continuously and automatically wound through the respective blocks.
The above description is given for the case where the present invention is applied to the coil bobbin for the high voltage winding of a fly-back transformer, but it will be clear that the present invention can be applied to other coil bobbins which require divided windings thereon with the same effects.
It will be apparent that many modifications and variations could be effected by one skilled in the art without departing from the spirits or scope of the novel concepts of the present invention, so that the spirits or scope of the invention should be determined by the appended claims only. 

 

 

SONY TRINITRON Raster distortion correcting circuit:

Side, or left and right pin-cushion distortions in the raster of a cathode ray tube, for example, of a color television receiver having an in-line arrangement of its electron beams, are corrected by connecting the horizontal deflection winding of the cathode ray tube, the collector-emitter path of a transistor and the output winding of a saturable reactor, in series, to a power supply source, and by applying to the base or control electrode of the transistor and to the input winding of the saturable reactor a correction signal having a parabolic waveform of the vertical scanning rate or frequency so that correction of the side pin-cushion distortions is effected satisfactorily at all portions of the raster.


1. A raster distortion correcting circuit for a television receiver including a cathode ray tube in which at least one electron beam is directed against a screen, a deflection yoke associated with said tube and having horizontal and vertical deflection windings, and horizontal and vertical deflection circuits for supplying horizontal and vertical deflection currents to said horizontal and vertical deflection windings, respectively, so that the resulting magnetic fields cause each said beam to scan horizontally and vertically for forming a raster on the screen: said raster distortion correcting circuit comprising a power supply source for supplying a power supply voltage to said horizontal deflection circuit; an active element having first and second electrodes and a control electrode for varying the effective resistance between said first and second electrodes in dependence on a control signal applied to said control electrode; a saturable reactor having input and output windings, means for generating a correction signal at the vertical scanning rate of said vertical deflection current; circuit means for connecting said first and second electrodes of the active element between said power supply source and said horizontal deflection circuit to connect said active element, said horizontal deflection winding of the yoke and said output winding of the saturable reactor in a series circuit connected to said power supply source; and circuit means for applying said correction signal to said control electrode of the active element as the control signal for the latter and to said input winding of said saturable reactor so that said active element and saturable reactor combine to correct a distortion of said raster over all portions of the latter. 2. A raster distortion correcting circuit according to claim 1; in which said correction signal is generated with a parabolic waveform at said vertical scanning rate for correcting side pin cushion distortions of said raster. 3. A raster distortion correcting circuit according to claim 1; in which said means for generating the correction signal includes a capacitor connected in series with said vertical deflection winding of the yoke, and said circuit means for applying said correction signal extends from between said vertical deflection winding and said capacitor. 4. A raster distortion correcting circuit according to claim 1; in which said active element is a transistor having collector, emitter and base electrodes which constitute said first, second and control electrodes, respectively. 5. A raster distortion correcting circuit according to claim 1; in which said horizontal deflection current is supplied from said horizontal deflection circuit to said series circuit at a location in the latter between said first and second electrodes of said active element and said horizontal deflection winding of the yoke. 6. In a horizontal deflection circuit for a cathode ray tube, including a horizontal deflection current output circuit for supplying horizontal deflection current to a horizontal deflection winding and a source of operating voltage adapted to be supplied to said output circuit, a raster distortion correcting circuit comprising a saturable reactor having an input winding to receive a correction signal whose frequency is equal to the vertical deflection frequency of said cathode ray tube, and an output winding connected in series with said horizontal deflection winding; an active element for supplying said operating voltage to said output circuit; and means for supplying said correction signal to said active element to thereby vary said operating voltage supplied to said output circuit as a function of said correction signal. 7. A raster distortion correcting circuit in accordance with claim 6 wherein said correction signal is generated with a parabolic waveform.

Description:

BACKGROUND OF THE INVENTION
1. Field of the Invention
This invention relates generally to a raster distortion correcting circuit for a cathode ray tube, for example, of a color television receiver.
2. Description of the Prior Art
In a television receiver having a cathode ray tube, a deflection yoke is positioned about the neck of the cathode ray tube, and deflection circuits associated with such deflection yoke cyclically vary currents which are made to flow through windings of the yoke so that the windings generate varying electromagnetic fields by which each electron beam of the cathode ray tube is deflected vertically and horizontally to scan a respective raster on the screen of the cathode ray tube. In general, the raster formed by each electron beam is desired to be substantially rectangular. However, various types of scanning distortions may occur so as to cause the configuration of the generated raster to deviate from the desired rectangular shape. One of the types of raster distortions that may occur is the so-called "pin-cushion" distortion which may appear in respect to the top and bottom or left- and right-hand sides of the raster, and this invention is particularly concerned with providing corrections for the side, or left and right pin-cushion distortions.
Heretofore, such side, or left and right pin-cushion distortions in the raster of a cathode ray tube have been corrected by one or the other of several methods. One of the most frequently employed methods for achieving correction of side pin-cushion raster distortion involves varying or modulating the power supply voltage for the horizontal deflection circuit of the cathode ray tube in accordance with a parabolic wave having the vertical scanning rate or frequency. Another frequently employed method for achieving the foregoing raster correction involves the use of a saturable reactor having an output winding connected in series with the horizontal deflection winding of the yoke and an input winding to which there is applied a correction signal in the form of a parabolic wave having the vertical scanning rate or frequency so that the horizontal deflection current is again varied or modulated by such parabolic wave. Each of the foregoing methods that are frequently employed for correcting side, or left and right pin-cushion distortions in the raster of a cathode ray tube has its inherent advantages and disadvatages, as hereinafter described.
In the case where the power supply voltage for the horizontal deflection circuit is varied or modulated, as aforesaid, the horizontal deflection current I _h flowing through the horizontal deflection winding of the yoke is expressed by the following equation: ##EQU1## in which V _cc is the power supply voltage, L is the inductance value of the deflection winding, and t is time.
It will be apparent from the above equation that, when the power supply voltage V _cc is varied or modulated in accordance with a correction signal having a parabolic waveform at the vertical scanning rate, the amplitude of the horizontal deflection current is varied in accordance with such parabolic waveform so that correction of side pin-cushion distortions in the raster is achieved. Such correction of side pin-cushion distortions in the raster is advantageous in that the circuit required therefor is very simple and inexpensive. However, with this method, the horizontal deflection current is varied only at the vertical scanning rate, and not within each horizontal or line scanning period, so that, if a single horizontal scanning line is considered, the same correction is effected adjacent the center and adjacent the opposite or left- and right-hand sides of the screen. The foregoing characteristic of the described method is disadvantageous, particularly when applied to a color cathode ray tube having a wide deflection angle, and the disadvantage, as hereinafter described, is most serious in the case of a color cathode ray tube having an electron gun structure with a so-called in-line arrangement of the plural electron beams issuing therefrom.
In a color cathode ray tube having an electron gun structure with an in-line arrangement of the plural electron beams issuing therefrom, it is desirable that the electromagnetic field for effecting horizontal deflection or scanning of the beams have a pin-cushion shape and that the electromagnetic field for effecting vertical deflection or scanning of the beams have a barrel shape, that is, that the horizontal and vertical deflection fields be non-uniform, so as to correct or compensate for misconvergence of the plural electron beams as the latter are deflected horizontally and vertically from the center of the screen, for example, as disclosed in U.S. Pat. No. 3,500,114, issued Mar. 10, 1970, and having a common assignee herewith. When such non-uniform deflection fields are employed so as to correct or compensate for misconvergence of the electron beams, it has been determined experimentally that correction of side pin-cushion distortions of the raster by means of varying the power supply voltage for the horizontal deflection circuit as mentioned above, is insufficient, particularly in respect to the extent of the correction effected at the central portion of each horizontal scanning line. Therefore, in the case being described, a side pin-cushion distortion may still remain adjacent the central portion of the raster.
On the other hand, when a saturable reactor is employed for correcting side pin-cushion distortions, as aforesaid, such distortions are fully eliminated even near the central portion of the raster by reason of the fact that the inductance value of the output winding of the saturable reactor is varied in response to the correction signal applied to the input winding of the reactor and having a parabolic waveform at the vertical scanning rate, and the inductance value of the output winding of the saturable reactor is also varied at the horizontal scanning rate in response to the horizontal deflection current flowing through such output winding. However, when the side pin-cushion distortions are corrected only by means of the described saturable reactor, the apparatus required for correction of side pin-cushion distortions becomes bulky, heavy and expensive, particularly when applied to a color cathode ray tube having a relatively large deflection angle. Further, when side pin-cushion distortions of the raster are corrected only by means of the described saturable reactor in the case of a color cathode ray tube having a relatively wide deflection angle, it has been found that the desired linearity of the horizontal deflection of the beam or beams if seriously deteriorated.
SUMMARY OF THE INVENTION
Accordingly, it is an object of this invention to provide an improved circuit for correcting side pin-cushion raster distortions which avoids the above mentioned disadvantages inherent in the arrangements previously employed for that purpose.
More specifically, it is an object of this invention to provide a side pin-cushion raster distortion correcting circuit which is relatively small in size and weight and also inexpensive, and which is effective to fully eliminate such distortions near the central portion of the raster as well as near the opposite sides thereof.
Another object is to provide an improved side pin-cushion raster distortion correcting circuit, as aforesaid, which is suitable for a color cathode ray tube with a relatively wide deflection angle.
Still another object is to provide an improved side pin-cushion raster distortion correcting circuit which is particularly adapted for use with a color cathode ray tube having an electron gun structure with a so-called in-line arrangement of the plural electron beams emitted thereby.
In accordance with an aspect of this invention, side pin-cushion distortions in the raster of a cathode ray tube are eliminated by varying or modulating the power supply voltage for the horizontal deflection circuit in accordance with a correction signal having a parabolic waveform at the vertical scanning rate, and by simultaneously applying such correction signal to the input winding of a saturable reactor which has its output winding connected in series with the horizontal deflection winding of the cathode ray tube. By reason of the foregoing arrangement, the horizontal deflection current is modulated in accordance with the parabolic waveform at the vertical scanning rate by the combined action of a transistor or other active element employed for varying or modulating the power supply voltage and of the saturable reactor, and the horizontal deflection current is further modulated in response to the flow of such current through the output winding of the saturable reactor which varies its inductance in accordance with the horizontal deflection current flowing therethrough.
The above, and other objects, features and advantages of the invention, will be apparent in the following detailed description of illustrative embodiments which is to be read in connection with the accompanying drawings.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a diagrammatic view illustrating the side pin-cushion distortion that may remain near the central portion of the raster on the screen of a cathode ray tube when correction for the side pin-cushion distortion is effected only by varying or modulating the power supply voltage in accordance with a correction signal having a parabolic waveform at the vertical scanning rate;
FIG. 2 is a schematic elevational view of a saturable reactor that may be used in a raster distortion correcting circuit according to this invention;
FIG. 3 is a circuit diagram of a basic or simplified raster distortion correcting circuit in accordance with an embodiment of this invention;
FIGS. 4 and 5 are graphical representations of characteristics of the saturable reactor shown on FIG. 2; and
FIG. 6 is a circuit diagram showing a practical application of a raster distortion correcting circuit according to this invention in association with horizontal and vertical deflection circuits of a typical cathode ray tube.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS
Referring to the drawings in detail, and initially to FIG. 1 thereof, it will be seen that, when side pin-cushion distortions in the raster of a cathode ray tube are sought to be corrected or removed only by the known method of varying or modulating the power supply voltage for the horizontal deflection circuit in accordance with a parabolic waveform at the vertical scanning rate, the distortion is not fully corrected or removed near the central portion of the raster. Generally, in accordance with this invention, the side pin-cushion distortion of the raster is, for the most part, removed or corrected by varying or modulating the power supply voltage of the horizontal deflection circuit in accordance with a parabolic waveform at the vertical scanning rate, while the side pin-cushion distortion which remains uncorrected near the central portion of the raster is corrected or removed by modulating the horizontal deflection current in accordance with the horizontal scanning rate. Generally, the horizontal deflection current is modulated or varied in accordance with the horizontal scanning rate for removing the side pin-cushion distortion remaining near the central portion of the raster by means of a saturable reactor 1 having its output winding connected in series with the horizontal deflection winding of the cathode ray tube so that the horizontal deflection current, in passing through the output winding of the saturable reactor, will vary the inductance value of such output winding in accordance with the horizontal scanning rate.
As shown on FIG. 2, the saturable reactor 1 employed in a raster distortion correcting circuit according to this invention may include an E-shaped core 2 and an I-shaped core 3 which are both formed of a magnetically saturable material, and which are arranged relative to each other so that core 3 extends across the free ends of the legs of core 2 with a small gap g therebetween. Saturable reactor 1 further is shown to include an output winding constituted by two windings Lh ₁ and LH ₂ which are respectively wound on the outer legs of core 2, and an input winding Lv wound on the central leg of core 2. The directions in which windings Lh ₁ and Lh ₂ are respectively wound on the outer legs of core 2 are selected so that, when such windings are connected in series to constitute the output winding, the compound magnetic fluxes generated by the windings Lh ₁ and Lh ₂ in the central leg of core 2 are effective to oppose or cancel each other. Since saturable reactors of the type shown on FIG. 2 are well known, the construction and operation thereof will not be further described. However, it may be noted that the satuable reactor 1 for use in a raster distortion correcting circuit according to this invention may, for example, have a length l of 20mm, 11 turns in each of windings Lh ₁ and Lh ₂ , 500 turns in input winding Lv, and a gap g between cores 2 and 3 of about 50 microns.
Referring now to FIG. 3 in which only the basic or essential components of a circuit according to this invention for correcting or compensating for side pin-cushion distortions of the raster are illustrated, it will be seen that, in such circuit, the windings Lh ₁ and Lh ₂ constituting the output winding of saturable reactor 1 are connected in series with the horizontal deflection winding or windings 4 of the deflection yoke of a cathode ray tube. A transistor 5, which forms the output of a horizontal deflection circuit, receives a horizontal driving signal at its base electrode, while the emitter electrode of transistor 5 is connected to ground. The collector electrode of transistor 5 is connected to ground through a parallel circuit of a damper diode 6 and a tuning capacitor 7, and further connected to ground through the series connection of horizontal deflection winding or windings 4, the output winding of saturable reactor 1 and a capacitor 8 which is provided to correct for S-letter distortions. In accordance wtih this invention, an active element 9, shown in the form of a transistor, has its collector-emitter path connected in series with the primary winding 10 of a horizontal output transformer (which is not otherwise illustrated) between a power supply source or terminal 11 to which a positive DC voltage +B is applied, and the collector electrode of transistor 5. The base electrode of transistor 9 is connected to a terminal 12 which receives a correction signal having a parabolic waveform of the vertical scanning rate or frequency, and which may be generated as hereinafter described in detail. Such correction signal having a parabolic waveform of the vertical scanning rate is also applied to input winding Lv of saturable reactor 1.
The raster distortion correcting circuit of FIG. 3 operates as follows:
The correction signal having a parabolic waveform of the vertical scanning rate, when applied to terminal 12, varies the conductance of the collector-emitter path of transistor 9 so that the voltage at the emitter electrode of transistor 9, that is, the power supply voltage for transistor 5 forming the output of the horizontal deflection circuit is varied in accordance with such parabolic waveform. Therefore, the amplitude of the horizontal deflection current attains maximum values during horizontal scanning across the center of the raster or screen, considered in the vertical direction, and is relatively reduced during horizontal scanning across the upper and lower edge portions of the screen so that a correction for side, or left and right pin-cushion distortion is effected.
While the above described correction is being effected by the operation of transistor 9, the correction signal having a parabolic waveform of the vertical scanning rate is being simultaneously applied to input winding Lv of saturable reactor 1 having its output winding Lh ₁ , Lh ₂ connected in series with horizontal deflection winding 4. As indicated by the line l ₁ on FIG. 4, the compound inductance Lh of the windings Lh ₁ and Lh ₂ of saturable reactor 1 has a substantially linear relationship to a DC current Iv which flows through the control or input winding Lv of the saturable reactor. Of course, the value of the compound inductance Lh will become saturated when the current Iv reaches a sufficiently high value (not shown). Therefore, when a correction signal having a parabolic waveform, as indicated at Iv' on FIG. 4, is applied to input winding Lv with an appropriate DC voltage, the value of the compound inductance Lh of output windings Lh ₁ and L h ₂ undergoes a corresponding parabolic variation, as indicated at Lh' on FIG. 4. Since horizontal deflection winding 4 is connected in series with windings Lh ₁ and Lh ₂ of the saturable reactor, the horizontal deflection current Ih flowing through horizontal deflection winding 4 is also varied in accordance with the parabolic waveform of the correction signal applied to input winding Lv. Therefore, saturable reactor 1 also operates to provide a correction for a side or left and right pin-cushion distortion, which correction is added to that provided by the transistor 9, as previously described.
In the raster distortion correcting circuit according to this invention, the correction for side or left and right pin-cushion distortion is effected mainly by transistor 9, and only to a relatively smaller extent by saturable reactor 1. For example, from 70 to 80% of the required correction may be provided by the operation of transistor 9, while the remaining 30 to 20% of the required correction is provided by saturable reactor 1.
Referring again to FIG. 4, it will be apparent that the linear relationship between the compound inductance Lh of the output windings of saturable reactor 1 and the signal Iv applied to the input winding, as represented by the line l ₁ , assumes that the current through windings Lh ₁ and Lh ₂ of the saturable reactor is constant. However, in the circuit according to this invention as shown on FIG. 3, the connection of windings Lh ₁ and Lh ₂ in series with horizontal deflection winding 4 causes the current through windings Lh ₁ and Lh ₂ to be equal to the horizontal deflection current Ih which, of course, varies at the horizontal scanning rate so that the Lh-Iv characteristic curve of the saturable reactor is also changed in accordance with the varying value or amplitude of the horizontal deflection current Ih. More specifically, when the horizontal deflection current Ih increases, the saturable reactor 1 is directed to a more saturated condition, so that the compound or total inductance L h decreases and the rate of change of the inductance Lh relative to the input current or signal Lv also decreases. For example, on FIG. 5, the lines l ₂ , l ₃ , l ₄ and l ₅ represent the Lh -Iv characteristics of the saturable reactor 1 for progressively increasing values of the horizontal deflection current Ih.
By reason of the foregoing, it will be apparent that the correction or compensation for side pin-cushion distortion provided by saturable reactor 1 is at its maximum when each electron beam is directed at the center of the screen, considered in the horizontal direction, and is relatively reduced when each electron beam is directed toward one or the other of the opposite side edges of the screen. Thus, the side pin-cushion distortion which remains near the central portion of the screen when the power supply voltage for the horizontal deflection circuit is varied in accordance with a parabolic waveform at the vertical scanning rate, as by the transistor 9, may be completely removed by suitably selecting the parabolic waveform and the DC voltage level of the signal Iv' applied to the input winding of saturable reactor 1.
From the foregoing, it will be apparent that, in accordance with the present invention, side pin-cushion distortions are removed by modulating the power supply voltage applied to the collector of transistor 5, as by transistor 9, in accordance with a parabolic waveform at the vertical scanning rate and by similarly modulating the input signal to the winding Lv of saturable reactor 1, while the side pin-cushion distortion near the central portion of the screen is corrected or removed by saturable reactor 1 by reason of the connection of its output winding Lh ₁ , Lh ₂ in series with the horizontal deflection winding 4.
In theory, the input winding Lv of saturable reactor 1 may have applied thereto merely an appropriate DC voltage, rather than the described correction signal having a parabolic waveform at the vertical scanning rate. In that case, transistor 9 of the circuit shown in FIG. 3 has to be relied upon to provide the full correction for side pin-cushion distortions, while saturable reactor 1 then operates only to correct the pin-cushion distortion which remains near the central portion of the screen or raster. However, it is preferred that the input winding Lv of saturable reactor 1 also receive the correction signal having a parabolic waveform, as explained above, so that the saturable reactor can also operate to provide at least a portion of the correction for the side pin-cushion distortion, and thereby relieve a portion of the load on transistor 9.
When saturable reactor 1 is employed to provide only a portion of the correction for side pin-cushion distortion and to remove the remaining distortion near the center of the screen or raster, as in accordance with this invention, such saturable reactor can be relatively small and light in weight. For example, it has been found that a saturable reactor having a length of 30mm and the other dimensions given above will operate satisfactorily to perform the stated functions in connection with a color cathode ray tube having a screen with a 17 inch diagonal dimension and a 114° deflection angle. On the other hand, if the side pin-cushion distortion for such a color cathode ray tube is to be corrected only by a saturable reactor, rather than by the latter in combination with the modulating transistor 9, as shown on FIG. 3, such saturable reactor would have to be substantially larger, for example, have a length of 50mm, and have a mass or weight that is approximately five times greater than the saturable reactor with a length of 30mm which may be used in accordance with this invention.
Referring now to FIG. 6 in which circuit elements corresponding to those described above with reference to FIG. 3 are identified by the same reference numerals, it will be seen that a raster distortion correcting circuit according to this invention is there shown associated with typical horizontal and vertical deflection circuits for a cathode ray tube. More specifically, on FIG. 6, the horizontal deflection circuit is shown to include an input terminal 13 which receives horizontal driving pulses, for example, from a horizontal oscillator (not shown), a horizontal driver transistor 14 having its base electrode connected to terminal 13, and a horizontal drive transformer 15 having its primary winding connected in series with the collector-emitter path of transistor 14 between a power supply terminal and ground. The secondary winding of transformer 15 is connected to the control electrode of a semiconductor device 16, for example, to the gate electrode of a GCS (gate controlled switch), as shown, which functions as a horizontal output switching device and corresponds to the transistor 5 on FIG. 3.
The typical vertical deflection circuit 17 illustrated on FIG. 6 is shown to include an input terminal 18 which receives a saw-tooth wave signal at the vertical scanning rate, that is, in synchronism with the vertical synchronizing signal, for example, from a vertical oscillator (not shown), and transistors 19A and 19B which are connected, as shown, to form a single ended, push-pull output amplifier. The vertical deflection winding 20 of the deflection yoke associated with the cathode ray tube and a charge-discharge capacitor 21 are connected in series between the emitter electrode of transistor 19A and ground so that the desired correction signal having a parabolic waveform at the vertical scanning rate is obtained across capacitor 21. Such correction signal is applied to the base electrode of transistor 9 by way of terminal 12 so that transistor 9 will correspondingly modulate the power supply voltage applied to the horizontal output switching device 16, and hence the horizontal deflection current supplied to horizontal deflection winding 4, in the same manner as in the circuit of FIG. 3. In the circuit of FIG. 6, a variable resistor 22 is provided for adjusting the DC bias voltage applied to the base electrode of transistor 9, and thereby controlling the width of the horizontal deflection.
The correction signal having a parabolic waveform obtained across capacitor 21 is also applied to the base electrode of a transistor 26 through a variable resistor 23 and capacitor 24, with the variable resistor 23 serving to control the amplitude of the parabolic waveform as applied to the base electrode of transistor 26. A variable resistor 25 is connected in the base circuit of transistor 26 for adjusting the DC voltage or level of the correction signal having a parabolic waveform as applied to the input winding Lv of saturable reactor 1. A variable resistor 27 and a capacitor 28 are connected in series between the movable tap of variable resistor 23 and the ground so as to form a wave-shaping circuit for the correction signal having a parabolic waveform.
The input winding Lv of the saturable reactor 1 is connected in the collector circuit of transistor 26 so as to receive the correction signal with a parabolic waveform, as adjusted by the elements 23-25, 27 and 28. The output winding Lh ₁ , Lh ₂ of saturable reactor 1 is connected in series with the horizontal deflection winding 4 and with a coil 29 which is provided to achieve horizontal linearity compensation.
Although the circuit according to this invention for correcting side pin-cushion distortions is shown on FIG. 6 in association with typical horizontal and vertical deflection circuits of a cathode ray tube, which circuits do not appear on FIG. 3, it will be understood that the raster distortion correcting circuit according to this invention, as shown on FIG. 6, operates in the same manner as has been described above with reference to FIG. 3.
Having described specific embodiments of the present invention with reference to the accompanying drawings, it is to be understood that the invention is not limited to those precise embodiments, and that various changes and modifications may be effected therein by one skilled in the art without departing from the scope or spirit of the invention as defined in the appended claims.

 

SONY TRINITRON CONVERGENCE DEFLECTING DEVICE FOR SINGLE-GUN, PLURAL-BEAM COLOR PICTURE TUBE In a color picture tube of the single-gun, plural-beam type in which a central beam and two side beams originate in a common horizontal plane and are all made to pass through the center of an electron lens for focussing the beams on the color screen with the central beam emerging from the lens along the optical axis of the latter and the side beams emerging from the lens along paths that are oppositely divergent from the axis, the divergent side beams are acted upon by an electrostatic convergence deflecting device constituted by pairs of horizontally spaced plates arranged along the divergent paths and having voltages applied thereacross to produce electric fields by which the divergent side beams passing therethrough are deflected to converge at a common spot with the central beam on the apertured grill or mask associated with the screen, and a main deflection yoke produces magnetic fields by which the beams are deflected horizontally and vertically, respectively, for causing the beams to scan the screen; the horizontal distances between the plates of each pair are varied in the vertical direction from a maximum at the common horizontal plane to minimums at the opposed edges of the plates remote from such common plane so as to correspondingly vary the strengths of the electric fields and thus correct distortions in the rasters of the side beams. 1. A single-gun, plural-beam color picture tube comprising a color screen, beam generating means directing a central beam and two side beams in a common horizontal plane toward said screen, electron lens means defining a focusing field having a center through which the beams pass and by which the bundle of electrons in each of the beams are focused on said color screen with the central beam emerging from said lens along the optical axis of the latter and said two side beams emerging from said lens along paths that are oppositely divergent from said central beam, electrostatic convergence deflecting means including a pair of horizontally spaced plates arranged along each of said divergent paths, said spaced plates of each pair being disposed at the inside and outside, respectively, of the side beam in the related divergent path and having voltages applied thereacross to produce an electric field by which the respective side beam passing therethrough is deflected horizontally to converge at a common spot with said central beam and the other of said side beams, and a main deflection yoke producing magnetic fields by which said beams are deflected horizontally and vertically respectively, for causing the beams to scan said screen and produce respective rasters on the latter; said convergence deflecting means being located within the field produced by said yoke to deflect said beams vertically so that said beams are similarly deflected vertically within said convergence deflecting means, and the horizontal distance between said plates of each of said pairs varying progressively in the vertical direction normal to said common horizontal plane from a maximum at said common horizontal plane to minimums at the opposed edges of the plates remote from said common plane so as to correspondingly vary the strength of the respective electric field for changing the rasters of said side beams with respect to the raster of said central beam and thereby compensating for deviations between said rasters as produced by said magnetic fields of the main deflection yoke. 2. A single gun, plural-beam color picture tube according to claim 1, in which the inner plate of each of said pairs which is closest to said central beam is flat, and the other plate of the respective pair is convex at the side thereof facing away from said inner plate. 3. A single-gun, plural-beam color picture tube according to claim 1, in which the plates of each of said pairs are convex at the sides thereof facing away from each other.

Description:

This invention relates generally to color picture tubes of the single-gun, plural-beam type, and particularly to tubes of that type in which the plural beams are passed through the optical center of a common electron lens by which the beams are focussed on the color phosphor screen. In single-gun, plural-beam color picture tubes of the described type, for example, as specifically disclosed in U.S. Pat. No. 3,448,316, issued June 3, 1969, and having a common assignee herewith, three laterally spaced electron beams are emitted by a beam generating or cathode assmebly and directed in a common substantially horizontal plane with the central beam coinciding with the optical axis of the single electron focussing lens and the two outer or side beams being converged to cross the central beam at the optical center of the lens and thus emerge from the latter along paths that are divergent from the optical axis. Arranged along such divergent paths are respective pairs of convergence deflecting plates constituting a convergence deflecting device and having voltages applied thereacross to produce electric fields which laterally deflect the divergent beams in a substantially horizontal plane for causing all beams to converge at a common spot on the apertured beam selecting grill or shadow mask associated with the color screen. Further, arranged between the convergence deflecting device and the screen is a main deflection yoke which, in response to its reception of horizontal and vertical sweep signals, produces horizontal and vertical magnetic deflection fields acting on all of the beams to cause the latter to scan the color screen in predetermined rasters. Since the beams are horizontally spaced and non-parallel during their passage through the horizontal deflection field, the distances that the side beams travel through such field will be respectively greater and less than the distance that the central beam travels through the field when the beams are deflected toward one side or the other side of the screen. If the horizontal deflection field has a uniform flux density thereacross, the side beam traveling therethrough for the greater distance will be deflected to a greater extent than the side beam traveling the shorter distance through the field and misconvergence of the beams will result. Even if the horizontal deflection field is given a non-uniform flux density thereacross, misconvergence of the beams can be thereby avoided only when the beams are deflected toward one side or the other of the screen midway between the top and bottom of the screen, that is, when the common plane of the beams passing through the horizontal deflection field is directed horizontally, that is, substantially perpendicular to the vertical lines of magnetic flux of the horizontal deflection field. However, when the common plane of the beams passing through the horizontal deflection field is substantially inclined from the horizontal, that is, when the vertical deflection field deflects the beams to cooperate with the horizontal deflection field in directing the beams toward an upper or lower corner of the screen, the difference between the distances traveled by the side beams through the horizontal deflection field is further increased and hence may not be compensated by the non-uniform flux density established across the horizontal deflection field. Thus, the rasters of the side beams may have shapes that are oppositely distorted with respect to the shape of the raster of the central beam. Accordingly, it is an object of this invention to provide a single-gun, plural-beam color picture tube in which the rasters of the several beams are free of distortion with respect to each other. Another object is to provide a single-gun, plural-beam color picture tube in which distortions of the rasters of the several beams are avoided by a particular construction of the convergence deflecting device. In accordance with an aspect of the invention, the described distortions of the rasters of the side beams with respect to the raster of the central beam are avoided by suitably varying, in the direction perpendicular to the common plane in which the beams originate, the distances between the paired plates of the convergence deflecting device, whereby to correspondingly vary the strengths of the electric fields between such plates by which the side beams are convergently deflected. The above, and other objects, features and advantages of this invention, will be apparent in the following detailed description of illustrative embodiments which is to be read in connection with the accompanying drawing, wherein: FIG. 1 is a schematic sectional view in a horizontal plane passing through the axis of a single-gun, plural-beam color picture tube of the type to which this invention is preferably applied; FIG. 2 is a diagrammatic view to which reference is hereinafter made in explaining the invention; FIG. 3 is a diagrammatic view showing the possible relative distortions of the rasters of the several beams, as seen from the viewer's side of the tube screen, and which are to be avoided by this invention; FIG. 4 is a diagrammatic, transverse sectional view through the convergence deflecting device of a color picture tube according to a first embodiment of this invention; and FIGS. 5-8 are views similar to FIG. 4, but showing other embodiments of the invention. Referring to the drawings in detail, and initially to FIG. 1 thereof, it will be seen that a single-gun, plural-beam color picture tube of the type to which this invention may be applied comprises a glass envelope (indicated in broken lines) having a neck N and cone C extending from the neck to a color screen S provided with the usual arrays of color phosphors S _R , S _G and S _B and with an apertured beam selecting grill or shadow mask G _P . Disposed within neck N is an electron gun A having cathodes K _R , K _G and K _B , each of which is constituted by a beam-generating source with the respective beam-generating surfaces thereof disposed as shown in a plane which is substantially perpendicular to the axis of the electron gun A. In the embodiment shown, the beam-generating surfaces are arranged in a straight line so that the respective beams B _R , B _G and B _B emitted therefrom are directed in a substantially horizontal plane containing the axis of the gun, with the central beam B _G being coincident with such axis. A first grid G ₁ is spaced from the beam-generating surfaces of cathodes K _R , K _G and K _B and has apertures g _1R , g _1G and g _1B formed therein in alignment with the respective cathode beam-generating surfaces. A common grid G ₂ is spaced from the first and grid G ₁ and has apertures g _2R , g _2G and g _{2B 1} . Successively arranged in the axial direction away from the common grid G ₂ are open-ended, tubular grids or electrodes G ₃ , G ₄ and G ₅ , respectively with cathodes K _R , K _G and K _B , grids G ₁ and G ₂ , and electrodes G ₃ , G ₄ and G ₅ being suitably maintained in the depicted, assembled positions thereof. formed therein in alignment with the respective apertures of the first grid G For operation of the electron gun A of FIG. 1, appropriate voltages are applied to the grids G _{1 2} and to the electrodes G ₃ , G ₄ and G ₅ . Thus, for example, a voltage of 0 to minus 400V is applied to the grid G ₁ , a voltage of 0 to 500V is applied to the grid G ₂ , a voltage of 13 to 20KV is applied to the electrodes G ₃ and G ₅ , and a voltage of 0 to 400V is applied to the electrode G ₄ , with all of these voltages being based upon the cathode voltage as a reference. As a result, the voltage distributions between the respective electrodes and cathodes, and the respective lengths and diameters thereof, may be substantially identical with those of a unipotential-single beam type electron gun which is constituted by a single cathode and first and second, single-apertured grids. and G With the applied voltage distribution as described hereinabove, an electron lens field will be established between grid G ₂ and the electrode G ₃ to form an auxiliary lens L' as indicated in dashed lines, and an electron lens field will be established around the axis of electrode G ₄ , by the electrodes G ₃ , G ₄ and G ₅ , to form a main lens L, again as indicated in dashed lines. In a typical use of electron gun A, bias voltages of 100V, 0V, 300V, 20KV, 200V and 20V may be applied respectively to the cathodes K _R , K _G and K _B , the first and second grids G ₁ and G ₂ and the electrodes G ₃ , G ₄ and G ₅ . Further included in the electron gun A of FIG. 1 and electron beam convergence deflecting means F which comprise inner shielding plates P and P' disposed in the depicted spaced, relationship at opposite sides of the gun axis, and axially extending, deflector plates Q and Q' which are disposed, as shown, in outwardly spaced, opposed relationship to shielding plates P and P', respectively. Although depicted as substantially straight, it is to be understood that the deflector plates Q and Q' may, alternatively, be somewhat curved or outwardly bowed, as is well known in the art. The shielding plates P and P' are equally charged and disposed so that the central electron beam B _G will pass substantially undeflected therebetween, while the deflector plates Q and Q' have negative charges with respect to the plates P and P' so that electron beams B _B and B _R will be convergently deflected as shown by the respective passages thereof between the plates P and Q and the plates P' and Q'. More specifically, a voltage V _P which is equal to the voltage applied to the electrode G ₅ , may be applied to both shielding plates P and P', and a voltage V _Q , which is some 200 to 300V lower than the voltage V _P , is applied to the plates Q and Q' to result in the plates P and P' being at the same potential, and in the application of a deflecting voltage difference or convergence deflecting voltages V _C between the plates P' and Q' and the plates P and Q and it is, of course, this convergence deflecting voltage V _C which will impart the requisite convergent deflection to the electron beams B _B and B _R . In operation, the electron beams B _R , B _G and B _B which emanate from the beam generating surfaces of the cathodes K _R , K _G and K _B will pass through the respective grid apertures g _1R , g _1G and g _1B , to be intensity modulated with what may be termed the "red", "green" and "blue" intensity modulation signals applied between the said cathodes and the first grid G ₁ . The electron beams will then pass through the common auxiliary lens L' to cross each other at the center of the main lens L. Thereafter, the central electron beam B _G will pass substantially undeflected between sheilding plates P and P' since the latter are at the same potential. Passage of the electron beam B _B between the plates P' and Q' and of the electron beam B _R between the plates P and Q will, however, result in the convergent deflections thereof as a result of the convergence deflecting voltage V _Q applied therebetween, and the system of FIG. 1 is so arranged that the electron beams B _B , B _G and B _R will desirably converge or cross each other at a common spot centered in an aperture of the beam selecting grill or mask G _P so as to diverge therefrom to strike the respective color phosphors of a corresponding array thereof on screen S. More specifically, it may be noted that the color phosphor screen S is composed of a large plurality of sets or arrays of vertically extending "red", "green" and "blue" phosphor stripes or dots S _R , S _{G B} with each of the arrays or sets of color phosphors forming a color picture element. Thus, it will be understood that the common spot of beam convergence corresponds to one of the thusly formed color picture elements. and S The voltage V _P may also be applied to the lens electrodes G ₃ and G ₅ and to the screen S as an anode voltage as well as to the aperture grill G _p . Electron beam scanning of the face of the color phosphor screen is effected in conventional manner, for example, main deflection yoke means indicated in broken lines at D and which receives horizontal and vertical sweep signals to produce horizontal and vertical deflection fields by which the beams are made to scan the screen for providing a color picture thereon. Since, with this arrangement, the respective electron beams are each passed, for focussing, through the center of the main lens L of the electron gun A, the beam spots formed by impingement of the beams on the color phosphor screen S will be substantially free from the effects of coma and/or astigmatism of the same main lens, whereby improved color picture resolution will be provided. In the color picture tube as illustrated on FIG. 1, plates P and P' and plates Q and Q' are shown flat and parallel with each other so that the electric fields between plates P and Q and plates P' and Q' are substantially uniform thereacross, that is, in the direction perpendicular to the common horizontal plane of beams B _B , B _G and B _B . Thus, as the beams are vertically deflected by the vertical deflection field of yoke D so as to be directed at the upper or lower portions of screen S and such vertical deflection field vertically displaces the beams within convergence deflecting device F, the deflecting effects on beams B _B and B _R of the fields between plates P and Q and plates P' and Q', respectively, are substantially unchanged. However, as shown on FIG. 2, when the horizontal deflection field of yoke D deflects the beams toward the left side of the screen as seen from the viewer's side of the latter, that is, downwardly as viewed on FIG. 2, the side beams B _B and B _R travel distances through such horizontal deflection field that are respectively greater than and smaller than the distance that the central beam B _G travels through the horizontal deflection field. Conversely, when the horizontal deflection field of yoke D deflects the beams toward the right side of the screen as viewed from the viewer's side, the distances traveled by the beams B _B and B _R through the horizontal deflection field are respectively smaller than and greater than the distance that the central beam B _G travels through such field. By reason of the foregoing differences between the distances that the beams travel through the horizontal deflection field when deflected by the latter toward one side or the other of the screen S, the raster of beam B _B and the raster of beam B _R would be displaced toward the left and toward the right, respectively, from the raster of the beam B _G , as seen from the viewer's side of the screen. If the horizontal deflection field of yoke D is given a non-uniform flux density thereacross, for example, a greater flux density at the sides than at the middle of the field, the described relative displacements of the rasters can be compensated for so long as the common plane of the beams is substantially horizontal, that is, so long as the beams are directed at the screen substantially midway between the top and bottom of the screen. However, when the horizontal deflection field of yoke D directs the beams toward one side or the other of the screen at a time when the vertical deflection field of yoke D deflects the beams vertically so that the common plane of the beams is substantially inclined from the horizontal to direct the beams toward a corner of the screen, the differences between the distances traveled by the beams through the horizontal deflection field are further increased, as compared with the differences in the distances traveled through the field when the common plane of the beams is horizontal, so that even the mentioned non-uniform flux density across the horizontal deflection field of yoke D would be ineffective to avoid distortions of the rasters of beams B _B and B _R relative to the raster of beam B _G . Assuming that the raster of central beam B _G has a rectangular shape, as indicated at L _G on FIG. 3, the raster L _B of beam B _B , as seen from the viewer's side of the screen, is distorted in the sense that its sides are convex toward the right, while the raster L _R of beam B _R is oppositely distorted, that is, its sides are convex toward the left. In accordance with the present invention, such distortions of the rasters of side beams B _B and B _R relative to the raster of central beam B _G are avoided by suitably varying, in the direction perpendicular to the common plane in which the beams originate, for example, in the vertical direction for the tube of FIG. 1, the distances by which plates P and Q and plates P' and Q' are spaced from each other. For example, in the embodiment shown by FIG. 4, plates P and P' are made flat or planar while plates Q and Q' are outwardly concave in the vertical direction or the direction across the plates, whereby the distances between plates P and Q and between plates P' and Q' are relatively small at the horizontal plane 21 containing the tube axis and such distances between the plates increase progressively in the direction of vertical plane 22 upwardly and downwardly from horizontal plane 21 in which the beams all originate. Since convergence deflecting device F is disposed adjacent the main deflecting yoke D (FIG. 1), it will be apparent that the vertical deflection field of yoke D will extend into device F, and thereby influence the vertical positions of the beams B _B , B _G and B _R in passing through device F. Thus, when the vertical and horizontal deflection fields of yoke D are effective to direct the beams toward a corner of the screen, the vertical deflection field of yoke D will vertically displace beams B _R , B _G and B _B either upwardly or downwardly from plane 21 within convergence deflection device F. By reason of the increased distance betweeen plates P and Q and plates P' and Q' at such displaced positions of beams B _B and B _R , the parts of the electric fields then acting on such beams will be of relatively reduced intensity thereby to similarly reduce the convergent deflections imparted to beams B _B and B _R . Thus, for example, when the beams are horizontally and vertically deflected by yoke D so as to be directed at the upper or lower left-hand corner of the screen, as seen from the viewer's side thereof, the left-ward deflection of beam B _B by the field between plates P and Q will be reduced and the right-ward deflection of beam B _R by the field between plates P' and Q' will be similarly reduced, whereby to bring the left-hand sides of the rasters L _B and L _R , as seen on FIG. 3, into agreement with the left-hand side of the raster L _G . Similarly, when the beams are horizontally and vertically deflected by yoke D so as to be directed at the upper or lower right-hand corner of the screen as viewed on FIG. 3, the left-ward and right-ward deflections of beams B _B and B _R , respectively, by the fields between plates P and Q and plates P' and Q' will be reduced whereby to bring the right-hand sides of rasters L _B and L _R into agreement with the right-hand side of raster L _G . Thus, distortions of the rasters L _B and L _R relative to the raster L _G can be effectively avoided by suitably selecting the position of convergence deflecting device F relative to yoke D and the shapes of plates Q and Q'. As shown on FIGS. 5 and 7, the effect described above may also be achieved by providing flat or planar outer plates Q and Q' and outwardly convex inner plates P and P' (FIG. 5), or by providing outer plates Q and Q' that are inwardly convex and inner plates P and P' that are outwardly convex (FIG. 7). In each of these modifictions, the distances between plates P and Q and between P' and Q' vary from a minimum at the horizontal plane passing through the tube axis to maximums at the upper and lower portions of the plates to conversely vary the strengths of the electrical fields between plates P and Q and plates P' and Q'. Since plates P and P' are at equal potential there is no electric field established therebetween, and thus the varying distance between plates P and P' in FIGS. 5 and 7 does not affect beam B _G as the latter is vertically deflected. Of course, in the foregoing, it has been assumed that the distortions of rasters L _B and L _R relative to raster L _G that are to be corrected are those shown on FIG. 3. However, a situation may arise, for example, by reason of a particular configuration of the horizontal deflection field produced by yoke D, in which the raster of beam B _B has the shape indicated at L _R on FIG. 3 while the raster of beam B _R has the shape indicated at L _R . In the latter case, the plates P and Q and the plates P' and Q' are shaped so that the distances therebetween are maximum at the horizontal plane containing the axis of the tube and decrease progressively therefrom in the vertical direction, that is, in the direction perpendicular to the common plane in which the beams originate. In achieving such variations in the distances between the plates, plates P and P' may be flat or planar and plates Q and Q' may be outwardly convex (FIG. 6), or plates P and P' may be inwardly convex and plates Q and Q' may be outwardly convex (FIG. 8). Further, in each of the above described embodiments of this invention, the convergence deflection device F consists of only a single pair of plates P and Q or P' and Q' acting on each of the beams B _B and B _R to deflect the respective beam in the direction for convergence with the central beam B _G . However, the invention can also be applied to color picture tubes, for example, as disclosed in the copending U.S. application Ser. No. 718,738, filed Apr. 4, 1968, and having a common assignee herewith, in which the beams following paths diverging from the tube axis upon emerging from the focussing lens are each successively acted upon by two pairs of deflecting plates, with the first pair of plates establishing an electric field therebetween by which the respective beam is further diverged from the tube axis and the second pair of plates establishing a field therebetween by which the beam is deflected in the direction for converging with the other beams. The foregoing arrangement makes it possible to increase the angles of incidence of the side beams B _B and B _R at the beam selecting apertured grill or mask G _P , whereby to permit a decrease in the distance of the latter from screen S for facilitating the accurate locating and mounting of the grill or mask G _P relative to the screen S. Where each of the side beams B _B and B _R is successively acted upon by two pairs of deflecting plates, as aforesaid, one or the other of such pairs of plates, and preferably the pair of plates closest to the location of the main deflection yoke, is provided with a distance between the plates that varies in the direction perpendicular to the common plane in which the beams originate so as to avoid distortion of the raster of the respective beam in accordance with this invention. Having described various embodiments of this invention, it is to be understood that the invention is not limited to those precise embodiments, and that various changes and modifications may be effected therein by one skilled in the art without departing from the scope or spirit of the invention.

 

 HITACHI 2SC1942 HORIZONTAL DEFLECTION TRANSISTOR.

 Material of Transistor: Si
Polarity: NPN
Maximum Collector Power Dissipation (Pc): 50 W
Maximum Collector-Base Voltage Vcb: 1500 V
Maximum Collector-Emitter Voltage Vce: 800 V
Maximum Emitter-Base Voltage Veb: 6 V
Maximum Collector Current Ic max: 3 A
Max. Operating Junction Temperature (Tj): 200 °C
Forward Current Transfer Ratio (hFE), MIN: 30

 

 

Power supply circuit:

 

ply circuit for supplying stabilized relatively low DC voltages to at least first and second load circuits as operating power voltages for the latter without the use of a power transformer, and in which the first and second load circuits are connected in series to each other across a DC voltage source supplying a relatively high DC voltage for dividing the latter between the first and second load circuits in accordance with their respective impedances; an additional DC voltage souce is connected between both load circuits to produce an additional DC voltage varying in response to variations in the voltage applied across the first load circuit, for example, as a result of variations in the impedance of the latter, and to apply the additional DC voltage to the second load circuit so as to compensate for the variations in the voltage applied across said second load circuit as a result of the variations in the voltage applied across the first load circuit, thereby to stabilize the relatively low DC voltages applied to both load circuits.
 
 1. A power supply circuit for supplying stabilized direct currents to at least first and second load circuits comprising:

A. a direct current source means having a pair of output terminals and being operative to produce a DC voltage between said terminals,

B. means connecting said first and second load circuits in series to each other between said pair of output terminals of said direct current source means so that said first and second load circuits have respective DC voltages applied thereacross in dependence on their respective impedances, and

C. control means connected to said first load circuit for producing an additional direct current varying in response to variations in the DC voltage applied across said first load circuit, said control means being connected to said second load circuit for applying said additional direct current in series with the direct current from said direct current source means to said second load circuit to compensate for the variations in the DC voltage applied across said second load circuit that result from said variations in said DC voltage applied across said first load circuit.


2. A power supply circuit according to claim 1, wherein said DC voltage source means includes means adapted to be connected to an AC voltage source, and means for rectifying the AC voltage from said AC voltage source.

3. A power supply circuit according to claim 2, wherein said first load circuit includes a transformer as an inductive load having primary and secondary windings, and said control means includes a rectifying device having an input connected to said secondary winding and an output connected to a power supply terminal of said second load circuit to supply part of the current to said second load circuit and to affect the voltage applied to said first load circuit in response to variations in operation of said second circuit.

4. A power supply circuit according to claim 3, wherein said first load circuit includes an output stage of a horizontal deflection circuit and said transformer acts as a flyback transformer of said horizontal deflection circuit and said second load circuit includes a vertical deflection circuit.

5. A power supply circuit according to claim 4, wherein said rectifying device in said control means is constituted by a diode connected between said secondary winding of the flyback transformer and said power supply terminal of said vertical deflection circuit.

Description:

BACKGROUND OF THE INVENTION

1. Field of the Invention

This invention relates generally to power supply circuits, and more particularly is directed to an improved circuit for supplying stabilized operating DC voltages to a plurality of load circuits.

2. Description of the Prior Art

In an electric apparatus, such as television receivers, a radio receivers or the like, having circuits of which include transistors and other semiconductor devices and which are supplied with operating power from a commercial electric power lines, the limits of the endurable voltage of the transistors and other semiconductor device and/or desired reductions of power consumption, make it necessary to produce relatively low DC voltages from the AC voltage of the commercial electric power lines for application, as operating DC voltages, to the circuits in which the transistors and other semiconductor devices are included. In such cases, the AC voltage, for example, having a value of 100 volts, of the commercial electric power lines is usually stepped down by a power transformer, and the stepped down AC voltage is rectified to provide the relatively low DC operating voltages, for example, in the range of 12 to 50 volts. However, in apparatus using transistorized circuits a relatively large current flows through a power supply circuit, and therefore a relatively large power transformer is required with the result that the weight of the apparatus and the cost of the latter are undesirably increased.

To avoid the above disadvantages, it has been proposed to provide a circuit arrangement without the power transformer in which plural load circuits, for example, a horizontal deflection circuit and a vertical deflection circuit of a television receiver, are connected in series between output terminals of a rectifier circuit which produces a relatively high DC voltage at its output terminals by directly rectifying the AC voltage of the commercial electric power lines. In this case, the load circuits are supplied with respective relatively low DC voltages by dividing the high DC voltage obtained at the output terminals of the rectifier circuit. However, there is still the disadvantage that variation in the impedance of one of the load circuits causes variations in the low DC voltages applies to the respective load circuits, because the low DC voltages applied to the load circuits are obtained by dividing the high DC voltage in proportion to the respective impedances of the series connected load circuits.

SUMMARY OF THE INVENTION

Accordingly, it is an object of the present invention to provide an improved power supply circuit for producing DC operating voltages without using a power transformer.

A more specific object of the invention is to provide an improved power supply circuit, in which a plurality of load circuits are connected in series between output terminals of a rectifier circuit which directly rectifies an AC voltage from a supply source to provide a relatively high DC voltage at such output terminals, as aforesaid, and in which the resulting low DC voltages respectively applied to the load circuits are stabilized, that is, are not subject to variations in response to variations in the impedance of either of the load circuits.

In accordance with an aspect of this invention, the low DC voltages applied to the load circuits by a power supply circuit, as aforesaid, are stabilized by providing a control means connected to the load circuits for producing an additional DC voltage varying in response to variations in the DC voltage applied across one of the load circuits, and by applying such additional DC voltage to another load circuit to compensate for the variations in the DC voltage applied across that other load circuit resulting from the variations in the DC voltage applied across the said one load circuit.

The above, and other objects, features and advantages of the present invention, will be apparent from the following description taken in conjunction with the accompanying drawing.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a circuit diagram showing an example of a power supply circuit according to the prior art; and

FIG. 2 is a circuit diagram showing one embodiment of a power supply circuit according to this invention.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

Referring initially to FIG. 1, it will be seen that a conventional power supply circuit without a power transformer for use in a television receiver may include commercial electric power supply lines or an AC voltage source 1 of, for example, 100 volts. The AC voltage source 1 is shown to be connected, at one side, with the anode of a diode 2 having its cathode connected to one end of a primary winding 3a of a flyback transformer 3. A connection point between diode 2 and primary winding 3a of the flyback transformer 3 is grounded through a smoothing capacitor 4, while the other end of primary winding 3a is grounded through a horizontal deflection circuit 5. The other side of AC voltage source 1 is grounded through a vertical deflection circuit 6, and a connection point between that other side of AC voltage source 1 and vertical deflection circuit 6 is grounded through a smoothing capacitor 7. Further, one end of a secondary winding 3b of flyback transformer 3 is connected through a high voltage rectifier circuit 8 to a high voltage output terminal 9, and the other end of secondary winding 3b is grounded. A high voltage output derived from high voltage output terminal 9 may be supplied to the anode of a television picture tube (not shown).

With the circuit arrangement as described above, an AC voltage at source 1 is rectified by diode 2 to electrically charge capacitors 4 and 7, respectively. The voltages to which capacitors 4 and 7 are charged are in inverse proportion to the capacities of these capacitors. However, since load circuits, that is, horizontal deflection circuit 5 and vertical deflection circuit 6, are connected in parallel with capacitors 4 and 7, respectively, the voltages E1 and E2 applied across capacitors 4 and 7 in their stationary state are respectively determined by the impedances of the parallel loads, that is, the impedances of horizontal and vertical deflection circuits 5 and 6. In other words, the AC voltage of 100 volts at source 1 is rectified by diode 2, and if it is assumed that the impedances of horizontal and vertical deflection circuits 5 and 6 are equal to each other, these horizontal and vertical deflection circuits 5 and 6 will have equal DC voltages applied thereto, for example, of 50 volts each, respectively.

Thus, the previously proposed circuit is capable of supplying low DC voltages from a commercial electric power source to respective load circuits without requiring the use of a heavy and costly power transformer. In such conventional power supply circuit, however, if a current flowing through one load circuit, for example, a current flowing through the secondary winding 3b of flyback transformer 3, that is, a beam current flowing through a picture tube, varies in accordance with the brightness of the image on the screen of the picture tube, the voltages applied to the horizontal deflection circuit 5 and the vertical deflection circuit 6 will be varied. More specifically, if the beam current increases according to the brightness of the image on the screen of the picture tube, the current flowing through primary winding 3a increases and hence the current flowing through the horizontal deflection circuit 5 also increases. As a result of the current increase in load circuit 5, the impedance of horizontal deflection circuit 5, as viewed from voltage source 1, is apparently lowered, with the result that the DC voltage applied across horizontal deflection circuit 5 is lowered. Since the DC output voltage of diode 2 is constant, and if the impedance of vertical deflection circuit 6 is assumed to be constant, the DC voltage applied across vertical deflection circuit 6 will be increased with the result that the vertical deflection angle of the beam is increased to lengthen the produced picture or image in the vertical direction.

In accordance with the present invention, a power supply circuit of the type described above is provided with additional means by which DC voltages applied to loads are prevented from being veried even though a load impedance varies. Thus, when a power supply circuit according to this invention is applied, for example, to a television receiver, variations in the beam current flowing through the television picture tube in response to changes in the brightness of the image on the screen of the picture tube will not substantially change the respective voltages applied across the horizontal deflection circuit and the vertical deflection circuit of the television receiver.

Referring now to FIG. 2, it will be seen that, in one embodiment of a power supply circuit arrangement according to this invention which is shown applied to a television receiver capable of employing an AC-DC voltage source, a commercial electric power source 11, for example, an AC voltage source of 100 volts, has one side connected through a rectifier circuit 12 and a resistor 20 to a movable contact of a voltage source switch SW1. The fixed contact of switch SW is connected through a ripple filter 21 and a choke coil 22 to a DC voltage terminal t1 of a first primary winding 13a of a flyback transformer 13, while a terminal t2 of the first primary winding 13a is grounded through a horizontal deflection circuit 15. A high voltage output terminal t4 of a second primary winding 13b of flyback transformer 13 is connected through a high voltage rectifier circuit 18 to an anode of a picture tube 19. Further, the other side of AC voltage source 11 is connected through a smoothing capacitor 23 to the fixed contact of voltage source switch SW1 so that an AC voltage of 100 volts is rectified and smoothed by rectifier circuit 12 and capacitor 23. A connection point between choke coil 22 and first primary winding 13a of the flyback transformer 13 is grounded through a smoothing capacitor 24.

The illustrated television receiver is further shown to include a tuner 25 and a video intermediate frequency amplifier circuit 26. One of the voltage source terminals of tuner 25 and of video intermediate frequency amplifier circuit 26, respectively, are connected to a movable contact of a connecting switch SW2 and the other voltage source terminals of the tuner 25 and of the circuit 26, respectively, are connected to a common or ground terminal. Further, a connection point between the first-mentioned voltage source terminals of tuner 25 and circuit 26 and the movable contact of connecting switch SW2 is connected, through smoothing capacitor 14, to the ground terminal. A video detector circuit 27, a sound intermediate frequency amplifier circuit 28, a synchronous separator circuit 29, an AFC(automatic frequency control) circuit 30, a horizontal synchronizing signal oscillator circuit 31 and a horizontal period drive circuit 32 are connected, as a voltage source system, in parallel with capacitor 14. One of the voltage source terminals of each of the respective circuits 27, 28, 29, 30, 31 and 32 is connected through a boosting diode 33 to an intermediate terminal t3 of the first primary winding 13a of flyback transformer 13. Further, a reverse-current preventing diode 34 is connected between one of the voltage source terminals of the synchronous separator circuit 29 and of the AFC circuit 30, respectively, and a starting diode 35 is connected between the connection point of the voltage source terminals of circuit 30 and 31 and the ground.

A connection point A between the other side of AC voltage source 11 and capacitor 23 is connected through a smoothing capacitor 36 to the ground terminal and is also connected through a resistor 37 and a parallel circuit, consisting of a sound output circuit 38 and a smoothing capacitor 39, to the ground terminal. Thus, an output signal obtained from sound output circuit 38 is adapted to be applied to a speaker 40. Further, the connection point A is connected to the ground terminal through a resistor 41 and a parallel circuit, consisting of a smoothing capacitor 17 and a vertical deflection circuit 16, and connection point A is also connected through a resistor 42 to one of the voltage source terminals of a video output circuit 43, while a smoothing capacitor 44 is connected between the voltage source terminals of video output circuit 43. The voltage source terminal of video output circuit 43 which is connected to resistor 42 is also connected to the first grid G1 of picture tube 19, and the other voltage source terminal of circuit 43 is connected through a resistor 45 to the second and fourth grids G2 and G4 of picture tube 19. An output terminal of video output circuit 43 is connected to the cathode of picture tube 19, and a conductive layer coated on the exterior wall surface of picture tube 19 is connected to connection point A. A connection point between the other voltage source terminal of video output circuit 43 and resistor 45 is connected to a connection point between ripple filter 21 and coil 22, while the connection point between coil 22 and ripple filter 21 is also connected through a capacitor 46 and diode 33 to intermediate terminal t3 of first primary winding 13a of the flyback transformer 13. The capacitor 46 is functions to supply a kick voltage to the circuits, such as, horizontal period drive circuit 32 and the like, and to form a boost circuit together with diode 33, in its stationary state.

A connection point between capacitor 14 and one of the voltage source terminals of video intermediate frequency amplifier circuit 26 is connected to a movable contact of a connecting switch SW3, while the fixed contact of switch SW3 is connected through a storage battery 47 to the ground terminal. The movable contacts of switches SW1 and SW3 are ganged or linked to each other. Further, the fixed contact of connecting switch SW2 is connected through a diode 48 to a terminal t5 of a secondary winding 13c of flyback transformer 13, at which terminal t5 there is produced a negative pulse which is rectified by diode 48 to obtain a low DC voltage.

In accordance with this invention, an intermediate terminal t6 of secondary winding 13c is grounded, and a terminal t7 of winding 13c is connected through a diode 49 to connection point A. A positive pulse generated at this terminal t7 is rectified by diode 49 to obtain a predetermined low DC voltage.

With the above described circuit, when ganged switches SW1 and SW3 are closed and switch SW2 is also closed, that is, when an AC voltage of 100 volts is supplied from AC voltage source 11, such AC voltage of 100 volts is rectified and smoothed by rectifier circuit 12 and capacitor 23, and hence the capacitor 23 has a DC voltage applied thereacross, for example, of +110 volts. This DC voltage of +110 volts is applied through ripple filter 21 to series-connected capacitors 24 and 36 and is divided thereby, with the voltages across capacitors 24 and 36, respectively, being applied to horizontal deflection circuit 15 and vertical deflection circuit 16, respectively. In this case, since a connection point between capacitors 24 and 36, and hence a connection point between circuits 15 and 16, is grounded, capacitors 24 and 36 have stored therein voltages which are of reversed polarity to each other. Thus, for example, horizontal deflection circuit 15 may have applied thereto a DC voltage of +70 volts, while vertical deflection circuit 16 may have applied thereto a DC voltage of -30 volts which is obtained from -40 volts by the voltage drop across resistor 41. Moreover, the negative pulse obtained at terminal t5 of secondary winding 13c of the flyback transformer 13 is rectified by diode 48 and smoothed by capacitor 14 to produce a low DC voltage of, for example, +12 volts. This voltage of +12 volts is applied to those circuits, such as, horizontal oscillator circuit 31, horizontal drive circuit 32 and the like, which are connected in parallel with capacitor 14 and which require a positive low DC voltage. Further, sound output circuit 38 has applied thereto a negative DC voltage of -30 volts which is obtained, by reason of the voltage drop across resistor 37, from the voltage of -40 volts across capacitor 36. The video output circuit 43 has applied thereto a voltage of, for example, 100 volts, which is lower than the voltage across capacitor 23 by reason of the voltage drop across resistor 42. Finally, the positive pulse obtained at terminal t7 of the secondary winding 13c of flyback transformer 13 is rectified by diode 49 and applied to sound output circuit 38 and vertical deflection circuit 16 together with the voltage across capacitor 36. Accordingly, the total voltage applied to vertical deflection circuit 16, in this case, is the sum of the voltage which is obtained by dividing the voltage across capacitor 23 in proportion to the respective impedances of horizontal and vertical deflection circuits 15 and 16, and of the voltage which is obtained as a result of the rectifying by diode 49 of the positive pulse derived from terminal t 7 of the secondary winding 13c of flyback transformer 13.

When switch SW2 is opened, the voltage of +12 volts from storage battery 47 is supplied through switch SW3 to those circuits, such as, horizontal oscillator circuit 31, horizontal drive circuit 32, and the like, which require a low positive DC voltage. In this case, a boost circuit is formed by the winding between terminals t1 and t3 of the first primary winding 13a of flyback transformer 13, capacitor 46 and diode 33. Thus, a voltage of 58 volts (70 - 12 = 58 volts) is produced across capacitor 46, and a voltage of +70 volts is obtained across capacitor 24.

In this case, a video signal received by an antenna 50 is supplied as indicated by arrows in FIG. 2. In other words, the video signal received by antenna 50 is supplied through tuner 25 and video intermediate frequency amplifier circuit 26 to video detector circuit 27. An audio signal derived from the video detector circuit 27 is supplied through sound intermediate frequency amplifier circuit 28 and sound output circuit 38 to speaker 40 for being reproduced thereby. A video signal derived from video detector circuit 27 is supplied to synchronous separator circuit 29 in which a horizontal period synchronizing signal and a vertical period synchronizing signal are separately obtained. The horizontal period synchronizing signal is supplied to AFC circuit 30 to control the oscillation frequency of horizontal signal oscillator circuit 31 and the output signal from the latter is supplied through horizontal drive circuit 32 to horizontal deflection circuit 15. The vertical period synchronizing signal is supplied from separator circuit 29 to the vertical deflection circuit 16. Further, the video signal derived from video detector circuit 27 is supplied to video output circuit 43 to derive therefrom an output signal which is supplied to the cathode of picture tube 19.

With the circuit according to this invention as described above, when the brightness of an image on the screen of picture tube 19 is increased, and hence the beam current of the picture tube 19 is increased, a current flowing through horizontal deflection circuit 15 is also increased and its impedance, as viewed from the voltage source side (hereinafter referred to as merely an impedance), is lowered, so that the voltage applied thereacross tends to be decreased. At such time, if the impedance of vertical deflection circuit 16 is assumed constant, the voltage applied across vertical deflection circuit 16 would be increased because the voltage across the capacitor 23 is constant. In the circuit according to the invention, however, the pulse voltage generated in flyback transformer 13 is reduced due to the drop in the impedance of horizontal deflection circuit 15, thereby to decrease the voltage which is obtained as a result of the rectifying, by diode 49, of the pulse from secondary winding 13c, and which is applied to vertical deflection circuit 16. Hence, the total voltage across vertical deflection circuit 16 is restrained from increasing, and the current flowing through vertical deflection circuit 16 as a result of the voltage applied thereto from diode 49 is decreased. Consequently, the current flowing through horizontal deflection circuit 15 is decreased from its increased value and hence its impedance is apparently increased or returned toward its original value. In other words, the ratio of the impedances of horizontal deflection circuit 15 and of vertical deflection circuit 16 is maintained substantially constant, and hence the DC operating voltages applied to these circuits 15 and 16 become stable.

Conversely, when the impedance of vertical deflection circuit 16 is varied, the respective DC voltages can be similarly stabilized. For example, if the current flowing through vertical deflection circuit 16 is increased for some reason to lower its apparent impedance with the result that the voltage applied across horizontal deflection circuit 15 will tend to be increased, the output voltage of diode 49 applied to vertical deflection circuit 16 is increased. In other words, the voltage applied from secondary winding 13c through diode 49 to the vertical deflection circuit 16 is increased to further increase the current flowing through circuit 16, and hence the current flowing through horizontal deflection circuit 15 is also increased, so that the apparent impedance of horizontal deflection circuit 15 is also lowered. Thus, the impedance ratio of horizontal deflection circuit 15 and vertical deflection circuit 16 is not varied with the result that the voltages applied across these circuits 15 and 16 become constant.

From the above, it will be apparent that in the circuit according to this invention, the pulse obtained at the output end of secondary winding 13c of the flyback transformer 13 is rectified by the diode 49 with the resulting voltage being applied to the vertical deflection circuit 16 so that the DC operating voltages respectively applied to horizontal deflection circuit 15 and vertical deflection circuit 16 are maintained constant.

Although an illustrative embodiment of the invention has been described in detail herein with reference to the accompanying drawings, it is to be understood that the invention is not limited to that precise embodiment, and that various changes and modifications may be effected therein by one skilled in the art without departing from the scope or spirit of the invention.

 

 

 

SONY KV-1842E  CHASSIS  SCC-191A-A AMBIENT LIGHT RESPONSIVE CONTROL OF BRIGHTNESS, CONTRAST AND COLOR SATURATION Gain control arrangement useful in a television signal processing system

In a color television receiver, first and second amplifiers are respectively included in the luminance and chrominance channels to permit control of contrast and saturation. The amplifiers have gain versus control voltage characteristics including linear portions extrapolated to cut off at predetermined voltages which may or may not be the same. A first potentiometer is coupled between a source of fixed voltage equal to the extrapolated cut off voltage of the first amplifier and a gain controlling voltage source. The gain controlling voltage may be produced by a circuit including an element responsive to ambient light. The wiper of the first potentiometer is coupled to the first amplifier to couple a voltage developed at a predetermined point of the first potentiometer to the first amplifier to control its gain. A second potentiometer is coupled between a source of voltage equal to the extrapolated cut off voltage of the second amplifier and the gain controlling voltage source to receive a portion of the gain controlling voltage in accordance with the ratio of the extrapolated cut off voltages of the first and second amplifiers. The wiper of the second potentiometer is coupled to the second amplifier to couple a voltage developed at a predetermined point of the second potentiometer to the second amplifier to control its gain. In this manner, the contrast of the receiver may be varied over a relatively wide range while saturation is maintained substantially constant.


1. In a color television signal processing system of the type including luminance and chrominance signal processing channels, apparatus comprising:
first and second amplifiers respectively included in said luminance and chrominance channels, said amplifiers having gain versus control voltage characteristics including linear portions extrapolated to cut-off at predetermined voltages which may or may not be the same voltage;
a gain controlling voltage source;
means for coupling said gain controlling voltage to said first amplifier to control its gain;
potentiometer means coupled between a fixed voltage substantially equal to the extrapolated cut-off voltage of said second amplifier and to said gain controlling voltage source to recieve a portion of said gain controlling voltage in accordance with the ratio of the extrapolated cut-off voltages of said first and second amplifiers; and
means for coupling a voltage developed at a predetermined point on said potentiometer means to said second amplifier to control its gain.
2. The apparatus recited in claim 1 wherein said means for coupling said gain controlling voltage to said first amplifier includes another potentiometer coupled between a source of fixed voltage substantially equal to the extrapolated cut-off voltage of said first amplifier and said gain controlling voltage source. 3. In a color television signal processing system of the type including luminance and chrominance signal processing channels, apparatus comprising:
first and second amplifiers respectively included in said luminance and chrominance channels, said amplifiers having gain control voltage characteristics including linear portions extrapolated to cut-off at substantially the same predetermined voltage;
a source of gain controlling voltage; and
means for coupling said gain controlling voltage to said first and second amplifiers.
4. Apparatus comprising:
first variable gain amplifying means for amplifying a first signal in response to a first DC control signal, said first amplifying means having a first gain versus DC control voltage characteristic including a linear region, said linear region having a gain substantially equal to 0 at a DC control voltage equal to V_O ;
second variable gain amplifying means for amplifying a second signal in response to a second DC control signal, said second amplifying means having a second gain versus DC control voltage characteristic including a linear region, said linear region having a gain substantially equal to 0 at a DC control voltage equal to AV_O, where A is a number greater than 0;
a first source of fixed voltage substantially equal to V_O ;
a second source of fixed voltage substantially equal to AV_O ;
means for developing a third DC control voltage v;
means for developing a portion Av of said third control voltage v;
first means for deriving said first control voltage including means for providing the difference between said third control voltage v and said fixed voltage V_O and means for adding a predetermined portion of the difference between said third control voltage v and said fixed voltage V_O to said DC control voltage v; and
second means for deriving said second control voltage including means for providing the difference between a portion Av of said third control voltage v and said fixed voltage AV_O and means for adding a predetermined portion of the difference between said portion Av and said fixed voltage AV_O to said DC control voltage v.
5. The apparatus recited in claim 4 wherein A is equal to 1. 6. The apparatus recited in claim 4 wherein said first amplifying means is included in a luminance channel of a televeision signal processing system and said second amplifying means is included in a chrominance channel of said television signal processing system. 7. The apparatus recited in claim 6 wherein means for developing said third control voltage includes means responsive to ambient light. 8. The apparatus recited in claim 4 wherein said first means includes first voltage divider means coupled between said fixed voltage V_O and said third DC control voltage v; and wherein said second means includes second voltage divider means coupled between said fixed voltage AV_O and said portion Av. 9. The apparatus recited in claim 8 wherein said first voltage divider means includes a first potentiometer, said first potentiometer having a wiper coupled to said first amplifying means; and wherein said second voltage divider means includes a second potentiometer, said second potentiometer having a wiper coupled to said amplifying means. 10. The apparatus recited in claim 4 wherein said second gain versus DC control voltage characteristic includes a region between said voltage AV_O and a voltage V_B where the gain is greater than 0, said voltage V_B being substantially equal to the voltage at which said second amplifying means has a gain substantially equal to 0; and wherein said second source of fixed voltage includes means for coupling said voltage V_B to said second amplifying means. 11. The apparatus recited in claim 10 wherein said second source of said voltage AV_O includes a third source of fixed voltage V_B ; potentiometer means coupled between said third source of fixed voltage V_B and said means for developing said third DC control voltage; and means coupled to said potentiometer means for developing said voltage AV_O at a point along said potentiometer means; said potentiometer means including a wiper coupled to said second amplifier means, said wiper being adjustable to couple a DC voltage V_FB and said third control voltage to said second amplifying means.

Description:

The present invention pertains to gain controlling apparatus and particularly to apparatus for controlling the gains of amplifiers included in the luminance and chrominance channels of a television signal processing system.
Recently, the maximum brightness available from television receivers has increased sufficiently so that a pleasing image may be reproduced under conditions of high ambient light as well as under conditions of low ambient light. Apparatus is known for automatically controlling the contrast and brightness properties of a television receiver in response to ambient light to provide a pleasing image over a range of ambient light conditions. Such apparatus is described in U.S. Pat. Nos. 3,027,421, entitled "Circuit Arrangements For Automatically Adjusting The Brightness And The Contrast In A Television Receiver," issued to H. Heijligers on Mar. 27, 1962 and 3,025,345, entitled "Circuit Arrangement For Automatic Readjustment Of The Background Brightness And The Contrast In A Television Receiver," issued to R. Suhrmann on Mar. 13, 1962.
Apparatus is also known for automatically controlling the contrast and saturation properties of a color television receiver by controlling the gains of luminance and chrominance channel amplifiers, respectively, in response to ambient light. Such apparatus is described in U.S. Pat. Nos. 3,813,686 entitled "Ambient Light Responsive Control Of Brightness, Contrast And Color Saturation," issued to Eugene Peter Mierzwinski, on May 28, 1974 and 3,814,852 entitled "Ambient Light Responsive Control Of Brightness, Contrast and Color Saturation," issued to Eugene P. Mierzwinski on June 4, 1974.
Also of interest is apparatus for manually controlling the gains of luminance and chrominance channel amplifiers. Such apparatus is described in U.S. Pat. Nos. 3,374,310, entitled "Color Television Receiver with Simultaneous Brightness and Color Saturation Controls," issued to G.L. Beers on Mar. 19, 1968; 3,467,770, entitled "Dual Channel Automatic Control Circuit," issued to DuMonte O. Voigt on June 7, 1966; and 3,715,463, entitled "Tracking Control Circuits Using a Common Potentiometer," issued to Lester Tucker Matzek, on Feb. 6, 1973.
When the gain of luminance channel is adjusted to control the contrast of an image, either manually or automatically, in response to ambient light, it is desirable to simultaneously control the gain of the chrominance channel in such a manner that the ratio of the gains of the luminance and chrominance channels is substantially constant over a wide range of contrast control to maintain constant saturation. If the proper ratio between the amplitudes of the chrominance and luminance signals is not maintained incorrect color reproduction may result. For instance, if the amplitude of the luminance signals are increased without correspondingly increasing the amplitude of the chrominance signals, colors may become desaturated, i.e., they will appear washed out or pastel in shade. Furthermore, it may be desirable to provide controls for presetting the gains of the luminance and chrominance channels to compensate for tolerance variations in other portions of the television signal processing apparatus.
In accordance with the present invention, apparatus is provided which may be utilized in a color television receiver to control contrast over a relatively wide range while maintaining constant saturation. The apparatus includes first and second amplifiers having gain versus control voltage characteristics including linear portions extrapolated to cut off at predetermined voltages which may or may not be the same. Means couple a gain controlling voltage source to the first amplifier to control its gain. Potentiometer means are coupled between a source of fixed voltage substantially equal to the extrapolated cut off voltage of the second amplifier and the source of gain controlling voltage to receive a portion of said gain controlling voltage in accordance with the ratio of the extrapolated cut off voltages of the amplifiers. A voltage developed at a predetermined point along the potentiometer means is coupled to the second amplifier to control its gain.
In accordance with another feature of the present invention, the means for coupling said gain controlling voltage to said first amplifier includes another potentiometer coupled between a source of fixed voltage substantially equal to the extrapolated cut off voltage of said first amplifier and said gain controlling voltage source.
In accordance with still another feature of the present invention the gain controlling voltage source includes an element responsive to ambient light .
These and other aspects of the present invention may best be understood by references to the following detailed description and accompanying drawing in which:
FIG. 1 shows the general arrangement, partly in block diagram form and partly in schematic diagram form, of a color television receiver employing an embodiment of the present invention;
FIG. 1A shows, in schematic form, a modification to the embodiment shown in FIG. 1;
FIG. 2 shows graphical representation of gain versus control voltage characteristics of amplifiers utilized in the embodiment shown in FIG. 1;
FIG. 3 shows graphical representations of gain versus control voltage characteristics of amplifiers which may be utilized in the receiver shown in FIG. 1;
FIG. 4 shows, in schematic form, another embodiment of the present invention which may be utilized to control the amplifiers whose gain versus control voltage characteristics are shown in FIG. 3;
FIG. 5 shows, in schematic form, an amplifier which may be utilized in the receiver shown in FIG. 1; and
FIG. 6 shows, in schematic form, another amplifier which may be utilized in the receiver shown in FIG. 1.
Referring now to FIG. 1, the general arrangement of a color television receiver employing the present invention includes a video signal processing unit 112 responsive to radio frequency (RF) television signals for generating, by means of suitable intermediate frequency (IF) circuits (not shown) and detection circuits (not shown), a composite video signal comprising chrominance, luminance, sound and synchronizing signals. The output of signal processing unit 112 is coupled to chrominance channel 114, luminance channel 116, a channel 118 for processing the synchronizing signals and a channel (not shown) for processing sound signals.

Chrominance processing channel 114 includes chrominance processing unit 120 which serves to remove chrominance signals from the composite video signal and otherwise process chrominance signals. Chrominance signal processing unit 120 may include, for example, automatic color control (ACC) circuits for adjusting the amplitude of the chrominance channels in response to amplitude variations of a reference signals, such as a color burst signal, included in the commposite video signal. Chrominance signal processing circuits of the type described in the U.S. Pat. No. 3,740,462, entitled "Automatic Chroma Gain Control System," issued to L.A. Harwood, on June 19, 1973 and assigned to the same assignee as the present invention are suitable for use as chrominance processing unit 120.
The output of the chrominance signal processing unit 120 is coupled to chrominance amplifier 122 which serves to amplify chrominance signals in response to a DC signal v_C generated by gain control network 142. As illustrated, chrominance amplifier 122 provides chrominance signals to a chroma demodulator 124. An amplifier suitable for use as chrominance amplifier 122 will subsequently be described with reference to FIG. 6.
Chroma demodulator 124 derives color difference signals representing, for example, R-Y, B-Y and G-Y information from the chrominance signals. Demodulator circuits of the general type illustrated by the chrominance amplifier CA 3067 integrated circuit manufactured by RCA Corporation are suitable for use as chrominance demodulator 124.
The color difference signals are applied to a video driver 126 where they are combined with the output signals -Y of luminance channel 116 to produce color signals of the appropriate polarity, representing for example, red (R), green (G) and blue (B) information. The color signals are coupled to kinescope 128.
Luminance channel 116 includes a first luminance signal processing unit 129 which relatively attenuates undesirable signals, such as chrominance or sound signals or both, present in luminance channel 116 and otherwise processes the luminance signals. The output of first luminance processing unit 129 is coupled to luminance amplifier 130 which serves to amplify the luminance signals in response to a DC control signal v_L generated by gain control unit 142 to thereby determine the contrast of a reproduced image. An amplifier suitable for use as luminance amplifier 130 will subsequently be described with reference to FIG. 5. The output of luminance amplifier 130 is coupled to second luminance signal processing unit 132 which serves to further process luminance signals. A brightness control unit 131 is coupled to luminance signal processing unit 132 to control the DC content of the luminance signals. The output -Y of luminance processing unit 132 is coupled to kinescope driver 126.
Channel 118 includes a sync separator 134 which separates horizontal and vertical synchronizing pulses from the composite video signal. The synchronizing pulses are coupled to horizontal deflection circuit 136 and vertical deflection circuit 138. Horizontal deflection circuit 136 and vertical deflection circuit 138 are coupled to kinescope 128 and to a high voltage unit 140 to control the generation and deflection of one or more electron beams generated by kinescope 128 in the conventional manner. Deflection circuits 136 and 138 also generate horizontal and vertical blanking signals which are coupled to luminance signal processing unit 132 to inhibit its operation during the horizontal and vertical retrace intervals.
Gain control unit 142 is coupled to luminance amplifier 130 and to chrominance amplifier 122 to control their gains. Gain control unit 142 includes a PNP transistor 152 arranged as an emitter-follower amplifier. The collector of transistor 152 is coupled to ground while its emitter is coupled through a series connection of a potentiometer 156 and fixed resistor 154 to a source of positive supply voltage V_O. The wiper of potentiometer 156 is coupled to luminance amplifier 130. The series connection of a potentiometer 158 and a variable resistor 159 is coupled between the source of positive supply voltage V_O and the emitter of transistor 152. The wiper of potentiometer 158 is coupled to chrominance amplifier 122.
The base of transistor 152 is coupled to the wiper of a potentiometer 146. One end of potentiometer 146 is coupled to the source of positive supply voltage V_O through a fixed resistor 144. The other end of potentionmeter 146 is coupled to ground through a light dependent resistor (LDR) 148. LDR 148 is a resistance element whose impedance varies in inverse relationship with light which impinges on it. LDR 148 may comprise a simple cadmium sulfide type of light dependent element or other suitable light dependent device. LDR 148 is desirably mounted to receive ambient light in the vicinity of the screen of kinescope 128.
A single pole double-throw switch 150 has a pole coupled to the junction of potentiometer 146 and LDR 148. A resistor 151 is coupled between the wiper of potentiometer 146 and the other pole of switch 150. The arm of switch 150 is coupled to ground.
The general arrangement shown in FIG. 1 is suitable for use in a color television receiver of the type shown, for example, in RCA Color Television Service Data 1973 No. C -8 for a CTC-68 type receiver, published by RCA Corporation, Indianapolis, Indiana.
In operation, gain control circuit 142 maintains the ratio of the gain of chrominance amplifier 122 to the gain of amplifier 130 constant in order to maintain constant saturation while providing for contrast adjustment either manually by means of potentiometer 146 or automatically by means of LDR 148. If the gain of luminance were adjusted to control the contrast of an image without a corresponding change in the gain of chrominance amplifier 122, the amplitudes of luminance signals -Y and color difference signals R-Y, B-Y and G-Y would not, in general, be in the correct ratio when combined by divider 126 to provide the desired color.
When switch 140 is in the MANUAL position, the gains of chrominance amplifier 122 and luminance amplifier 130 are controlled by adjustment of the position of potentiometer 146. When switch 150 is in the AUTO position the gain of the chrominance amplifier 122 and luminance amplifier is automatically controlled by the response of LDR 148 to ambient light conditions. The voltage developed at the wiper of potentiometer 146 (base of transistor 152) when switch 150 is in the AUTO position is inversely related to the ambient light recieved by LDR 148. It is noted that the values of resistors 114, potentiometer 146, LDR 148 and resistor 151 are desirably selected such that the adjustment of the wiper arm of potentiometer 146 when switch 150 is in the MANUAL position does not substantially affect the voltage developed at the base of transister 152 when switch 150 is placed in the AUTO position.
The control voltage v developed at the wiper arm of potentiometer 146 is coupled through emitter-follower transistor 152 to the common junction of potentiometer 156 and variable resistor 159. A control voltage v_L comprising v plus a predetermined portion of the difference V_O -v developed across the series connection of fixed resistor 154 and potentiometer 156, depending on the setting of potentiometer 156, is coupled to luminance amplifier 130 to control its gain. Similarly, a control voltage v_C comprising v plus a predetermined portion of the difference voltage V_O -v developed across the series connection of potentiometer resistor 158 and variable resistor 159, depending on the setting of the wiper of potentiometer 158, is coupled to chrominance amplifier 122 to control its gain.
The gain of luminance amplifier 130 may be pre-set to a desired value by the factory adjustment of potentiometer 156. Similarly, variable resistor 159 is provided to allow factory pre-set of the gain of the chrominance amplifier 122. Potentiometer 158 is provided to allow customer control of saturation.
Referring to FIG. 2, the gain versus voltage characteristics of chroma amplifier 122 (g_C) and luminance amplifier 130 (g_L) are shown. The characteristic g_C has a reversed S-shape including a linear portion 214. Extrapolated linear portion 214 of g_C intersects the GAIN axis at G_C and intersects the CONTROL VOLTAGE axis at V_O. Similarly, the characteristics g_L has a reverse S-shape characteristic including a linear portion 212. Extrapolated linear portion 214 of g_L intersects the GAIN axis at G_L and intersects the CONTROL VOLTAGE axis at V_O.
From FIG. 2, the expression for linear portion 212 of g_L is ##EQU1## The expression for linear portion 214 of g_C is ##EQU2## From FIG. 1, the expression for v_L is v_L = v + (V_O -v) K₁ [3]
where K₁ is determined by the voltage division of fixed resistor 154 and potentiometer 156 at the wiper of potentiometer 156. When the wiper of potentiometer 156 is at the emitter of transistor 152, K₁ =0. The expression for v_C is v_C = v + (V_O -v)K₂ [4]
where K₂ is determined by the voltage division of potentiometer 158 and fixed resistor 159 at the wiper of potentiometer 158. By combining equations [1] and [3], the equation for g_L becomes ##EQU3## By combining equations [2] and [4], the equation for g_C becomes ##EQU4## The ratio of g_L to g_C is thus ##EQU5## It is noted that this ratio is independent of DC control voltage v. Thus, although DC control voltage v may be varied either manually or in response to ambient light to control the contrast of an image reproduced by kinescope 128, the saturation remains constant.
With reference to FIG. 2, it is noted that although the linear portion 214 of g_C has an extrapolated gain equal to 0 at a control voltage equal to V_O, the non-linear portion of g_C does not attain a gain equal to 0 until a control voltage equal to V_B. That is, a control voltage of V_O will not cut-off chrominance amplifier 122.
In FIG. 1A there is shown, in schematic form, a modification to the arrangement of gain control network 142 of FIG. 1 with provisions which allow a viewer to cut off chrominance amplifier 122 to produce a more pleasing image under conditions of poor color reception due, for example, to noise or interference. The modifications to gain control unit 142 shown in FIG. 1A include coupling potentiometer resistor 158 between a source of positive supply voltage V_B, the value of V_B being greater than the value of V_O, and coupling a resistor 160 from a tap-off point 162 along potentiometer 158 to ground. The value of potentiometer 158 and resistor 160 and the location of tap 162 are selected so that voltage V_O is developed at tap 162.
The arrangement shown in FIG. 1A allows for the adjustment of contrast while constant saturation is maintained and additionally allows a viewer, by adjusting the wiper of potentiometer 158 to voltage V_B, to cut off chrominance amplifier 122.
Referring to FIG. 3 there are shown gain versus DC control voltage characteristics of chrominance and luminance amplifiers which do not have the same extrapolated linear cut off control voltage. The gain versus control voltage characteristic g_L ' of the luminance amplifier has a reverse S-shape characteristic including a linear portion 312. Extrapolated linear portion 312 of g_L ' intersects the GAIN axis at a gain G_L ' and intersects the CONTROL VOLTAGE axis at a voltage V_O '. The gain versus control voltage characteristic g_C ' of the chrominance amplifier has a reverse S-shape characteristic having a linear portion 314. Extrapolated linear portion 314 of g_C ' intersects the GAIN axis at a gain G_C ' and intersects the CONTROL VOLTAGE axis at a voltage AV_O ', where A is a number greater than zero.
From FIG. 3, the expression for linear portion 312 of g_L ' is ##EQU6## where v_L ' is the DC conrol voltage coupled to the luminance amplifier. The expression for linear portion 314 of g_C ' is ##EQU7## where v_C ' is the DC control voltage coupled to the chrominance amplifier.
A modified form of the control network 142 of FIG. 1 suitable for controlling the gain of a chrominance and a luminance amplifier having characteristics such as shown in FIG. 3 is shown in FIG. 4. Similar portions of FIGS. 1 and 4 are identified by reference numbers having the same last two significant digits and primed (') designations. The modified portions of FIG. 1 shown in FIG. 4 include the series connection resistors 460 and 462 coupled between the emitter of transistor 452 to ground. The values of resistors 460 and 462 are selected so that a portion Av' of the DC control voltage v' developed at the emitter of transistor 452 is developed at the junction of resistors 460 and 462. Furthermore, the series connection of potentiometer 458 and variable resistor 459 is coupled between the junction of resistor 460 and 462 and a source of positive supply voltage AV_O '.
From FIG. 4, the expression for control voltage v_L ' developed at the wiper of potentiometer 456 is v_L ' = v' + (v_O '-v')K₁ ' [10]
where K₁ ' is determined by the voltage division at the wiper of potentiometer 456. The expression for control voltage v_C ' developed at the wiper of potentiometer 458 is V_C ' = Av' + (AV_O ' - Av')K ₂ ' [11]
where K₂ ' is determined by the voltage division at the wiper of potentiometer 458. By combining equations [8] and [10], ##EQU8## By combining equations [9] and [11], ##EQU9## The ratio of g_L ' to g_C ' is given by the expression ##EQU10## It is noted that this ratio is independent of DC control voltage v'. Therefore, gain control network 442 of FIG. 4 also allows for the adjustment of contrast while maintaining constant saturation.
It is noted that if A were made equal to 1, the arrangement gain control unit 442 would be suitable to control the gains of chrominance and luminance amplifiers having the characteristics shown in FIG. 2.
In FIG. 5, there is shown an amplifier suitable for use as luminance amplifier 130 of FIG. 1. The amplifier includes a differential amplifier comprising NPN transistors 532 and 534. The commonly coupled emitters of transistors 532 and 534 are coupled to the collector of an NPN transistor 528. The emitter of transistor 528 is coupled via a resistor 530 to ground. The collector of transistor 532 and the collector of transistor 534, via load resistor 536, is coupled to a bias voltage provided by bias supply 546, illustrated as a series connection of batteries. The bases of transistors 532 and 534 are respectively coupled to a lower bias voltage through resistors 533 and 535 respectively.
An input signal, such as, for example, the output signal provided by first luminance processing circuit 129 of FIG. 1 is coupled to the base of transistor 532 via terminal 542. The output signal of the amplifier is developed at the collector of transistor 534 and coupled to output terminal 544.
A DC control voltage, such as v_L provided by gain control unit 142 of FIG. 1, is coupled to the base of an NPN transistor 514, arranged as an emitter-follower, via terminal 512. The collector of transistor 514 is coupled to bias supply 546. The emitter of transistor 514 is coupled to ground through the series connection of resistor 516, a diode connected transistor 518 and resistor 520.
The anode of diode 520 is coupled to the base of an NPN transistor 538. The collector of transistor 538 is coupled to the collector of transistor 534 while its emitter is coupled to ground through resistor 540. Transistor 538, resistor 540, diode 518 and resistor 520 are arranged in a current mirror configuration.
The emitter of transistor 514 is coupled to the base of a PNP transistor 522. The emitter of transistor 522 is coupled to bias supply 546 while its collector is coupled to the base of transistor 528 and to ground through the series connection of a diode connected transistor 524 and resistor 526. Transistor 528, resistor 530, diode 524 and resistor 526 are arranged in a current mirror configuration
In operation, the DC control voltage coupled to terminal 512 is coupled in inverted fashion to the anode of diode 524 by transistor 522. As a result, current directly related to the voltage developed at the anode of diode 524 flows through diode 524 and resistor 526. Due to the operation of the current mirror arrangement of diode 524, resistor 526, transistor 528 and resistor 530, a similar current flows through the emitter circuit of transistor 528. The gain of the differential amplifier comprising transistors 532 and 534 is directly related to this current flowing in the emitter circuit of transistor 528, and therefore is inversely related to the DC control voltage at terminal 512. The gain versus DC control voltage characteristics of the differential is similar to g_L shown in FIG. 2.
Further, a current is developed through the series connection of resistor 516, diode 518 and resistor 520 in direct relationship to the DC control coupled to terminal 512. A similar current is developed through resistor 540 due to the operation of the current mirror comprising diode 518, resistor 520, transistor 538 and resistor 540. This current is of the opposite sense to that provided by the current mirror arrangement of diode 524, resistor 526, transistor 528 and resistor 530 and is coupled to the collector of transistor 534 so that the DC voltage at output terminal 544 does not substantially vary with the DC control voltage.
In FIG. 6, there is shown an amplifier suitable for use as chroma amplifier 120 of FIG. 1. The amplifier shown in FIG. 6 is of the type described in U.S. patent application Ser. No. 530,405 entitled "Controllable Gain Signal Amplifier," fled by L.A. Harwood et al. on Dec. 6, 1974.
The amplifier comprises a differential amplifier including NPN transistors 624 and 625 having their bases coupled to terminal 603 via a resistor 626. Chrominance signals, provided by a source of chrominance signals such as chrominance processing unit 120 of FIG. 1, are coupled to terminal 603. The current conduction paths between the collectors and emitters of transistors 624 and 625 are respectively coupled to ground via resistors 628, 629 and 630.
A current splitter circuit comprising an NPN transistor 632 and a diode 634 is coupled to the collector of transistor 624. Diode 634 and the base-emitter junction of transistor 632 are poled in the same direction with respect to the flow of collector current in transistor 624. It desirable that conduction characteristics of transistor 632 and diode 635 be substantially matched. Similarly, the collector of transistor 625 is coupled to a second current splitter comprising a transistor 633 and a diode 635.
An output load circuit comprising series connected resistors 636 and 638 is coupled between the collector of transistor 632 and a source of operating voltage provided by bias supply 610. Amplified chrominance signals are provided at output terminal 640 for coupling, for example, to a chroma demodulator such as chroma demodulator 124 of FIG. 1. Similarly, series connected load resistors 637 and 639 are coupled between the collector of transistor 633 and bias supply 610. An output terminal 641 at the junction of resistors 637 ad 639 provides oppositely phased chrominance signals to those provided at terminal 640. The gain associated with the cascode combination of transistors 624 and 632 is controlled in response to a DC control voltage, such as, for example, v_C provided by gain control unit 142 of FIG. 1, coupled to the base of an NPN transistor 646 via terminal 602. Direct control current is supplied from the emitter of transistor 646 to diode 634 and 635 via a series resistor 652. A signal by-pass circuit comprising a series resonant combination 654 of inductance and capacitance is coupled from the anode of diode 634 to ground. Resonant circuit 654 is tuned, for example, to 3.58 MHz to provide a low impedance path to ground for color subcarrier signals.
Bias voltages and currents are supplied to the amplifier arrangement by bias supply 610, illustrated as a series connection of batterys. A voltage B+ is coupled to the collector of transistor 646. A lower bias voltage is coupled to the load circuits of transistors 632 and 633. The bases of transistors 632 and 633 are coupled in common to a still lower bias voltage. The bases of transistors 624 and 625 are coupled to a still lower bias voltage via substantially equal in value resistors 658 and 659. A resistor 694 is coupled from the common junction of resistors 658 and 659 to ground.
I

n operation, a quiescent operating current is provided through resistor 630. In the absence of an input signal at terminal 603, this current will divide substantially equally between the similarly biased transistors 624 and 625. If the DC control voltage at terminal 602 is near ground potential, transistor 646 will be effectively cut off and no current will flow in resistor 652 and diodes 634 and 635. In that case, neglecting the normally small difference betweeen collector and emitter currents of NPN transistors, the collector currents of transistors 624 and 625 will flow, respectively, in transistors 632 and 633. The transistors 632 and 633 are operated in common base mode and form cascode signal amplifiers with respective transistors 624 and 625. With the DC control voltage near ground potential, one-half of the quiescent current from resistor 630 flows in each of the load circuits and maximum gain for chrominance signals supplied from terminal 603 is provided.
Transistor 646 will conduct when the DC control voltage approaches the bias voltage supplied to the bases of transistors 632 and 633 of the current splitters. By selection of the circuit parameters, diodes 634 and 635 may be arranged to operate in a range between cut off to the conduction of all of the quiescent operating current supplied via resistor 630, thereby cutting off transistors 632 and 633 to provide no output signals at terminals 640 and 641.
At a DC control voltage intermediate to that corresponding to cut off of transistors 632 and 633 on the one hand and cut off of diodes 634 and 635 on the other hand, the voltage gain of the illustrated amplifier will vary in a substantially linear manner with the DC control voltage.
It is noted that although the characteristics shown in FIGS. 2 and 3 were reversed S-shaped characteristics, the characteristics could have other shapes including linear portions. For example, the characteristics could be substantially linear. Furthermore, with reference to FIG. 3, although g_C ' was shown as having a linear portion that had a cut off control voltage lower than the cut off control voltage of the linear portion of g_L ', the cut off control voltage of the linear portion of g_C ' could be greater than the cut off voltage for the linear region of g_L '. In addition, the gain control units and associate amplifiers could be arranged to utilize voltages opposite in polarity to those shown. These and other modifications are intended to be within the scope of the invention.