Richtige Fernseher haben Röhren!

Richtige Fernseher haben Röhren!

In Brief: On this site you will find pictures and information about some of the electronic, electrical and electrotechnical technology relics that the Frank Sharp Private museum has accumulated over the years .

Premise: There are lots of vintage electrical and electronic items that have not survived well or even completely disappeared and forgotten.

Or are not being collected nowadays in proportion to their significance or prevalence in their heyday, this is bad and the main part of the death land. The heavy, ugly sarcophagus; models with few endearing qualities, devices that have some over-riding disadvantage to ownership such as heavy weight,toxicity or inflated value when dismantled, tend to be under-represented by all but the most comprehensive collections and museums. They get relegated to the bottom of the wants list, derided as 'more trouble than they are worth', or just forgotten entirely. As a result, I started to notice gaps in the current representation of the history of electronic and electrical technology to the interested member of the public.


Following this idea around a bit, convinced me that a collection of the peculiar alone could not hope to survive on its own merits, but a museum that gave equal display space to the popular and the unpopular, would bring things to the attention of the average person that he has previously passed by or been shielded from. It's a matter of culture. From this, the Obsolete Technology Tellye Web Museum concept developed and all my other things too. It's an open platform for all electrical Electronic TV technology to have its few, but NOT last, moments of fame in a working, hand-on environment. We'll never own Colossus or Faraday's first transformer, but I can show things that you can't see at the Science Museum, and let you play with things that the Smithsonian can't allow people to touch, because my remit is different.

There was a society once that was the polar opposite of our disposable, junk society. A whole nation was built on the idea of placing quality before quantity in all things. The goal was not “more and newer,” but “better and higher" .This attitude was reflected not only in the manufacturing of material goods, but also in the realms of art and architecture, as well as in the social fabric of everyday life. The goal was for each new cohort of children to stand on a higher level than the preceding cohort: they were to be healthier, stronger, more intelligent, and more vibrant in every way.

The society that prioritized human, social and material quality is a Winner. Truly, it is the high point of all Western civilization. Consequently, its defeat meant the defeat of civilization itself.

Today, the West is headed for the abyss. For the ultimate fate of our disposable society is for that society itself to be disposed of. And this will happen sooner, rather than later.

OLD, but ORIGINAL, Well made, Funny, Not remotely controlled............. and not Made in CHINA.

How to use the site:

- If you landed here via any Search Engine, you will get what you searched for and you can search more using the search this blog feature provided by Google. You can visit more posts scrolling the left blog archive of all posts of the month/year,
or you can click on the main photo-page to start from the main page. Doing so it starts from the most recent post to the older post simple clicking on the Older Post button on the bottom of each page after reading , post after post.

You can even visit all posts, time to time, when reaching the bottom end of each page and click on the Older Post button.

- If you arrived here at the main page via bookmark you can visit all the site scrolling the left blog archive of all posts of the month/year pointing were you want , or more simple You can even visit all blog posts, from newer to older, clicking at the end of each bottom page on the Older Post button.
So you can see all the blog/site content surfing all pages in it.

- The search this blog feature provided by Google is a real search engine. If you're pointing particular things it will search IT for you; or you can place a brand name in the search query at your choice and visit all results page by page. It's useful since the content of the site is very large.

Note that if you don't find what you searched for, try it after a period of time; the site is a never ending job !

Every CRT Television saved let revive knowledge, thoughts, moments of the past life which will never return again.........

Many contemporary "televisions" (more correctly named as displays) would not have this level of staying power, many would ware out or require major services within just five years or less and of course, there is that perennial bug bear of planned obsolescence where components are deliberately designed to fail and, or manufactured with limited edition specificities..... and without considering........picture......sound........quality........

..............The bitterness of poor quality is remembered long after the sweetness of todays funny gadgets low price has faded from memory........ . . . . . .....
Don't forget the past, the end of the world is upon us! Pretty soon it will all turn to dust!

Have big FUN ! !
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©2010, 2011, 2012, 2013, 2014 Frank Sharp - You do not have permission to copy photos and words from this blog, and any content may be never used it for auctions or commercial purposes, however feel free to post anything you see here with a courtesy link back, btw a link to the original post here , is mandatory.
All sets and apparates appearing here are property of
Engineer Frank Sharp. NOTHING HERE IS FOR SALE !

Wednesday, August 1, 2012

ITT DIGIVISION 3486 OSCAR CHASSIS DIGI 3 90° FST INTERNAL VIEW.
























































































SOUND BOARD 450918-9 6611 37 27










ITT DIGIVISION 3486 OSCAR CHASSIS DIGI 3 90° FST / IFB-285/2 INTERNAL VIEW AND Case Study: Digital TV- Digivision ITT DIGIT 2000 VLSI Digital TV System

Case Study: Digital TV - Digivision ITT
I will explain why this system is the father of all digital video/audio modern application field.

Today there is a race to design interoperable video systems for basic digital computer functions, involving multimedia applications in areas such as media information, education, medecine and entertainment, to name but a few. This chapter provides an overview of the current status in industry of digitized television including techniques used and their limitations, technological concerns and design methodologies needed to achieve the goals for highly integrated systems. Digital TV functions can be optimized for encoding and decoding and be implemented in silicon in a more dedicated way using a kind of automated custom design approach allowing enough flexibility.

Significance of VLSI for Digital TV Systems

When, at the 1981 Berlin Radio and TV Exhibition, the ITT Intermetall company exhibited to the public for the first time a digital television VLSI concept [1], [2], opinions among experts were by no means unanimously favourable. Some were enthusiastic, while others doubted the technological and economic feasibility. Today, after 13 years, more than 30 million TV sets worldwide have already been equipped with this system. Today, the intensive use of VLSI chips does not need a particular justification, the main reasons being increased reliability mainly because of the long-term stability of the color reproduction brought about by digital systems, and medium and long-term cost advantages in manufacturing which are essential for ensuring international competitiveness.
Digital signal processing permits solutions that guarantee a high degree of compatibility with future developments, whether in terms of quality improvements or new features like intermediate picture storage or adaptive comb filtering for example. In addition to these benefits, a digital system offers a number of advantages with regard to the production of TV sets:
- Digital circuits are tolerance-free and are not subject to drift or aging phenomena. These well-known properties of digital technology considerably simplify factory tuning of the sets and even permit fully automated, computer-controlled tuning.
- Digital components can be programmable. This means that the level of user convenience and the features offered by the set can be tailored to the manufacturer's individual requirements via the software.
- A digital system is inherently modular with a standard circuit architecture. All the chips in a given system are compatible with each other so that TV models of various specifications, from the low-cost basic model to the multi-standard satellite receiver, can be built with a host of additional quality and performance features.
- Modular construction means that set assembly can be fully automated as well. Together with automatic tuning, the production process can be greatly simplified and accelerated.

Macro-function Processing

The modular design of digital TV systems is reflected in its subdivision into largely independent functional blocks, with the possibility having special data-bus structures. It is useful to divide the structure into a data-oriented flow and control-oriented flow, so that we have four main groups of components:
1.- The control unit and peripherals, based on well-known microprocessor structures, with a central communication bus for flexibility and ease to use. An arrangement around a central bus makes it possible to easily expand the system constantly and thereby add on further quality-enhancing and special functions for the picture, text and/or sound processing at no great expense. A non-volatile storage element, in which the factory settings are stored, is associated to this control processor.
2.- The video functions are mainly the video signal processing and some additional features like for example deflection, a detailed description follows in the paper. However, the key point for VLSI implementations is a well-organized definition of the macro-blocks. This serves to facilitate interconnection of circuit components, and minimizes power consumption, which can be considerable at the processor speeds needed.
3.- The digital concept facilitates the decoding of today’s new digital sound broadcasting standards as well as the input of external signal sources, such as Digital Audio Tape (DAT) and Compact Disk (CD). Programmability permits mono, stereo, and multilingual broadcasts; the compatibility with other functions in the TV system is resolved with the common communication bus. This leads us to part two which is dedicated to the description of this problem.
4.- With a digital system, it is possible to add some special or quality-enhancing functions simply by incorporating a single additional macro-function or chip. Therefore, standards are no longer so important due to the high level of adaptability of digital solutions. For example adaptation to a 16:9 picture tube is easy.

In this chapter we first discuss the digitization of TV functions by analyzing general concepts based on existing systems. The second section deals with silicon technologies and, in particular design methodologies concerns. The intensive use of submicron technologies associated with fast on chip clock frequencies and huge numbers of transistors on the same substrate affects traditional methods of designing chips. As this chapter only outlines a general approach of the VLSI integration techniques for Digital TV



The idea of digitization of TV functions is not new. The time some companies have started to work on it, silicon technology was not really adequate for the needed computing power so that the most effective solutions were full custom designs. This forced the block-oriented architecture where the digital functions introduced were the one to one replacement of an existing analog function. In Figure 2 there is a simplified representation of the general concept.









Fig.2: Block Diagram of first generation digital TV set
The natural separation of video and audio resulted in some incompatibilities and duplication of primary functions. The emitting principle is not changed, redundancy is a big handicap, for example the time a SECAM channel is running, the PAL functions are not in operation. New generations of digital TV systems should re-think the whole concept top down before VLSI system partitioning.
In today’s state-of-the-art solution one can recognize all the basic functions of the analog TV set with, however, a modularity in the concept, permitting additional features becomes possible, some special digital possibilities are exploited, e.g. storage and filtering techniques to improve signal reproduction (adaptive filtering, 100 Hz technology), to integrate special functions (picture-in-picture, zoom, still picture) or to receive digital broadcasting standards (MAC, NICAM). The Figure 3 shows the ITT Semiconductors solution which was the first on the market in 1983 !! !!











Fig.3: The DIGIT2000 TV receiver block diagram

Description:
This invention relates generally to digital television receivers and, particularly, to digital television receivers arranged for economical interfacing with a plurality of auxiliary devices.

With the proliferation of low cost microprocessors and microprocessor controlled devices, television (TV) receivers are being designed to utilize digitized signals and controls. There are many advantages associated with digital TV receivers, including uniformity of product, precise control of signal parameters and operating conditions, elimination of mechanical switches and a potential for reliability that has been heretofore unknown. Digital television receivers include a high speed communication bus for interconnecting a central control unit microprocessor (CCU) with various TV function modules for processing a TV signal. These modules include a deflection processing unit (DPU), a video processing unit (VPU), an automatic phase control (APC), a video codec unit (VCU), an audio analog to digital converter (ADC) and an audio processing unit (APU). The CCU has associated with it a non-volatile memory, a hardware-generated clock signal source and a suitable interface circuit for enabling the CCU to control processing of the TV signal throughout the various TV function modules. The received TV signal is in analog form and suitable analog to digital (A/D) converters and digital to analog (D/A) converters are provided for converting the digital and analog signals for signal processing and for reconverting them after processing for driving a cathode ray tube (CRT) and suitable speakers. The CCU microprocessor is heavily burdened because of the high speed timing required to control the various TV function modules.
To further complicate matters, modern TV receivers are increasingly being used with auxiliary devices for other than simple processing of TV signals. For example, the video cassette recorder (VCR) has enabled so-called "time-shifting" of program material by recording TV signals for later, more convenient viewing. The VCR is also extensively used with prerecorded material and with programs produced by users having access to a video camera. Other auxiliary devices providing features such as "Space Phone" whereby the user is enabled to make and receive telephone calls through his TV receiver, are desirable options. Additionally, a source selector auxiliary device enables a host of different signal sources, such as cable, over-the-air antenna, video disk, video games, etc. to be connected for use with the signal processing circuitry of the TV. In addition, all of these many auxiliary devices are preferably controllable from a remote position. A great deal of flexibility is available since each of the above auxiliary devices includes a microprocessor for internally controlling functioning of the device.
In the digital TV system described, the CCU microprocessor and the microprocessors in the auxiliary devices may be conventionally arranged to communicate over the main communication bus. Such a system would entail a specialized microprocessor with a hardware-generated clock signal in each auxiliary device in order to communicate at the high speeds used on the main communication bus. A specialized microprocessor, that is, one that is hardware configured, is significantly more expensive than an off-the-shelf microprocessor. Also, the auxiliary devices may not be required, or even desired, by all users and their low volume production cost becomes very important. It would therefore be desirable to provide a digital TV in which such auxiliary devices utilized off-the-shelf microprocessors for their control.



A digital TV system includes a CCU that is interconnected by a three-wire, high speed bus to a plurality of TV signal function modules for controlling operation thereof by means of a high speed hardware generated clock signal. A software generated clock signal in the CCU is supplied on a low speed two-wire auxiliary device bus which is connected to microprocessors in a plurality of auxiliary devices for performing functions ancillary to TV signal processing. The microprocessor in each auxiliary device is an off-the-shelf type that does not require any special hardware because the timing on the auxiliary device bus is sufficiently slow to enable software monitoring of the line and data transfer.
As mentioned, the three-wire IM bus 21 is a high speed bidirectional bus in which CCU 20 functions as the master and all of the interconnected TV signal processing function modules are slaves that communicate with the CCU in accordance with the protocol established for the system. CCU 20 is also indicated as including a software generated clock which supplies a two-wire auxiliary device bus 50. Two-wire bus 50 includes a clock lead 51 and a data lead 52 coupled to a plurality of auxiliary devices. A VCR 54, including an off-the-shelf microprocessor 55, is coupled to bus 50. A Source Selector 56, including an off-the-shelf microprocessor 57, is also coupled to bus 50. Source Selector 56 has access to four RF inputs, two baseband video and audio inputs and one separate baseband audio input. It will be appreciated that Source Selector 56 may have a greater or lesser number of signal sources to which it has access. Source Selector 56 outputs are coupled to VCR 54 and also to tuner 10 and supply, under control of CCU 20 and keyboard 44, the signal from the signal source selected by keyboard 44 or IR transmitter 46 for use with the digital TV. Auxiliary device bus 50 is also coupled to a Space Phone 58 which includes an off-the-shelf microprocessor 59 and a modem 60 that is connectable to a conventional telephone terminal.
Two-wire auxiliary device bus 50 is a relatively low speed bus and there is no need for separate hardware generated clock signals to be developed by the auxiliary device microprocessors. As mentioned above, this feature involves a significant savings in the cost and complexity of the auxiliary devices.
The protocol used on the two-wire auxiliary device bus consists of a 16 bit sequence, the first eight bits of which are used for bus address commands for the auxiliary devices. Each auxiliary device may respond to 16 addresses which allows the CCU to write into or read from various storage registers in the devices which are used for control or data storage. Thus, with this low cost system, as many as 16 auxiliary devices may be connected to the auxiliary device bus. The second eight bits of the 16 bit sequence contain data which is either transferred from the CCU to the auxiliary device addressed, or transferred from the auxiliary device to the CCU, based upon the bus address used. Thus, the various bus addresses to which a given auxiliary device will respond determine whether the auxiliary device will receive data from the CCU or send data to the CCU. The clock line timing, generated by software in CCU 20, is slow enough to permit software monitoring of the line and data reception by simple auxiliary device microprocessors that are not equipped with an external interrupt feature. The timing on the auxiliary device bus is made sufficiently fast to avoid too many instruction steps or the need for special registers in CCU 20. In the system described, data is clocked every 82.5 microseconds, thus permitting a 16 bit word to be clocked in 1.32 milliseconds. A pause of 277.5 microseconds between the first 8 bits and the second 8 bits permits the slave auxiliary device to process the bus address data contained in the first 8 bits. This timing fits into the 2 millisecond timing block structure used for the CCU in controlling the DIGIT 2000 digital TV. Two-2 millisecond timing blocks have been established in the CCU, which has a 20 millisecond timing loop divided into ten-2 millisecond timing blocks. Thus, two control words may be sent to an auxiliary device every 20 milliseconds, or a request by the CCU to receive data and the actual receipt of that data may take place in that time period.



Referring to the drawing, a digital TV includes a tuner 10 coupled to an IF/Detector 12 which has a pair of outputs 13 and 14 supplying video and audio signals, respectively. Control signals for tuner 10 are supplied through an interface circuit 16 from a CCU microprocessor 20 which functions as a single master control unit for the system. Microprocessor 20 is interconnected by means of a bidirectional three-wire IM (Intermetal) bus 21 to a DPU 22, a VPU 26, an APC 30, a TTX (teletext processor) 38, an APU 36, an ADC 32 and a non-volatile memory 24. A serial control line 29 interconnects a hardware generated clock 28, VPU 26 and VCU 34. VPU 26 and VCU 34 are also interconnected by a seven wire cable and TTX 38 is interconnected with a DRAM 42. DRAM 42 is a dynamic RAM in which TTX information is stored for display. VCU 34 is supplied with video signal and supplies a digitized 7 bit grey coded video signal to VPU 24 for processing and RGB color signals to a Video Drive 40 which, in turn, supplies a cathode ray tube (not shown). A keyboard 44 is coupled to CCU 20 and includes an IR detector that is responsive to coded IR signals supplied from an IR transmitter (IRX) 46. A resident microprocessor in keyboard 44 decodes the received IR signals and generated control commands and supplies appropriate outputs to CCU 20. The diagram, as described, is substantially identical to that for a "DIGIT" 2000 VLSI Digital TV System developed by ITT Intermetal and published in Edition 1984/85 Order No. 6250-11-2E

--------------------------
By its very nature, computer technology is digital, while consumer electronics are geared to the analog world. Starts have been made only recently to digitize TV and radio broadcasts at the transmitter end (in form of DAB, DSR, D2-MAC, NICAM etc). The most difficult technical tasks involved in the integration of different media are interface matching and data compression [5].
After this second step in the integration of multimedia signals, an attempt was made towards standardization, namely, the integration of 16 identical high speed processors with communication and programmability concepts comprised in the architecture !

Many solutions proposed today (for MPEG 1 mainly) are derived from microprocessor architectures or DSPs, but there is a gap between today’s circuits and the functions needed for a real fully HDTV system. The AT&T hybrid codec [29], for instance, introduces a new way to design multimedia chips by optimizing the cost of the equipment considering both processing and memory requirements.
The concept is to provide generic architectures that can be applied to a wide variety of systems taking into account that certain functions have to be optimized and that some other complex algorithms have to be ported to generic processors.

Basics of current video coding standards

Compression methods take advantage of both data redundancy and the non-linearity of human vision. They exploit correlation in space for still images and in both space and time for video signals. Compression in space is known as intra-frame compression, while compression in time is called inter-frame compression. Generally, methods that achieve high compression ratios (10:1 to 50:1 for still images and 50:1 to 200:1 for video) use data approximations which lead to a reconstructed image not identical to the original.
Methods that cause no loss of data do exist, but their compression ratios are lower (no better than 3:1). Such techniques are used only in sensitive applications such as medical imaging. For example, artifacts introduced by a lossy algorithm into a X-ray radiograph may cause an incorrect interpretation and alter the diagnosis of a medical condition. Conversely, for commercial, industrial and consumer applications, lossy algorithms are preferred because they save storage and communication bandwidth.
Lossy algorithms also generally exploit aspects of the human visual system. For instance, the eye is much more receptive to fine detail in the luminance (or brightness) signal than in the chrominance (or color) signals. Consequently, the luminance signal is usually sampled at a higher spatial resolution. Second, the encoded representation of the luminance signal is assigned more bits (a higher dynamic) than are the chrominance signals. The eye is less sensitive to energy with high spatial frequency than with low spatial frequency [7]. Indeed, if the images on a personal computer monitor were formed by an alternating spatial signal of black and white, the human viewer would see a uniform gray instead of the alternating checkerboard pattern. This deficiency is exploited by coding the high frequency coefficients with fewer bits and the low frequency coefficients with more bits.
All these techniques add up to powerful compression algorithms. In many subjective tests, reconstructed images that were encoded with a 20:1 compression ratio are hard to distinguish from the original. Video data, even after compression at ratios of 100:1, can be decompressed with close to analog videotape quality.
Lack of open standards could slow the growth of this technology and its applications. That is why several digital video standards have been proposed:
  • JPEG (Joint Photographic Expert Group) for still pictures coding
  • H.261 at p times 64 kbit/s was proposed by the CCITT (Consultative Committee on International Telephony and Telegraphy) for teleconferencing
  • MPEG-1 (Motion Picture Expert Group) up to 1,5 Mbit/s was proposed for full motion compression on digital storage media
  • MPEG-2 was proposed for digital TV compression, the bandwith depends on the chosen level and profile [33].
Another standard, the MPEG-4 for very low bit rate coding (4 kbit/s up to 64 kbit/s) is currently being debated.

Digitalization of the fundamental TV functions is of great interest since more than 30 years. Several million of TV sets have been produced containing digital systems. However, the real and full digital system is for the future. A lot of work is done in this field today, the considerations are more technical than economical which is a normal situation for an emerging technology. The success of this new multimedia technology will be given by the applications running with this techniques.
The needed technologies and methodologies were discussed to emphasize the main parameters influencing the design of VLSI chips for Digital TV Applications like parallelization, electrical constraints, power management, scalability and so on............................... 

  ITT DIGIVISION 3486 OSCAR  CHASSIS DIGI 3 90° FST  Integrated circuit kit with a phase-locked loop for color television receivers:

 An improved kit of two integrated circuits having a phase-locked loop clock oscillator for use in a color television receiver utilizing digital signal processing is described. The voltage controlled oscillator and the phase discriminator of the phase-locked loop are each located in a different one of the two integrated circuits. The control signals for the voltage controlled oscillator are transferred into it via a two-wire digital bus. External discrete components are eliminated.
 1. An IC kit for a color-television receiver with digital signal processing, comprising first and second integrated circuits which jointly contain a clock oscillator realized in the form of a phase-locked loop for producing two-phase clock signals;
said first integrated circuit containing a phase-comparison stage said second integrated circuit containing a voltage-controlled oscillator of the phase-locked loop, control signals for said phase-locked loop being applied via a maximum of two connecting lines between said first and said second integrated circuits, characterized in that:
said phase-comparison stage comprises a digital low-pass filter at its output;
said control signals are digital signals applied to said second integrated circuit by said first integrated circuit via a first one of said connecting lines; and
said second integrated circuit comprising a first counter for counting a first one of said clock signals, and having a reset input connected to a second one of said connecting lines for receiving data clock signals during said control signals,
a shift register having parallel outputs and having a clock input to said second one connecting line, and having a serial input connected to the said first connecting line;
a storage device having parallel inputs connected to said parallel outputs of said shift register and having an enable input connected to a single counter-reading output of said first counter said single counter-reading output being selected in accordance with the relationship X/F≥1/Fd where X is the numerical value corresponding to said output, F indicates the frequency of said clock signals, and Fd indicates the frequency of said data clock signals; and
a digital-to-analog converter having inputs connected to the outputs of said storage device, and having an output connected to the control input of the oscillator.


2. An IC kit in accordance with claim 1, wherein said oscillator is a crystal oscillator selectively operable with at least two crystals, each of said crystals having a rated frequency being an integer multiple of the reference carrier frequency of a different television standard, said oscillator includes a switching stage for selecting one of said at least two crystals to be used by said oscillator,
said control signals comprise switching signals for selecting one of said crystals, and
said parallel outputs of said storage device includes outputs associated with said switching signals which are connected directly to said switching stage.


3. An IC kit in accordance with claim 2, wherein said said integer multiple frequency of each of said crystals is four times the reference carrier frequency of the corresponding television standard.

4. An IC kit for a color-television receiver with digital signal processing, comprising first and second integrated circuits which jointly contain a clock oscillator realized in the form of a phase-locked loop for producing two-phase clock signals; said first integrated circuit containing a phase-comparison stage said second integrated circuit containing a voltage-controlled oscillator of the phase-locked loop, control signals for said phase-locked loop being applied via one connecting line between said first and said second integrated circuits, characterized in that:
said phase-comparison stage comprises a digital low-pass filter at its output;
said control signals are digital signals applied to said second integrated circuit by said first integrated circuit via said one connecting line; and
said second integrated circuit comprising a first counter for counting a first one of said clock signals, and having a reset input connected to said connecting line for receiving data clock signals during said control signals,
a shift register having parallel outputs, a clock input, and having a serial input connected to the said connecting line,
a flip-flop having a first control input coupled to said single counter-reading output, a second control input coupled to said connecting line, a first output coupled to said electronic switch control input, and a second output coupled to a reset input of said second counter;
a storage device having parallel inputs connected to said parallel outputs of said shift register and having an enable input connected to a single counter-reading output of said first counter said single counter-reading output being selected in accordance with the relationship X/F≥1/Fd where X is the numerical value corresponding to said output, F indicates the frequency of said clock signals, and Fd indicates the frequency of said data clock signals; and
a second counter having a counting capacity equal to the ratio of the frequency of said clock signals to said data clock frequency,
an electronic switch having a control input and responsive to a first control signal at said first control input for connecting said first clock signal to said second counter,
a pulse shaping circuit having inputs coupled to the outputs of said second counter and having an output coupled to said shift register clock input, said pulse shaping circuit adjusting the pulse/no pulse ratio of signals at its output in accordance with the outputs of said second counter,
a digital-to-analog converter having inputs connected to the outputs of said storage device, and having an output connected to the control input of the oscillator.


5. An IC kit in accordance with claim 4, wherein said oscillator is a crystal oscillator selectively operable with at least two crystals, each of said crystals having a rated frequency being an integer multiple of the reference carrier frequency of a different television standard, said oscillator includes a switching stage for selecting one of said at least two crystals to be used by said oscillator,
said control signals comprise switching signals for selecting one of said crystals, and
said parallel outputs of said storage device includes outputs associated with said switching signals which are connected directly to said switching stage.

6. An IC kit in accordance with claim 5, wherein said integer multiple frequency of each of said crystals is four times the reference carrier frequency of the corresponding television standard.

7. An IC kit in accordance with claim 4, wherein said flip-flop is an RS flip-flop said first control input is the R input thereof, said second control input is the S input thereof, said first output is the non-inverting output thereof, and said second output is the inverting output thereof.

Description:
BACKGROUND OF THE INVENTION

This invention pertains to a kit of two integrated circuits (IC's) for a color television receiver with digital signal processing. The IC's contain a clock oscillator realized in the form of a phase-locked loop, for producing two phase clock signals. The phase comparator stage of the phase-locked loop is arranged in the first integrated circuit, and the voltage-controlled oscillator thereof is arranged in the second integrated circuit. The voltage control signals thereof are supplied via a maximum of two connecting lines from the first to the second integrated circuit. Such an IC kit is described on pages 1-3 and 1-4, 4-1 to 4-14 and 8-1 to 8-5 of the Intermetall book, "DIGIT 2000 VLSI-Digital-TV-System", March 1982. The pages beginning with the numeral 4 refer to the integrated circuit MAA 2200, and the pages beginning with the numeral 8 refer to the integrated circuit MEA 2600. On page 8-2 it is stated that the voltage-controlled oscillator as integrated into the integrated circuit MEA 2600 forms part of a phase-locked loop, with the other part thereof, i.e., the phase comparator, being integrated into the integrated circuit MAA 2200. The phase comparator stage supplies control signals to an external low-pass filter, i.e., one which is arranged outside the two integrated circuits, with this low-pass filter deriving the tuning voltages for the voltage-controlled oscillator from the control signals.
From an integration point of view, it is desirable to eliminate external discrete components. In addition thereto, the spatial separation of both the IC kit and the external low-pass filter presents a possible source of faults with respect to the DC and AC voltage behaviour of the control loop due to noise coupling owing to the voltage drop, as well as a capacitive or inductive coupling.

SUMMARY OF THE INVENTION:
It is one object of the invention to further embody the conventional IC kit in such a way that the external components required for the low-pass filter can be omitted, hence with the function thereof being replaced by integrated circuit portions. Relative thereto, care is to be taken that no more than the two already existing lines are required for transmitting the voltage control signals to the second integrated circuit. The solution to the problem resides in the digital transmission of the phase comparison signals and, additionally, in the inclusion of the low-pass filter in the IC kit as a digital low-pass filter.
In accordance with the principles of the invention, a first integrated circuit includes a phase-comparison stage with a digital low-pass filter and a second integrated circuit includes a voltage-controlled oscillator of a phase-locked loop. Digital control signals are applied to the second integrated circuit via a connecting line. The second integrated circuit contains a counter for counting the clock signal and the reset input thereof is connected to a second connecting line conducting the corresponding data clock signals. The second integrated circuit further includes a shift register having its clock input connected to the second connecting line, and having its serial input connected to the first connecting line. The second integrated circuit contains a storage device having a parallel input connected to the parallel output of the shift register, and having an enable input connected to a counter-reading output of the counter selected such that the following relationship applies: X/F≥1/Fd (X=numerical value of x; F=frequency of the clock signals and Fd=frequency of the data clock signals.) The second integrated circuit includes a digital-to-analog converter having its input connected to the parallel output of the storage device and its output connected to the control input of the oscillator.
In accordance with the invention, it is possible to do without an external integrating circuit for the control signals, and at most only two connecting lines are required for transmitting the control signals; in a modified arrangement in accordance with the invention even only one such connecting line is required.

BRIEF DESCRIPTION OF THE DRAWINGS
The invention will now be better understood from a reading of the following detailed description in greater detail with reference to FIGS. 1 to 5 of the accompanying drawing, in which:
FIG. 1 schematically and by way of a block diagram, shows the partial circuits of the IC kit essential to the invention;
FIG. 2 shows time diagrams of three signals appearing in the arrangement according to FIG. 1;
FIG. 3 shows a further embodiment of the arrangement according to FIG. 1;
FIG. 4, schematically and by way of a block diagram, shows a modification of the IC kit according to the invention employing only one connecting line for the control signals; and
FIG. 5 shows signal curves as a function of time relating to various signals appearing in the arrangement according to FIG. 4.

DETAILED DESCRIPTION
FIG. 1, schematically and in the form of a block diagram, shows a portion of the IC kit consisting of the first integrated circuit ic1 and of the second integrated circuit ic2. It is important with respect to the invention, that these parts together form a phase-locked loop (PLL), with the two essential partial circuits thereof being divided among the two integrated circuits. Thus, in the first integrated circuit ic1 there is contained, amongst others, the phase-comparison stage p with a digital low-pass filter tp at its output, whereas the voltage-controlled oscillator vc is arranged in the second integrated circuit ic2. According to the invention, the control signals cs for controlling the voltage of the oscillator vc, are transmitted via the first connecting line 11 from the first integrated circuit ic1 to the second integrated circuit ic2. These control signals, in accordance with a feature of the invention, are digital signals shown as curve c in FIG. 2.
Data clock signals fd are transmitted on the second connecting line l2 for the time-coordinated transmission of the digital control signals cs. Curve b of FIG. 2 shows the data clock signals fd as a function of time. The data clock signals fd can be formed in a suitable stage within the first integrated circuit ic1, to the input of which there is fed one of the clock signals f1, f2 produced by the voltage-controlled oscillator vc and, consequently by the clock oscillator. The signal curves as functions of time as shown in FIG. 2 require that the data clock signal fd is derived from the first clock signal f1 by means of a frequency divider 1 whose divisional number is 4.
The second integrated circuit ic2 contains a first counter z1 having a counting input ez for receiving the first clock signal f1 of the voltage-controlled oscillator vc. Accordingly, the first counter z1 counts these clock pulses. Counter z1 has a reset input er connected to the second connecting line l2, so that the data clock signals fd are fed thereto.
Moreover, the second integrated circuit ic2 contains a shift register sr having its series input es coupled to the first connecting line 11 for receiving the digital control signals cs. The clock input et of the shift register sr is connected to the second connecting line l2 and is thus supplied with the data clock signals fd. The parallel output of the shift register sr is connected to the parallel input of a storage device s. Storage device s has a parallel output connected to the parallel input of a digital-to-analog converter da. Storage device s has an enable input eu connected to the single counter-reading output x of the first counter z1. This counter-reading output x chosen such that with respect to the numerical value X associated therewith, as well as with respect to the frequency F of the clock signals f1, f2 and the frequency Fd of the data clock signals fd, there applies the following relationship: X/F≥1/Fd.
If the arrangement according to the invention as shown in FIG. 1 operates with a positive logic, the reset input er of the first counter z1 is reset in response to the more negative level L of two binary signal levels H, L. The enable input eu of the storage device s, however, responds to a corresponding H-level at the counter-reading output x. In other words, in response to the appearance of an H-level, the digital signals appearing at the parallel output of the shift register sr, are written into the storage device s. This writing only takes place at the end of a data word transmitted on the first connecting line 11, with this end being determined by the disconnection of the data clock signal fd and the clamping of the second connecting line 12 to a permanent H level as shown in curve b of FIG. 2. Owing to this permanent H level, the first counter z1 can reach the counter reading X, and consequently, the writing of the data into the storage device s can begin. During the appearance of the data clock signal fd, the first counter z1 cannot reach the counter reading X due to the L level periodically contained in the data clock signal fd which resets first counter z1. Due to the temporal assignment of the data clock signals fd to the duration of the control signals cs, as well as due to the partial circuits z1, da, sr, as provided for the second integrated circuit ic2, the two connecting lines are sufficient in the case of a digital control signal transmission.
The (time) relation between the digital control signals cs and the data clock signals fd can be chosen such that the shifting of the data within the shift register sr only takes place after a stationary state has been reached again in the individual stages compared with the preceding shifting cycle, that is when the data are stable. In FIG. 2, this is indicated on the left by the edges of the data clock signal fd marked by two small circles. At the falling edge the data are read in at the input, and at the rising edge they are shifted.
FIG. 3 shows a further embodiment of the arrangement according to FIG. 1 for multi-standard color television receivers. In this case, the clock oscillator is designed as a crystal oscillator comprising several crystals q1, q2. The rated frequency of the crystals is an integer multiple, preferably four times of the reference carrier frequency of different television standards. Thus, for example, the crystal q1 may be provided for the PAL standard, and the crystal q2 may be provided for the NTSC standard. Arrangements employing a third crystal for the SECAM standard are also possible. In this further embodiment, not only are the digital control signals cs transmitted over the first connecting line 11, but also the corresponding switching signals ss which are produced in the first integrated circuit ic1 with the corresponding selecting stage w.
Both the shift register sr and the storage device s have a sufficient number of stages to hold a data word sc' consisting of the digital control signals cs and of the switching signals ss. The outputs for the stages of the storage device s which are provided for the switching signals ss, are connected to the inputs of the switching stage vs of the oscillator vc. As in the arrangement of FIG. 1, the output of the digital-to-analog converter da is connected to the control input ec of the oscillator vc.
FIG. 4 shows such a modification of the invention wherein one single connecting line 1 is required for transmitting the digital control signals cs. Relative thereto, in the integrated circuit ic1 the control signals cs are correspondingly combined with the data clock signals fd.
In integrated circuit ic2, in addition to the stages provided for in the arrangements according to FIGS. 1 and 3, there is provided a second counter z2, an electronic switch s1 and a RS storage flip-flop ff. The first clock signal f1 is applied to the counting input ez of the second counter z2 via the switching section of the electronic switch s1. The counting capacity of the second counter z2 is equal to the ratio of the clock frequency F to the data clock frequency Fd.
The connecting line 1 is connected to the serial input es of the shift register sr and to the reset input er of the first counter z1. The single counter-reading output x of counter z1 is connected to the enable input eu of the storage device s and to the R input of the RS storage flip-flop ff. The inverting S input of flip-flop ff is connected to the connecting line 1. The Q output of flip-flop ff is connected to the reset input of the second counter z2, and the Q output is connected to the control input of the electronic switch s1.
FIG. 5 shows the signal waveforms occurring in the arrangement according to FIG. 4. Curve 5a shows the first clock signal f1, and curve 5b shows the combined control-data clock signal cs+fd. The hatchlined portions of curve 5b represent the digital data. Curve 5c shows the signal ts as applied to the clock input et of the shift register sr. Signal ts is produced by the pulse shaper is of FIG. 4 in such a way that the trailing edge thereof occurs in about the center of the second half of its pulse duration. For this purpose, the highest counter-reading output of the second counter z2 together with the preceding counter-outputs are applied to the input of the pulse-forming circuit adjusting the pulse / no pulse ratio.
Curve 5d shows the signal as appearing at the counter-reading output x of the first counter z1. The numerals 1 through 6 in FIG. 5b indicate various time positions which are essential to the mode of operation. The reference numeral 1 indicates the beginning of the shifting operation in the shift register sr. At the time position 2, the data are loaded into the shift register. At the time position 3, the last shifting pulse of the shown data word is started, and 4 indicates the last load time position. 5 and 6 indicate the time positions of writing the data word into the storage device s, with the writing being carried out, depending on the last data state, at one of these two time positions.
An advantage of the invention is that no external discrete components are required for producing the control voltage for the oscillator vc. Moreover, a considerable reduction in both the level dependence and the sensitivity to noise is achieved.
It should be apparent to those skilled in the art that the modification according to FIG. 4 may also be applied to the embodiment as shown in FIG. 3



  


ITT DIGIVISION 3486 OSCAR CHASSIS DIGI 3 90° FST / IFB-285/2 Synchronized switch-mode power supply:
In a switch mode power supply, a first switching transistor is coupled to a primary winding of an isolation transformer. A second switching transistor periodically applies a low impedance across a second winding of the transformer that is coupled to an oscillator for synchronizing the oscillator to the horizontal frequency. A third winding of the transformer is coupled via a switching diode to a capacitor of a control circuit for developing a DC control voltage in the capacitor that varies in accordance with a supply voltage B+. The control voltage is applied via the transformer to a pulse width modulator that is responsive to the oscillator output signal for producing a pulse-width modulated control signal. The control signal is applied to a mains coupled chopper transistor for generating and regulating the supply voltage B+ in accordance with the pulse width modulation of the control signal.



Description:
The invention relates to switch-mode power supplies.
Some television receivers have signal terminals for receiving, for example, external video input signals such as R, G and B input signals, that are to be developed relative to the common conductor of the receiver. Such signal terminals and the receiver common conductor may be coupled to corresponding signal terminals and common conductors of external devices, such as, for example, a VCR or a teletext decoder.
To simplify the coupling of signals between the external devices and the television receiver, the common conductors of the receiver and of the external devices are connected together so that all are at the same potential. The signal lines of each external device are coupled to the corresponding signal terminals of the receiver. In such an arrangement, the common conductor of each device, such as of the television receiver, may be held "floating", or conductively isolated, relative to the corresponding AC mains supply source that energizes the device. When the common conductor is held floating, a user touching a terminal that is at the potential of the common conductor will not suffer an electrical shock.
Therefore, it may be desirable to isolate the common conductor, or ground, of, for example, the television receiver from the potentials of the terminals of the AC mains supply source that provide power to the television receiver. Such isolation is typically achieved by a transformer. The isolated common conductor is sometimes referred to as a "cold" ground conductor.
In a typical switch mode power supply (SMPS) of a television receiver the AC mains supply voltage is coupled, for example, directly, and without using transformer coupling, to a bridge rectifier. An unregulated direct current (DC) input supply voltage is produced that is, for example, referenced to a common conductor, referred to as "hot" ground, and that is conductively isolated from the cold ground conductor. A pulse width modulator controls the duty cycle of a chopper transistor switch that applies the unregulated supply voltage across a primary winding of an isolating flyback transformer. A flyback voltage at a frequency that is determined by the modulator is developed at a secondary winding of the transformer and is rectified to produce a DC output supply voltage such as a voltage B+ that energizes a horizontal deflection circuit of the television receiver. The primary winding of the flyback transformer is, for example, conductively coupled to the hot ground conductor. The secondary winding of the flyback transformer and voltage B+ may be conductively isolated from the hot ground conductor by the hot-cold barrier formed by the transformer.
It may be desirable to synchronize the operation of the chopper transistor to horizontal scanning frequency for preventing the occurrence of an objectionable visual pattern in an image displayed in a display of the television receiver.
It may be further desirable to couple a horizontal synchronizing signal that is referenced to the cold ground to the pulse-width modulator that is referenced to the hot ground such that isolation is maintained.
A synchronized switch mode power supply, embodying an aspect of the invention, includes a transfromer having first and second windings. A first switching arrangement is coupled to the first winding for generating a first switching current in the first winding to periodically energize the second winding. A source of a synchronizing input signal at a frequency that is related to a deflection frequency is provided. A second switching arrangement responsive to the input signal and coupled to the second winding periodically applies a low impedance across the energized second winding that by transformer action produces a substantial increase in the first switching current. A periodic first control signal is generated. The increase in the first switching current is sensed to synchronize the first control signal to the input signal. An output supply voltage is generated from an input supply voltage in accordance with the first control signal.

















ITT DIGIVISION 3486 OSCAR CHASSIS DIGI 3 90° FST / IFB-285/2 CIRCUIT ARRANGEMENT IN A PICTURE DISPLAY DEVICE UTILIZING A STABILIZED SUPPLY VOLTAGE CIRCUIT:
Line synchronized switch mode power supply:

A stabilized supply voltage circuit for a picture display device comprising a chopper wherein the switching signal has the line frequency and is duration-modulated. The coil of the chopper constitutes the primary winding of a transformer a secondary winding of which drives the line output transistor so that the switching transistor of the chopper also functions as a driver for the line output stage. The oscillator generating the switching signal may be the line oscillator. In a special embodiment the driver and line output transistor conduct simultaneously and in order to limit the base current of the line output transistor a coil shunted by a diode is incorporated in the drive line of the line output transistor. Other secondary windings of the transformer drive diodes which conduct simultaneously with the efficiency diode of the chopper so as to generate further stabilized supply voltages.



1. An electrical circuit arrangement for a picture display device operating at a given line scanning frequency, comprising a source of unidirectional voltage, an inductor, first switching transistor means for periodically energizing said inductor at said scanning frequency with current from said source, an electrical load circuit coupled to said inductor and having applied thereto a voltage as determined by the ratio of the ON and OFF periods of said transistor, means for maintaining the voltage across said load circuit at a given value comprising means for comparing the voltage of said load circuit with a reference voltage, means responsive to departures of the value of the load circuit voltage from the value of said reference voltage for varying the conduction ratio of the ON and OFF periods of said transistor thereby to stabilize said load circuit voltage at the given value, a line deflection coil system for said picture display device, means for energizing said line deflection coil system from said load voltage circuit means, means for periodically interrupting the energization of said line deflection coil comprising second switching means and means coupled to said inductor for deriving therefrom a switching current in synchronism with the energization periods of said transistor and applying said switching current to said switching means thereby to actuate the same, and means coupled to said switching means and to said load voltage circuit for producing a voltage for energizing said 2. A circuit as claimed in claim 1 wherein the duty cycle of said switching 3. A circuit as claimed in claim 1 further comprising an efficiency first 4. A circuit as claimed in claim 3 further comprising at least a second diode coupled to said deriving means and to ground, and being poled to 5. A circuit as claimed in claim 1 wherein said second switching means comprises a second transistor coupled to said deriving means to conduct simultaneously with said first transistor, and further comprising a coil coupled between said driving means and said second transistor and a third diode shunt coupled to said coil and being poled to conduct when said 6. A circuit as claimed in claim 1 further comprising a horizontal oscillator coupled to said first transistor, said oscillator being the 7. A circuit as claimed in claim 1 further comprising means coupled to said inductor for deriving filament voltage for said display device.

Description:
The invention relates to a circuit arrangement in a picture display device wherein the input direct voltage between two input terminals, which is obtained be rectifying the mains alternating voltage, is converted into a stabilized output direct voltage by means of a switching transistor and a coil and wherein the transistor is connected to a first input terminal and an efficiency diode is connected to the junction of the transistor and the coil. The switching transistor is driven by a pulsatory voltage of line frequency which pulses are duration-modulated in order to saturate the switching transistor during part of the period dependent on the direct voltage to be stabilized and to cut off this transistor during the remaining part of the period. The pulse duration modulation is effected by means of a comparison circuit which compares the direct voltage to be stabilized with a substantially constant voltage, the coil constituting the primary winding of a transformer.

Such a circuit arrangement is known from German "Auslegeschrift" 1.293.304. wherein a circuit arrangement is described which has for its object to convert an input direct voltage which is generated between two terminals into a different direct voltage. The circuit employs a switch connected to the first terminal of the input voltage and periodically opens and closes so that the input voltage is converted into a pulsatory voltage. This pulsatory voltage is then applied to a coil. A diode is arranged between the junction of the switch and the coil and the second terminal of the input voltage whilst a load and a charge capacitor in parallel thereto are arranged between the other end of the coil and the second terminal of the input voltage. The assembly operates in accordance with the known efficiency principle i.e., the current supplied to the load flows alternately through the switch and through the diode. The function of the switch is performed by a switching transistor which is driven by a periodical pulsatory voltage which saturates this transistor for a given part of the period. Such a configuration is known under different names in the literature; it will be referred to herein as a "chopper." A known advantage thereof, is that the switching transistor must be able to stand a high voltage or provide a great current but it need not dissipate a great power. The output voltage of the chopper is compared with a constant reference voltage. If the output voltage attempts to vary because the input voltage and/or the load varies, a voltage causing a duration modulation of the pulses is produced at the output of the comparison arrangement. As a result the quantity of the energy stored in the coil varies and the output voltage is maintained constant. In the German "Auslegeschrift" referred to it is therefore an object to provide a stabilized supply voltage device.

In the circuit arrangement according to the mentioned German "Auslegeschrift" the frequency of the load variations or a harmonic thereof is chosen as the frequency for the switching voltage. Particularly when the load fed by the chopper is the line deflection circuit of a picture display device, wherein thus the impedance of the load varies in the rhythm of the line frequency, the frequency of the switching voltage is equal to or is a multiple of the line frequency.

It is to be noted that the chopper need not necessarily be formed as that in the mentioned German "Auslegeschrift." In fact, it is known from literature that the efficiency diode and the coil may be exchanged. It is alternatively possible for the coil to be provided at the first terminal of the input voltage whilst the switching transistor is arranged between the other end and the second terminal of the input voltage. The efficiency diode is then provided between the junction of said end and the switching transistor and the load. It may be recognized that for all these modifications a voltage is present across the connections of the coil which voltage has the same frequency and the same shape as the pulsatory switching voltage. The control voltage of a line deflection circuit is a pulsatory voltage which causes the line output transistor to be saturates and cut off alternately. The invention is based on the recognition that the voltage present across the connections of the coil is suitable to function as such a control voltage and that the coil constitutes the primary of a transformer. To this end the circuit arrangement according to the invention is characterized in that a secondary winding of the transformer drives the switching element which applies a line deflection current to line deflection coils and by which the voltage for the final anode of a picture display tube which forms part of the picture display device is generated, and that the ratio between the period during which the switching transistor is saturated and the entire period, i.e., the switching transistor duty cycle is between 0.3 and 0.7 during normal operation.

The invention is also based on the recognition that the duration modulation which is necessary to stabilize the supply voltage with the switching transistor does not exert influence on the driving of the line output transistor. This resides in the fact that in case of a longer or shorter cut-off period of the line output transistor the current flowing through the line deflection coils thereof is not influenced because of the efficiency diode current and transistor current are taken over or, in case of a special kind of transistor, the collector-emitter current is taken over by the base collector current and conversely. However, in that case the above-mentioned ratios of 0.3 : 0.7 should be taken into account since otherwise this take-over principle is jeopardized.

As will be further explained the use of the switching transistor as a driver for the line output transistor in an embodiment to be especially described hereinafter has the further advantage that the line output transistor automatically becomes non-conductive when this switching transistor is short circuited so that the deflection and the EHT for the display tube drop out and thus avoid damage thereof.

Due to the step according to the invention the switching transistor in the stabilized supply functions as
a driver for the line deflection circuit. The circuit arrangement according to the invention may in addition be equipped with a very efficient safety circuit so that the reliability is considerably enhanced, which is described in the U.S. Pat. No. 3,629,686. The invention is furthermore based on the recognition of the fact that the pulsatory voltage present across the connections of the coil is furthermore used and to this end the circuit arrangement according to the invention is characterized in that secondary windings of the transformer drive diodes which conduct simultaneously with the efficiency diode so as to generate further stabilized direct voltages, one end of said diodes being connected to ground.

In order that the invention may be readily carried into effect, a few embodiments thereof will now be described in detail by way of example with reference to the accompanying diagrammatic drawings in which:

FIG. 1 shows a principle circuit diagram wherein the chopper and the line deflection circuit are further shown but other circuits are not further shown.

FIGS. 2a, 2b and 2c show the variation as a function of time of two currents and of a voltage occurring in the circuit arrangement according to FIG. 1.

FIGS. 3a 3b, 3c and 3d show other embodiments of the chopper.

FIGS. 4a and 4b show modifications of part of the circuit arrangement of FIG. 1.

In FIG. 1 the reference numeral 1 denotes a rectifier circuit which converts the mains voltage supplied thereto into a non-stabilized direct voltage. The collector of a switching transistor 2 is connected to one of the two terminals between which this direct voltage is obtained, said transistor being of the npn-type in this embodiment and the base of which receives a pulsatory voltage which originates through a control stage 4 from a modulator 5 and causes transistor 2 to be saturated and cut off alternately. The voltage waveform 3 is produced at the emitter of transistor 2. In order to maintain the output voltage of the circuit arrangement constant, the duration of the pulses provided is varied in modulator 5. A pulse oscillator 6 supplies the pulsatory voltage to modulator 5 and is synchronized by a signal of line frequency which originates from the line oscillator 6' present in the picture display device. This line oscillator 6' is in turn directly synchronized in known manner by pulses 7' of line frequency which are present in the device and originate for example from a received television signal if the picture display device is a television receiver. Pulse oscillator 6 thus generates a pulsatory voltage the repetition frequency of which is the line frequency.

The emitter of switching transistor 2 is connected at one end to the cathode of an efficiency diode 7 whose other end is connected to the second input voltage terminal and at the other end to primary winding 8 of a transformer 9. Pulsatory voltage 3 which is produced at the cathode of efficiency diode 7 is clamped against the potential of said second terminal during the intervals when this diode conducts. During the other intervals the pulsatory voltage 3 assumes the value V i . A charge capacitor 10 and a load 11 are arranged between the other end of winding 8 and the second input voltage terminal. The elements 2,7,8,10 and 11 constitute a so-called chopper producing a direct voltage across charge capacitor 10, provided that capacitor 10 has a sufficiently great value for the line frequency and the current applied to load 11 flowing alternately through switching transistor 2 or through efficiency diode 7. The output voltage V o which is the direct voltage produced across charge capacitor 10 is applied to a comparison circuit 12 which compares the voltage V o with a reference voltage. Comparison circuit 12 generates a direct voltage which is applied to modulator 5 so that the duration of the effective period δ T of switching transistor 2 relative to the period T of pulses 3 varies as a function of the variations of output voltage V 0 . In fact, it is readily evident that output voltage V o is proportional to the ratio δ :

V o = V i . δ

Load 11 of the chopper consists in the consumption of parts of the picture display device which are fed by output voltage V 0 . In a practical embodiment of the circuit arrangement according to FIG. 1 wherein the mains alternating voltage has a nominal effective value of 220 V and the rectified voltage V i is approximately 270 V, output voltage V o for δ = 0.5 is approximately 135 V. This makes it also possible, for example, to feed a line deflection circuit as is shown in FIG. 1 wherein load 11 then represents different parts which are fed by the chopper. Since voltage V o is maintained constant due to pulse duration modulation, the supply voltage of this line deflection circuit remains constant with the favorable result that the line amplitude(= the width of the picture displayed on the screen of the picture display tube) likewise remains constant as well as the EHT required for the final anode of the picture display tube in the same circuit arrangement independent of the variations in the mains voltage and the load on the EHT generator (= variations in brightness).

However, variations in the line amplitude and the EHT may occur as a result of an insufficiently small internal impedance of the EHT generator. Compensation means are known for this purpose. A possibility within the scope of the present invention is to use comparison circuit 12 for this purpose. In fact, if the beam current passes through an element having a substantially quadratic characteristic, for example, a voltage-dependent resistor, then a variation for voltage V o may be obtained through comparison circuit 12 which variation is proportional to the root of the variation in the EHT which is a known condition for the line amplitude to remain constant.

In addition this facilitates smoothing of voltage V o since the repetition frequency of pulsatory voltage 3 is many times higher than that of the mains and a comparatively small value may be sufficient for charge capacitor 10. If charge capacitor 10 has a sufficiently high value for the line frequency, voltage V o is indeed a direct voltage so that a voltage having the same form as pulsatory voltage 3 is produced across the terminals of primary winding 8. Thus voltages which have the same shape as pulsatory voltage 3 but have a greater or smaller amplitude are produced across secondary windings 13, 14 of transformer 9 (FIG. 1 shows only 2 secondary windings but there may be more). The invention is based on the recognition that one end of each secondary winding is connected to earth while the other end thereof drives a diode, the winding sense of each winding and the direction of conductance of each diode being chosen to be such that these diodes conduct during the same period as does efficiency diode 7. After smoothing, stabilized supply voltages, for example, at terminal 15 are generated in this manner at the amplitudes and polarities required for the circuit arrangements present in the picture display device. In FIG. 1 the voltage generated at terminal 15 is, for example, positive relative to earth. It is to be noted that the load currents of the supply voltages obtained in this manner cause a reduction of the switching power which is economized by efficiency diode 7. The sum of all diode currents including that of diode 7 is in fact equal to the current which would flow through diode 7 if no secondary winding were wound on transformer 9 and if no simultaneous diode were used. This reduction may be considered an additional advantage of the circuit arrangement according to the invention, for a diode suitable for smaller powers may then be used. However, it will be evident that the overall secondary load must not exceed the primary load since otherwise there is the risk of efficiency diode 7 being blocked so that stabilization of the secondary supply voltages would be out of the question.

It is to be noted that a parabola voltage of line frequency as shown at 28 is produced across the charge capacitor 10 if this capacitor is given a smaller capacitance so that consequently the so-called S-correction is established.

In FIG. 1 charge capacitors are arranged between terminals 15 etc. and earth so as to ensure that the voltages on these points are stabilized direct voltages. If in addition the mean value of the voltage on one of these terminals has been made equal to the effective value of the alternating voltage which is required for heating the filament of the picture display tube present in the picture display device, this voltage is suitable for this heating. This is a further advantage of the invention since the cheap generation of a stabilized filament voltage for the picture display tube has always been a difficult problem in transistorized arrangements.

A further advantage of the picture display device according to the invention is that transformer 9 can function as a separation transformer so that the different secondary windings can be separated from the mains and their lower ends can be connected to ground of the picture display device. The latter step makes it possible to connect a different apparatus such as, for example, a magnetic recording and/or playback apparatus to the picture display device without earth connection problems occurring.

In FIG. 1 the reference numeral 14 denotes a secondary winding of transformer 9 which in accordance with the previously mentioned recognition of the invention can drive line output transistor 16 of the line deflection circuit 17. Line deflection circuit 17 which is shown in a simplified form in FIG. 1 includes inter alia line deflection coils 18 and an EHT transformer 19 a secondary winding 20 of which serves for generating the EHT required for the acceleration anode of the picture display tube. Line deflection circuit 17 is fed by the output voltage V o of the chopper which voltage is stabilized due to the pulse duration modulation with all previously mentioned advantages. Line deflection circuit 17 corresponds, for example, to similar arrangements which have been described in U.S. Pat. No. 3,504,224 issued Mar. 31, 1970 to J.J. Reichgelt et al., U.S. patent application Ser. No. 737,009 filed June 14, 1968 by W. H. Hetterscheid and U.S. application Ser. No. 26,497 filed April 8, 1970 by W. Hetterscheid et al. It will be evident that differently formed lined deflection circuits are alternatively possible.

It will now be shown that secondary winding 14 can indeed drive a line deflection circuit so that switching transistor 2 can function as a driver for the line deflection. FIGS. 2a and b show the variation as a function of time of the current i C which flows in the collector of transistor 16 and of the drive voltage v 14 across the terminals of secondary winding 14. During the flyback period (0, t 1 ) transistor 16 must be fully cut off because a high voltage peak is then produced at its collector; voltage v 14 must then be absolutely negative. During the scan period (t 1 , t 4 ) a sawtooth current i C flows through the collector electrode of transistor 16 which current is first negative and then changes its direction. As the circuit arrangement is not free from loss, the instant t 3 when current i C becomes zero lies, as is known, before the middle of the scan period. At the end t 4 of the scan period transistor 16 must be switched off again. However, since transistor 16 is saturated during the scan period and since this transistor must be suitable for high voltages and great powers so that its collector layer is thick, this transistor has a very great excess of charge carriers in both its base and collector layers. The removal of these charge carriers takes a period t s which is not negligible whereafter the transistor is indeed switched off. Thus the fraction δ T of the line period T at which v 14 is positive must end at the latest at the instant (t 4 - t s ) located after the commencement (t = 0) of the previous flyback.

The time δ T may be initiated at any instant t 2 which is located between the end t 1 of the flyback period and the instant t 3 when collector current i C reverses its direction. It is true that emitter current flows through transistor 16 at the instant t 2 , but collector current i C is not influenced thereby, at least not when the supply voltage (= V o ) for line deflection circuit 17 is high enough. All this has been described in the U.S. Pat. No. 3,504,224. The same applies to line deflection circuits wherein the collector base diode does not function as an efficiency diode as is the case in the described circuit 17, but wherein an efficiency diode is arranged between collector and emitter of the line output transistor. In such a case the negative part of the current i C of FIG. 2a represents the current flowing through the said efficiency diode.

After the instant t 3 voltage v 14 must be positive. In other words, the minimum duration of the period T when voltage v 14 must be positive is (t 4 - t s ) - t 3 whilst the maximum duration thereof is (t 4 - t s ) - t 1 . In a television system employing 625 lines per raster the line period t 4 is approximately 64 μus and the flyback period is approximately 12 μus. Without losses in the circuit arrangement instant t 3 would be located approximately 26 μus after the instant t 1 , and with losses a reasonable value is 22 μus which is 34 μus after the commencement of the period. If for safety's sake it is assumed that t s lasts approximately 10 μus, the extreme values of δ T are approximately 20 and 42 μus and consequently the values for δ are approximately 0.31 and 0.66 at a mean value which is equal to approximately 0.49. It was previously stated that a mean value of δ = 0.5 was suitable. Line deflection circuit 17 can therefore indeed be used in combination with the chopper in the manner described, and the relative variation of δ may be (0.66 - 0.31) : 0.49 = 71.5 percent. This is more than necessary to obviate the variations in the mains voltage or in the various loads and to establish the East-West modulation and ripple compensation to be described hereinafter. In fact, if it is assumed that the mains voltage varies between -15 and +10 percent of the nominal value of 220 V, while the 50 Hz ripple voltage which is superimposed on the input voltage V i has a peak-to-peak value of 40 V and V i is nominally 270 V, then the lowest occurring V i is:

0.85 × 270 V - 20 V = 210 V and the highest occurring V i is

1.1 × 270 V + 20 V = 320 V. For an output voltage V o of 135 V the ratio must thus vary between

δ = 135 : 210 = 0.64 and δ = 135 : 320 = 0.42.

A considerable problem presenting itself is that of the simultaneous or non-simultaneous drive of line output transistor 16 with switching transistor 2, it being understood that in case of simultaneous drive both transistors are simultaneously bottomed, that is during the period δ T. This depends on the winding sense of secondary winding 14 relative to that of primary winding 8. In FIG. 1 it has been assumed that the drive takes place simultaneously so that the voltage present across winding 14 has the shape shown in FIG. 2b. This voltage assumes the value n(V i - V o ) in the period δ T and the value -nVo in the period (1 - δ )T, wherein n is the ratio of the number of turns on windings 14 and 8 and wherein V o is maintained constant at nominal mains voltage V o = δ V inom . However, if as a result of an increase or a decrease of the mains voltage V i increases or decreases proportionally therewith, i.e., V i = V i nom + Δ V, the positive portion of V 14 becomes equal to n(V i nom - V o +Δ V) = n [(1 -δ)V i nom +ΔV] = n(0.5 V inom +ΔV) if δ = 0.5 for V i = V i nom. Relatively, this is a variation which is twice as great. For example, if V i nom = 270 V and V o = 135 V, a variation in the mains voltage of from -15 to +10 percent causes a variation of V i of from -40.5 V to +27 V which ranges from -30 to +20 percent of 135 V which is present across winding 8 during the period δ T. The result is that transistor 16 can always be bottomed over a large range of variation. If the signal of FIG. 2b would be applied through a resistor to the base of transistor 16, the base current thereof would have to undergo the same variation while the transistor would already be saturated in case of too low a voltage. In this case it is assumed that transformer 9 is ideal (without loss) and that coil 21 has a small inductance as is explained in the U.S. patent application Ser. No. 737,009 above mentioned. It is therefore found to be desirable to limit the base current of transistor 16.

This may be effected by providing a coil 22 having a large value inductance, approximately 100 μH, between winding 14 and the small coil 21. The variation of said base current i b is shown in FIG. 2c but not to the same scale as the collector current of FIG. 2a. During the conducting interval δ T current i b varies as a linear function of time having a final value of wherein L represents the inductance of coil 22. This not only provides the advantage that this final value is not immediately reached, but it can be shown that variation of this final value as a function of the mains voltage has been reduced, for there applies at nominal mains voltage that: If the mains voltage V i = V i nom +Δ V, then ##SPC1## because V i nom = 2 V o . Thus this variation is equal to that of the mains voltage and is not twice as great.

During switching off, t 2 , of transistor 16 coil 22 must exert no influence and coil 21 must exert influence which is achieved by arranging a diode 23 parallel to coil 22. Furthermore the control circuit of transistor 16 in this example comprises the two diodes 24 and 25 as described in U.S. application Ser. No. 26,497 above referred to, wherein one of these diodes, diode 25 in FIG. 1, must be shunted by a resistor.

The control circuit of transistor 16 may alternatively be formed as is shown in FIG. 4. In fact, it is known that coil 21 may be replaced by the parallel arrangement of a diode 21' and a resistor 21" by which the inverse current can be limited. To separate the path of the inverse current from that of the forward current the parallel arrangement of a the diode 29' and a resistor 29" must then be present. This leads to the circuit arrangement shown in the upper part of FIG. 4. This circuit arrangement may now be simplified if it is noted that diodes 25 and 21' on the one hand and diodes 23 and 29' on the other hand are series-arranged. The result is shown in the lower part of FIG. 4 which, as compared with the circuit arrangement of FIG. 1, employs one coil less and an additional resistor.

FIG. 3 shows possible modifications of the chopper. FIG. 3a shown in a simplified form the circuit arrangement according to FIG. 1 wherein the pulsatory voltage present across the connections of windings 8 has a peak-to-peak amplitude of V i - V o = 0.5 V i for δ = 0.5, As has been stated, the provision of coil 22 gives a relative variation for the base current of transistor 16 which is equal to that of the mains voltage. In the cases according to FIG. 3b, 3c and 3d the peak-to-peak amplitude of the voltage across winding 8 is equal to V i so that the provision of coil 22 results in a relative variation which is equal to half that of the mains voltage which is still more favorable than in the first case.

Transistors of the npn type are used in FIG. 3. If transistors of the pnp type are used, the relevant efficiency diodes must of course be reversed.

In this connection it is to be noted that it is possible to obtain an output voltage V o with the aid of the modifications according to FIGS. 3b, c and d, which voltage is higher than input voltage V i . These modifications may be used in countries such as, for example, the United of America or France where the nominal mains voltage is 117 or 110 V without having to modify the rest of the circuit arrangement.

The above-mentioned remark regarding the sum of the diode currents only applies, however, for the modifications shown in FIGS. 3a and d.

If line output transistor 16 is not simultaneously driven with switching transistor 2, efficiency diodes 7 conducts simultaneously with transistor 16 i.e., during the period which is denoted by δ T in FIGS. 1 and 2b. During that period the output voltage V o of the chopper is stabilized so that the base current of transistor 16 is stabilized without further difficulty. However, a considerable drawback occurs. In FIG. 1 the reference numeral 26 denotes a safety circuit the purpose of which is to safeguard switching transistor 2 when the current supplied to load 11 and/or line deflection circuit 17 becomes to high, which happens because the chopper stops. After a given period output voltage V o is built up again, but gradually which means that the ratio δ is initially small in the order of 0.1. All this is described in U.S. patent No. 3,629,686. The same phenomenon occurs when the display device is switched on. Since δ = 0.1 corresponds to approximately 6 μs when T = 64 μs, efficiency diode 7 conducts in that case for 64 - 6 = 58 μus so that transistor 16 is already switched on at the end of the scan or at a slightly greater ratio δ during the flyback. This would cause an inadmissibly high dissipation. For this reason the simultaneous drive is therefore to be preferred.

The line deflection circuit itself is also safeguarded: in fact, if something goes wrong in the supply, the driver voltage of the line deflection circuit drops out because the switching voltage across the terminals of primary winding 8 is no longer present so that the deflection stops. This particularly happens when switching transistor 2 starts to constitute a short-circuit between emitter and collector with the result that the supply voltage V o for the line deflection circuit in the case of FIG. 1 becomes higher, namely equal to V i . However, the line output transformer is now cut off and is therefore also safe as well as the picture display tube and other parts of the display device which are fed by terminal 15 or the like. However, this only applies to the circuit arrangement according to FIG. 1 or 3a.

Pulse oscillator 6 applies pulses of line frequency to modulator 5. It may be advantageous to have two line frequency generators as already described, to wit pulse oscillator 6 and line oscillator 6' which is present in the picture display device and which is directly synchronized in known manner by line synchronizing pulses 7'. In fact, in this case line oscillator 6' applies a signal of great amplitude and free from interference to pulse oscillator 6. However, it is alternatively possible to combine pulse oscillator 6 and line oscillator 6' in one single oscillator 6" (see FIG. 1) which results in an economy of components. It will be evident that line oscillator 6' and oscillator 6" may alternatively be synchronized indirectly, for example, by means of a phase discriminator. It is to be noted neither pulse oscillator 6, line oscillator 6' and oscillator 6" nor modulator 5 can be fed by the supply described since output voltage V o is still not present when the mains voltage is switched on. Said circuit arrangements must therefore be fed directly from the input terminals. If as described above these circuit arrangements are to be separated from the mains, a small separation transformer can be used whose primary winding is connected between the mains voltage terminals and whose secondary winding is connected to ground at one end and controls a rectifier at the other end.

Capacitor 27 is arranged parallel to efficiency diode 7 so as to reduce the dissipation in switching transistor 2. In fact, if transistor 2 is switched off by the pulsatory control voltage, its collector current decreases and its collector-emitter voltage increases simultaneously so that the dissipated power is not negligible before the collector current has becomes zero. If efficiency diode 7 is shunted by capacitor 27 the increase of the collector-emitter voltage is delayed i.e., this voltage does not assume high values until the collector current has already been reduced. It is true that in that case the dissipation in transistor 2 slightly increases when it is switched on by the pulsatory control voltage but on the other hand since the current flowing through diode 7 has decreased due to the presence of the secondary windings, its inverse current is also reduced when transistor 2 is switched on and hence its dissipation has become smaller. In addition it is advantageous to delay these switching-on and switching-off periods to a slight extent because the switching pulses then contain fewer Fourier components of high frequency which may cause interferences in the picture display device and which may give rise to visible interferences on the screen of the display tube. These interferences occupy a fixed position on the displayed image because the switching frequency is the line frequency which is less disturbing to the viewer. In a practical circuit wherein the line frequency is 15,625 Hz and wherein switching transistor 2 is an experimental type suitable for a maximum of 350 V collector-emitter voltage or 1 A collector current and wherein efficiency diode 7 is of the Philips type BA 148 the capacitance of capacitor 27 is approximately 680 pF whilst the load is 70 W on the primary and 20 W on the secondary side of transformer 9. The collector dissipation upon switching off is 0.3 W (2.5 times smaller than without capacitor 27) and 0.7 W upon switching on.

As is known the so-called pincushion distortion is produced in the picture display tubes having a substantially flat screen and large deflection angles which are currently used. This distortion is especially a problem in color television wherein a raster correction cannot be brought about by magnetic means. The correction of the so-called East-West pincushion distortion i.e., in the horizontal direction on the screen of the picture display tube can be established in an elegant manner with the aid of the circuit arrangement according to the invention. In fact, if the voltage generated by comparison circuit 12 and being applied to modulator 5 for duration-modulating pulsatory voltage 3 is modulated by a parabola voltage 28 of field frequency, pulsatory voltage 3 is also modulated thereby. If the power consumption of the line deflection circuit forms part of the load on the output voltage of the chopper, the signal applied to the line deflection coils is likewise modulated in the same manner. Conditions therefore are that the parabola voltage 28 of field frequency has a polarity such that the envelope of the sawtooth current of line frequency flowing through the line deflection coils has a maximum in the middle of the scan of the field period and that charge capacitor 10 has not too small an impedance for the field frequency. On the other hand the other supply voltages which are generated by the circuit arrangement according to the invention and which might be hampered by this component of field frequency must be smoothed satisfactorily.

A practical embodiment of the described example with the reference numerals given provides an output for the supply of approximately 85 percent at a total load of 90 W, the internal resistance for direct current loads being 1.5 ohms and for pulsatory currents being approximately 10 ohms. In case of a variation of ± 10 percent of the mains voltage, output voltage V o is stable within 0.4 V. Under the nominal circumstances the collector dissipation of switching transistor 2 is approximately 2.5 W.

Since the internal resistance of the supply is so small, it can be used advantageously, for example, at terminal 15 for supplying a class-B audio amplifier which forms part of the display device. Such an amplifier has the known advantages that its dissipation is directly proportional to the amplitude of the sound to be reproduced and that its output is higher than that of a class-A amplifier. On the other hand a class-A amplifier consumes a substantially constant power so that the internal resistance of the supply voltage source is of little importance. However, if this source is highly resistive, the supply voltage is modulated in the case of a class-B amplifier by the audio information when the sound intensity is great which may detrimentally influence other parts of the display device. This drawback is prevented by means of the supply according to the invention.

The 50 Hz ripple voltage which is superimposed on the rectified input voltage V i is compensated by comparison circuit 12 and modulator 5 since this ripple voltage may be considered to be a variation of input voltage V i . A further compensation is obtained by applying a portion of this ripple voltage with suitable polarity to comparison circuit 12. It is then sufficient to have a lower value for the smoothing capacitor which forms part of rectifier circuit 1 (see FIG. 3). The parabola voltage 28 of field frequency originating from the field time base is applied to the same circuit 12 so as to correct the East-West pincushion distortion.





ITT DIGIVISION CHIPSET FUNCTIONS.Digivision ITT DIGIT 2000 VLSI Digital TV System:

ITT DIGIT 2000 VLSI Digital TV System Set of three integrated circuits for digital video signal processing in color-television receivers: ITT DIGIVISION.

ITT DIGIVISION 3486 OSCAR CHASSIS DIGI 3 90° FST / IFB-285/2
Digivision ITT DIGIT 2000 VLSI Digital TV System DIGITAL CRT TUBE Cathode RAY CURRENT CONTROL / Cut OFF / Drive and processing.
In this IC set, the dark currents and the white levels of the three electron guns, the leakage currents of the cathodes, and a light-detector current are measured during four successive vertical blanking intervals. The cathode leakage currents and the dark currents are measured in the first half of the vertical blanking interval, and the light-detector current and the white level currents are measured at the end of this interval. From these measured data and alignment data stored in a reprogrammable memory (ps), a microprocessor (mp) contained together with the memory (ps) in an integrated circuit (ic2) derives operating data for the picture tube (b) as well as further data. These operating data are transferred over a wire of a chroma bus (cb), over which chroma signals are transferred during the vertical sweep, into a shift register (sr) of a further integrated circuit (ic3) at the beginning of each vertical blanking interval, from where they are passed on to the picture tube (b) in groups via digital-to-analog converters and analog amplifiers. By the use of the chroma bus for a dual purpose, and the successive measurements of the above-mentioned picture-tube data, a saving of external terminals of the integrated circuits (ic1, ic2, ic3) is achieved.

1. Set of three integrated circuits(ic1, ic2, ic3) for digital video-signal processing in color-television receivers,
wherein the first integrated circuit (ic1) contains an analog-to-digital converter (ad) followed by a first bus interface circuit (if1) for a serial data bus (sb), and a first multiplexer (mx1) following the first bus interface circuit (if1), the analog-to-digital converter (ad) being fed with measured data corresponding to the cathode currents of the picture tube (b) flowing at "black" (="dark current") and "white" (="white level") in each of the three electron guns, and with the signal of an ambient-light detector (ls) via a second multiplexer (mx) in the vertical blanking interval, and the first multiplexer (mx1) being fed with the processed digital chrominance signals (cs),
wherein the second integrated circuit (ic2) contains a microprocessor (mp), an electrically reprogrammable memory (ps), and a second serial-data-bus interface circuit (if2) corresponding to the first bus interface circuit (if1), the memory (ps) holding alignment data and nominal dark-current/white-level data of the picture tube used (b) which were entered by the manufacturer of the color-television receiver and, together with the measured data, are used by the microprocessor (mp) to generate video-signal-independent operating data for the picture tube (b), and
wherein the third integrated circuit (ic3) contains a demultiplexer (dx), an analog RGB matrix (m), and three analog amplifiers (vr, vg, vb) each designed to drive one of the electron guns via an external video output stage (ve), the dark current of the picture tube (b) being adjusted via the operating point of the respective analog amplifier, and the white level of the picture tube (b) being adjusted by adjusting the gain of the respective amplifier after digital-to-analog conversion, and with the demultiplexer (dx) connected to the first multiplexer (mx1)of the first integrated circuit (ic1) via a chroma bus (cb),
Characterized by the Following Features:
The first multiplexer (mx1) consists of three electronic switches (s1, s2, s3),
the first of which (s1) has its input grounded through a first resistor (r1) and connected to the collectors of external transistors (tr, tg, tb) which are each associated with one of the electron guns and the base of each of which is driven by the associated video output stage, while the emitter is connected to the associated electron gun system, and the output of the first switch (s1) is connected to the input of the analog-to-digital converter (ad);
the second of which (s2) has its input connected to the light detector (ls), while its output is coupled with the input of the analog-to-digital converter (ad), and
the third of which (s3) has its input connected to the input of the first electronic switch (s1) via a second resistor (r2), and its output is grounded, the value of the second resistor (r2) being about one order of magnitude smaller than that of the first resistor (r1);
the three electronic switches (s1, s2, s3) have the following positions:
______________________________________
s1 s2 s3
______________________________________


during vertical

closed open closed

sweep

during vertical

closed/open

open/closed

open/closed

retrace: for

leakage/light-

det. current meas.

for white level

closed open closed

measurement

for dark current

closed open open

measurement

______________________________________
the measurements of the dark current together with the white level of each electron gun and the measurements of the light-detector current together with the cathode leakage currents are performed in four successive vertical blanking intervals;
to this end, the cathodes are connected at one end to a voltage for blacker than black (us), and at the other end to a voltage for black (ud) and then to a voltage for white (uw) in accordance with the following table:
______________________________________
Measurement in the first at about the Vertical half of the end of the blanking vertical vertical interval blanking blanking Cathode No. interval interval red green blue
______________________________________


1 Leakage cur-

Light-detect-

us us us

rents of the

or current

cathodes

2 Dark current

White level

ud/uw us us

red red

3 Dark current

White level

us ud/uw us

green green

4 Dark current

White level

us us ud/uw

blue blue

______________________________________
the measured data are transferred from the analog-to-digital converter (ad) to the microprocessor (mp) of the second integrated circuit (ic2) via the two interface circuits (if1, if2) and the data bus (sb) at an appropriate instant, and
the video-signal-independent operating data for the picture tube (b), which are generated by the microprocessor (mp), are transferred from the second integrated circuit (ic2) via the two interface circuits (if1, if2) and a line (db) to the first multiplexer (mx1) of the first integrated circuit (ic1) at an appropriate instant, and from there over a wire of the chroma bus (cb) into a shift register (sr) of the third integrated circuit (ic3) shortly after the beginning of the next vertical blanking interval, the parallel outputs of which shift register (sr) are combined in groups each assigned to one type of operating value, and each of the groups is connected to one digital-to-analog converter (dh, ddr, ddg, ddd, dwr, dwg, dwb) which drives the RGB matrix (m) or the respective analog amplifier (vr, vg, vb).
. 2. An integrated-circuit set as claimed in claim 1, characterized in that the voltage for blacker than black (us) is applied to the cathodes of the picture tube (b) during the data transfer to the shift register (sr). 3. An integrated-circuit set as claimed in claim 2, characterized in that the microprocessor (mp) determines the appropriate instant for the measured-data transfer, and that, if a measurement has not yet been finished at that instant, the measured data of the corresponding earlier measurement are transferred. 4. An integrated-circuit set as claimed in claim 3, characterized in that the measurement performed in a vertical blanking interval is not enabled until the data of the preceding measurement have been transferred to the microprocessor (mp). 5. An integrated-circuit set as claimed in claim 2, characterized in that the measurement performed in a vertical blanking interval is not enabled until the data of the preceding measurement have been transferred to the microprocessor (mp). 6. An integrated-circuit set as claimed in claim 1, characterized in that the microprocessor (mp) determines the appropriate instant for the measured-data transfer, and that, if a measurement has not yet been finished at that instant, the measured data of the corresponding earlier measurement are transferred. 7. An integrated-circuit set as claimed in claim 6, characterized in that the measurement performed in a vertical blanking interval is not enabled until the data of the preceding measurement have been transferred to the microprocessor (mp). 8. An integrated-circuit set as claimed in claim 1, characterized in that the measurement performed in a vertical blanking interval is not enabled until the data of the preceding measurement have been transferred to the microprocessor (mp).
Description:
The present invention relates to a set of three integrated circuits for digital video signal processing in color-television receivers as is set forth in the preamble of claim 1. An IC set of this kind is described in a publication by INTERMETALL entitled "Eine neue Dimension-VLSI-Digital-TV-System", Freiburg im Breisgau, September 1981, on pages 6 to 11 (see also the corresponding English edition entitled "A new dimension-VLSI Digital TV System", also dated September 1981).
The first integrated circuit, designated in the above-mentioned publications by "MAA 2200" and called "Video Processor Unit" (VPU), includes an analog-to-digital converter followed by a first serial-data-bus interface circuit which, in turn, is followed by a first multiplexer. During the vertical blanking interval, the analog-to-digital converter is fed, via a second multiplexer, with measured data corresponding to the cathode currents of the picture tube flowing at "black" (="dark current") and "white" ("white level") in each of the three electron guns, and with the signal of an ambient-light detector. The processed digital chrominance signals are applied to the first multiplexer.
The second integrated circuit, designated by "MAA 2000" and called "central control unit" (CCU) in the above publications, contains a microprocessor, an electrically reprogrammable memory, and a second serial-data-bus interface circuit. The memory holds alignment data and nominal dark-current/white-level data entered by the manufacturer of the color-television receiver. From these data and the measured data, the microprocessor derives video-signal-independent operating data for the picture tube.
The third integrated circuit, designated by "MAA 2100" and called "video-codec unit" (VCU) in the above publications, includes a demultiplexer, an analog RGB matrix, and three analog amplifiers each designed to drive one of the electron guns via an external video output stage. After digital-to-analog conversion, the dark current of the picture tube is adjusted via the operating point of the respective analog amplifier, and the white level of the picture tube is adjusted by adjusting the gain of the respective analog amplifier. The demultiplexer is connected to the first multiplexer of the first integrated circuit via a chroma bus.
As to the prior art concerning such digital color-television receiver systems, reference is also made to the journal "Elektronik", Aug. 14, 1981 (No. 16), pages 27 to 35, and the journal "Electronics", Aug. 11, 1981, pages 97 to 103.
During the further development of the prior art system following the above-mentioned publication dates, the developers were faced with the problem of how to accomplish the dark-current/white-level control of the picture tube within the existing system, particularly with respect to measured-data acquisition and transfer and to the transfer of the operating data to the picture tube.
Another requirement imposed during the further development of the prior art system was that the leakage currents of the electron guns of the picture tube be measured and processed within the existing system. The solution of these problems is to take into account the requirement that the number of external terminals of the individual integrated circuits be kept to a minimum.
The object of the invention as claimed is to solve the problems pointed out. The essential principles of the solution, which directly give the advantages of the invention, are, on the one hand, the division of the measurement to four successive vertical blanking intervals and, on the other hand, the utilization of one wire of the chroma bus at the beginning of the next vertical blanking interval as well as the measurement of the ambient light by means of the light detector and the measurement of the leakage currents during a single vertical blanking interval.
The invention will now be explained in more detail with reference to the accompanying drawing, which is a block diagram of one embodiment of the IC set in accordance with the invention. It shows the first, second, and third integrated circuits ic1, ic2, and ic3, which are drawn as rectangles bordered by heavy lines. The first integrated circuit ic1 includes the analog-to-digital converter ad, which converts the measured dark-current, white-level, ambient-light, and leakage-current data into digital signals, which are fed to the first bus interface circuit if1. The latter is connected via the line db to the first multiplexer mx1, which interleaves data from the first bus interface circuit if1 with digital chrominance signals cs produced in the first integrated circuit ic1, and places the interleaved signals on the chroma bus cb. The generation of the digital chrominance signals cs is outside the scope of the present invention and is disclosed in the references cited above.
The first integrated circuit ic1 further includes the second multiplexer mx2, which consists of the three electronic switches s1, s2, s3, and represents a subcircuit which is essential for the invention. The input of the first switch s1 is grounded through the first resistor r1, and connected to the collectors of the external transistors tr, tg, tb, each of which is associated with one of the electron guns. Via the base-emitter paths of these transistors, the cathodes of the three electron guns are driven by the video output stages ve. The final letters r, g, and b in the reference characters tr, tg, and tb and in the reference characters explained later indicate the assignment to the electron gun for RED (r), GREEN (g), and BLUE (b), respectively. The output of the first switch s1 is connected to the input of the analog-to-digital converter ad.
The input of the second switch s2 is connected to the light detector ls, which has its other terminal connected to a fixed voltage u and combines with the grounded resistor r3 to form a voltage divider. The input of the second switch s2 is thus connected to the tap of this voltage divider, while the output of this switch, too, is coupled to the input of the analog-to-digital converter ad.
The input of the third switch s3 is connected to the input of the first switch s1 via the second resistor r2, while the output of the third switch s3 is grounded. The value of the resistor r1 is about one order of magnitude greater than that of the resistor r2.
For the whole duration of the picture shown on the screen of the picture tube b, and throughout the vertical sweep, the first switch s1 and the third switch s3 are closed, and the second switch s2 is open. During the vertical retrace interval, for the white-level measurement, the switches s1, s3 are closed, and the switch s2 is open; for the dark-current measurement and the leakage-current measurement, the switch s1 is closed, and the switches s2, s3 are open, and for the light-detector-current measurement, the switches s2, s3 are closed, and the switch s1 is open.
The measurements of the dark current and the white level of each electron gun and the measurements of the light-detector current and the leakage currents are made in four successive vertical blanking intervals. One end of the respective cathode is connected to a voltage us for blacker-than-black, and the other end is connected to a voltage ud for black and then to a voltage uw for white, in accordance with the following table:
______________________________________
Measurement in the first at about the Vertical half of the end of the blanking vertical vertical interval blanking blanking Cathode No. interval interval red green blue
______________________________________


1 Leakage cur-

Light-detect-

us us us

rents of the

or current

cathodes

2 Dark current

White level

ud/uw us us

red red

3 Dark current

White level

us ud/uw us

green green

4 Dark current

White level

us us ud/uw

blue blue

______________________________________
The voltage ud for black is, as usual, a voltage which just causes no brightness on the screen of the picture tube b, i.e., a voltage just below the dark threshold of the picture tube. The voltage us for blacker-than-block is then a cathode voltage lying further in the black direction than the voltage for black. The voltage for white is the voltage for the screen brightness to be measured; the brightness of the screen is generally below the maximum permissible value.
Thus, two measurements are made during each vertical blanking interval, namely one in the first half, preferably at one-third of the pulse duration of the vertical blanking interval, and the other at about the end of the first half. During the four successive vertical blanking intervals, the first measurement determines the leakage currents of the cathodes and the dark currents for red, green, and blue. The second measurements determine the light-detector current and the white levels for red, green, and blue. During the measurement of the cathode leakage currents and the light-detector current, all three cathodes are at the voltage us. During the measurements of the dark current and the white level of the respective cathode, the latter is connected to the respective dark-current cathode voltage ud and white-level cathode voltage uw, respectively, while the cathodes of the two other electron guns, which are not being measured, are at the voltage us.
The second integrated circuit circuit ic2 contains the microprocessor mp, the electrically reprogrammable memory ps, and the second bus interface circuit if2, which is associated with the serial data bus sb in this integrated circuit and also connects the microprocessor mp and the memory ps with one another and with itself. The memory ps holds alignment data and nominal dark-current/white-value data of the picture tube used, which were entered by the manufacturer. From this alignment and nominal data and from the measured data obtained via the second multiplexer mx2 and the analog-to-digital converter ad of the first integrated circuit ic1, the microprocessor mp derives video-signal-independent operating data for the picture tube.
The derivation of these operating data is also outside the scope of the invention; it should only be mentioned that with respect to the operating data of the picture tube, the microprocessor performs a control function in accordance with a predetermined control characteristic.
The third integrated circuit ic3 includes the demultiplexer dx, which is connected to the first multiplexer mx1 of the first integrated circuit ic1 via the chroma bus cb and separates the chrominance signals cs and the operating data of the picture tube from the interleaved signals transferred over the chroma bus. While the transfer of measured data from the analog-to-digital converter ad to the microprocessor mp of the second integrated circuit ic2 takes place via the two interface circuits if1, if2 and the data bus sb at an appropriate instant, the video-signal-independent operating data for the picture tube b, which are derived by the microprocessor mp, are transferred from the second integrated circuit ic2 via the two interface circuits if1, if2 and the line db to the first multiplexer mx1 at an appropriate instant, and from the first multiplexer mx1 over a wire of the chroma bus cb into the shift register sr of the third integrated circuit ic3 shortly after the beginning of the next vertical blanking interval. To accomplish this, the first interface circuit if1 also includes a shift register from which the operating data are read serially.
During this data transfer into the shift register sr, the cathodes of the picture tube b are preferably at the voltage us in order that this data transfer does not become visible on the screen.
The appropriate instant for the transfer of measured data to the microprocessor mp is determined by the latter itself, i.e., depending on the program being executed in the microprocessor, and on the time needed therefor, the measured data are called for from the interface circuits not at the time of measurement but at a selectable instant within the working program of the microprocessor mp. If the measurement currently being performed should not yet be finished at the instant at which the measured data are called for, in a preferred embodiment of the invention, the stored data of the previous measurement will be transferred to the microprocessor mp.
As mentioned previously, the operating data for the picture tube b are transferred into the shift register sr at the beginning of a vertical blanking interval. The parallel outputs of this shift register are combined in groups each assigned to one operating value, and each group has one of the digital-to-analog converters dh, ddr, ddg, ddb, dwr, dwg, dwb associated with it. In the figure, the division of the shift register into groups is indicated by broken lines. The shift register sr performs a serial-to-parallel conversion in the usual manner, and the operating data are entered by the demultiplexer dx into the shift register in serial form and are then available at the parallel outputs of the shift register.
The digital-to-analog converter dh provides the analog brightness control signal, which is applied to the RGB matrix m in the integrated circuit ic3. Also applied to the RGB matrix m are the analog color-difference signals r-y, b-y and the luminance signal y. The formation of these signals is outside the scope of the invention and is known per se from the publications cited at the beginning.
The three analog-to-digital converters ddr, ddg, ddb provide the dark-current-adjusting signals for the three cathodes, which are currents and are applied to the inverting inputs--of the analog amplifiers vr, vg, vb. Also connected to these inputs is a resistor network which is adjustable in steps in response to the digital white-level-adjusting signals at the respective group outputs of the shift register sr. The resistors serve as digital-to-analog converters dwr, dwg, dwb and establish the connection between the inverting inputs--and the outputs of the analog amplifiers vr, vg, vb.
In an arrangement according to the invention which has proved good in practice, each of the three dark-current-adjusting signals is a seven-digit signal, and each of the three white-level-adjusting signals and the brightness control signal are five-digit signals. The voltages us and ud/uw of the three cathodes are assigned a three-digit identification signal in accordance with the above table, which signal is also fed into the shift register sr in the implemented circuit. Finally, a three-digit contrast control signal is provided in the implemented circuit for the Teletext mode of the color-television receiver. These nine data blocks are transferred in the implemented circuit from the demultiplexer dx to the shift register sr in the following order, with the least significant bit transmitted first, and with the specified number of blanks: identification signal, white-level signal blue, three blanks, white-level signal green, three blanks, white-level signal red, one blank, dark-current signal blue, one blank, dark-current signal green, one blank, dark-current signal red, contrast signal Teletext, and brightness control signal. These are seven eight-digit data blocks which are assigned to 56 pulses of a 4.4-MHz clock frequency, which is the frequency of the shift clock signal of the shift register sr.
It should be noted that the data sequence just described does not correspond to the order of the groups of the shift register sr in the figure. The order in the figure was chosen only for the sake of clarity.
The outputs of the three analog amplifiers vr, vg, vb are coupled to the inputs of the video output stage ve, whose outputs, as explained previously, are connected to the bases of the transistors pr, tg, td, so that the cathodes of the picture tube b are driven via the base-emitter paths of these transistors.
In another preferred embodiment of the invention, the measurement performed during a vertical blanking interval is not enabled until the data of the previous measurement has been transferred into the microprocessor mp. In this manner, no measurement will be left out.
It is also possible to omit the digital-to-analog converter dh if the analog RGB matrix m is replaced with a digital one.
One advantage of the invention is that the use of the chroma bus for the transfer of operating data facilitates the implementation of the third integrated circuit ic3 using bipolar technology, because an additional bus interface circuit, which could be used there, would occupy too much chip area.

ITT DIGIVISION 3486 OSCAR CHASSIS DIGI 3 90° FST / IFB-285/2 Digivision ITT DIGIT 2000 VLSI Digital TV System
Color-television receiver having integrated circuit for the luminance signal and the chrominance signals:

VIDEO CODEC UNIT (VCU).



The invention permits an n bit resolution to be achieved with an n-1 bit converter. In a color television receiver the analog-to-digital converter is a parallel analog-to-digital converter with p=2r -1 differential amplifiers as comparators, where r is the number of binary digits of the output signal of the analog-to-digital converter minus one. The composite color signal is then applied as the input signal to the noninverting (or inverting) inputs of all p differential amplifiers and the inverting (noninverting) inputs of the differential amplifiers being connected successively to the taps of a resistive voltage divider which contains equal-value resistors and is fed with a reference voltage (Ur).
For the duration of every second line, either the reference voltage or the input signal is shifted by ΔU=0.5 Ur/2r.
1. A color-television receiver comprising at least one integrated circuit for separating and conditioning the luminance signal and the chrominance signals from the composite color signal, said integrated circuit containing:
a chrominance-subcarrier oscillator,
a chrominance-subcarrier band-pass filter,
a synchronous demodulator,
a PAL switch,
a color matrix, and, if necessary,
an R--G--B matrix, and being characterized by the following subcircuits for conditioning digital signals:
the chrominance-subcarrier oscillator is a squarewave clock generator providing four clock signals the first of which has four times the chrominance-subcarrier frequency and the second to fourth of which have the chrominance-subcarrier frequency, with the first and second clock signals having a pulse duty factor of 0.5, and the third and fourth clock signals each consisting of two consecutive, T/2-long pulses separated by T/2 within each 4T-long period (T=period of the first clock signal);
an analog-to-digital converter clocked by the first clock signal, whose analog input is presented with the composite color signal, and which forms as its output signal a parallel binary word from the amplitude of the composite color signal (F) at the instants the respective amplitudes of the undemodulated chrominance signal are equal to the amplitudes of the respective color-difference signal;
a first binary arithmetic stage which multiplies the output signal of the analog-to-digital converter by a binary overall-contrast control signal;
a two-stage delay line which delays the output signal of the first binary arithmetic stage by T/2;
a second binary arithmetic stage which forms the arithmetic mean of the delayed and undelayed output signals of the first binary arithmetic stage;
a third binary arithmetic stage which subtracts the output signal of the second binary arithmetic stage from the output signal of the first delay stage;
a buffer-memory arrangement which temporarily stores the output signal of the third binary arithmetic stage, and whose enable input is fed with the third clock signal;
a shift-register arrangement consisting of n parallel shift registers (n=number of bits at the output of the third binary arithmetic stage) each of which provides a delay of one line period and whose serial inputs are connected to the parallel outputs of the buffer-memory arrangement, while their clock inputs are fed with the fourth clock signal;
a fourth binary arithmetic stage which forms the arithmetic mean of the input and output signals of the shift-register arrangement;
a fifth binary arithmetic stage which subtracts the input signal of the shift-register arrangement from the output signal of this arrangement and then divides the difference by two;
a sixth binary arithmetic stage which, controlled by the PAL switch, either leaves the output signal of the fifth binary arithmetic stage unchanged or forms its absolute value;
a seventh binary arithmetic stage which forms the green color-difference signal from the output signals of the fourth and sixth binary arithmetic stages;
the outputs of the second, fourth, sixth and seventh binary arithmetic stages are connected to the binary R-G-B matrix each of whose outputs is coupled to one of three digital-to-analog converters for deriving the analog signals for controlling the R-G-B values of the picture tube, or
the outputs of the second, fourth, sixth and seventh binary arithmetic stages are each connected to one of four digital-to-analog converters for deriving the analog signals for controlling the color-difference value of the picture tube;
the improvement wherein
the analog-to-digital converter is a parallel analog-to-digital converter with p=2r -1 differential amplifiers as comparators, where r is the number of binary digits of the output signal of the analog-to-digital converter minus one, the composite color signal being applied as the input signal to one of the noninverting or inverting inputs of all p differential amplifiers and the other of the inverting or noninverting inputs of the differential amplifiers being connected successively to the taps of a resistive voltage divider which contains equal-value resistors and is fed with a reference voltage (Ur), and
for the duration of every second line, either the reference voltage (Ur) or the input signal (F) is shifted by ΔU=0.5 Ur/2r.
Description:
FIELD OF THE INVENTION
Color-television receivers comprising at least one integrated circuit for separating and conditioning the luminance signal and the chrominance signals from the composite color signal are known in the art. The particular color-television receiver of such a known type comprises at least one integrated circuit for separating and conditioning the luminance signal and the chrominance signals from the composite color signal. This integrated circuit contains a chrominance-subcarrier oscillator, a chrominance-subcarrier band-pass filter, a synchronous demodulator, a PAL switch, a color matrix, and, if necessary, an R-G-B matrix. Additionally, such a color-television receiver contains the following subcircuits for conditioning digital signals; (1) the chrominance-subcarrier oscillator is a square-wave clock generator providing four clock signals the first of which has four times the chrominance-subcarrier frequency and the second to fourth of which have the chrominance-subcarrier frequency, with the first and second clock signals having a pulse duty factor of 0.5, and the third and fourth clock signals each consisting of two consecutive, T/2-long pulses separated by T/2 within each 4T-long period (T=period of the first clock signal); (2) an analog-to-digital converter clocked by the first clock signal, whose analog input is presented with the composite color signal, and which forms as its output signal a parallel binary word from the amplitude of the composite color signal (F) at the instants the respective amplitudes of the undemodulated chrominance signal are equal to the amplitudes of the respective color-difference signal; (3) a first binary arithmetic stage which multiplies the output signal of the analog-to-digital converter by a binary overall-contrast control signal; (4) a two stage delay line which delays the output signal of the first binary arithmetic stage by T/2; (5) a second binary arithmetic stage which forms the arithmetic means of the delayed and undelayed output signals of the first binary arithmetic stage; (6) a third binary arithmetic stage, which subtracts the output signal of the second binary arithmetic stage from the output signal of the first delay stage; (7) a buffer-memory arrangement which temporarily stores the output signal of the third binary arithmetic stage, and whose enable input is fed with the third clock signal; (8) a shift-register arrangement consisting of n parallel shift registers (n=number of bits at the output of the third binary arithmetic stage) each of which provides a delay of one line period and whose serial inputs are connected to the parallel outputs of the buffer-memory arrangement, while their clock inputs are fed with the fourth clock signal; (9) a fourth binary arithmetic stage which forms the arithmetic mean of the input and output signals of the shift-register arrangement; (10) a fifth binary arithmetic stage which subtracts the input signal of the shift-register arrangement from the output signal of this arrangement and then divides the difference by two; (11) a sixth binary arithmetic stage which, controlled by the PAL switch, either leaves the output signal of the fifth binary arithmetic stage unchanged or forms its absolute value; (12) a seventh binary arithmetic stage which forms the green color-difference signal from the output signals of the fourth and sixth binary arithmetic stages; (13) the outputs of the second, fourth, sixth and seventh binary arithmetic stages are connected to the binary R-G-B matrix each of whose outputs is coupled to one of three digital-to-analog converters for deriving the analog signals for controlling the R-G-B values of the picture tube, or (14) the outputs of the second, fourth, sixth and seventh binary arithmetic stages are each connected to one of four digital-to-analog converters for deriving the analog signals for controlling the color-difference values of the picture tube. An essential feature of such a receiver is the use of an analog-to-digital converter whose analog input is presented with the composite color signal and which is clocked by a clock signal at four times the chrominance-subcarrier frequency, so that a parallel binary word is obtained from the amplitudes of the composite color signal at the instants the respective amplitudes of the undemodulated chrominance signal are equal to the amplitudes of the respective color-difference signal.
Thus, because of the high frequencies to be be processed, a parallel analog-to-digital converter is needed. Such fast parallel analog-to-digital converters are well known (cf. D. F. Hoeschele, "Analog-to-Digital/Digital-to-Analog Conversion Techniques", New York, 1968, p. 10) and contain 2 s -1 differential amplifiers as comparators, where s is the number of binary digits of the digital converter output signal. The noninverting (or inverting) inputs of all differential amplifiers are presented with the composite color signal, while the inverting (or noninverting) inputs are connected successively to the taps of a resistive voltage divider inserted between a constant reference voltage and ground and consisting of 2 s or 2 s -1 equal-value resistors.
A 6-bit parallel analog-to-digital converter thus has 63 comparators and 63 resistors. A 7-bit converter has 127 comparators and resistors, and an 8-bit converter even has 255 comparators and resistors. It is readily apparent that as the number of digits increases, the implementation of such converters using integrated circuit techniques quickly becomes uneconomical. In particular, a reduction by one digit would result in the component count being halved.
SUMMARY OF THE INVENTION
Accordingly, the object of the invention is to reduce the number of comparators and resistors in an arrangement as set forth hereinbefore to one half without adversely affecting the digital resolution. In other words, the invention is to permit a 6-bit resolution, for example, to be achieved with a 5-bit converter. This is done by using the means set forth above recourse being had to the principle described in the above-cited book on pp. 413 to 415 as follows: In color-television receiver described above, the analog-to-digital converter is a parallel analog-to-digital converter with p=2 r -1 differential amplifiers as comparators, where r is the number of binary digits of the output signal of the analog-to-digital converter minus one. The composite color signal is then applied as the input signal to the noninverting (or inverting) inputs of all p differential amplifiers and the inverting (noninverting) inputs of the differential amplifiers being connected successively to the taps of a resistive voltage divider which contains equal-value resistors and is fed with a reference voltage (Ur). For the duration of every second line, either the reference voltage or the input signal is shifted by ΔU=0.5 Ur/2 r .
BRIEF DESCRIPTION OF THE DRAWINGS
The invention will now be explained in more detail with reference to the accompanying drawings, in which:
FIG. 1 shows the block diagram of a color-television receiver of a known type.
FIGS. 2a-h, k, l, and p-t show various waveforms occurring in the arrangement of FIG. 1, and, in tabular form, signals occurring at given points of the circuit at given times, and
FIG. 3 is a block diagram of a preferred embodiment of the invention.
DESCRIPTION OF THE BEST MODE
At the outset, FIG. 1, will be explained to permit a better understanding of the invention.
In the block diagrams shown in FIGS. 1 to 3, like parts are designated by like reference characters. In addition to interconnections indicated by solid lines as is usual in circuit diagrams, these figures contain interconnections indicated by stripes. These stripes mark connections between digital parallel outputs of the delivering portion of the circuit and digital parallel inputs of the receiving portion. The interconnections indicated by stripes, therefore, consist of at least as many wires as there are bits in the binary word to be transferred. Thus, the signals transferred over the lines indicated by stripes in FIGS. 1 to 3 are all binary signals whose instantaneous binary value corresponds to the instantaneous analog value of the composite color signal and of other signals.
Like in conventional color-television receivers, the composite color signal F, derived in the usual manner controls the chrominance-subcarrier oscillator, which, according to the invention, is designed as a squarewave clock generator 1. By means of the so-called burst contained in the composite color signal F, the clock generator 1 is synchronized to the transmitted chrominance-subcarrier frequency. The clock generator 1 generates the clock signal F1, whose frequency is four times the chrominance-subcarrier frequency, i.e. about 17.73 MHz (precisely 17.734475 MHz) in the case of the CCIR standard.
The clock generator 1 also generates the square-wave clock signal F2 having the frequency of the chrominance subcarrier. The first and second clock signals F1, F2 have a pulse duty factor of 0.5 (cf. FIGS. 2a and 2b). In addition, the clock generator 1 generates the third clock signal F3 and the fourth clock signal F4, each of which consists of two consecutive, T/2-long pulses separated by T/2 within each 4T-long period, where T is the period of the first clock signal F1. The third and fourth clock signals F3, F4 are shown in FIGS. 2b and 2g.
The individual clock signals are generated within the clock generator 1 in the usual manner using conventional digital techniques. The clock signal F1, for instance, may be generated by means of a suitable 17.73--MHz crystal, and the clock signals F2, F3, F4 may be derived therefrom by frequency division and suitable elimination of pulses. Like in conventional color-television receivers, the clock generator 1 is also fed with a pulse Z from the horizontal output stage during which the clock generator 1 is sychronized by the burst.
The composite color signal F is also applied to the analog input of the analog-to-digital converter 2, which is clocked by the first clock signal F1 and, (at the beginning of each pulse of the first clock signal F1) forms from the amplitude of this pulse a parallel binary word and delivers it as an output signal. These leading edges of the pulses of the first clock signal F1 thus occur at the instants the respective amplitudes of the undemodulated chrominance signal contained in the composite color signal are equal to the amplitudes of the respective color-difference signal.
These parallel binary words then remain unchanged for the respective period T of the first clock signal F1, i.e., they are held like in a sample-and-hold circuit. The signals appearing at the output of the analog-to-digital converter 2 are given in tabular form in FIG. 2c, where the vertical lines symbolize the respective clock periods of the first clock signal F1. The letter c of FIG. 2 is also shown in FIG. 1 (encircled).
According to FIG. 2c, successive signals Y+V, Y-U, Y-V, and Y+U are obtained in a line m during one period of the second clock signal F2, where U, V and Y have the formal meanings given in the above-mentioned book, namely U=B-Y, V=R-Y, B=blue chrominance signal, R=red chrominance signal, and Y=luminance signal, but designate here the corresponding digitized signals, i.e., the corresponding binary words. The second line in the Table of FIG. 2c gives the corresponding binary signals in the line m+1, namely the signals Y-V, Y-U and Y+U, occurring during that period of the clock signal F2 which is under consideration.
This output signal of the analog-to-digital converter 2 is applied to one of the two inputs of the first binary arithmetic stage 10, which multiplies this output signal by a binary overall-contrast control signal GK. This overall contrast control signal thus corresponds to the analog overall-contrast control signal present in conventional color-television receivers. In present day color-television receivers, the binary overall contrast control signal GK, just as the binary color-saturation control signal FK and the binary brightness control signal H to be explained below, is available in digital form, because remote-control units and digital controls are usually present which provide these signals.
An advantage of the present application is, therefore, seen in the fact that these signals need no longer be conditioned in analog form in their place of action.
The output signal of the first binary arithmetic stage 10 is fed to the second binary arithmetic stage 20 and to the two-stage delay line 3, which delays this output signal by T/2. The second binary arithmetic stage 20 forms the arithmetic mean of the delayed and undelayed signals. The underlying idea is that if a sinusoidal signal, namely the chrominance subcarrier, is sampled at double frequency, the mean of two successive sample values will always be zero. Thus, by forming the arithmetic means in the second binary arithmetic stage 20, the chrominance subcarrier is suppressed and the luminance signal Y is obtained in digital form.
The output signal of the first binary arithmetic stage 10, delayed in the first stage 31 of the delay line 3 by half the delay provided by this stage, i.e., by T/4, and the output signal of the second binary arithmetic stage 20 are then fed to the third binary arithmetic stage 30, which subtracts the latter signal, i.e., the Y signal, from the former signal. As a result, the output of the third binary arithmetic stage 30 provides the color-difference signal, made up of the successive components B-Y, R-Y, -(B-Y) and -(R-Y), as shown in FIG. 2d in tabular form for the lines m and m+1.
These signals are fed to the buffer-memory arrangement 4, whose enable input is fed with the third clock signal F3, which is shown in FIG. 2e. This buffer memory operates in such a manner that the binary word fed to the input at the beginning of each pulse of the third clock signal F3 appears at the output when the next clock pulse occurs. Thus, the instantaneous output signals given in FIG. 2f in tabular form for the lines m and m+1 are obtained. The individual stages of the buffer-memory arrangement may be so-called D flip-flops, for example.
The output signal of the buffer-memory arrangement 4 is applied to the shift-register arrangement 5, which consists of n parallel shift registers, where n is the number of bits at the ouput of the third binary arithmetic stage 30. The delay provided by the n parallel shift registers is equal to the duration of one line, i.e., 64 μs in the case of PAL television sets. The clock inputs of the n parallel shift registers are fed with the fourth clock signal F4, which is shown in FIG. 2g. The output signal of the shift-register arrangement is given in tabular form in FIG. 2h for the lines m and m+1.
This output signal, together with the input signal of the shift-register arrangement 5 is fed to the fourth binary arithmetic stage 40, which forms the arithmetic means of the two signals, so that its output provides the signal B-Y in digital form, which is given in tabular form in FIG. 2k. The input and output signals of the shift-register arrangement 5 are also fed to the fifth binary arithmetic stage 50, which subtracts the input signal from the output signal and divides the difference by two. By the division, a sort of averaging is performed as well.
The output signal of the fifth binary arithmetic stage 50 is given in tabular form in FIG. 21, again for the lines m and m+1. This output signal is fed to the sixth binary arithmetic stage 60, which, in response to the output signal of the PAL switch 12, leaves it unchanged in one line and forms its absolute value in the other. "To form the absolute value" is used here first of all in the mathematical sense i.e., the negative sign of a negative number is suppressed and only the positive value of this negative number is taken into account. Within the scope of the present invention, however, "absolute value" also means "value with respect to a constant number". By this it is meant that for a number A below the constant X, the "absolute value with respect to X" is 2X-A. Thus, for the number 30, the "absolute value with respect to 50" is 70. The output of the sixth binary arithmetic stage 60 thus provides the PAL compensated signal R-Y in digital form, i.e., the red color-difference signal, which is given in tabular form in FIG. 2p for the lines m and m+1.
The output signals of the fourth binary arithmetic stage 40 and of the sixth binary arithmetic stage 60 are fed to the seventh binary arithmetic stage 70, which forms the green color-difference signal G-Y by the well-known formula Y=0.3R+0.59G+0.11B.
The subcircuits 5, 40, 50, 60 and 70, together with the PAL switch 12, represent the portion for correcting the phase of the received signal by the PAL method.
The output signals of the second, fourth, sixth and seventh binary arithmetic stages 20, 40, 60, 70, i.e., the luminance signal Y and the color-difference signals B-Y, R-Y, and G-Y, are then fed to the binary R-G-B matrix 6, which forms therefrom the binary chrominance signals R, G, B by the above formula. Each of these binary chrominance signals is then fed to one of the three digital-to-analog converters 7, 8, 9, which convert the binary chrominance signals to the analog chrominance signals R', G', B' necessary for R-G-B control of the picture tube.
In the embodiment of FIG. 1, each of thes digital-to-analog converters is also fed with the color-saturation control signal FK and the brightness control signal H, both in binary form. The PAL switch 12 is fed with the second clock signal F2, i.e., a signal having the chrominance-subcarrier frequency locked to the burst, with the composite color signal F, and with the reference pulse Z from the horizontal output stage.
FIG. 3 shows the block diagram of an embodiment of the invention. The analog-to-digital converter 2 is designed as a parallel analog-to-digital converter 2' and contains the differential amplifiers D1, D2, D3, Dp-1, Dp which are used as comparators, the resistors R1, R2, R3, Rp-1, Rp, RO, connected in series to form a voltage divider, and the decoder 21, which changes the output signals of the comparators into corresponding binary words. That portion of FIG. 3 located on the right-hand side of the decoder 21 is a greatly simplified representation of the units designated by like reference characters in FIG. 1.
The parallel analog-to-digital converter 2' contains p=2 r -1 differential amplifiers and a corresponding number of resistors, where r is the number of binary digits of the output signal of the analog-to-digital converter 2 of FIG. 1 minus one. If the analog-to-digital converter is to provide 8 bits, for example, then r is 7. The resistors R2 to Rp are alike and have a value of R, while the resistors RO, R1 have a value of 0.5 R.
According to the invention, the reference voltage applied to the comparators, in the embodiment of FIG. 3 to all inverting inputs, is shifted by ΔU=0.5 Ur/2 r during every second line as electronic switches S1 and S2 in parallel with resistors R1 and RO, respectively, are opened and closed alternately. Their control signal comes from one of the outputs Q, Q of the binary divider BT, which is fed with the horizontal synchronizing or horizontal flyback pulses Z.
Instead of shifting the reference voltage Ur as described, the amount of change ΔU may be added to the composite color signal in an analog adding stage during every second line. The reference voltage UR then remains constant.
By influencing the reference voltage Ur during every second line, and with the fourth or fifth binary arithmetic stage 40, 50 and the shift-register arrangement 5, which acts as a delay stage providing a delay of exactly one line period, the intended effect is produced, i.e., the number of comparators required is reduced to one half, while the resolution corresponds to that achieved with an additional binary digit since the average of the signals of two successive lines is taken at the output of the fourth or fifth binary arithmetic stage 40,50.
The principle explained with the aid of FIG. 3 can also be applied to the luminance channel if a comb filter and a delay arrangement providing a delay of one line period are provided in this channel.

Digivision ITT DIGIT 2000 VLSI Digital TV System
Digital integrated chrominance-channel circuit with gain control:

(Pal) Video Processing Unit (VPU - PVPU)

An improved digital integrated chrominance-channel circuit having gain control for color-television receivers includes at least one integrated circuit for digitally processing the composite color signal. The circuit includes a first limiter inserted between a parallel multiplier and a burst-amplitude-measuring stage, and a control stage including a parallel subtracter whose minuend input is fed with a reference signal, and whose subtrahend input is connected to the output of the burst-amplitude-measuring stage. A digital accumulator whose enable input is presented with a signal derived from the trailing edge of a burst gating signal is used as an integrator.

1. A digital integrated chrominance-channel circuit with gain control for color-television receivers, comprising:
at least one integrated circuit for digitally processing the composite color signal, wherein a digital chrominance signal appearing at an output of a digital chroma filter is applied to a first input of a parallel multiplier, and a digital gain control signal is applied to a second input of the parallel multiplier, the output of the parallel multiplier is connected to an input of a digital chroma demodulator with a color killer stage and to an input of a burst-amplitude-measuring stage whose output signal is compared with a reference signal in a control stage, the output signal of the control stage passes through an integrator whose output signal is the gain control signal;
a square-wave clock generator used as a chrominance subcarrier oscillator generates at least a first clock signal, whose frequency is four times that of the chrominance subcarrier, and a second clock signal, whose frequency is equal to that of the chrominance subcarrier; and
a first limiter is inserted between the parallel multiplier and the burst-amplitude-measuring stage, the control stage is a parallel subtracter whose minuend input is presented with the reference signal, and whose subtrahend input is connected to the output of the burst-amplitude-measuring stage and the integrator is a digital accumulator whose enable input is fed with a signal derived from the trailing edge of a burst gating signal.
2. A chrominance-channel circuit as claimed in claim 1, wherein the output signal from the first limiter is applied to the input of a first buffer memory and, through a delay element which provides a delay equal to the period of the first clock signal, to the input of a second buffer memory, the second clock signal being applied to the enable inputs of the first and second buffer memories during the burst gating signal, the output signals from the first buffer memory and the second buffer memory are fed, respectively, to a first absolute-value former and a second absolute-value former which have their outputs connected to the first and the second input, respectively, of a first parallel adder, the output of the first parallel adder is connected via a second limiter to the input of a third buffer memory and to the minuend input of a parallel comparator whose minuend-greater-than-subtrahend output is coupled to the enable input of the third buffer memory through the first input-output path of an AND gate whose second input is fed with the second clock signal, and the output of the third buffer memory is coupled to the subtrahend input of the parallel comparator, the output of the third buffer memory is connected to the input of a fourth buffer memory whose output is coupled to the subtrahend input of the parallel subtracter, and whose enable input is fed with a signal derived from the leading edges of horizontal-frequency pulses not coinciding with the burst gating signal, and the clear input of the third buffer memory is fed with a signal derived from the trailing edges of the pulses not coinciding with the burst gating signal. 3. A chrominance-channel circuit as claimed in claim 1, wherein the output signal from the parallel subtracter is applied to the first input of a second parallel adder having its output connected via a third limiter to the input of a fifth buffer memory whose output is coupled to the second input of the second parallel adder, and which has normalizing-data inputs and the enable input of the accumulator. 4. A chrominance-channel circuit as claimed in claim 2, wherein the output signal from the parallel subtracter is applied to the first input of a second parallel adder having its output connected via a third limiter to the input of a fifth buffer memory whose output is coupled to the second input of the second parallel adder, and which has normalizing-data inputs and the enable input of the accumulator. 5. A chrominance-channel circuit as claimed in claim 1, additionally comprising:
a first bus switch having its path from the break-contact input to the output inserted between the output of the chroma filter and the associated input of the parallel multiplier and its make-contact input connected to the input of the chroma filter;
a second bus switch having its path from the break-contact input to the output inserted between the output of the first limiter and the input of the chroma demodulator and its make-contact input connected to the input of the chroma filter;
a first test enable signal and a second test enable signal, which does not overlap the first test enable signal, being applied to the control input of the first bus switch and to the control input of the second bus switch, respectively;
an actuating signal being applied to the input of the color killer stage during the second test enable signal;
a normalizing signal being applied to the enable input of the fifth buffer memory during a third test enable signal; and
in addition to the usual contact pads, there is a contact pad via which the test-result signals of the individual subcircuits are accessible.
6. A chrominance-channel circuit as claimed in claim 2, additionally comprising:
a first bus switch having its path from the break-contact input to the output inserted between the output of the chroma filter and the associated input of the parallel multiplier and its make-contact input connected to the input of the chroma filter;
a second bus switch having its path from the break-contact input to the output inserted between the output of the first limiter and the input of the chroma demodulator and its make-contact input connected to the input of the chroma filter;
a first test enable signal and a second test enable signal, which does not overlap the first test enable signal, being applied to the control input of the first bus switch and to the control input of the second bus switch, respectively;
an actuating signal being applied to the input of the color killer stage during the second test enable signal;
a normalizing signal being applied to the enable input of the fifth buffer memory during a third test enable signal; and
in addition to the usual contact pads, there is a contact pad via which the test-result signals of the individual subcircuits are accessible.
7. A chrominance-channel circuit as claimed in claim 3, additionally comprising:
a first bus switch having its path from the break-contact input to the output inserted between the output of the chroma filter and the associated input of the parallel multiplier and its make-contact input connected to the input of the chroma filter;
a second bus switch having its path from the break-contact input to the output inserted between the output of the first limiter and the input of the chroma demodulator and its make-contact input connected to the input of the chroma filter;
a first test enable signal and a second test enable signal, which does not overlap the first test enable signal, being applied to the control input of the first bus switch and to the control input of the second bus switch, respectively;
an actuating signal being applied to the input of the color killer stage during the second test enable signal;
a normalizing signal being applied to the enable input of the fifth buffer memory during a third test enable signal; and
in addition to the usual contact pads, there is a contact pad via which the test-result signals of the individual subcircuits are accessible.
8. A chrominance-channel circuit as claimed in claim 4, additionally comprising:
a first bus switch having its path from the break-contact input to the output inserted between the output of the chroma filter and the associated input of the parallel multiplier and its make-contact input connected to the input of the chroma filter;
a second bus switch having its path from the break-contact input to the output inserted between the output of the first limiter and the input of the chroma demodulator and its make-contact input connected to the input of the chroma filter;
a first test enable signal and a second test enable signal, which does not overlap the first test enable signal, being applied to the control input of the first bus switch, respectively;
an actuating signal being applied to the input of the color killer stage during the second test enable signal;
a normalizing signal being applied to the enable input of the fifth buffer memory during a third test enable signal; and
in addition to the usual contact pads, there is a contact pad via which the test-result signals of the individual subcircuits are accessible.
9. A method of testing a chrominance-channel circuit as claimed in claim 5, characterized by the following features:
in a first step, the chroma demodulator is tested by applying the second test enable signal to the control input of the second bus switch, the actuating signal to the input of the color killer stage, and a known data sequence to the input of the chroma filter;
in a second step, the parallel multiplier is tested by applying the first test enable signal to the control input of the first bus switch, the third test enable signal and the normalizing signal to the enable input of the accumulator, the normalizing data to the normalizing-data input of the accumulator, and a known data sequence to the input of the chroma filter;
in further steps, the absolute-value formers the first adder, and the parallel comparator are tested by applying the first test enable signal to the control input of the first bus switch, and known data sequences to the input of the chroma filter, and
in the last step, the accumulator is tested by applying the first test enable signal to the control input of the first bus switch, the third test enable signal and the normalizing signal to the enable input of the accumulator, the normalizing data to the normalizing data to the normalizing-data input of the accumulator, a trigger signal to the second limiter, and known data sequences to the minuend input of the parallel sub- tracter.
10. A method of testing a chrominance-channel circuit as claimed in claim 6, characterized by the following features:
in a first step, the chroma demodulator is tested by applying the second test enable signal to the control input of the second bus switch, the actuating signal to the input of the color killer stage, and a known data sequence to the input of the chroma filter;
in a second step, the parallel multiplier is tested by applying the first test enable signal to the control input of the first bus switch, the third test enable signal and the normalizing signal to the enable input of the accumulator, the normalizing data to the normalizing-data input of the accumulator, and a known data sequence to the input of the chroma filter;
in further steps, the absolute-value formers the first adder, and the parallel comparator are tested by applying the first test enable signal to the control input of the first bus switch, and known data sequences to the input of the chroma filter, and
in the last step, the accumulator is tested by applying the first test enable signal to the control input of the first bus switch, the third test enable signal and the normalizing signal to the enable input of the accumulator, the normalizing data to the normalizing-data input of the accumulator, a trigger signal to the second limiter, and known data sequences to the minuend input of the parallel subtracter.
11. A method of testing a chrominance-channel circuit as claimed in claim 7, characterized by the following features:
in a first step, the chroma demodulator is tested by applying the second test enable signal to the control input of the second bus switch, the actuating signal to the input of the color killer stage, and a known data sequence to the input of the chroma filter;
in a second step, the parallel multiplier is tested by applying the first test enable signal to the control input of the first bus switch, the third test enable signal and the normalizing signal to the enable input of the accumulator, the normalizing data to the normalizing-data input of the accumulator, and a known data sequence to the input of the chroma filter;
in further steps, the absolute-value formers the first adder, and the parallel comparator are tested by applying the first test enable signal to the control input of the first bus switch, and known data sequences to the input of the chroma filter, and
in the last step, the accumulator is tested by applying the first test enable signal to the control input of the first bus switch, the third test enable signal and the normalizing signal to the enable input of the accumulator, the normalizing data to the normalizing-data input of the accumulator, a trigger signal to the second limiter, and known data sequences to the minuend input of the parallel subtracter.
12. A method of testing a chrominance-channel circuit as claimed in claim 8, characterized by the following features:
in a first step, the chroma demodulator is tested by applying the second test enable signal to the control input of the second bus switch, the actuating signal to the input of the color killer stage, and a known data sequence to the input of the chroma filter;
in a second step, the parallel multiplier is tested by applying the first test enable signal to the control input of the first bus switch, the third test enable signal and the normalizing signal to the enable input of the accumulator, the normalizing data to the normalizing-data input of the accumulator, and a known data sequence to the input of the chroma filter;
in further steps, the absolute-value formers the first adder, and the parallel comparator are tested by applying the first test enable signal to the control input of the first bus switch, and known data sequences to the input of the chroma filter, and
in the last step, the accumulator is tested by applying the first test enable signal to the control input of the first bus switch, the third test enable signal and the normalizing signal to the enable input of the accumulator, the normalizing data to the normalizing-data input of the accumulator, a trigger signal to the second limiter, and known data sequences to the minuend input of the parallel subtracter.
Description:
BACKGROUND OF THE INVENTION
1. Field of the Invention
The present invention relates to a digital integrated chrominance-channel circuit with gain control for color-television receivers containing at least one integrated circuit for digitally processing the composite color signal.
2. Description of the Prior Art
A chrominance-channel circuit is disclosed in the published patent application EP 51075 Al. (U.S. application Ser. No. 311,218, Oct. 11, 1981).
Practical tests of color-television receivers with digital signal processing circuitry have shown that the prior art chrominance-channel circuit still has a few disadvantages. For example, the burst-amplitude-measuring circuit is not yet optimal because it is possible in the prior art arrangement that the burst signals are sampled, i.e., measured, near or at the zero crossing. As these measured values are small, so that the digitized values formed therefrom are small numbers, the measurement error is large.
Another disadvantage of the prior art arrangement is that it has two set points for the gain control, namely a lower and an upper threshold level in the form of corresponding numbers entered into two read-only memories. Finally, the integration of the control signal is implemented with two counters, so that the time constant of this "integrator" is determined only by the clock signals for the counters and by the count lengths of these counters. As to the prior art, reference is also made to the journal "Fernseh- und Kino-Technik", 1981, pages 317 to 323, particularly FIG. 9 on page 321. However, the digital chrominance-channel circuit shown there works on the principle of feed-forward control, while both the invention and the above-mentioned prior art use a feedback control system, so that the arrangement disclosed in that journal lies further away from the present invention, the more so since in that prior art arrangement, the set point is implemented only with the concrete circuit (hardware).
SUMMARY OF THE INVENTION
The invention as claimed eliminates the above disadvantages and, thus, has for its object to improve the prior art digital integrated chrominance-channel circuit with gain control in such a way that error-free burst amplitude measurement is ensured, that a single set point can be generated, and that the integration of the control signal is implemented in optimum fashion. Another object of the invention is to modify the chrominance-channel circuit so that the automatic control system can be opened for measuring purposes.
DESCRIPTION OF THE DRAWINGS
FIG. 1 is a block diagram of the chrominance channel in accordance with the invention.
FIG. 2 is a block diagram of a preferred embodiment of the burst-amplitude-measuring stage and the digital accumulator.
FIG. 3 is a block diagram of another embodiment of the invention with the aforementioned measuring facility.
DESCRIPTION OF THE INVENTION
The block diagram of FIG. 1 includes a digital chroma filter cf, which derives a digital chrominance signal cs from a digitized composite color signal. The digital chrominance signal cs is applied to a first input of a parallel multiplier m, whose second input is fed with a digital gain control signal st. The output of the parallel multiplier m is connected to an input of a first limiter b1, which limits the output signals from the parallel multiplier m to a predetermined value. This can be done by arranging, for example, that at least one of the high-order digits of the output signal from the parallel multiplier is indicated by the interconnecting lead between these two subcircuits in FIG. 1.
In the figures of the accompanying drawing, the lines interconnecting the signal inputs and outputs of the individual subcircuits are shown as stripelike connections (buses), while the solid lines commonly used to indicate interconnections in discrete-component circuits are used for interconnections over which only individual bits or clock and/or noise signals are transferred. The stripelike lines thus interconnect parallel inputs and parallel outputs, i.e., inputs to which complete binary words are applied, which are transferred in parallel into the subcircuit at a given time, and outputs which provide complete binary words.
An output signal bs of the first limiter b1 is applied to the input of a burst-amplitude-measuring stage bm, which has its output coupled to a subtrahend input (-) of a parallel subtracter sb, while its minuend input (+) is fed with the reference signal rs, i.e., the set point. The output of the parallel subtracter sb is connected to the input of a digital accumulator ak, which provides the digital gain control signal st, which is applied to the second input of the parallel multiplier m, as mentioned above. A signal rb derived from the trailing edge of the burst gating signal (keying pulse) is applied to an enable input eu of the accumulator ak.
It is also indicated in FIG. 1 that a square-wave clock generator os, used as a chrominance-subcarrier oscillator, forms part of the invention. It provides at least the first clock signal f1, whose frequency is four times that of the chrominance subcarrier, and a second clock signal f2, having the same frequency as the chrominance subcarrier.
FIG. 2 is a block diagram of a preferred embodiment of the burst-amplitude-measuring stage bm and the digital accumulator ak of FIG. 1. The burst-amplitude-measuring stage in FIG. 2 comprises all subcircuits ahead of the subtrahend input (-) of the parallel subtracter sb, while the accumulator consists of the subcircuits following the output of the parallel subtracter sb.
The output signal bs from the first limiter b1 of FIG. 1 is applied in FIG. 2 to the input of a first buffer memory p1 and, through a delay element v, which provides a delay equal to the period of the first clock signal f1, i.e., to one quarter or 90° of the chrominance-subcarrier frequency, to an input of a second buffer memory p2.
The second clock signal f2 is applied to the enable inputs eu of these two buffer memories p1, p2 during the burst gating signal ki, which is indicated in FIG. 2 by the logical term f2.ki. During the keying pulse ki, whose duration usually equals about 10 periods of the chrominance-subcarrier frequency, a corresponding number of digital values are thus transferred successively from the first limiter b1 into the two buffer memories p1, p2, the values transferred into the second buffer memory p2 differing in phase from those transferred into the first buffer memory p1 by the aforementioned 90°; thus, two zero-crossing values are never evaluated at the same time.
The outputs of the two buffer memories p1, p2 are connected to the inputs of a first absolute-value former bb1 and a second absolute-value former bb2, respectively, whose outputs are coupled to a first and a second input, respectively, of a first adder a1. The absolute-value formers bb1, bb2 provide digital values without the sign of the input value, i.e., without the sign bit, for example. They thus contain a subcircuit which converts negative numbers in one's or two's complement notation into the corresponding positive number, i.e., they include complement reconverters.
The first adder a1 is followed by the second limiter b2, whose limiting action is controlled by at least one of the high-order digits of the first adder a1.
The output signal from the second limiter b2 is applied to the input of a third buffer memory p3 and to a minuend input a of a parallel comparator k, which has its subtrahend input b connected to the output of the third buffer memory p3.
In the present description, the two inputs of the parallel comparator k, too, are referred to as "minuend input" and "subtrahend input", respectively, which is considered justifiable in view of the fact that, purely formally, the arithmetic operation performed by comparators is more closely related to subtraction than to addition by means of an adder, even though the internal circuit of a comparator resembles that of an adder more than that of a subtracter, cf. the corresponding mathematical operations a-b and a b as opposed to a+b.
The minuend-greater-than-subtrahend output a>b of the parallel comparator k is connected to the enable input eu of the third buffer memory p3 via the first input-output path of the AND gate u, while the second clock signal f2 is applied to the second input of the AND gate u. The output of the third buffer memory p3 is also connected to an input of a fourth buffer memory p4, which has its output coupled to the subtrahend input (-) of the parallel subtracter sb. The enable input eu of the fourth buffer memory p4 is presented with a signal vz derived from the trailing edges of horizontal-frequency pulses zf, which, however, do not coincide with the burst gating signal ki, while a signal rz derived from the trailing edges of the horizontal-frequency pulses zf not coinciding with the burst gating signal ki is applied to the clear input el of the third buffer memory p3.
The derivation of the two signals rz, vz from the horizontal-frequency pulses zf is indicated in FIG. 2 by a pulse-shaper stage if. The section consisting of the two buffer memories p3, p4, the parallel comparator k, the AND gate u, and the pulse shaper if determines, for each line of the television picture, the maximum value of the burst amplitude from the--possibly limited--output signal of the first adder a1, and feeds this maximum value to the subtrahend input (-) of the parallel subtracter sb. This is achieved essentially by transferring only those words of the output signal of the second limiter b2 into the third buffer memory p3 which are greater than any word already stored in the third buffer memory p3. This is done line by line during the keying pulse ki.
As mentioned, a preferred embodiment of the accumulator ak of FIG. 1 is shown in the lower portion of FIG. 2. The output signal from the parallel subtracter sb is applied to a first input of a second parallel adder a2, which has its output connected to an input of a fifth buffer memory p5 through the third limiter b3. To realize the adding function, the output of the fifth buffer memory p5 is connected to the second input of the second adder a2. The buffer memory p5 has, in addition to the enable input eu, which is the enable input of the accumulator ak of FIG. 1, the normalizing-data inputs ne, through which normalizing data nd, i.e., known data, can be entered if necessary. The enable input eu is presented with the signal rb derived from the trailing edge of the burst gating signal ki. With the trailing edge of the keying pulse, the output signal from the third limiter b3 is thus transferred into the fifth buffer memory p5 and simultaneously transferred to the output. With the trailing edge of each keying pulse, the sum of the value from the preceding line and the set-point deviation calculated in the measured line by the parallel subtracter sb is thus produced line by line as the control signal st.
Thus, the essential advantages of the invention follow directly from the solution of the problem, namely particularly the line-by-line subtraction of the maximum burst amplitude, which is integrated in the accumulator ak to form the control signal st for the automatic control system, from the reference signal rs.
FIG. 3, a block diagram like FIGS. 1 and 2, shows a preferred embodiment of the invention which makes it possible to test the digital automatic control system after the fabrication of the integrated circuit, and to make the test-result signals accessible. The testing is necessary because the automatic control system contains several subcircuits each of which may be faulty. The test procedure and the design of the overall circuit must therefore be adapted to one another in such a way that all subcircuits of the automatic control system can be tested with little additional circuitry.
To this end, the path from a break-contact input to an output of a first bus switch bu1, whose make-contact input is connected to the input of the chroma filter cf, is interposed between the output of this chroma filter and the associated input of the parallel multiplier m, as shown in the block diagram of FIG. 3. For the graphic representation of the bus switch bu1, the symbol of a mechanical transfer switch has been chosen, with the above mentioned stripelike interconnecting lines, i.e., buses, connected to the signal inputs and the output of the switch. It is thus clear that the bus switch consists of as many individual electronic switches as there are wires in the buses.
Inserted between the output of the first limiter b1 and the input of the chroma demodulator cd, which is also present in FIG. 1, where it "demodulates" the output signal bs of the first limiter b1 into the chroma signal cs, is a path from a break-contact input to an output of a second bus switch bu2, which has its make-contact input am connected to the input of the chroma filter cf. Viewed in the direction of signal flow, the second bus switch bu2 lies behind the junction point where the signal bs for the burst-amplitude-measuring circuit is taken off. What was said on the circuit design and the graphic representation of the first bus switch bu1 applies analogously to the second bus switch bu2.
The first test enable signal t1 and the second test enable signal t2, which does not overlap the first test enable signal t1, are applied to the control input of the first bus switch bu1 and to the control input of the second bus switch bu2, respectively. Thus, when the second bus switch bu2 is in its "make" position, the first bus switch bu2 is in its "break" position, and vice versa.
During the first test enable signal t1, an actuating signal db is applied to the input ec of the color killer stage ck of the chroma demodulator cd, so that the latter is active during the testing of the automatic control system although the circuit is not in its normal mode of operation but only in a test mode.
The enable input eu of the accumulator ak, i.e., the enable input eu of the fifth buffer memory p5 in FIG. 3, may be fed with a normalizing signal ns during the third test enable signal t3. During testing and measurement, instead of the signal rb, derived from the trailing edge of the keying pulse and applied in the normal mode of operation, the normalizing signal ns is applied to the enable input eu of the fifth buffer memory p5 and causes the normalizing data nd to be transferred into this buffer.
In addition to the usual contact pads of the integrated circuit, through part of which the output signal cs of the chroma demodulator cd is coupled out, a contact pad is provided via which test-result signals of individual subcircuits are accessible, i.e., transferred out of the integrated circuit. These test-result signals are advantageously coupled to this additional contact pad through transfer transistors which, in turn, are driven by the above-mentioned test enable signals or corresponding additional signals of this kind or by signals derived by performing simple logic operations on the signals just mentioned. In this manner, only the respective subcircuit to be tested is connected to the additional contact pad.
An advantageous method of testing the chrominance-channel circuit according to the invention consists in the following time sequence of test steps. In the first step, the chroma demodulator cd is tested. This is necessary because, throughout the testing of the chrominance-channel circuit, signals are transferred out through the chroma demodulator cd and must not be falsified by the latter.
This first test step is performed by applying the second test enable signal t2 to the control input of the second bus switch bu2, the actuating signal db to the input ec of the color killer stage ck, and a known data sequence, i.e., a test-data sequence, to the input of the chroma filter cf. The application of the actuating signal db to the input ec of the color killer stage ck is necessary because an actual actuating signal coming from other stages of the chrominance-channel circuit is applied to the color killer only during normal operation of the chrominance-channel circuit, cf. the above-mentioned printed publication EP 0 051 075 Al.
In response to the application of the second test enable signal t2 to the second bus switch bu2, the input signals of the chroma filter cf are transferred directly to the input of the chroma demodulator cd, so that, if a known test-data sequence is used, the performance of the chroma demodulator cd can be checked by means of the output signals.
In the second step, the parallel multiplier m is tested. This is done by applying the first test enable signal t1 to the control input of the first bus switch bu1, the third test enable signal t3 and the normalizing signal ns to the enable input of the accumulator ak, i.e., to the enable input of the fifth buffer memory p5, for example; the normalizing data nd are applied to the normalizing-data input ne of the fifth buffer memory p5, and a known data sequence, i.e., a test-data sequence, is applied to the input of the chroma filter cf.
As in the first test, the first test enable signal t1 causes the test-data sequence to bypass the chroma filter cf, so that the test data are applied directly to one input of the parallel multiplier m. This bypassing of the chroma filter cf is necessary because the chroma filter is generally a dynamic subcircuit, which is not suitable for being included in the individual tests for this reason alone.
As a result of the entry of normalizing data into the accumulator ak or into the fifth buffer memory p5 as a subcircuit of the accumulator, known data are also applied to the second input of the parallel multiplier m, so that the output signal of the latter is predeterminable, which makes it possible to check the correct functioning of the multiplier. Since the chroma demodulator cd was tested already in the first test step, the data appearing at its output during the second test step are the unchanged output data of the parallel multiplier m if the chroma demodulator cd was found to operate correctly.
Further tests may now be performed on the absolute-value formers bb1, bb2, the first adder a1, and the parallel comparator k. To do this, the first test enable signal t1 is applied to the control input of the first bus switch bu1, and known data sequences are applied to the input of the chroma filter cf, the individual test results being accessible via the above-mentioned additional contact pad and being generally present in the form of a go/no-go decision.
The last test to be performed is that of the accumulator ak. To this end, the first test enable signal t1 is applied to the control input of the first bus switch bu1; the third test enable signal t3 and the normalizing signal ns are applied to the enable input of the accumulator ak, i.e., to the corresponding input of the fifth buffer memory p5, for example; a trigger signal is applied to the second limiter b2, and known data sequences are fed to the minuend input (+) of the parallel subtracter sb. With the second limiter sb2 triggered, one of the input signals of the accumulator is predetermined and, thus, known because the output data of the subtracter sb are known as well. The accumulator ak can thus be tested by varying the reference data rs.
The reference data rs, the above-mentioned various test-data sequences, and the normalizing data nd may come from a microprocessor.


ITT DIGIVISION 3486 OSCAR CHASSIS DIGI 3 90° FST / IFB-285/2 Digivision ITT DIGIT 2000 VLSI Digital TV System Digital horizontal-deflection circuit:

Digital deflection Processor (DPU)
Instead of fine-controlling the horizontal deflection signal in a digital television receiver by means of two phase-locked loops and gate-delay stages as is done in prior art arrangements, in the horizontal-deflection circuit according to the invention, a first digital word delivered by a first phase-locked loop and representative of the horizontal frequency is added in an adder to a suitably amplified third digital word delivered by a phase comparator of a second phase-locked loop. The output of the adder is fed to the control input of a digital sine-wave generator which drives a frequency divider. The latter delivers the horizontal deflection signal, which drives the horizontal output stage. The phase comparator is fed with the horizontal flyback signal, which is derived from the horizontal deflection signal, and a second digital word generated by the first phase-locked loop and representative of the desired phase position of the flyback signal.

What is claimed is: 1. A digital horizontal-deflection circuit for generating an analog horizontal deflection signal driving the horizontal output stage of a digital television receiver clocked with a system clock, comprising:
a first digital phase-locked loop which synchronizes the horizontal deflection signal with the horizontal synchronizing signal separated from the composite color signal and delivers for each line of video signal a first digital word representative of the horizontal frequency and a second digital word representative of the desired phase position of the horizontal flyback signal;
a second phase-locked loop which uses a digital phase comparator to generate a third digital word representative of the phase deviation of the horizontal flyback signal from the desired position and shifts the horizontal deflection signal in time so that the horizontal flyback signal takes up the desired phase position;
an adder having a first input to which said first digital word is fed and a second input to which said third digital word is fed via a multiplier serving as an amplifier;
a digital sine-wave generator having a control input to which the output of said adder is fed; and
a frequency divider to which the output of said digital sine-wave generator is supplied, the output of said frequency divider providing the horizontal deflection signal.
2. A horizontal-deflection circuit as defined in claim wherein said first digital word is representative of the period of the horizontal deflection signal, and additionally comprising a digital period-to-frequency converter connected between said first phase-locked loop and said first input of said adder. 3. A horizontal-deflection circuit as defined in claims 1 or 2, additionally comprising a protection circuit coupled between the output of said digital sine-wave generator and the input of said frequency divider, said protection circuit providing a sine-wave signal of a desired frequency if the frequency of said sine-wave generator departs from a desired-value range. 4. A horizontal-deflection circuit as defined in claim 3, wherein said protection circuit is an analog phase-locked loop.
Description:
BACKGROUND OF THE INVENTION
The present invention relates to a digital horizontal-deflection circuit for generating an analog horizontal deflection signal driving the horizontal output stage of a digital television receiver clocked with a system clock. A digital horizontal-deflection circuit of this kind is described in a data book of Intermetall, "DIGIT 2000 VLSI Digital TV System," 1984/5, pages 112 to 114, which deal with the integrated circuit DPU 2500.
In the prior art arrangement, the phase variation which is necessary for the digital generation of the horizontal deflection signal and must be stepped in fractions of the period of the system clock is achieved essentially by the use of gate-delay stages or chains as are described, for example, in the European Patent Applications EP-A Nos. 0,059,802; 0,080,970; and 0,116,669, which essentially utilize the inherent delay of inverters. It turned out, however, that with these arrangements, it is not possible to completely control all operating conditions which may occur.
SUMMARY OF THE INVENTION
It is, therefore, the object of the invention to modify and improve the digital horizontal-deflection circuit described in the above prior art in such a way that the gate-delay stages can be dispensed with.
DETAILED DESCRIPTION OF A PREFERRED EMBODIMENT
The invention will now be explained in more detail with reference to the single FIGURE of the accompanying drawing, which is a block diagram of an embodiment of the invention. The block diagram shows that portion of a digital television receiver, i.e., of a television receiver in which the analog signal received via the antenna is processed digitally, which is of interest in connection with the invention. Thus, all subcircuits for digital-to-analog conversion, sync separation, chrominance-signal and luminance-signal processing or sound-signal processing have been omitted; the overall circuit concept of digital television receivers has been well known for some time.
The first digital phase-locked loop (PLL) p1 is supplied with the (digital) horizontal synchronizing signal hs, which was separated from the composite color signal, and the system clock st, and derives therefrom, in the manner described in the prior art, the first digital word d1, which is representative of the horizontal frequency, and the second digital word d2, which is representative of the desired phase position of the horizontal flyback signal fy. The signal fy comes from the receiver's horizontal output stage ps, which supplies the necessary sawtooth current to the deflection coil 1. The phase position of the flyback signal fy relative to the horizontal deflection signal ps is dependent on the switching properties of the horizontal output stage ps and is also influenced by the video signal applied to the picture tube.
By means of the second PLL p2, indicated in the FIGURE by the large rectangle bounded by a broken line, these dependences are compensated in the manner described in the prior art. The phase comparator pv generates the third digital word d3, which is representative of the phase deviation of the flyback signal fy from its desired position, and the second PLL p2 shifts the horizontal deflection signal ds in time so that the flyback signal fy takes up the desired phase position.
The first digital word d1 is fed to the first input of the adder ad, and the third digital word d3 is fed to the second input of this adder via the multiplier m, which serves as an amplifier. The second input of the multiplier m is fed with the signal k determining the gain of the second PLL p2, so that the transient response of the latter can be optimally adjusted by the manufacturer of the television receiver.
The output of the adder ad is fed to the control input of the digital sine-wave generator s, which may be designed as an accumulator followed by a sine looker table (ROM). If an n-bit word d4 is applied to its control input, this arrangement, which is known in principle, delivers a sine-wave of frequency (d4)fs/2 n , where fs is the frequency of the system clock st.
The output of the digital sine-wave generator sg is fed to the frequency divider ft, which provides the horizontal deflection signal ds, a square-wave signal as usual. The frequency divider ft thus not only divides the frequency of the signal delivered by the sine-wave generator sg, but also converts the sine-wave signal into the above-mentioned square-wave signal; this can be done in a suitable sine-to-square wave converter stage at the input of the frequency divider ft.
Two stages which can be added to the arrangement singly or in combination are indicated in the FIGURE by rectangles bounded by broken lines. The period-to-frequency converter fw between the output of the first PLL pl for the first digital word d1 and the corresponding input of the adder ad is necessary if the first digital word d1, generated by the first PLL p1, represents the period of the horizontal deflection signal ds (if this word represents the frequency of the horizontal deflection signal, the stage fw is not necessary).
Between the output of the digital sine-wave generator sg and the input of the frequency divider ft, the protection circuit sc may be inserted. It is preferably an analog phase-locked loop which provides a sine-wave signal of the desired frequency if the frequency of the sine-wave generator sg departs from a predetermined desired-value range. This may be to advantage during the start-up phase after the turning on of the television receiver or may serve to afford protection in the event of a failure of one or both of the PLL's p1, p2.
In the FIGURE, the stripe-like connecting leads represent signal paths over which digital signals are transferred in parallel, i.e., on these buses, the individual (parallel) digital words follow one after the other at the pulse repetition rate of the system clock st. The fact that the individual stages of the second PLL p2--where necessary and appropriate--and the period-to-frequency converter fw are clocked with the system clock st, too, is indicated by the respective clock input lines.
The digital horizontal-deflection circuit in accordance with the invention is preferably realized using monolithic integrated circuit techniques, particularly MOS technology. It may form part of a larger integrated circuit but can also be implemented as a separate integrated circuit.

Digivision ITT DIGIT 2000 VLSI Digital TV System Digital circuit for steepening color-signal transitions:

Digital Transient Improvement Processor (DTI)

This circuit arrangement is designed for use in digital color-television receivers or the like and contains for each of the two digital color-difference signals a slope detector to which both a digital signal defining an amplitude threshold value and a digital signal defining a time threshold value are applied. At least one intermediate value occurring during an edge to be steepened is stored, and at the same time value of the steepened edge, it is "inserted" into the latter. This is done by means of memories switches, output registers, and a sequence controller.

What is claimed is: 1. A circuit arrangement for steepening color-signal transitions, comprising:
first and second circuit branches, said first branch receiving a first color difference digital signal from a first color difference channel and said second branch receiving a second color difference digital signal from a second color difference channel, each of said branches comprising:
a digital slope detector for generating a control signal at an output when the respective one of said first or second color difference digital signals has a predetermined relationship to predetermined amplitude and time thresholds;
a first delay element receiving and delaying said respective one color difference digital signal by a time equal to the delay of said digital slope detector;
at least one memory having its input connected to the output of said first delay element;
a switch having first and second inputs connected to the outputs of said delay element and said at least one memory, respectively; and
an output register having its input connected to the output of said switch;
and
a sequence controller coupled to the outputs of said digital slope detectors in said first and second circuit branches, and receiving a clock signal having a predetermined frequency relationship to a chrominance subcarrier frequency, and receiving a digital signal determining the hold time equal to the known system rise time of said first and second color difference channels, said sequence controller providing sequence control signals for controlling said at least one memory, said switch and said output register in both of said first and second circuit branches such that:
a color difference signal value occurring at an intermediate value of said hold time is read into said memory, said color difference signal value stored in said memory is read via said switch into said output rergister at the corresponding intermediate value of the steepened leading edge of said color-signal, the input of said output register being connected to the output of said delay element at all times except at said intermediate value of said steepened leading edge.
2. A circuit arrangement in accordance with claim 1, wherein each said slope detector comprises:
a first digital differentiator receiving the respective color difference digital signal;
a digital absolute value stage coupled to said first digital differentiator output;
a first digital comparator having a minuend input coupled to said digital absolute value stage output, a subtrahend input supplied with a digital signal corresponding to said amplitude threshold value, and an output;
a second digital differentiator having an input coupled to said comparator output;
a counter for counting pulses of said clock signal, said counter having an enable input coupled to said comparator output, and having a reset input coupled to the output of said second digital differentiator;
a fifth memory having its inputs coupled to the count outputs of said counter and an enable input coupled to said second digital differentiator output;
a second digital comparator having a minuend input coupled to the output of said fifth memory, a subtrahend input supplied with a digital signal corresponding to said time threshold value; and
gate means for combining the output of said comparator and the output of said second digital differentiator to provide said control signal when the output of said comparator and the output of said second digital differentiator are both active.
3. A circuit arrangement in accordance with claim 2,
wherein each of said first and second circuit branches further comprises a second memory having its input connected to said first delay element output, said switch having a third input coupled to said second memory output; and
wherein said sequence controller comprises:
a counter for counting pulses of said clock signal; and
a decoder for decoding the count output of said counter to provide said sequence control signals, said sequence control signals also controlling each said second memory,
said sequence controller operating such that color difference signal values occurring at the end of the first third of said hold time are written into said at least one memory, and color difference signal values occurring at the end of the second third of said hold time are written into said second memory; and
wherein:
in said first circuit branch the contents of said at least one memory and said second memory are written via said switch into said output register at the end of the first third and at the end of the second third, respectively, of said steepened leading edge;
in said second circuit branch the contents of said at least one memory and said second memory are written via said switch into said output register at the end of the first third and the second third, respectively, of said steepened leading edge; and
the input of the respective output register of each of said first and second circuit branches is connected to the output of the respective first delay element at all times except at the end of said first third and said second third or said steepened leading edge.
4. A circuit arrangement in accordance with claim 1,
wherein each of said first and second circuit branches further comprises a second memory having its input connected to said first delay element output, said switch having a third input coupled to said second memory output;
wherein said sequence controller comprises:
a counter for counting pulses of said clock signal; and
a decoder for decoding the count output of said counter to provide said sequence control signals, said sequence control signals also controlling each said second memory,
said sequence controller operating such that color difference signal values occurring at the end of the first third of said hold time are written into said at least one memory, and color difference signal values occurring at the end of the second third of said hold time are written into said second memory; and
wherein:
in said first circuit branch the contents of said at least one memory and said second memory are written via said switch into said output register at the end of the first third and at the end of said second third, respectively, of said steepened leading edge;
in said second circuit branch the contents of said at least one memory and said second memory are written via said switch into said output register at the end of the first third and the second third, respectively, of said steepened leading edge; and
the input of the respective output register of each of said first and second circuit branches is connected to the output of the respective first delay element at all times except at the end of said first third and said second third of said steepened leading edge.
Description:
BACKGROUND OF THE INVENTION
The invention pertains to a circuit for steepening color-signal transitions in color television receivers or the like.
A circuit arrangement of this kind includes a slope detector which, when a predetermined amplitude threshold value is exceeded, delivers a switching signal which causes a substitute signal to appear at the respective output of the two color-difference channels for the duration of the system rise time of said channels. One circuit arrangement of this kind, which provides a chroma transient improvement, is described in a publication by VALVO entitled "Technische Information 840228 (Feb. 28, 1984): Versteilerung von Farbsignalsprungen and Leuchtdichtesignal-Verzogerung mit der Schaltung TDA 4560".
The bandwidth of the color-difference channel is very small compared with the bandwidth of the luminance channel, namely only about 1/5 that of the luminance channel in the television standards now in use. This narrow bandwidth leads to blurred color transitions ("color edging") in case of sudden color-signal changes, e.g., at the edges of the usual color-bar test signal, because, compared with the associated luminance-signal transition, an approximately fivefold duration of the color-signal transition results from the narrow transmission bandwidth.
In the prior circuit arrangement, the relatively slowly rising color-signal edges are steepened by suitably delaying the color-difference signals and the luminance signal and steepening the edges of the color-difference signals at the end of the delay by suitable analog circuits. The color-difference signals and the luminance signal are present and processed in analog form as usual.
The problem to be solved by the invention is to modify the principle of the prior art analog circuits in such a way that it can be used in known color-television receivers with digital signal-processing circuitry (cf. "Electronics", Aug. 11, 1981, pages 97 to 103), with the slope detector responding not only to one criterion, namely a predeterminable amplitude threshold value as in the prior art arrangement, but to an additional criterion.
SUMMARY OF THE INVENTION
In accordance with the invention a circuit arrangement provides a fully digital solution for chroma transient improvement. The circuit arrangement contains a slope detector, a memory, a switch-over switch and a timing control stage for the processing of each color difference signal. A time period threshold signal and an amplitude threshold signal are fed to the slope detector. If the amplitude threshold is exceeded and the time threshold is not being reached, the slope is improved.
This circuit arrangement is designed for use in digital color-television receivers or the like and contains for each of the two digital color-difference signals a slope detector to which both a digital signal defining an amplitude threshold value and a digital signal defining a time threshold value are applied. At least one intermediate value occurring during an edge to be steepened is stored, and at the same time value of the steepened edge, it is "inserted" into the latter. This is done by means of memories, switches, output registers, and a sequence controller.
BRIEF DESCRIPTION OF THE DRAWING
The invention will be better understood from a reading of the following detailed description in conjunction with the drawing in which:
FIG. 1 is a block diagram of a first embodiment of the invention;
FIG. 2 is a block diagram of a second form of the arrangement of FIG. 1;
FIG. 3 is a block diagram of an embodiment of the slope detectors of FIGS. 1 and 2;
FIGS. 4a-c shows various waveforms to explain the basic operation of the invention; and
FIGS. 5a and 5b shows waveforms to explain the operation of the improved arrangement of FIG. 2.
DETAILED DESCRIPTION
In the block diagram of FIG. 1, the digital color-difference signals yr, yb are present in the baseband at the frequency of the clock signal f, which is four times the chrominance-subcarrier frequency, i.e., the individual data words appear one after the other at this frequency. If a subharmonic of the clock signal f, i.e., the chrominance-subcarrier frequency itself, for example, is chosen for the color-difference-signal demodulation as may be the case in known digital color-television receivers, these digital signals must be brought to the aforementioned repetition frequency of the clock signal f by digital interpolation.
In FIG. 1, there are two branches for the two color-difference signal yr and yb, respectively. They are of the same design, with the branch z1 assigned to the red-minus-luminance channel, and the branch z2 to the blue-minus-luminance channel. In the branch z1, the red-minus-luminance signal yr is applied to the inputs of the first delay element v1 and the first digital slope detector fs1. The output of the first delay element v1 is fed to the input of the first memory s1 and to one of the inputs of the first switch us1, whereas the output of the first memory s1 is connected to the other input of the first switch us1, whose output is coupled to the input of the first output register r1.
The second branch z2, to which the blue-minus-luminance signals yb are applied, is of the same design as the first branch z1 as far as the individual circuits and their interconnections are concerned, and contains the second digital slope detector fs2, the second delay element v2, the second memory s2, the second switch us2, and the second output register r2.
The output signals of the two slope detectors fs1, fs2 are applied, respectively, to the first and second inputs of the OR gate og, whose output is connected to the first input of the sequence controller ab. The second input of the latter is presented with the clock signal f, and the third input with the digital signal hz, by which the hold time equal to the system rise time of the color-difference channels can be preset. The outputs of the sequence controller ab are connected to the enable inputs en of the first and second memories s1, s2 and of the first and second output registers r1, r2 and to the control inputs of the two switches us1, us2.
The sequence controller ab controls these subcircuits as follows. A red-minus-luminance signal value yr1 and a blue-minus-luminance signal value yb1 occurring at an intermediate value of the hold time are read into the memories s1 and s2, respectively. This intermediate value of the hold time lies preferably in the middle of the hold time. Furthermore, the sequence controller causes the contents of the memories s1 and s2 to be transferred via the associated switches us1 and us2 into the associated output registers r1 and r2, respectively, at the corresponding intermediate value, preferably one-half, of the steepened leading edge, while at all times other than the instant of the intermediate value of the steepened leading edge, the inputs of the associated output registers are connected to the outputs of the delay elements v1 and v2, respectively.
The block diagram of FIG. 2 shows an improved version of the arrangement of FIG. 1. The improvement is that the first and second memories s1 and s2 of FIG. 1 have been supplemented with the third and fourth memories s3 and s4, respectively, each of which is connected in parallel with the associated memory, and that the two switches us1 and us2 of FIG. 1 have been expanded into multiposition switches us1' and us2' each having one additional input connected to the output of the third memory s3 and the output of the fourth memory s4, respectively.
This improved portion of FIG. 2 concerns the sequence controller ab of FIG. 1. In FIG. 2, the latter consists of the counter c2, which counts the pulses of the clock signal s, the decoder dc, and the AND gate u2. The start input st of the counter c2 is connected to the output of the OR gate og, whereas the stop input sp is controlled by the decoder dc. The digital signal hz is fed to the decoder dc, cf. FIG. 1.
The counts of the counter c2 are decoded by reading the red- and blue-minus-luminance signal values occurring at the end of the first third of the hold time, i.e., the values yr1' and yb1', into the first memory s1 and the second memory s2, respectively, and the red- and blue-minus-luminance signal values occurring at the end of the second third of the hold time, i.e., the values yr2 and yb2, into the third memory s3 and the fourth memory s4, respectively. At the end of the first third and second third, respectively, of the steepened leading edge, the contents of the memories s1 and s3, respectively, are transferred through the switch us1' into the output register r1, and at the end of the first third and second third, respectively of that edge, the contents of the memories s2 and s4, respectively, are transferred through the switch us2' into the output register r2. The inputs of the two outputs registers are connected to the outputs of the first and second delay elements v1 and v2, respectively, except at the end of the first and second thirds, respectively, of the steepened leading edge.
The clock signal f is applied to one of the inputs of the AND gate u2, whose other input is connected to one of the outputs of the decoder dc, and whose output is coupled to the enable inputs en of the first and second output registers r1, r2.
The block diagram of FIG. 3 shows a preferred embodiment of the circuit of the slope detectors fs1, fs2. The input for the color-difference signal yr, yb is followed by the series combination of the first digital differentiator d1, the digital absolute-value stage bb, and the minuend input m of the first digital comparator k1. The subtrahend input s of the latter is presented with the digital signal corresponding to the amplitude threshold value, the signal ta.
The absolute-value stage bb delivers digital values which are unsigned, i.e., which have no sign bit, for example.
Accordingly, the absolute-value stage bb contains a subcircuit which changes negative binary numbers in, e.g., one's or two's complement representation into the corresponding positive binary number, i.e., a recomplementer.
The term "comparator" as used herein means a digital circuit which compares the two digital signals appearing at the two inputs to determine which of the two signals is greater. Since, purely formally, such a comparison is closer to the arithmetic operation of subtraction than to that of addition although the concrete internal circuitry of such comparators is more similar to that of adders than to that of subtracters, the two inputs of the comparator are called "minuend input" and "subtrahend input" as in the case of a subtracter. The three logic output signals are "minuend greater than subtrahend", "subtrahend greater than minuend", and "minuend equal to subtrahend". Thus, in positive logic, the more positive logic level will appear at the minuend-greater-than-subtrahend output of a comparator if and as long as the minuend is greater than the subtrahend. If needed, the more negative logic level appearing at this output may serve to signal the "minuend-smaller-than-subtrahend" function, i.e., it is also possible to use negative logic.
In the slope detector of FIG. 3, the enable input eb of the first clock-pulse counter c1 and one of the inputs of the second digital differentiator d2 are connected to the minuend-greater-than-subtrahend output ms of the first comparator k1. The count outputs of the first counter c1 are coupled to the input of the fifth memory s5, which has its output connected to the minuend input m of the second digital comparator k2. The subtrahend input s of the latter is presented with a digital signal corresponding to the time threshold value, the signal tt.
The reset input re of the first counter c1, the enable input en of the fifth memory s5, and the first input of the first AND gate u1 are connected to the output of the second differentiator d2. The subtrahend-greater-than-minuend output sm of the second comparator k2 is connected to the second input of the second AND gate u2, whose output is fed to the OR gate of FIGS. 1 or 2. The subcircuits d1, bb, k1, d2, and, as mentioned above, c1 are clocked by the clock signal f.
FIGS. 4a-c and 5a and b serve to illustrate the operation of the circuit arrangement in accordance with the invention. FIG. 4a shows the assumed shape of one of the two color-difference signals yr, yb; it should be noted that, in those figures, the representation commonly used for analog signals has been chosen for simplicity.
FIG. 4b shows the output signal of the absolute-value stage bb and the amplitude threshold value corresponding to the digital signal ta. Also shown is the time threshold value corresponding to the digital signal tt. FIG. 4c shows the shape of the assumed color-difference signal of FIG. 4a as it appears at the output of the output register r1, r2 of FIG. 1 or FIG. 2. A comparison between FIGS. 4a and 4c shows that the last edge on the right has been steepened since, during this edge, both the amplitude threshold value is exceeded and the time threshold value is not reached (cf. the use of the subtrahend-greater-than-minuend output sm of the second comparator k2), the steepening function becomes effective. The first comparator k1 provides a signal at the minuend-greater-than-subtrahend output ms as long as the output signal of the absolute-value stage bb is greater than the amplitude threshold value. During that time, the first counter c1 can count the clock pulses until it is reset by a signal derived by the second differentiator d2 from the trailing edge of the output signal of the first comparator k1. The previous count of the counter c1 is transferred into the fifth memory s5 and compared with the time threshold value by the second comparator k2. If the time threshold value is greater than the period measured by the counter c1, the above-mentioned function will be initiated.
FIGS. 5a and 5b serve to explain how the steepened edge is formed. Curve a of FIG. 5a shows a slowly rising edge used for the explanation. The distances between the points in curves a and b of FIG. 5a are to illustrate the period of the clock signal f. FIG. 5b shows the waveform at the enable inputs en of the output registers r1, r2. At the arrow shown on the left between curves a and b of FIG. 5a, the signal periodically applied to these inputs at the repetition rate of the clock signal f is stopped, so to speak, so that no signals are transferred to the output registers r1, r2 over several clock periods, but the signal read in at the "clocking" of the enable inputs en is retained in those registers. After the "clocking" of the enable inputs of the output registers r1, r2 has resumed at the beginning of the edge to be steepened, the signal values yr1', yb1' and yr2, yb2 read into the memories s1, s2 and s3, s4 at the end of the first third and the second third, respectively, of the slowly rising edge of curve a of FIG. 5a are transferred into the output registers r1, r2 at the end of the first third and the second third, respectively, of this edge. The arrow shown on the right between curves a and b of FIG. 5a is to indicate that, at the end of the slowly rising edge of curve a, the steepened edge of curve b has reached the desired signal value.
The period for which the "clocking" of the enable inputs en of the output registers r1, r2 is "interrupted" is equal to the duration of the digital signal hz fed to the sequence controller ab of FIG. 1 or to the decoder dc of FIG. 2.
The circuit arrangement in accordance with the invention can be readily implemented in monolithic integrated form. As it uses exclusively digital circuits, it is especially suited for integration using insulated-gate field-effect transistors, i.e., MOS technology.


VCU 2133 Video Codec UNIT


High-speed coder/decoder IC for analog-to-digital and di-
gital-to-analo
g conversion of the video signal in digital TV
receivers based on the DIGIT 2000 concept. The VCU 2133
is a VLSI circuit in Cl technology, housed in a 40-pin Dil
plastic package. One single silicon chip combines the fol-
lowing functions and circuit details (Fig. 1):
- two input video amplifiers
- one A/D converter for the composite video signal
- the noise inverter
- one D/A converter for the luminance signal
- two D/A converters for the color difference signals
- one RGB matrix for converting the color difference sig-
nals and the luminance signal into RGB signals
- three RGB output amplifiers
- programmable auxiliary circuits for blanking, brightness
adjustment and picture tube alignment
- additional clamped RGB inputs for text and other analog
RGB signals
- programmable beam current limiting
1. Functional Description
The VCU 2133 Video Codec is intended for converting the
analog composite video signal from the video demodulator
into a digital signal. The latter is further processed

digitally
in the VPU 2203 Video Processor and in the DPU 2553 De-
flection Processor. After processing in the VPU 2203 (color
demodulation, PAL compensation, etc.), the VPU‘s digital
output signals (luminance and color difference) are recon-
verted into analog signals in the VCU 2133. From these an-
alog signals are derived the RGB signals by means of the
RGB matrix, and, after amplification in the integrated RGB
amplifiers, the RGB signals drive the RGB output amplifiers
of the color T\/ set.
For TV receivers using the NTSC standard the VPU 2203
may be replaced by the CVPU 2233 Comb Filter Video Pro-
cessor which is pin-compatible with the VPU 2203, but of-
fers better video performance. In the case of SECAM, the
SPU 2220 SECAM Chroma Processor must be connected
in parallel to the VPU 2203 for chroma processing, while
the luma processing remains inthe VPU 2203.
In a more sophisticated CTV receiver according to the Dl-
GIT 2000 concept, after the VPU Video Processor may be
placed the DTI 2223 Digital Transient Improvement Proces-
sor which serves for sharpening color transients on the
screen. The output signals of the DTI are fed to the VCU’s
luma and chroma inputs. To achieve the desired transient
improvement, the R-Y and B-Y D/A converters of the VCU
must be stopped for a certain time which is done by the
hold pulse supplied by the DTI and fed to the Reset pin 23
of the VCU. The pulse detector following this pin seperates
the (capacitively-coupled) hold pulse from the reset signal.
In addition, the VCU 2133 carries out the functions:
- brightness adjustment
- automatic CRT spot-cutoff control (black level)
- white balance control and beam current limiting
Further, the VCU 2133 offers direct inputs for text or other
analog RGB signals including adjustment of brightness and
contrast for these signals.
The RGB matrix and RGB amplifier circuits integrated in
the VCU 2133 are analog. The CRT spot-cutoff control is
carried out via the RGB amplifiers’ bias, and the white bal-
ance control is accomplished by varying the gain of these
amplifiers. The VCU 2133 is clocked by a 17.7 or 14.3 MHz
clock signal supplied by the MCU 2632 Clock Generator IC.
1.1. The A/D Converter with Input Amplifiers and Bit
Enlargement
The video signal is input to the VCU 2133 via pins 35 and 37
which are intended for normal TV video signal (pin 35) and
for VCR or SCART video signal (pin 37) respectively. The
video amplifier whose action is required, is activated by the
CCU 2030, CCU 2050 or CCU 2070 via the IM bus by soft-
ware. The amplification of both video amplifiers is doubled
during the undelayed horizontal blanking pulse (at pin 36)
in order to obtain a higher digital resolution of the color
synchronization signal (burst). At D 2-MAC reception, the
doubled gain is switched off by means of bit p = 1 (Fig. 8).

The A/D converter is of the flash type, a circuit of 2" com-
parators connected in parallel. This means that the number
of comparators must be doubled if one additional bit is
needed. Thus it is important to have as few bits as possi-
ble. For a slowly varying video signal, 8 bits are required.

ln
order to achieve an 8-bit picture resolution using a 7-bit
converter, a trick is used: during every other line the refer-
ence voltage of the A/D converter is changed by an
amount corresponding to one half of the least significant
bit. ln this procedure, a grey value located between two 7-
bit steps is converted to the next lower value during one
line and to the next higher value during the next line. The
two grey values on the screen are averaged by the viewer’s
eye, thus producing the impression of grey values with
8-bit resolution. Synchronously to the changing reference
voltage of the A/D converter, to the output signal of the Y
D/A converter is added a half-bit step every second line.
The bit enlargement just described must be switched off in
the case of using the D2-MAC standard (q = 1 and r = 1
in Fig. 8). ln the case of using the comb filter CVPU instead
of the VPU, the half-bit adding in the Y D/A converter must
be switched off (r = 1 in Fig. 8).
The A/D converter’s sampling frequency is 17.7 MHZ for
PAL and 14.3 MHz for NTSC, the clock being supplied by
the MCU 2632 Clock Generator IC which is common to all
circuits for the digital T\/ system. The converter’s resolu-
tion is 1/2 LSB of 8 bits. Its output signal is Gray-coded to
eliminate spikes and glitches resulting from different com-
parator speeds or from the coder itself. The output is fed to
the VPU 2203 and to the DPU 2553 in parallel form.
1.2. The Noise Inverter
The digitized composite video signal passes the noise in-
verter circuit before it is put out to the VPU 2203 and to the
DPU 2553. The noise inverter serves for suppressing bright
spots on the screen which can be generated by noise
VCU 2133
pulses, p. ex. produced by ignition sparks of cars etc. The
function of the noise inverter can be seen in Fig. 2. The
maximum white level corresponds with step 126 of the A/D
converter’s output signal (that means a voltage of 7 V at
pin 35). lf, due to an unwanted pulse on the composite
video signal, the voltage reaches 7.5 V (what means step
127 in digital) or more, the signal level is reduced by such
an amount, that a medium grey is obtained on the screen
(about 40 lFiE). The noise inverter circuit can be switched
off by software (address 16 in the VPU 2203, see there).
1.3. The Luminance D/A Co
nverter (Y)
After having been processed in the VPU 2203 (color de-
modulation, PAL compensation, etc.), the different parts of
the digitized video signal are fed back to the VCU 2133 for
further processing to drive the RGB output amplifiers. The
luminance signal (Y) is routed from the VPU’s contrast mul-
tiplier to the Y D/A converter in the VCU 2133 in the form of
a parallel 8-bit signal with a resolution of 1/2 LSB of 9

bits.
This bit range provides a sufficient signal range for contrast
as well as positive and negative overshoot caused by the
peaking filter (see Fig. 3 and Data Sheet VPU 2203).


The luminance D/A converter is designed as an R-2R lad-
der network. lt is clocked with the 17.7 or the 14.3 MHz
clock signal applied to pin 22. The cutoff frequency of the
luminance signal is determined by the clock frequency.
1.4. The D/A Converters for the Color Difference Signals
R-Y and B-Y
ln order to save output pins at the VPU 2203 and input pins
at the VCU 2133 as well as connection lines, the two digital
color difference signals R-Y and B-Y are transferred in time
multiplex operation. This is possible because these signals’
bandwidth is only 1 MHZ and the clock is a 17.7 or 14.3
MHz signal.
The two 8-bit D/A converters R-Y and B-Y are also built as
R-2R ladder networks. They are clocked with ‘A clock fre-
quency, but the clock for the multiplex data transfer is 17.7
or 14.3 MHz. Four times 4 bits are transferred sequentially,
giving a total of 16 bits. A sync signal coordinates the

multi-
plex operations in both the VCU 2133 and the VPU 2203.
Thus, only four lines are needed for 16 bits. Fig. 4 shows
the timing diagram of the data transfer described.
ln a CTV receiver with digital transient improvement (DTI
2223), the R-Y and B-Y D/A converters are stopped by the
hold pulse supplied by the DTI, and their output signal is
kept constant for the duration of the hold pulse. Thereafter,
the output signal jumps to the new value, as described in
the DTl’s data sheet.
Fig. 4:
Timing diagram of the multiplex data transfer of the chroma
channel between VPU 2203, VCU 2133 and SPU 2220
a) main clock signal QSM
b) valid data out of the VCU 2133’s video A/D converter.
AIAD is the delay time of this converter, about 40 ns.
c) valid data out of the VPU 2203.
d) MUX data transfer of the chroma signals from VPU 2203
to VCU 2133, upper li
ne: sync pulse from pin 27 VPU to
pin 21 VCU during sync time in vertical blanking time,
see Fig. 8; lower line: valid data from pins 27 to 30
(VPU) to pins 18 to 21 (VCU)
1.5. The RGB Matrix and the RGB Output Amplifiers
ln the RGB matrix, the signals Y, R-Y and B-Y are dema-
trixed, the reduction coefficients of 0.88 and 0.49 being tak-
en into account. In addition, the matrix is supplied with a
signal produced by an 8-bit D/A converter for setting the
brightness of the picture. The brightness adjustment range
corresponds to 1/2 of the luminance signal range (see Fig.
3). It can be covered in 255 steps. The brightness is set by
commands fed from the CCU 2030, CCU 2050 or CCU 2070
Central Control Unit to the VPU 2203 via the IM bus.
There are available four different matrices: standard PAL,
matrix 2, 3 and 4, the latter for foreign markets. 'The re-
quired matrix must be mask-programmed during produc-
tion. The matrices are shown in Table 1, based on the for-
mulas:
R = r1~(R-Y)+ l'2~(B-Y) +Y
G = Q1-(Ft-Y)+ Q2 - (B-Y) +Y
B = b1-(Ft-Y)+ bg - (B-Y) +Y
The three RGB output amplifiers are impedance converters
having a low output impedance, an output voltage swing of
6 V (p-p), thereof 3 V for the video part and 3 V for bright-
ness and dark signal. The output current is 4 mA. Fig. 5
shows the recommended video output stage configuration.

For the purpose of white-balance control, the amplification
factor of each output amplifier can be varied stepwise in
127 steps (7 bits) by a factor of 1 to 2. Further, the CRT
spot-cutoff control is accomplished via these amplifiers’ bi-
as by adding the output signal of an 8-bit D/A converter to
the intelligence signal. The amplitude of the output signal
corresponds to one half of the luminance range. The eight
bits make it possible to adjust the dark voltage in 0.5 %
steps. By means of this circuit, the factory-set values for
the dark currents can be maintained and aging of the pic-
ture tube compensated.
1.6. The Beam Current and Peak Beam Current Limiter
The principle of this circuitry may be explained by means of
Fig. 6. Both facilities are carried out via pin 34 of the VCU
2133. For beam current limiting and peak beam current li-
miting, contrast and brightness are reduced by reducing
the reference voltages for the D/A converters Y, Ft-Y and
B-Y. At a voltage of more than +4 V at pin 34, contrast and
brightness are not affected. In the range of +4 V to +3 V,
the contrast is continuously reduced. At +3 V, the original
contrast is reduced to a programmable level, which is set
by the bits of address 16 of the VPU as shown in Table 2. A
further decrease of the voltage merely reduces brightness,
the contrast remains unchanged. At 2 V, the brightness is
reduced to zero. At voltages lower than 2 V, the output
goes to ultra black. This is provided for security purposes.
The beam current limiting is sensed at the ground end of
the EHT circuit, where the average value of the beam cur-
rent produces a certain voltage drop across a resistor in-
serted between EHT circuit and ground. The peak beam
current limiting can be provided additionally to avoid
“blooming” of white spots or letters on the screen. For
this, a fast peak current limitation is needed which is
sensed by three sensing transistors inserted between the
RGB amplifiers and the cathodes of the picture tube. One
of these three transistors is shown in Fig. 6. The sum of the
picture tube’s three cathode currents produces a voltage
drop across resistor R1. If this voltage exceeds that gen-
erated by the divider R2, B3 plus the base emitter voltage
of T2, this transistor will be turned on and the voltage at

pin
34 of the VCU 2133 sharply reduced. Time constants for
both beam current limiting and peak beam current limiting
can be set by the capacitors C1 and C2.
1.7. The Blanking Circuit
The blanking circuit coo
rdinates blanking during vertical
and horizontal flyback. During the latter, the VCU 2133's
output amplifiers are switched to “ultra black”. Such
switching is different during vertical f
lyback, however, be-
cause at this time the measurements for picture tube align-
ment are Carried out. During vertical flyback, only the ca-
thode to be measured is switched to “black” during mea-
suring time, the other two are at ultra black so that only the
dark current of one cathode is measured at the same time.
For measuring the leakage current, all three cathodes are
switched to ultra black.
The sequence described is controlled by three code bits
contained in a train of 72 bits which is transferred from the
VPU 2203 to the VCU 2133 during each vertical blanking in-
terval. This transfer starts with the vertical blanking pulse.
During the transfer all three cathodes of the picture tube
are biased to ultra black. In the same manner, the white-
balance control is done.
The blanking circuit is controlled by two pulse combina-
tions supplied by the DPU 2553 Deflection Processor
(“sandcastle pulses"). Pin 39 of the VCU 2133 receives the
combined vertical blanking and delayed horizontal blanking


pulse from pin 22 of the DPU (Fig. 7 b), and pin 36 of the
VCU gets the combined undelayed horizontal blanking and
color key pulse from pin 19 of the DPU (Fig. 7 a). The two
outputs of the DPU are tristate-controlled, supplying the
output levels max. 0.4 V (low), min. 4.0 V (high), or high-im-
pedance, whereby the signal level in the high-impedance
mode is determined by the VCU’s input configuration, a
voltage divider of 3.6 KS! and 5 KQ between the +5 V sup-
ply and ground, to 2_8 V. The VCU’s input amplifier has two
thresholds of 2.0 V and 3.4 V for detecting the three levels
of the combined pulses. ln this way, two times two pulses
are transferred via only two lines.
1.8. The Circuitry for Picture Tube Alignment
During vertical flyback, a number of measurements are tak-
en and data is exchanged between the VCU 2133, the VPU
2203 and the CCU 2030 or CCU 2050. These measure-
ments deal with picture tube alignment, as white level and
dark current adjustment, and with the photo current sup-
plied by a photo resistor (Fig. 5) which serves for adapting
Fig. 8:
Data sequence during the transfer of test results from the
VPU 2203 to the VCU 2133. Nine Bytes are transferred, in
each case the LSB first. These 9 Bytes, 8 bits each, coin-
cide with the 72 pulses of 4.4 MHz that are transferred dur-
ing vertical flyback from pin 27 of the VPU 2203 to pin 21 of
the VCU 2133 (see Fig. 9).
l and mi beam current limiter range
l<: noise inverter on/off
n: video input switching bit
S: SECAM chroma sync bit; S = 1 means that the chroma
demultiplexer is synchronized every line. The switch-over
time from C0 to demux counter begins with the end of the
undelayed horizontal blanking pulse and remains valid for a
time of 12 Q M clock periods.
6
the contrast of the picture to the light in the room where
the TV set is operated. The circuitry for transferring the

pic-
ture tube alignment data, the sensed beam currents and
the photo current is clocked in compliance with the VPU
2203 by the vertical blanking pulse and the color key pulse.
To carry out the measurements, a quadruple cycle is pro-
vided (see Table 3). The timing of the data transfer during
the vertical flyback is shown in Fig. 9, and Fig. 8 shows the
data sequence during that data transfer.
Ft, G, B: code bits
p=1; no doubled gain in the input amplifier during horizon-
tal blanking (see section 1.1.)
q=1: no changing of the A/D converter’s reference vol-
tage during every other line (see section 1.1.)
r=1: when operating with the DMA D2-MAC decoder or
the CVPU comb filter video processor, the adding of
a step of ‘/2 LSB to the output signal of the Y D/A
converter is switched off (see section 1.1.).
s=1; the blankirig pulse in the analog video output signal
at pins 26 to 28 is switched off, as is required in
stand-alone applications.


1.9. The Additional RGB Inputs
The three additional analog RGB inputs are provided for
inputting text or other analog RGB signals. They are con-
nected to fast voltage-to-current converters whose output
current can be altered in 64 steps (6 bits) for contrast set-
ting between 100 % and 30 %. The three inputs are
clamped to a DC black level which corresponds to the level
of 31 steps in the luminance channel, by means of the color
key pulse. So, the same brightness level is achieved for
normal and for external RGB signals. The output currents
ofthe converters are then fed to the three RGB output am-
plifiers. Switchover to the external video signal is also

fast.
1.10. The Reset Circuit and Pulse Detector
The reset pulse produced by the external reset RC network
in common for the whole DIGIT 2000 system, switches the
RGB outputs to ultra black during the power-on routine of
the TV set. At other times, high level must be applied to the
reset input pin 23.
There is an additional facility with pin 23 which is used only
in conjunction with the DTl 2223 Digital Transient Improve-
ment Processor. The hold pulse produced by the latter
which serves for stopping the R-Y and B-Y D/A converters,
is also fed to pin 23, capacitively-coupled. The pulse detec-
tor responds on positive pulses which exceed the 5 V sup-
ply by about 1 V. The two DACs are stopped as long as the
hold pulse lasts, and supply a constant output signal of the
amplitude at the begin of the hold pulse.


5. Description of the Connections and the Signals
Pins 1, 9, and 25 - Supply Voltage, +5 V
The supply voltage is +5 V. Pins 1 and 25 supply the ana-
log part and must be filtered separately.
Pins 2 to 8 - Outputs V0 to V6

Via these pins the VCU 2133 supplies the digitized video
signal in a parallel 7-bit Gray code to the VPU 2203 and the
DPU 2553. The output configuration is shown in Fig. 16.
Pins 10 to 17 - Inputs L7 to L0
Fig. 17 shows these inputs’ configuration. Via these pins,
the VCU 2133 receives the digital luminance signal from the
VPU 2203 in a paraliel 8-bit code.
Pins 18 to 21 - Inputs C0 to C3
Via these inputs, whose circuitry and data correspond to
those of pins 10 to 17, the VCU 2133 is fed with the digi-
tized color difference signals R-Y and B-Y and with the
control and alignment signals described in section 1.8., in
multiplex operation. Pin 21 is additionally used for the

multi-
plex sync signal.
Pin 22 - QSM Main Clock Input
Via this pin, whose circuitry is shown in Fig. 18, the VCU
2133 is supplied with the clock signal QSM produced by the
MCU 2600 or MCU 2632 Clock Generator IC. The clock fre-
quency is 17.7 MHz for PAL and SECAM and 14.3 MHz for
NTSC. The clock signal must be DC-coupled.
Pin 23 - Reset and Hold Pulse Input (Fig. 19)
Via this pin, the VCU 2133 is supplied with the reset and
hold signals which are supplied by pin 21 of the DTI 2223
Digital Transient Improvement Processor for stopping the
R-Y and B-Y D/A converters, and for Reset.
Pins 24 and 29 - Analog Ground, 0
These pins serve as ground connections for the supply and
for the analog signals (GND pin 24 for RGB).
Pins 26 to 28 - RGB Outputs
These three analog outputs deliver an analog signal suit-
able for driving the RGB output transistors. Their diagram
is shown in Fig. 20. The output voltage swing is 6 V total,
3 V for the black-to-white signal and 3 V for adjusting
the brightness and the black level.
Pins 30 to 32 - Additional Analog Inputs R, G and B
Fig. 21 shows the configuration of these inputs. They serve
to feed analog RGB signals, for example for Teletext or si-
milar applications, and they are clamped during the color
key pulse. At a 1 V input, full brightness is reached. The
bandwidth extends from 0 to 8 MHz.
Pin 33 - Fast Switching Input
This input is connected as shown in Fig. 22. It ser\/es for
fast switchover of the video channel between an internally-
produced video signal and an externally-applied video sig-
nal via pins 30 to 32. With 0 V at pin 33, the RGB outputs
will supply the internal video signal, and at a 1 V input

level,
the RGB outputs are switched to the external video signal.
Bandwidth is 0 to 4 MHz, and input impedance 1 KQ mini-
mum.
Pin 34 - Beam Current Limiter Input
The diagram of pin 34 is shown in Fig. 25. The input voltage
may be between +5 V and 0 V. The input impedance is 100
kQ. The function of pin 34 is described in section 1.6.
Pin 35 - Composite Video Signal Input 1
To fully drive the video A/D converter the following ampli-
tudes are required at pin 35: +5 V = sync pulse top level,
all bits low; +7 V = peak white, all bits high. Fig. 24 shows
the configuration of pin 35.
Pin 36 - Undelayed Horizontal Blanking and Color Key
Pulse Input
The circuitry of this pin is shown in Fig. 23. Pin 36 receives
the combined undelayed horizontal blanking and color key
pulse which are “sandcastled” and are supplied by pin 19
of the DPU 2553 Deflection Processor. During the undelay-
ed horizontal blanking pulse, the input amplifiers’ gain is
doubled, and the bit enlargement circuit is also switched
by this pulse, and the counter for the data transmission
gap started. The color key pulse is used for clamping the
RGB inputs pins 30 to 32.
Pin 37 - Composite Video Signal Input 2
This pin has the same function and properties as pin 35,
except the gain of the input amplifier which is twice the
gain as that of the amplifier at pin 35. This means an input
voltage range of +5 V to +6 V.
Pin 38 - Supply Voltage, +12 V »
The 12 V supply is needed for certain circuit parts to obtain
the required input or output voltage range, as the video in-
put and the RGB outputs (see Figs. 20 and 24).
Pin 39 - Vertical Blanking and Delayed Horizontal Blanking
Input
This pin receiv
es the combined vertical blanking and delay-
ed horizontal blanking. pulse from pin 22 of the DPU 2553
Deflection Processor. Both pulses are “sandcastled” so
that only one connection is needed for the transfer of two
pulses. These two pulses are separated in the input circui-
try of the VCU 2133, and are used for blanking the picture
during vertical and horizontal flyback. Fig. 23 shows the cir-
cuitry of pin 39.
Pin 40 - Digital Ground, O
This pin is used as GND connection in conjunction with the
pins 2 to 8 and 10 to 21 which carry digital signals.




DPU 2553, DPU
2554 Deflection Processors UNIT

Note: lf not otherwise designated, the pin numbers
mentioned refer to the 40-pin Dil package.

1. Introduction
These programmable VLSI circuits in n-channel mOS
technology carry out the deflection functions in digital
colorTV receivers based onthe
DiGiT 2000 system and
are also suitable for text and D2~mAC application. The
three types are basically identical, but are modified ac-
cording to the intended application:

DPU 2553
normal-scan horizontal deflection, standard CTV re-
ceivers, also equipped with Teletext and D2-mAC fa-
cility
DPU 2554
double-scan horizontal deflection, for CTV receivers
equipped with double-frequency horizontal deflection
and double-~frequency vertical deflection for improved
picture quality. At power-up, this version starts with
double horizontal frequency.

1.1. General Description
The DPU 2553/54 Deflection Processors contain the fol-
lowing circuit functions on one single silicon chip:
- video clamping
- horizontal and vertical sync separation
~ horizontal synchronization
- normal horizontal deflection
-east-west correction, also for flat-screen picture
tubes
- vertical synchronization
- normal vertical deflection
~ sawtooth generation
-text display mode with increased deflection frequen-
cies (18.7 kHz horizontal and 60 Hz vertical)
- D2-mAC operation mode

and for DPU 2554 only:
- double-scan horizontal deflection
- normal and double-scan vertical deflection
ln this data sheet, all information given for double~scan
mode is available with the DPU 2554 only. Type DPU
2553 starts the horizontal deflection with 15.5 kHz ac-
cording to the normal TV standard, whereas type DPU
2554 starts with 31 kHz according to the double-scan
system.
The following characteristics are programmable:
~ selection ofthe TV standard (PAL, D2-mAC or NTSC)
- selection ofthe deflection standard (Teletext, horizon-
tal and vertical double-scan, and normal scan)
- filter time»constant for horizontal synchronization
- vertical amplitude, S correction, and vertical position
for in-line, flat-screen and Trinitron picture tubes
- east-west parabola, horizontal width, and trapezoidal
correction for in-line, flat-screen and Trinitron picture
tubes
- switchover characteristics between the different syn-
chronization modes
~characteristic of the synchronism detector for PLL
switching and muting

1.2. Environment
Fig. 1-1 showsthe simplified block diagram ofthe video
and deflection section of a digital TV receiver based on
the DIGIT 2000 system. The analog video signal derived
from the video detector is digitized in the VCU 2133,
VCU 2134 or VCU 2136 Video Codec and supplied in a
parallel 7 bit Gray code. This digital video signal is fed to
the video section (PVPU, CVPU, SPU and DmA) and to
the DPU 2553/54 Deflection Processorwhich carries out
all functions required in conjunction with deflection, from
sync separation to the control of the deflection power
stages, as described in this data sheet.




3. Functional Description
3.1. Block Diagram
The DPU 2553 and DPU 2554 Deflection Processors
perform all tasks associated with deflection in TV sets;
- sync separation
- generation and synchronization of the horizontal and
the vertical deflection frequencies
-the various eastevvest corrections
- vertical savvtooth generation including S correction
as described hereafter. The DPU communicates, viathe
bidirectional serial lm bus, with the CCU 2050 or CCU
2070 Central Control Unit and, via this bus, is supplied
with the picture-correction alignment information stored
in the mDA 2062 EEPROM during set production, vvhen
the set is turned on. The DPU is normally clocked with
a trapezoidal 17.734 mHz (PAL or SECAm), or 14.3 mhz
(NTSC) or 20.25 mHz (D2-mAC) clock signal supplied
by the mCU 2600 or mCU 2632 Clock Generator IC.

The functional diagram of the DPU is shovvn in Fig. 3-1.
3.2. The Video Clamping Circuit and the Sync Pulse
Separation Circuit

The digitized composite video signal delivered as a 7»bit
parallel signal by the VCU 2133, VCU 2134 or VCU 2136
Video Codec is first noise-filtered by a 1 mHz digital lovv-
pass filter and, to improve the noise immunity ofthe
clamping circuit, is additionally filtered by a 0.2 mHz low-
pass filter before being routed to the minimum and back
porch level detectors (Fig. 3-3).
The DPU has tvvo different clamping outputs, no. 1 and
No. 2, one of vvhich supplies the required clamping
pulses to the video input of the VCU as shovvn in Fig.
3-1. The following values forthe clamping circuit apply
for Video Amp. l. since the gain of Video Amp. ll istwice
th at of Video Amp l, all clamping and signal levels of Vid-
eo Amp ll are halt those of Video Amp l referred to +5 V.
Afterthe TV set is switched on,thevideo clamping circuit
first of all ensures by means of horizontal-frequency
current pulses from the clamping output of the DPU to
the coupling capacitor of the analog composite video
signal, that the video signal atthe VCU’s input is optimal-
ly biased for the operation range of the A/D converter of
5 to 7 V. For this, the sync top level is digitally measured
and set to a constant level of 5.125 V by these current
pulses. The horizontal and vertical sync pulses are novv
separated by a fixed separation level of 5.250 V so that
the horizontal synchronization can lock to the correct
phase (see section 3.3. and Figs. 3-2 and 3-3).
vvith the color key pulse which is now present in syn-
chronism with the composite video signal, the video
clamping circuit measures the DC voltage level of the
porch and by means of the pulses from pin 21 (or pin4),
sets the DC level ofthe porch at a constant 5.5 V (5.25 V
for Video Amp ll). This level is also the reference black
to Video Processorffeletext Processor, D2-MAC Processor tc.


level for the PVPU 2204 or CvPU 2270 Video Proces-
sors.
When horizontal synchronization is achieved, the slice
level for the sync pulses is set to 50 % of the sync pulse
amplitude by averaging sync top and black level. This
ensures optimum pulse separation, even with small
sync pulse amplitudes (see application notes, section
4).


3.3. Horizontal Synchronization
Two operating modes are provided for in horizontal syn-
chronization. The choice of mode depends on whether
or not the Tv station is transmitting a standard PAL or
NTSC signal, in which there is a fixed ratio between color
subcarrier frequency and horizontal frequency. ln the
first case we speak of “color-locked” operation and in
the second case of “non-color-locked” operation (e.g.
black-and-white programs). Switching between thetwo
modes is performed automatically by the standard sig-
nal detector.


3.3.1. Non-Color-Locked Operation
ln the non»locked mode,which is needed in the situation
where there is no standard fixed ratio between the color
subcarrier frequency and the horizontal frequency ofthe
transmitter, the horizontal frequency is produced by subdemding the clock frequency (1 7.7 mHz for PAL and SECAM, 14.3
mHz for NTSC) in the programmable fre-
quency dmder (Fig. 3-4) until the correct horizontal
frequency is obtained. The correct adjustment of fre-
quency and phase is ensured by phase comparator l.
This determines the frequency and phase deviation by
means of a digital phase comparison between the sepa-
rated horizontal sync pulses and the output signal of the
programmable dmder and corrects the d
mder accordingly. For
optimum adjustment of phase iitter, capture
behavior and transient response of the horizontal PLL
circuit, the measured phase deviation is filtered in a digi-
lowpass filter (PLL phase filter). ln the case of non-
OZMH synchronized horizontal PLL, this filter is set to
wideband PLL response with a pull-in range of 1800 Hz. if the
- sync sync PLL circuit is locked, the PLL filter is
automatically switched to narrow-band response by an internal
synchronism detector in order to limit the phase jitter to a
minimum, even in the case of weak and noisy signals.

A calculator circuit in phase comparator , which analyzes the
edges of the horizontal sync pulses, increases
the resolution of the phase measurement from 56 ns at
Fig. 3-3: Principle ofvideo clamping and pulse separa- 17.7

mHz clock frequency to approx. 6 ns, or from 70 ns
NON at 14.3 MHz clock frequency to approx. 2.2 ns.
The various key and gating pulses such as the color key
pulse (tKe(,), the normal-scan (1 H) and double-scan
(2H) horizontal blanking pulse (tAZ(/) and the 1 H hori-
zontal undelayed gating pulse (t/(Z) are derived from the
output signals ofthe programmable dmder and an addi-
tional counter forthe2H signals and the 1 H and 2H skew
data output. These pulses retain a fixed phase position
with respect to the 1 H inputvideo signal andthe double-
scan output video signal from the CvPU 2270 Video Pro-
cessor
Forthe purpose of equalizing phase changes in the hori-
zontal output stage due to switching response toler-
ances or video influence, a second phase control loop
is used which generates the horizontal output pulse at
pin 31 to drivethe horizontal output stage. ln phase com-
parator li (Fig. 3~4), the phase difference between the
output signal of the programmable dmder and the lead-
ing edge (or the center) of the horizontal flyback pulse
(pin 23) is measured by means of a balanced gate delay
line. The deviation from the desired phase difference is
used as an input to an adder. ln this, the information on
the horizontal frequency derived from phase com-
parator l is added to the phase deviation originating form
phase comparator ll. The result of this addition controls
a digital on-chip sinewave generator (about 1 mHz)
which acts as a phase shifter with a phase resolution of
1/128 of one main clock period m_
By means of control loop ll the horizontal output pulse
(pin 31) is shifted such that the horizontal flyback pulse
(pin 23) acquiresthe desired phase position with respect
to the output signal of the programmable dmder which,
in turn, due to phase comparator l, retains a fixed phase
position with respect to the video signal. The horizontal
output pulse itself is generated by dmding the frequency
ofthe 1 mHz sinewave oscillator by a fixed ratio of 64 in
the case of norm al scan and of 32 in the case of double-
scan operation.


3.3.2. Color-Locked Operation
When in the color~locked operating mode, after the
phase position has been set in the non-color-locked
mode, the programmable dmder is set to the standard
dmsion ratio (1135:1 for PAL, 91O:1 for NTSC) and
phase comparator is disconnected so that interfering
pulses and noise cannot influence the horizontal deflec-
tion. Because phase comparator ll is still connected,
phase errors ofthe horizontal output stage are also cor-
rected in the color»locKed operating mode. The stan-
dard signal detector is so designed that it only switches
to color-locked operation when the ratio between color
subcarrier frequency and horizontal frequency deviates
no more than 1O'7 from the standard dmsion ratio. To
ascertain this requires about 8 s (NTSC). Switching off
color-locked operation takes place automatically, in the
_ case of a change of program for example, within approx-
imately 67 ms (e.g. two NTSC fields, 60 Hz).


3.3.3. Skew Data Output and Field Number Informa-
tion
with non-standard input signals, the TPU 2735 or TPU
2740 Teletext Processor produce a phase error vvith re-
spect to the deflection phase.
The DPU generates a digital data stream (skevv data,
pin 7 ofthe DPU), which informs the PSP and TPU on
the amount of phase delay (given in 2.2 ns increments)
used in the DPU for the 1H and 2h output pulse com-
pared With the Fm main clock signal of 17.7 mHz (PAL
or SECAm) or 14.3 mhz (NTSC), see also Figs. 3-6 to
3-8. The skew data is used by the PSP and by the TPU
to adjust the double-scan video sign
al to the 1 H and 2H
phase of the horizontal deflection to correct these phase
errors.
For the vmC processor the skew data contains three additional

bits for information about frame number, 1 V
sync and 2 V sync start.


3.3.4. Synchronism Detector for PLL and Muting
Signal
To evaluate locking ofthe horizontal PLL and condition
of the signal, the DPU’s HSP high-speed processor
(Fig. 3~1) receives two items of information from the hor-
izontal PLL circuit (see Fig. 3-11).
a) the overall pulsevvidth of the separated sync pulses
during a 6.7 us phase window centered to the horizontal
sync pulse (value A in Fig. 3-11).
b) the overall pulsevvidth of the separated sync pulse
during one horizontal line but outside the phase window
(value B in Fig. 3-11).
Based on a) and b) and using the selectable coefficients
KS1 and KS2 and a digital lovi/pass filter, the HSP pro-
cessor evaluates an 8-bit item of information “SD” (see
Fig. 3-12). By means of a comparator and a selectable
level SLP, the switching threshold for the PLL signal
“UN” is generated. UN indicates Whether the PLL is in
the synchronous or in the asynchronous state.
To produce a muting signal in the CCU, the data SD can
be read by the CCU. The range ot SD extends from O
(asynchronous) to +127 (synchronous). Typical values
torthe comparator levels and their hysteresis B1 = 30/20
and for muting 40/30 (see also HSP Bam address Table
5-6).



DPU 2553, DPU 2554

3.4. Start Oscillator and Protection Circuit
To protect the horizontal output stage of the TV set dur-
ing changing the standard and for using the DPU as a
low power start oscillator, an additional oscillator is pro-
vided on-chip (Fig. 3-4), with the output connected to
pin 31. This oscillator is controlled by a 4 mHz signalin-
dependent trom the Fm main clock produced by the
MCU 2600 or mCU 2632 Clock Generator IC and is pow-
ered by a separate supply connected to pin 35. Thefunc-
tion ofthis circuitry depends on the external standard se-
lection input pin 33 and on the start oscillator select input
pin 36, as described in Table 3-3. Using the protection
circuit as a start oscillator, the following operation modes
are available (see Table 3-3).
With pin 33 open-circuit, pin 36 at high potential (con-
nected to pin 35) and a 4 mHz clock applied to pin 34,
the protection circuit acts as a start oscillator. This pro-
duces a constant-frequency horizontal output pulse of
15.5 kHz in the case of DPU 2553, and of 31 khz in the
case of DPU 2554 while the Beset input pin 5 is at low
potential. The pulsewidth is 30 us with DPU 2553, and
16 us with DPU 2554. main clock at pin 2 or main power
supplies at pins 8, 32 and 40 are not required for this start
oscillator After the main power supply is stabilized and
the main clock generator has started, the reset input pin
5 must be switched to the high state. As long as the start
values from the CCU are invalid, the start oscillator will
continuously supply the output pulses of constant fre-
quency to pin 31 _ By means of the start values given by
the CCU via the lm bus, the register FL must be set to
zero to enable the stan oscillator to be triggered by the
horizontal PLL circuit. After that, the output frequency
and phase are controlled by the horizontal PLL only.
It the external standard selection input pin 33 is con-
nected to ground or to +5 V, the start oscillator is
switched off as soon as it ls in phase with PLL circuit. Pin
33to ground selects PAL or SECAm standard (17.7 mHz
main clock), and pin 33 to +5 V selects NTSC standard
(14.3 MHz main clock). After the main power supplies to
pins 8, 32 and 40 are stabilized, the start oscillator can
be used as a separate horizontal oscillator with a con-
stant frequency of 15.525 khz. For this option, pin 33
must be unconnected. By means ofthe lm bus register
SC the start oscillator can be switched on (SC = 0) or oft
(SC = 1). Setting SC =1 is recommended.
By means of pin 29 (horizontal output polarity selectin-
put and start oscillator pulsewidth select input), the out-
put pulsewidth and polarity ofthe start oscillator and pro-
tection circuit can be hardware-selected. Pin 29 at low
potential gives 30 us for DPU 2553 and 16 us for DPU
2554,with positive output pulses. Pin 29 at high potential
gives 36 us for DPU 2553 and 18 its for DPU 2554, with
negative output pulses. Both apply forthetime period in
which no start values are valid from the CCU. If pin 29
is intended to be in the high state, it must be connected
to pin 35 (standby power). Pin 29 must be connected to
ground or to +5 V in both cases.
Table 3-3: Operation modes ofthe start oscillator and
protection circuit


Operation Mode Pins
33 34 35 36
Horizontal output stage protected not connected 4 mHz Clock at

+5 V at ground
during main clock frequency changing
(for PAL and NTSC)
Horizontal output stage protected not connected 4 MHz Clock +5

V with connected to
and start oscillator function start oscilla- pin 35
(for PAL and NTSC) tor supply
Only start oscillator function with at +5 V 4 mHz Clock +5 V

with connected to
NTSC standard after Beset start oscilla- pin 35
tor supply
Only start oscillator function with at ground 4 mHz Clock +5 V

with connected to
PAL or SECAM standard after Beset start oscilla~ pin 35
5 tor supply
_ with 17.7 mHz clock at ground at ground at +5 V at ground
without protection.



3.5. Blanking and Color Key Pulses

Pin 19 supplies a combination ofthe color key pulse and
the undelayed horizontal blanking pulse in the form of a
three-level pulse as shown in Fig. 3-13. The high level
(4 V min.) and the low level (0.4 V max.) are controlled
by the DPU. During the low time of the undelayed hori-
zontal blanking pulse, pin 19 of the DPU i sin the high--
impedance mode and the output level at pin 19 is set to
2.8 V by the VCU.
At pin 22, the delayed horizontal blanking pulse in com-
bination with the vertical blanking pulse i
s available as
athree-level pulse as shown in Fig. 3-13. Output pin 22
is in high-impedance mode during the delayed horizon-
tal blanking pulse.
ln double-scan operation mode (DPU 2554), pin 22 sup-
plies the double-scan (2H) horizontal blanking pulse in-
stead ofthe 1H blanking pulse (DPU 2553). ln text dis-
play mode with increased deflection frequencies (see
section 1.), pin 22 ofthe respective DPU (DPU 2553, as
defined by register ZN) delivers the horizontal blanking
pulse with 18.7 kHz and the vertical blanking pulse with
60 Hz according to the display. At pin 24 the undelayed
horizontal blanking pulse is output.
normally,pin3suppliesthe samevertical blanking pulse
as pin 22. However, with“DVS” = 1, pin 3 will be in the
single-scan mode also with double-scan operation of
the system. The pulsewidth of the single-scan vertical
blanking pulse at pin 3 will be the same as.that of the
double-scan vertical blanking pulse at pin 22. The out-
put pulse of pin 3 is only valid if the COU register “VBE”
is set to 1 . The default value is set to 0 (high-impedance
state of pin 3).

Fig. 3-13: Shape of the output pulses at pins 19 and 22
*) The output level is externally defined
3.6. Output for Switching the Horizontal Power
Stage Between 15.6 kHz (PAL/NTSC) and 18 kHz
(Text Display)
This output (pin 37) is designed as a tristate output. High
levels (4 V mln.) and low levels (0.4 V max.) are con-
trolled bythe DPU. During high-impedance state an ex-
ternal resistor network defines the output level,
For changing the horizontal frequency from 15 kHz to
18 kHz, the following sequence of output levels is
derived at pin 37 (see Fig. 3-14).
After register ZN is set from ZN = 2 (15 kHz) to ZN = 0
(18 kHz) by the CCU, pin 37 is switched from High level
to high-impedance state synchronously with the fre-
quency change at pin 31. Following a delay of 20ms, pin
37 is set to Low level and remains in this state forthetime
the horizontal frequency remains 18 kHz (with ZN == 0).
This 20 ms delay is required for switching-over the hori-
zontal power stage.
To change the horizontal frequency in the opposite di-
rection, from 18 kHz to 15.6 kHz, the sequence de-
scribed is reversed.


3.7. Text Display Mode with Increased Deflection
Frequencies
As already mentioned, the DPU 2553 provides the fea-
ture of increased deflection frequencies for text display
for improved picture quality in this mode of operation. To
achieve this, the processor acting as deflection proces-
sor has its register Zn set to 0. The horizontal output fre-
quency at pin 31 is then switched to a frequency of
18746.802 Hz which is generated by dmding the Fm
main clock frequency by 946 i 46. The horizontal PLL is
then able to synchronize to an external composite sync
signal offH = 18.746 kHzi 46. The horizontal PLL isthen
able to synchronizeto an external composite sync signal
of fH = 18.746 kHzi 5 % and f\, = 60 Hz i 10 % and can
be set to an independent horizontal and vertical sync
generator by setting register VE = 1 and register VB = 0.
That means a constant dmder of 946 for horizontal fre-
quency and constant 312 lines per frame.

The DPU working in this mode supplies the TPU 2740
Teletext Processor or the respective Viewdata Proces-
sor with the 18.7 kHz horizontal blanking pulses form pin
24 and the 60 Hz vertical blanking pulses form pin 22
(see Fig. 3-8).
To be able to receive and store data from an IF video sig-
nal at the same time, the Teletext or Viewdata Processor
requires horizontal and vertical sync pulses from this IF
signal. Therefore, the second DPU provides video
clamping and sync separation forthe external signal and
supplies the horizontal sync pulses (pin 24) and the ver-
tical sync pulses (pin 22) to the Teletext or viewdata Pro-
cessor. For this, the second DPU is set to the PAL stan-
dard by register ZN = 2, and the clamping pulses of the
other DPU are disabled by CLD = 1.
To change the output frequency ofthe DPU acting as de-
flection processor from 18.7 kHz to 15.6 kHz, the control
switch output pin 37 prepares the horizontal output
stage for 15.6 khz operation (pin 37 is in the high-impe-
dance state) beforethe DPU changesthe horizontal out-
put frequencyto 15.6 kHz, after a minimu
m delay of one
vertical period. Switching the horizontal deflection fre-
quency from 15.6 kHzto 18.7 kHz is done in the reverse
sequence. Firstly, the horizontaloutput frequency of pin
31 is switched to 1 8.7 khz, and after a delay of one verti-
cal period, pin 37 is set low.
3.8. D2-MAC Operation Mode
When receiving Tv signals having the D2-mAC stan-
dard (direct satellite reception), register ZN is set to 3.
The programmable dmder is set to a dmsion ratio of
1296 i48 to generate a horizontal frequency of 15.625
khz with the clock rate of 20.25 mHz used in the
D2-mAC standard. ln this operation mode, pin 6 acts as
input forthe composite sync signal supplied by the DmA
2271 D2-mAC Decoder. The DPU is synchronized to
this sync signal, and after locking-in (status register
UN = 0), the CCU switches the DPU to a clock-locked
mode between clock signal and horizontal frequency
(fm main
clock by 1024, during the vertical sync signal separated
from the received video signal. To use an 8-bit register,
the result of the count is dmded by 2 and given to the
DPU status register. ln the CCU, the vertical frequency
can be evaluated using the following equation:

fv I __lL1’_l\
1024- vP- 2
with
fm), = 17.734475 mHz with PAL and SECAm
fq,M =14.31818 mHz with NTSC
rw = 2o_25 MHZ with D2-mAc
VP = status value, read from DPU.

The interlace control output pin 39 supplies a 25 Hz (for
PAL and SECAm) or 80 Hz (for NTSC) signal for control-
ling an external interlace-off switch, which is required
with A.C.-coupled vertical output stages, becausethese
are not able to handle the internal interlace-off proce-
dure using register “ZS”.
For operation with the vmC Processor the DPU 2554
hasthree interlace control modes in double vertical scan
mode (DVS = 1). These options can be selected with the
register “IOP” and can be used together with the control
output pin 39 only. This output has to be connected to the
vertical output stage, so that the vertical phase can be
shifted by 16 us (or 32 us with DPU 2553).









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