CGE M3788 TVC 15" CHASSIS F860
Power supply Description based on TDA4601d (SIEMENS)
TDA4601 Operation. * The TDA4601 device is a single in line, 9 pin chip. Its predecessor was the TDA4600 device, the TDA4601 however has improved switching, better protection and cooler running. The (SIEMENS) TDA4601 power supply is a fairly standard parallel chopper switch mode type, which operates on the same basic principle as a line output stage. It is turned on and off by a square wave drive pulse, when switched on energy is stored in the chopper transformer primary winding in the form of a magnetic flux; when the chopper is turned off the magnetic flux collapses, causing a large back emf to be produced. At the secondary side of the chopper transformer this is rectified and smoothed for H.T. supply purposes. The advantage of this type of supply is that the high chopping frequency (20 to 70 KHz according to load) allows the use of relatively small H.T. smoothing capacitors making smoothing easier. Also should the chopper device go short circuit there is no H.T. output. In order to start up the TDA4601 I.C. an initial supply of 9v is required at pin 9, this voltage is sourced via R818 and D805 from the AC side of the bridge rectifier D801, also pin 5 requires a +Ve bias for the internal logic block. (On some sets pin 5 is used for standby switching). Once the power supply is up and running, the voltage on pin 9 is increased to 16v and maintained at this level by D807 and C820 acting as a half wave rectifier and smoothing circuit. PIN DESCRIPTIONS Pin 1 This is a 4v reference produced within the I.C. Pin 2 This pin detects the exact point at which energy stored in the chopper transformer collapses to zero via R824 and R825, and allows Q1 to deliver drive volts to the chopper transistor. It also opens the switch at pin 4 allowing the external capacitor C813 to charge from its external feed resistor R810. Pin 3 H.T. control/feedback via photo coupler D830. The voltage at this pin controls the on time of the chopper transistor and hence the output voltage. Normally it runs at Approximately 2v and regulates H.T. by sensing a proportion of the +4v reference at pin 1, offset by conduction of the photo coupler D830 which acts like a variable resistor. An increase in the conduction of transistor D830 and therefor a reduction of its resistance will cause a corresponding reduction of the positive voltage at Pin 3. A decrease in this voltage will result in a shorter on time for the chopper transistor and therefor a lowering of the output voltage and vice versa, oscillation frequency also varies according to load, the higher the load the lower the frequency etc. should the voltage at pin 3 exceed 2.3v an internal flip flop is triggered causing the chopper drive mark space ratio to extend to 244 (off time) to 1 (on time), the chip is now in over volts trip condition. Pin 4 At this pin a sawtooth waveform is generated which simulates chopper current, it is produced by a time constant network R810 and C813. C813 charges when the chopper is on and is discharged when the chopper is off, by an internal switch strapping pin 4 to the internal +2v reference, see Fig 2. The amplitude of the ramp is proportional to chopper drive. In an overload condition it reaches 4v amplitude at which point chopper drive is reduced to a mark-space ratio of 13 to 1, the chip is then in over current trip. The I.C. can easily withstand a short circuit on the H.T. rail and in such a case the power supply simply squegs quietly. Pin 4 is protected by internal protection components which limit the maxi
mum voltage at this pin to 6.5v. Should a fault occur in either of the time constant components, then the chopper transistor will probably be destroyed. Pin 5 This pin can be used for remote control on/off switching of the power supply, it is normally held at about +7v and will cause the chip to enter standby mode if it falls below 2v. Pin 6 Ground. Pin 7 Chopper switch off pin. This pin clamps the chopper drive voltage to 1.6v in order to switch off the chopper. Pin 8 Chopper base current output drive pin. Pin 9 L.T. pin, approximately 9v under start-up conditions and 16v during normal running, Current consumption of the I.C. is typically 135mA. The voltage at this pin must reach 6.7v in order for the chip to start-up.
Semiconductor circuit for supplying power to electrical equipment, comprising a transformer having a primary winding connected, via a parallel connection of a collector-emitter path of a transistor with a first capacitor, to both outputs of a rectifier circuit supplied, in turn, by a line a-c voltage; said transistor having a base controlled via a second capacitor by an output of a control circuit acted upon, in turn by the rectified a-c line voltage as actual value and by a reference voltage; said transformer having a first secondary winding to which the electrical equipment to be supplied is connected; said transformer having a second secondary winding with one terminal thereof connected to the emitter of said transistor and the other terminal thereof connected to an anode of a first diode leading to said control circuit; said transformer having a third secondary winding with one terminal thereof connected, on the one hand, via a series connection of a third capacitor with a first resistance, to the other terminal of said third secondary winding and connected, on the other hand, to the emitter of said transistor, the collector of which is connected to said primary winding; a point between said third capacitor and said first resistance being connected to the cathode of a second diode; said control circuit having nine terminals including a first terminal delivering a reference voltage and connected, via a voltage divider formed of a third and fourth series-connected resistances, to the anode of said second diode; a second terminal of said control circuit serving for zero-crossing identification being connected via a fifth resistance to said cathode of said second diode; a third terminal of said control-circuit serving as actual value input being directly connected to a divider point of said voltage divider forming said connection of said first terminal of said control circuit to said anode of said second diode; a fourth terminal of said control circuit delivering a sawtooth voltage being connected via a sixth resistance to a terminal of said primary winding of said transformer facing away from said transistor; a fifth terminal of said control circuit serving as a protective input being connected, via a seventh resistance to the cathode of said first diode and, through the intermediary of said seventh resistance and an eighth resistance, to the cathode of a third diode having an anode connected to an input of said rectifier circuit; a sixth terminal of said control circuit carrying said reference potential and being connected via a fourth capacitor to said fourth terminal of said control circuit and via a fifth capacitor to the anode of said second diode; a seventh terminal of said control circuit establishing a potential for pulses controlling said transistor being connected directly and an eighth terminal of said control circuit effecting pulse control of the base of said transistor being connected through the intermediary of a ninth resistance to said first capacitor leading to the base of said transistor; and a ninth terminal of said control circuit serving as a power supply input of said control circuit being connected both to the cathode of said first diode as well as via the intermediary of a sixth capacitor to a terminal of said second secondary winding as well as to a terminal of said third secondary winding.
Such a blocking oscillator switching power supply is described in the German periodical, "Funkschau" (1975) No. 5, pages 40 to 44. It is well known that the purpose of such a circuit is to supply electronic equipment, for example, a television set, with stabilized and controlled supply voltages. Essential for such switching power supply is a power switching transistor i.e. a bipolar transistor with high switching speed and high reverse voltage. This transistor therefore constitutes an important component of the control element of the control circuit. Furthermore, a high operating frequency and a transformer intended for a high operating frequency are provided, because generally, a thorough separation of the equipment to be supplied from the supply naturally is desired. Such switching power supplies may be constructed either for synchronized or externally controlled operation or for non-synchronized or free-running operation. A blocking converter is understood to be a switching power supply in which power is delivered to the equipment to be supplied only if the switching transistor establishing the connection between the primary coil of the transformer and the rectified a-c voltage is cut off. The power delivered by the line rectifier to the primary coil of the transformer while the switching transistor is open, is interim-stored in the transformer and then delivered to the consumer on the secondary side of the transformer with the switching transistor cut off.
In the blocking converter described in the aforementioned reference in the literature, "Funkschau" (1975), No. 5, Pages 40 to 44, the power switching transistor is connected in the manner defined in the introduction to this application. In addition, a so-called starting circuit is provided. Because several diodes are generally provided in the overall circuit of a blocking oscillator according to the definition provided in the introduction hereto, it is necessary, in order not to damage these diodes, that due to the collector peak current in the case of a short circuit, no excessive stress of these diodes and possibly existing further sensitive circuit parts can occur.
Considering the operation of a blocking oscillator, this means that, in the event of a short circuit, the number of collector current pulses per unit time must be reduced. For this purpose, a control and regulating circuit is provided. Simultaneously, a starting circuit must bring the blocking converter back to normal operation when the equipment is switched on, and after disturbances, for example, in the event of a short circuit. The starting circuit shown in the literature reference "Funkschau" on Page 42 thereof, differs to some extent already from the conventional d-c starting circuits. It is commonly known for all heretofore known blocking oscillator circuits, however, that a thyristor or an equivalent circuit replacing the thyristor is essential for the operation of the control circuit.
It is accordingly an object of the invention to provide another starting circuit. It is a further object of the invention to provide a possible circuit for the control circuit which is particularly well suited for this purpose. It is yet another object of the invention to provide such a power supply which is assured of operation over the entire range of line voltages from 90 to 270 V a-c, while the secondary voltages and secondary load variations between no-load and short circuit are largely constant.
With the foregoing and other objects in view, there is provided, in accordance with the invention, a blocking oscillator-type switching power supply for supplying power to electrical equipment wherein a primary winding of a transformer, in series with an emitter-collector path of a first bipolar transistor, is connected to a d-c voltage obtained by rectification of a line a-c voltage fed-in via two external supply terminals, a secondary winding of the transformer being connectible to the electrical equipment for supplying power thereto, the first bipolar transistor having a base controlled by the output of a control circuit acted upon, in turn, by the rectified a-c line voltage as actual value and by a set-point transmitter, and including a starting circuit for further control of the base of the first bipolar transistor, including a first diode in the starting circuit having an anode directly connected to one of the supply terminals supplied by the a-c line voltage and a cathode connected via a resistor to an input serving to supply power to the control circuit, the input being directly connected to a cathode of a second diode, the second diode having an anode connected to one terminal of another secondary winding of the transformer, the other secondary winding having another terminal connected to the emitter of the first bipolar transmitter.
In accordance with another feature of the invention, there is provided a second bipolar transistor having the same conduction type as that of the first bipolar transistor and connected in the starting circuit with the base thereof connected to a cathode of a semiconductor diode, the semiconductor diode having an anode connected to the emitter of the first bipolar transistor, the second bipolar transistor having a collector connected via a resistor to a cathode of the first diode in the starting circuit, and having an emitter connected to the input serving to supply power to the control circuit and also connected to the cathode of the second diode which is connected to the other secondary winding of the transformer.
In accordance with a further feature of the invention, the base of the second bipolar transistor is connected to a resistor and via the latter to one pole of a first capacitor, the anode of the first diode being connected to the other pole of the first capacitor.
In accordance with an added feature of the invention, the input serving to supply power to the control circuit is connected via a second capacitor to an output of a line rectifier, the output of the line rectifier being directly connected to the emitter of the first bipolar transistor.
In accordance with an additional feature of the invention, the other secondary winding is connected at one end to the emitter of the first bipolar transistor and to a pole of a third capacitor, the third capacitor having another pole connected, on the one hand, via a resistor, to the other end of the other secondary winding and, on the other hand, to a cathode of a third diode, the third diode having an anode connected via a potentiometer to an actual value input of the control circuit and, via a fourth capacitor, to the emitter of the first bipolar transistor.
In accordance with yet another feature of the invention, the control circuit has a control output connected via a fifth capacitor to the base of the first bipolar transistor for conducting to the latter control pulses generated in the control circuit.
In accordance with a concomitant feature of the invention, there is provided a sixth capacitor shunting the emitter-collector path of the first transistor.
Other features which are considered as characteristic for the invention are set forth in the appended claim.
The construction and method of operation of the invention, however, together with additional objects and advantages thereof will be best understood from the following description of specific embodiments when read in connection with the accompanying drawings, in which:
FIGS. 1 and 2 are circuit diagrams of the blocking oscillator type switching power supply according to the invention; and
FIG. 3 is a circuit diagram of the control unit RS of FIGS. 1 and 2.
Referring now to the drawing and, first, particularly to FIG. 1 thereof, there is shown a rectifier circuit G in the form of a bridge current, which is acted upon by a line input represented by two supply terminals 1' and 2'. Rectifier outputs 3' and 4' are shunted by an emitter-collector path of an NPN power transistor T1 i.e. the series connection of the so-called first bipolar transistor referred to hereinbefore with a primary winding I of a transformer Tr. Together with the inductance of the transformer Tr, the capacitance C1 determines the frequency and limits the opening voltages of the switch embodied by the first transistor T1. A capacitance C2, provided between the base of the first transistor T1 and the control output 7,8 of a control circuit RS, separates the d-c potentials of the control or regulating circuit RS and the switching transistor T1 and serves for addressing this switching transistor T1 with pulses. A resistor R1 provided at the control output 7,8 of the control circuit RS is the negative-feedback resistor of both output stages of the control circuit RS. It determines the maximally possible output pulse current of the control circuit RS. A secondary winding II of the transformer Tr takes over the power supply of the control circuit, in steady state operation, via the diode D1. To this end, the cathode of this diode D1 is directly connected to a power supply input 9 of the control circuit RS, while the anode thereof is connected to one terminal of the secondary winding II. The other terminal of the secondary winding II is connected to the emitter of the power switching transistor T1.
The cathode of the diode D1 and, therewith, the power supply terminal 9 of the control circuits RS are furthermore connected to one pole of a capacitor C3, the other pole of which is connected to the output 3' of the rectifier G. The capacitance of this capacitor C3 thereby smoothes the positive half-wave pulses and serves simultaneously as an energy storage device during the starting period. Another secondary winding III of the transformer Tr is connected by one of the leads thereof likewise to the emitter of the first transistor T1, and by the other lead thereof via a resistor R2, to one of the poles of a further capacitor C4, the other pole of which is connected to the first-mentioned lead of the other secondary winding III. This second pole of the capacitor C4 is simultaneously connected to the output 3' of the rectifier circuit G and, thereby, via the capacitor C3, to the cathode of the diode D1 driven by the secondary winding II of the transformer Tr as well as to the power supply input 9 of the control circuit RS and, via a resistor R9, to the cathode of a second diode D4. The second pole of the capacitor C4 is simultaneously connected directly to the terminal 6 of the control circuit RS and, via a further capacitor C 6, to the terminal 4 of the control circuit RS as well as, additionally, via the resistor R6, to the other output 4' of the rectifier circuit G. The other of the poles of the capacitor C4 acted upon by the secondary winding II is connected via a further capacitor C5 to a node, which is connected on one side thereof, via a variable resistor R4, to the terminals 1 and 3 of the control circuit RS, with the intermediary of a fixed resistor R5 in the case of the terminal 1. On the other side of the node, the latter and, therefore, the capacitor C5 are connected to the anode of a third diode D2, the cathode of which is connected on the one hand, to the resistor R2 mentioned hereinbefore and leads to the secondary winding III of the transformer Tr and, on the other hand, via a resistor R3 to the terminal 2 of the control circuit RS.
The nine terminals of the control circuit RS have the following purposes or functions:
Terminal 1 supplies the internally generated reference voltage to ground i.e. the nominal or reference value required for the control or regulating process;
Terminal 2 serves as input for the oscillations provided by the secondary winding III, at the zero point of which, the pulse start of the driving pulse takes place;
Terminal 3 is the control input, at which the existing actual value is communicated to the control circuit RS, that actual value being generated by the rectified oscillations at the secondary winding III;
Terminal 4 is responsive to the occurrence of a maximum excursion i.e. when the largest current flows through the first transistor T1 ;
Terminal 5 is a protective input which responds if the rectified line voltage drops too sharply; Terminal 6 serves for the power supply of the control process and, indeed, as ground terminal;
Terminal 7 supplies the d-c component required for charging the coupling capacitor C2 leading to the base of the first transistor T1 ;
Terminal 8 supplies the control pulse required for the base of the first transistor T1 ; and
Terminal 9 serves as the first terminal of the power supply of the control circuit RS.
Further details of the control circuit RS are described hereinbelow.
The capacity C3 smoothes the positive half-wave pulses which are provided by the secondary winding II, and simultaneously serves as an energy storage device during the starting time. The secondary winding III generates the control voltage and is simultaneously used as feedback. The time delay stage R2 /C4 keeps harmonics and fast interference spikes away from the control circuit RS. The resistor R3 is provided as a voltage divider for the second terminal of the control circuit RS. The diode D2 rectifies the control pulses delivered by the secondary winding III. The capacity C5 smoothes the control voltage. A reference voltage Uref, which is referred to ground i.e. the potential of terminal 6 is present at the terminal 1 of the control circuit RS. The resistors R4 and R5 form a voltage divider of the input-difference control amplifier at the terminal 3. The desired secondary voltage can be set manually via the variable resistor R4. A time-delay stage R6 /C6 forms a sawtooth rise which corresponds to the collector current rise of the first bipolar transistor T1 via the primary winding I of the transformer Tr. The sawtooth present at the terminal 4 of the control circuit RS is limited there between the reference voltage 2 V and 4 V. The voltage divider R7 /R8 (FIG. 2), brings to the terminal 5 of the control circuit RS the enabling voltage for the drive pulse at the output 8 of the control circuit RS.
The diode D4, together with the resistor R9 in cooperation with the diode D1 and the secondary winding II, forms the starting circuit provided, in accordance with the invention. The operation thereof is as follows:
After the switching power supply is switched on, d-c voltages build up at the collector of the switching transistor T1 and at the input 4 of the control circuit RS, as a function in time of the predetermined time constants. The positive sinusoidal half-waves charge the capacitor C3 via the starting diode D4 and the starting resistor R9 in dependence upon the time constant R9.C3. Via the protective input terminal 5 and the resistor R11 not previously mentioned and forming the connection between the resistor R9 and the diode D1, on the one hand, and the terminal 5 of the control circuit RS, on the other hand, the control circuit RS is biased ready for switching-on, and the capacitor C2 is charged via the output 7. When a predetermined voltage value at the capacitor C3 or the power supply input 9 of the control circuit RS, respectively, is reached, the reference voltage i.e. the nominal value for the operation of the control voltage RS, is abruptly formed, which supplies all stages of the control circuit and appears at the output 1 thereof. Simultaneously, the switching transistor T1 is switched into conduction via the output 8. The switching of the transistor T1 at the primary winding T of the transformer Tr is transformed to the second secondary winding II, the capacity C3 being thereby charged up again via the diode D1. If sufficient energy is stored in the capacitor C3 and if the re-charge via the diode D1 is sufficient so that the voltage at a supply input 9 does not fall below the given minimum operating voltage, the switching power supply then remains connected, so that the starting process is completed. Otherwise, the starting process described is repeated several times.
In FIG. 2, there is shown a further embodiment of the circuit for a blocking oscillator type switching power supply, according to the invention, as shown in FIG. 1. Essential for this circuit of FIG. 2 is the presence of a second bipolar transistor T2 of the type of the first bipolar transistor T1 (i.e. in the embodiments of the invention, an npn-transistor), which forms a further component of the starting circuit and is connected with the collector-emitter path thereof between the resistor R9 of the starting circuit and the current supply input 9 of the control circuit RS. The base of this second transistor T2 is connected to a node which leads, on the one hand, via a resistor R10 to one electrode of a capacitor C7, the other electrode of which is connected to the anode of the diode D4 of the starting circuit and, accordingly, to the terminal 1' of the supply input of the switching power supply G. On the other hand, the last-mentioned node and, therefore, the base of the second transistor T2 are connected to the cathode of a Zener diode D3, the anode of which is connected to the output 3' of the rectifier G and, whereby, to one pole of the capacitor C3, the second pole of which is connected to the power supply input 9 of the control circuit RS as well as to the cathode of the diode D1 and to the emitter of the second transistor T2. In other respects, the circuit according to FIG. 2 corresponds to the circuit according to FIG. 1 except for the resistor R11 which is not necessary in the embodiment of FIG. 2, and the missing connection between the resistor R9 and the cathode of the diode D1, respectively, and the protective input 5 of the control circuit RS.
Regarding the operation of the starting circuit according to FIG. 2, it can be stated that the positive sinusoidal half-wave of the line voltage, delayed by the time delay stage C7, R10 drives the base of the transistor T2 in the starting circuit. The amplitude is limited by the diode D3 which is provided for overvoltage protection of the control circuit RS and which is preferably incorporated as a Zener diode. The second transistor T2 is switched into conduction. The capacity C3 is charged, via the serially connected diode D4 and the resistor R9 and the collector-emitter path of the transistor T2, as soon as the voltage between the terminal 9 and the terminal 6 of the control circuit RS i.e. the voltage U9, meets the condition U9 <[UDs -UBE (T2)].
Because of the time constant R9.C3, several positive half-waves are necessary in order to increase the voltage U9 at the supply terminal 9 of the control circuit RS to such an extent that the control circuit RS is energized. During the negative sine half-wave, a partial energy chargeback takes place from the capacitor C3 via the emitter-base path of the transistor T2 of the starting circuit and via the resistor R10 and the capacitor C7, respectively, into the supply network. At approximately 2/3 of the voltage U9, which is limited by the diode D3, the control circuit RS is switched on. At the terminal 1 thereof, the reference voltage Uref then appears. In addition, the voltage divider R5 /R4 becomes effective. At the terminal 3, the control amplifier receives the voltage forming the actual value, while the first bipolar transistor T1 of the blocking-oscillator type switching power supply is addressed pulsewise via the terminal 8.
Because the capacitor C6 is charged via the resistor R6, a higher voltage than Uref is present at the terminal 4 if the control circuit RS is activated. The control voltage then discharges the capacitor C6 via the terminal 4 to half the value of the reference voltage Uref, and immediately cuts off the addressing input 8 of the control circuit RS. The first driving pulse of the switching transistor T1 is thereby limited to a minimum of time. The power for switching-on the control circuit RS and for driving the transistor T1 is supplied by the capacitor C3. The voltage U9 at the capacitor C3 then drops. If the voltage U9 drops below the switching-off voltage value of the control circuit RS, the latter is then inactivated. The next positive sine half-wave would initiate the starting process again.
By switching the transistor T1, a voltage is transformed in the secondary winding II of the transformer Tr. The positive component is rectified by the diode D1, recharing of the capacitor C3 being thereby provided. The voltage U9 at the output 9 does not, therefore, drop below the minimum value required for the operation of the control circuit RS, so that the control circuit RS remains activated. The power supply continues to operate in the rhythm of the existing conditions. In operation, the voltage U9 at the supply terminal 9 of the control circuit RS has a value which meets the condition U9 >[UDs -UBE (T2)], so that the transistor T2 of the starting circuit remains cut off.
For the internal layout of the control circuit RS, the construction shown, in particular, from FIG. 3 is advisable. This construction is realized, for example, in the commercially available type TDA 4600 (Siemens AG).
The block diagram of the control circuit according to FIG. 3 shows the power supply thereof via the terminal 9, the output stage being supplied directly whereas all other stages are supplied via Uref. In the starting circuit, the individual subassemblies are supplied with power sequentially. The d-c output voltage potential of the base current gain i.e. the voltage for the terminal 8 of the control circuit RS, and the charging of the capacitor C2 via the terminal 7 are formed even before the reference voltage Uref appears. Variations of the supply voltage U9 at terminal 9 and the power fluctuations at the terminal 8/terminal 7 and at the terminal 1 of the control circuit RS are leveled or smoothed out by the voltage control. The temperature sensitivity of the control circuit RS and, in particular, the uneven heating of the output and input stages and input stages on the semiconductor chip containing the control circuit in monolithically integrated form are intercepted by the temperature compensation provided. The output values are constant in a specific temperature range. The message for blocking the output stage, if the supply voltage at the terminal 9 is too low, is given also by this subassembly to a provided control logic.
The outer voltage divider of the terminal 1 via the resistors R5 and R4 to the control tap U forms, via terminal 3, the variable side of the bridge for the control amplifier formed as a differential amplifier. The fixed bridge side is formed by the reference voltage Uref via an internal voltage divider. Similarly formed are circuit portions serving for the detection of an overload short circuit and circuit portions serving for the "standby" no-load detection, which can be operated likewise via terminal 3.
Within a provided trigger circuit, the driving pulse length is determined as a function of the sawtooth rise at the terminal 4, and is transmitted to the control logic. In the control logic, the commands of the trigger circuit are processed. Through the zero-crossing identification at input 2 in the control circuit RS, the control logic is enabled to start the control input only at the zero point of the frequency oscillation. If the voltages at the terminal 5 and at the terminal 9 are too low, the control logic blocks the output amplifier at the terminal 8. The output amplifier at the terminal 7 which is responsible for the base charge in the capacitor C2, is not touched thereby.
The base current gain for the transistor T1 i.e. for the first transistor in accordance with the definition of the invention, is formed by two amplifiers which mutually operate on the capacitor C2. The roof inclination of the base driving current for the transistor T1 is impressed by the collector current simulation at the terminal 4 to the amplifier at the terminal 8. The control pulse for the transistor T1 at the terminal 8 is always built up to the potential present at the terminal 7. The amplifier working into the terminal 7 ensures that each new switching pulse at the terminal 8 finds the required base level at terminal 7.
Supplementing the comments regarding FIG. 1, it should also be mentioned that the cathode of the diode D1 connected by the anode thereof to the one end of the secondary winding II of the transformer Tr is connected via a resistor R11 to the protective input 5 of the control circuit RS whereas, in the circuit according to FIG. 2, the protective input 5 of the control circuit RS is supplied via a voltage divider R8, R7 directly from the output 3', 4' of the rectifier G delivering the rectified line a-c voltage, and which obtains the voltage required for executing its function. It is evident that the first possible manner of driving the protective input 5 can be used also in the circuit according to FIG. 2, and the second possibility also in a circuit in accordance with FIG. 1.
The control circuit RS which is shown in FIG. 3 and is realized in detail by the building block TDA 4600 and which is particularly well suited in conjunction with the blocking oscillator type switching power supply according to the invention has 9 terminals 1-9, which have the following characteristics, as has been explained in essence hereinabove:
Terminal 1 delivers a reference voltage Uref which serves as the constant-current source of a voltage divider R5.R4 which supplies the required d-c voltages for the differential amplifiers provided for the functions control, overload detection, short-circuit detection and "standby"-no load detection. The dividing point of the voltage divider R5 -R4 is connected to the terminal 3 of the control circuit RS. The terminal 3 provided as the control input of RS is controlled in the manner described hereinabove as input for the actual value of the voltage to be controlled or regulated by the secondary winding III of the transformer Tr. With this input, the lengths of the control pulses for the switching transistor T1 are determined.
Via the input provided by the terminal 2 of the control circuit RS, the zero-point identification in the control circuit is addressed for detecting the zero-point of the oscillations respectively applied to the terminal 2. If this oscillation changes over to the positive part, then the addressing pulse controlling the switching transistor T1 via the terminal 8 is released in the control logic provided in the control circuit.
A sawtooth-shaped voltage, the rise of which corresponds to the collector current of the switching transistor T1, is present at the terminal 4 and is minimally and maximally limited by two reference voltages. The sawtooth voltage serves, on the one hand as a comparator for the pulse length while, on the other hand, the slope or rise thereof is used to obtain in the base current amplification for the switching transistor T1, via the terminal 8, a base drive of this switching transistor T1 which is proportional to the collector current.
The terminal 7 of the control circuit RS as explained hereinbefore, determines the voltage potential for the addressing pulses of the transistor T2. The base of the switching transistor T1 is pulse-controlled via the terminal 8, as described hereinbefore. Terminal 9 is connected as the power supply input of the control circuit RS. If a voltage level falls below a given value, the terminal 8 is blocked. If a given positive value of the voltage level is exceeded, the control circuit is activated. The terminal 5 releases the terminal 8 only if a given voltage potential is present.
DE2417628A1 1975-10-23 363/37
DE2638225A1 1978-03-02 363/49
Grundig Tech. Info. (Germany), vol. 28, No. 4, (1981).
IBM Technical Disclosure Bulletin, vol. 19, No. 3, pp. 978, 979, Aug. 1976.
German Periodical, "Funkschau", (1975), No. 5, pp. 40 to 44.
The TDA8185I is a monolithic integrated circuit in
24 pins dual in line plastic package intended for TV
signal processing and driving Horizontal and Vertical
output stages. It was specially designed for
VCR working conditions.
503kHz REFERENCE OSCILLATOR .5.5V SUPPLY VOLTAGE INTERNALLY
REGULATED .VERY SOPHISTICATED SYNC. SEPARATOR .
COUNTDOWN TIMING LOGIC .ADAPTS AUTOMATICALLY TO 625
LINE/50Hz AND 525 LINE/60Hz STANDARDS .50/60 Hz IDENTIFICATION OUTPUT .AUTOMATIC VERTICAL AMPLITUDE CORRECTION
50/60Hz .CRT PROTECTION CIRCUIT .PHASE-CORRECTED HORIZONTAL OUTPUT
WITH CONSTANT DUTY CYCLE
- Video chrominance and Luminance with TDA3562A
.CHROMINANCE SIGNALPROCESSOR .LUMINANCE SIGNAL PROCESSING WITH
CLAMPING .HORIZONTAL AND VERTICAL BLANKING .LINEAR TRANSMISSION OF INSERTED
RGB SIGNALS .LINEAR CONTRAST AND BRIGHTNESS
CONTROL ACTING ON INSERTED AND MATRIXED
SIGNALS .AUTOMATIC CUT-OFF CONTROL .NTSC HUE CONTROL
The TDA3562A is a monolithic IC designed as
decode PAL and/or NTSC colour television standards
and it combines all functions required for the
identification and demodulation of PAL and NTSC
TDA8140 HORIZONTAL DEFLECTION POWER DRIVER
The TDA 8140 is a monolithic integrated circuit designed
to drive the horizontal deflectionpower transistor.
The current source characteristic of this device is
adapted to the on-linear current gain behaviour of
the power transistor providing a minimum power
dissipation. The TDA8140 is internally protected
against short circuit and thermal overload.
During the active deflection phase the collector
current of the power transistor is linearly rising and
the driving circuitry mustbe adaptedto the required
base current in order to ensure the power transistor
According to the limited components number the
typical approach of the present TVs provides only
a rough approximation of this objective ; in Figure 5
wegive a comparisonbetweenthe typical real base
current and the ideal base current waveform and
the collector waveform.
The marked area represents a useless base current
which gives an additional power dissipation on
the power transistor.
Furthermoreduring the turn-ONand turn-OFFtransient
phase of the chassis the power transistor is
extremely stressed when the conventionalnetwork
cannot guarantee the saturation ; for this reason,
generally, the driving circuit must be carefully designed
and is different for each deflection system.
The new approach, using the TDA 8140, overcomes
these restrictions by means of a feedback
As shown in Figure 5, at each instant of time the
ideal base current of the power transistor results
from its collector current divided by such current
gain which ensure the saturation ; thus the required
base current Ib can be easily generated by a feedback
transconductanceamplifier gm which senses
the deflection current across the resistor Rs at the
emitter of the power transistor and delivers :
Ib = RS . gm . Ie
The transconductance must only fulfill the condition
1 + bmin V 1
Where bmin is the minimum current gain of the
transistor. This method always ensures the correct
base current and acts time independent on principle.For the turn-OFF, the base of the power transistor
must be discharged by a quasi linear time decreasing
current as given in Figure 6.
Conventional driver systems inherently result into
a stable condition with a constant peak current
This is due to the constant base charge in the
turn-ON phase independent from the collector current
; hence a high peak current results into a low
storage time of the transistor because the excess
base charge is a minimum and vice versa. In the
active deflection the required function, high peak
current-fast switch-OFF and low peak current-slow
switch-OFF, is obtained by a controlled base discharge
current for the power transistor ; the negative
slope of this ramp is proportional to the actual
As a result, the active driving system even improves
the sharpnessof vertical lines on the screen
compared with the traditional solution due to the
increasedstability factor of the loop representedas
the variation of the storagetime versus the collector
TDA8170 TV VERTICAL DEFLECTION OUTPUT CIRCUITThe functions incorporated are :
The TDA8170 is a monolithic integrated circuit in
HEPTAWATTTM package. It is a high efficiency
power booster for direct driving of verticalwindings
of TV yokes. It is intended for use in Colour and B
&Wtelevision receivers as well as in monitorsand